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FP7-ICT-2009-5 WiserBAN (257454) D3.3v0.2

This project is funded by the European Commission under the 7th Research Framework Programme.

WiserBAN

Project Acronym: WiserBAN

Project Title: Smart miniature low-power wireless microsystem for Body Area Networks

Call: FP7-ICT-2009-5, Collaborative project

Grant Agreement no.: 257454

Project Duration: 36 months

Coordinator: CSEM

Beneficiaries:

CSEM Centre Suisse D’Electronique et de Microtechnique SA –

Recherche et Development

CSEM CH

Commissariat a L’Energie Atomique et aux Energies Alternatives CEA FR

Fraunhofer-Gesellschaft zur Foerderung der Angewandten

Forschung E.V.

FRAUNHOFER DE

Valtion Teknillinen Tutkimuskeskus VTT FI

Technische Universitat Berlin TUB DE

Alma Mater Studiorum-Universita di Bologna UNIBO IT

Sorin CRM SAS SORIN FR

EPCOS SAS EPCOS FR

MED-EL Elektromedizinische Geraete GmbH MED-EL AT

Siemens Audiologische Technik GmbH DE-SAT DE

Debiotech S.A. DEBIOTECH CH

SignalGenerix Ltd SG CY

RTD TALOS Ltd TALOS CY

FP7-ICT-2009-5 WiserBAN (257454) D3.3.v0.2

2

WiserBAN

Smart miniature low-power wireless microsystem for

Body Area Networks

WP Number: 3

Deliverable identifier: D3.3

Deliverable title: Final Active and Passive Antennas Ready for BAN Node Packaging and

Integration

Due date of the deliverable:

Actual submission date to the EC:

Organization name of lead partner for this Document (partner name): VTT

Author(s): J. Aurinsalo (VTT), A. Lamminen (VTT), A. Rantala (VTT), A. Vorobyov

(CSEM), R. D’Errico (CEA), K. Michaelides (SG)

Project funded by the European Commission within the Seventh Framework

Programme

Dissemination Level

PU Public

PP Restricted to other programme participants (including the Commission Services)

RE Restricted to a group specified by the consortium (including the Commission Services)

CO Confidential, only for members of the consortium (including the Commission Services) X


3

Revision History

Version Date Changed page(s) Cause of change Partner

0.1 26.06.2014 Template created VTT

0.2

02.09.2014 Contributions of VTT,

CEA, CSEM and SG

added

VTT

Disclaimer: The information in this document is subject to change without notice. Company

or product names mentioned in this document may be trademarkers or registered trademarks

of their respective companies.

All rights reserved.

The document is proprietary of the WiserBAN consortium members. No copying or

distributing, in any form or by any means is allowed without the prior written agreement of

the owner of the property rights.

Τhis document reflects the authors’ view. The European Community is not liable for any use

that may be made of the information contained herein.


4

Contents

Executive Summary ................................................................................................................... 7

1 Introduction ........................................................................................................................ 8

2 L antenna for BTE hearing aid ............................................................................................. 9

2.1 Antenna in free space .................................................................................................. 9

2.2 Active antenna in the BTE hearing aid ....................................................................... 11

2.3 Head influence on the active antenna in the BTE hearing aid ................................... 15

2.4 Active antenna with realistic matching network ....................................................... 19

3 Antenna impedance sensing and tuning ........................................................................... 21

Introduction ..................................................................................................................... 21

3.1 AIST hardware ............................................................................................................ 21

3.1.1 Antenna impedance tuning ................................................................................. 21

3.1.2 Antenna impedance sensing ............................................................................... 23

3.1.3 AIST realisation.................................................................................................... 24

3.2 AIST characterization ................................................................................................. 25

3.2.1 Characterization platform ................................................................................... 25

3.2.2 Experimental results ........................................................................................... 27

3.3 AIST software ............................................................................................................. 30

3.3.1 Introduction ........................................................................................................ 30

3.3.2 Principle of Operation ......................................................................................... 31

3.3.3 AIST Tuning Algorithms ....................................................................................... 31

4 Miniature frequency agile antenna .................................................................................. 38

4.1 Reminder of the antenna design ............................................................................... 38

4.1.1 Design and performances ................................................................................... 38

4.1.2 Considerations on antenna miniaturization ....................................................... 40

4.2 Antenna Impedance characterization ........................................................................ 42


5

4.2.1 Cable effect on impedance characteristics ......................................................... 42

4.2.2 Impedance measurement methodology and results .......................................... 48

4.3 Antenna radiation characterization ........................................................................... 51

4.3.1 Cable effect on radiation characteristics ............................................................ 52

4.3.2 Radiation measurement methodology and results ............................................ 54

4.4 Antenna integration ................................................................................................... 60

4.4.1 Modified antenna design for varactor integration ............................................. 60

4.4.2 SiP integration ..................................................................................................... 62

4.4.3 Integration in the ITE hearing aid ....................................................................... 65

5 Passive antenna for micro-SD card ................................................................................... 67

5.1 Reminder of the antenna structure and its performances ........................................ 67

5.2 Optimized micro SD antenna design .......................................................................... 70

5.2.1 Isolated antenna impedance performance......................................................... 71

5.2.2 Micro-SD integration into a tablet ...................................................................... 73

6 Passive loop antenna for cochlear implant ....................................................................... 75

6.1 Introduction ............................................................................................................... 75

6.2 Antenna matching circuit ........................................................................................... 76

6.3 Antenna S-parameter characterization in a different environment .......................... 77

6.4 Conclusion on antenna matching .............................................................................. 80

6.5 Antenna radiation performance ................................................................................ 80

6.5.1 Antenna measurement setup and installation ................................................... 81

6.5.2 Conclusion on radiation patterns ........................................................................ 85

6.6 Path loss ..................................................................................................................... 85

6.6.1 Theoretical link budget ....................................................................................... 85

6.6.2 Estimation of the communication range ............................................................ 87

6.6.3 Measurement of path loss .................................................................................. 88

6.6.4 Conclusion on path loss ...................................................................................... 88


6

7 Conclusions ....................................................................................................................... 90


7

Executive Summary

The WiserBAN project will create an ultra-miniature and ultra-low power RF micro-system for wireless Body Area Networks (BAN) targeting primarily wearable and implanted devices for healthcare, biomedical and lifestyle applications. The proposed research concerns the extreme miniaturization of the BAN with primarily the areas of ultra-low power radio SoC (System on Chip), RF and Low-frequency MEMS, miniature reconfigurable antennas, miniaturized SiP (System in Package), sensor signal processing and flexible communication protocols. In WiserBAN project an extensive antenna development has been done. Two active

(tunable) and two passive antennas were developed. A tunable L- antenna for BTE

(behind-the-ear) hearing aid was developed. This antenna was also integrated in BTE

hearing aid device. In addition, a tunable miniature frequency agile antenna was

developed for ITE (in-the-ear) hearing aid application. For a cochlear implant a

passive loop antenna was developed. This loop antenna was also integrated and

characterized in a cochlear implant. The passive micro-SD antenna was implemented

on a micro-SD platform. The micro-SD module is used as the remote node of the

WiserBAN micro-system.

This document has two main goals. At first the document describes the final antenna

structures and their EM simulation results. Secondly the document describes the

final test results for each developed antenna. Part of the test results are measured

with stand-alone antennas. Part of the performance testing has been performed as

system measurements where the antenna is integrated into the respective end user

device. This document describes the final results of WP3 of the WiserBAN project.


8

1 Introduction

This WiserBAN document describes the final antenna designs and their simulation

and measurement results. A part of the measurements have been performed as

stand-alone antenna measurements and a part as system measurements in which

the antenna has been integrated in the end user device. There are two types of end

user devices: wearable and implanted. For L- antenna an antenna impedance sensing

and tuning system (AIST) has been developed and implemented on WiserBAN SoC

(System on Chip). The AIST system (hardware and related software) and its test

results are also described in the document.

The key components of WiserBAN antenna-to-radio interface include the RF front-

end (transceiver), RF SAW filter and various antennas. The RF front-end and RF SAW

filters have same 50 Ω interfaces impedance in all use cases. Instead, various

antennas are employed in different use cases. Consequently, the antenna-to-radio

interface varies somewhat in different use cases. Excluding L- antenna all the other

antennas can be connected directly to the radio 50 Ω antenna port. The L- antenna is

not internally matched to 50 Ω. It is connected to the AIST tuner port of SoC where

the tunable CMOS capacitor bank locates. After going through the CMOS capacitor

bank the antenna signal is routed to the radio 50 Ω antenna port.

In WiserBAN project two active (tunable) and two passive antennas have been

developed. For BTE (behind-the-ear) hearing aid a tunable L- antenna has been

developed. This antenna includes also an antenna impedance sensing and tuning

system (AIST) which is able to measure the antenna resonant frequency and tune the

antenna automatically to the correct frequency. Changes in the antenna

environment may alter the antenna tuning. The antenna itself includes L type

radiating element and a matching coil. The rest part of the matching circuit (series

and shunt capacitor) locates on CMOS SoC. Both the series and shunt capacitors are

switchable and they form a digitally controlled capacitor bank.

For the ITE (in-the-ear) hearing aid a miniature frequency agile antenna has been

developed. This antenna is also tunable. A varactor has been integrated on the

antenna radiating structure. The antenna can be tuned by controlling the varactor

bias voltage. The antenna is internally matched and it can be connected directly to

50 Ω radio port. Both hearing aid antennas are targeted to the wearable end user

case.

Furthermore, two passive antennas have been developed. For a cochlear implant a

loop antenna has been developed and also integrated into a cochlear implant. For

WiserBAN micro-system remote control node a micro-SD antenna has been

developed. This compact planar antenna has been integrated on the micro-SD

platform.


9

2 L antenna for BTE hearing aid

An L-antenna topology was chosen for the final antenna prototype which is integrated in a

behind-the-ear (BTE) hearing aid device provided by Siemens Audio Technology (SAT). Due

to unavailability of a functional 3D system-in-package (SiP) module, a so called dummy SiP

was integrated in the final antenna design to mimic the 3D SiP. The dummy SiP acts only as a

spacer between the antenna and the hearing aid PCB. In the antenna demonstrator module,

all the components required for a radio transmitter including the WiserBAN system-on-chip

(SoC), were placed on a hearing-aid PCB.

The design procedure of the final antenna prototype was as follows. First, an antenna with

the dummy SiP was designed for operating in free space. Thereafter, the antenna was

simulated in the hearing aid device. Due to shift in center frequency induced by the

proximity of the hearing aid dielectrics and the metal parts, a few design iterations had to be

done to obtain a desired center frequency for the antenna. Due to the possible

discrepancies between the design and the realisation, three versions (“A”, “B”, “C”) of the

final antenna prototype were designed, each having a slightly different center frequency.

2.1 Antenna in free space

The simulation model of an antenna with the dummy SiP is shown in Figure 1. The size of the

antenna is 4 mm × 8 mm × 0.5 mm. The dummy SiP has a size of 4 mm × 4 mm × 1 mm. The

whole structure was manufactured in a two-layered Rogers RO4003 substrate having

electrical parameters of εr = 3.38 and tan δ = 0.0027. Due to the manufacturing processing

guidelines, some layout adjustments had to be done in order to drill the vias through the

two-layered PCB and to bring the RF and ground signals from the bottom of the dummy SiP

to the antenna.

Figure 1 Simulation model of the final L-antenna with integrated the dummy SiP.


10

Figure 2 Reflection coefficients and normalized input impedances on a Smith chart of the final L-

antenna prototypes in free space.

The center frequencies of the three antenna versions in free space are 2.85 GHz (“A”), 2.39

GHz (“B”), and 2.56 GHz (“C”) (see Figure 2). The different versions were optimized by

varying the spiral size in the radiator. Simulated radiation efficiencies as a function of

frequency are shown in Figure 3. The values range between -7 dB (20%) and -6 dB (25%). A

typical radiation pattern of the antenna in free space is also presented. The maximum

directivity is about 1.7 dBi. Photographs of a manufactured L-antenna prototype with the

dummy SiP are presented in Figure 4. A solder mask with openings for solder balls having a

diameter of 0.25 mm was applied on the bottom side of the dummy SiP.

Figure 3 Radiation efficiencies and a typical radiation pattern of the final L-antenna prototypes in

free space. The maximum directivity is about 1.7 dBi.


11

Figure 4 Photographs of a manufactured final L-antenna prototype integrated with the dummy 3D

SiP: (a) top view, (b) bottom view, (c) side view. Bottom side is covered with a solder mask.

2.2 Active antenna in the BTE hearing aid

The simulation model of an L-antenna integrated in the BTE hearing aid is shown in Figure 5.

It is seen that the top side of the hearing aid PCB is fully metallized and acts as a ground

plane. In the demonstrator module, majority of the radio components were placed on the

bottom side of the PCB to prevent disturbances to the antenna. Only a “slimstack”

connector had to be placed on the top side of the PCB due to space restrictions in order to

program the WiserBAN SoC. The radio components or the connector were not included in

the EM simulations done by using the CST Microwave Studio.

Figure 5 Simulation model of the final L-antenna integrated in the BTE hearing aid: (a) antenna and

hearing aid PCB, (b) metal parts and half of the housing added, (c) complete hearing aid model.


12

Figure 6 Reflection coefficients and normalised input impedances on the Smitch chart of the final

hearing aid antennas.

Simulated reflection coefficients and input impedances of the hearing aid antennas are

presented in Figure 6. When comparing to Figure 2, it is seen that the center frequencies are

about 300 MHz ‒ 340 MHz lower than in free space. This is due to increased ground plane

and dielectric housing of the hearing aid device. Additional resonances are also excited

around 1.9 GHz. Those lower-frequency resonances are due to the hearing aid PCB and are

quite identical for the three antenna versions. The PCB resonances are also seen in the

radiation efficiency curves in Figure 7. The radiation efficiencies of the antennas vary

between -6.5 dB (22%) and -3.3 dB (47%). The maximum directivity of the hearing aid

antennas is about 2.0 dBi.

Figure 7 Radiation efficiencies and a typical radiation pattern of the final hearing aid antennas. The

maximum directivity is about 2.0 dBi.

The impedance matching network used for the L-antenna in the CST Microwave Studio is

shown in Figure 8. The matching network consists of tunable series and shunt capacitors (Cs

and Cp). The tunable capacitors are implemented in the WiserBAN SoC as discrete capacitor

banks. The series resistors are also included (Rs = 1 Ω) to implement realistic losses of the

capacitor banks. The tuning ranges for the capacitor banks are: Cs: 1.06 pF ‒ 4.24 pF with

0.212 pF steps and Cp: 0.515 pF ‒ 2.06 pF or 1.545 pF ‒ 3.09 pF with 0.103 pF steps. One of

the two ranges for Cp is selected by using an RF switch.


13

Figure 8 Matching network for the L-antenna. The tunable series and shunt capacitors (Cs and Cp)

are implemented as discrete capacitor banks in the WiserBAN SoC.

Simulations were done for the three antenna versions by sweeping the Cs and Cp from 0.5 pF

to 10 pF. The center frequency as a function of capacitor values for the antenna “C” is shown

in Figure 9. The minimum reflection coefficient is presented in Figure 10. Capacitance values

obtainable with the WiserBAN capacitor banks are marked with black dots. From Figure 9 it

is observed that the tuning range of the antenna is from 2.27 GHz up to 2.36 GHz. The center

frequency is tuned with Cs but not with Cp. In addition, the center frequency is more

sensitive to Cs at low capacitance values than at high values.

Figure 9 Center frequency of the hearing aid antenna version “C” integrated with the matching

network. Capacitance values obtainable with the WiserBAN capacitor banks are shown with black

dots.

From Figure 10 it can be seen that the reflection coefficient of the antenna is mainly

dependent on Cp and not Cs. The optimum value for Cp would be about 4.5 pF. With the

WiserBAN capacitor banks, the reflection coefficient for the hearing aid antenna version “C”

is between -9 dB and -16 dB when Cp = 3.09 pF. It should be mentioned that the PCB

resonance is not matched and the antenna efficiency is low around 1.9 GHz.

2.22

2.22

2.24

2.24

2.24

2.2

6

2.26

2.262.26

2.2

8

2.2

8

2.28

2.282.28

2.3

2.3

2.3 2.32.32

2.32 2.322.342.34 2.342.362.36 2.362.382.38 2.38

2.42.4 2.4

2.422.42 2.422.44

Cp (pF)

Cs (

pF

)

Center frequency of antenna version C (GHz)

1 2 3 4 5 6 7 8 9 10

1

2

3

4

5

6

789

10


14

Figure 10 Minimum reflection coefficient of the hearing aid antenna version “C” integrated with the

matching network. Capacitance values obtainable with the WiserBAN capacitor banks are shown

with black dots.

The reflection coefficient as a function of frequency and the series capacitance for the

antenna version “C” is shown in Figure 11. The impedance bandwidth, i.e., the frequency

range at which the |S11| < -6 dB, is between 23 MHz and 25 MHz. Respectively, the realized

gain for the antenna “C” is presented in Figure 12. Similar figures could be drawn also for the

antenna versions “A” and “B”. For brevity, the CST simulation results for the other antenna

versions are summarized in Table 2. The antenna version “C” was favored for the

demonstration due to frequency restrictions of the WiserBAN SoC transmitter which

operates at frequencies between 2.0 ‒ 2.5 GHz. An intermediate antenna version for the 3D

SiP operating around 2.45 GHz was also designed, but it was considered to be risky to use

that antenna in the demonstration phase due to possibility of antenna to tune out of the

transmitter frequency band.

-30

-30

-20

-20

-20

-20

-20

-20

-16

-16

-16

-16

-16

-16

-12

-12

-12

-12

-12

-12

-10

-10

-10

-10

-10

-10

-8

-8-8

-8

-8-8

-6

-6-6

-6

-6

-6-6

-4

-4-4

-4

-4

-2-2

-2

Cp (pF)

Cs (

pF

)

Reflection coefficient of antenna version C (dB)

1 2 3 4 5 6 7 8 9 10

1

2

3

4

5

6

789

10


15

Figure 11 Reflection coefficient of the hearing aid antenna version “C” integrated with the matching

network. Cp = 3.09 pF and Cs is swept from 1.06 pF (highest frequency) to 4.24 pF (lowest frequency)

with 0.212 pF (16) steps.

Figure 12 Realized gain of the hearing aid antenna version “C” integrated with the matching

network. Cp = 3.09 pF and Cs is swept from 1.06 pF (highest frequency) to 4.24 pF (lowest frequency)

with 0.212 pF (16) steps. Red circles indicate the center frequencies.

2.3 Head influence on the active antenna in the BTE hearing aid

Influence of a human head on the active hearing aid antenna was investigated with EM

simulations. A simplified head model consisting of “skin”, “bone” and “brain” layers with

electrical parameters shown in Table 1 was provided by the CST corporation. Pictures of the

head model are shown in Figure 13. The hearing aid device including the active antenna was

placed behind an ear as seen in Figure 13d.

2.2 2.22 2.24 2.26 2.28 2.3 2.32 2.34 2.36 2.38 2.4-20

-19

-18

-17

-16

-15

-14

-13

-12

-11

-10

-9

-8

-7

-6

-5

Frequency (GHz)

Realis

ed g

ain

(dB

i)

Realised gain of antenna version C (dBi)


16

Table 1 Electrical parameters of the head model at 2.45 GHz.

Relative permittivity ε'r Conductivity σ (S/m)

skin 38 1.49

cartilage 38.7 1.79

bone 11.4 0.4

brain 48.8 1.84

fat 5.3 0.11

Figure 13 Head model used in EM simulations: (a) vertical cut of ear, (b) head, (c) vertical cut of

head, (d) BTE hearing aid device with active antenna behind an ear.

The center frequency and reflection coefficient as a function of capacitor values of the

matching network are presented in Figure 14 and Figure 15. With the WiserBAN capacitor

banks the tuning range is from about 2.22 GHz up to 2.33 GHz. So, it seems that the antenna

is tuned downwards in frequency due to head proximity. However, the difference is small,

only about 40 MHz (2%), and the results may include some numerical errors due to very

large simulation structure. From Figure 15 it can be seen that impedance matching of the

antenna is improved by the head proximity. This is due to increased antenna resistance near

the head model. The reflection coefficient as a function of frequency and the series

capacitance is shown in Figure 16. The impedance bandwidth is between 40 MHz and 30

MHz.


17

Figure 14 Center frequency of the hearing aid antenna version “C” integrated with the matching

network and placed behind an ear of a head model. Capacitance values obtainable with the

WiserBAN capacitor banks are shown with black dots.

Figure 15 Minimum reflection coefficient of the hearing aid antenna version “C” integrated with the

matching network and placed behind an ear of a head model. Capacitance values obtainable with

the WiserBAN capacitor banks are shown with black dots.

2.18

2.182.2

2.2

2.2

2.2

2

2.22

2.222.22

2.24

2.24

2.242.24

2.26

2.26

2.262.28

2.28 2.282.32.3 2.32.32

2.32 2.322.342.34 2.342.362.36 2.36

2.382.38 2.38

2.42.4

Cp (pF)

Cs (

pF

)

Center frequency of antenna version C (GHz)

1 2 3 4 5 6 7 8 9 10

1

2

3

4

5

6

789

10

-30

-30

-30

-20

-20

-20

-20

-20

-16

-16

-16

-16

-16

-16

-14

-14

-14

-14

-14

-14

-12

-12

-12

-12

-12

-12

-10

-10

-10

-10

-10

-10

-8

-8-8

-8

-8-8

-6

-6-6

-6

-6-6

-4

-4-4

-4

-4

-2

-2

Cp (pF)

Cs (

pF

)

Reflection coefficient of antenna version C (dB)

1 2 3 4 5 6 7 8 9 10

1

2

3

4

5

6

789

10


18

Figure 16 Reflection coefficient of the hearing aid antenna version “C” integrated with the matching

network and placed behind an ear of a head model. Cp = 3.09 pF and Cs is swept from 1.06 pF

(highest frequency) to 4.24 pF (lowest frequency) with 0.212 pF (16) steps.

The head proximity has a bigger effect on the radiation pattern and the antenna efficiency

than on the center frequency. The antenna directivity is increased up to 5.0 dBi near head

model which can be seen in Figure 17. Although the antenna efficiency is decreased down to

6 ‒ 8 %, the maximum antenna gain remains between -7 and -6 dBi. A comparison was done

by placing the hearing aid antenna behind the left or the right ear. The center frequencies

were the same, and the radiation patterns were symmetrical, but a 1 dB difference in gain

was observed. The difference was probably due to small asymmetry of the simulation setup

due to manual alignment of the hearing aid. The CST simulation results of the hearing aid

antennas are summarized in Table 2.

Figure 17 Radiation pattern of the final hearing aid antenna placed behind an ear of a head model.

The maximum directivity is 5.0 dBi.


19

Table 2 Summary of the CST simulation results of the final hearing aid antennas tuned with the

WiserBAN capacitor banks. Cp was fixed to 3.09 pF.

Freq. (MHz) BW (MHz) min |S11| (dB) max Gain (dBi) Tot. eff. (%)

Antenna “A” 2500 ‒ 2600 34 ‒ 40 -16 ‒ -26 -5 ‒ -6 16 ‒ 20

Antenna “B” 2145 ‒ 2225 19 ‒ 20 -8 ‒ -14 -6 ‒ -7 13 ‒ 16

Antenna “C” 2272 ‒ 2362 23 ‒ 25 -9 ‒ -16 -5.5 ‒ -6.5 13 ‒ 18

Antenna “C” and

head model

2225 ‒ 2325 40 ‒ 30 -13 ‒ -24 -7.0 ‒ -6.1 6 ‒ 8

2.4 Active antenna with realistic matching network

In order to investigate hearing aid antenna tuning with a more realistic matching network,

the S-parameters of the antennas were imported in the Cadence circuit design software

which includes also the parasitic impedances of the capacitor banks. The reflection

coefficient as a function of frequency and the series capacitance for the antenna version “C”

is presented in Figure 18. It can be observed that the tuning range is smaller (48 MHz) than

with an ideal matching network (90 MHz). Also, the input matching is better with low values

of Cs than high values, which indicates an additional parasitic shunt capacitance of about 1

pF in the more realistic matching network. The Cadence simulation results for the three

antenna versions are summarized in Table 3.

Figure 18 Reflection coefficient of the hearing aid antenna version “C” integrated with the realistic

matching network in Cadence. Cp = 3.09 pF and Cs is swept from 1.06 pF (highest frequency) to 4.24

pF (lowest frequency) with 0.212 pF (16) steps.

2.2 2.22 2.24 2.26 2.28 2.3 2.32 2.34 2.36 2.38 2.4

x 109

-20

-18

-16

-14

-12

-10

-8

-6

-4

-2

0

Frequency (Hz)

|S11|

Reflection coefficent of antenna version C (dB)


20

Table 3 Summary of the Cadence simulation results of the final hearing aid antennas tuned with the

WiserBAN capacitor banks. Cp was fixed to 3.09 pF.

Freq. (MHz) BW (MHz) min |S11| (dB)

Antenna “A” 2467 ‒ 2521 20 ‒ -10 ‒ -5

Antenna “B” 2118 ‒ 2161 16 ‒ 6 -22 ‒ -8

Antenna “C” 2242 ‒ 2290 17 ‒ 5 -17 ‒ -7


21

3 Antenna impedance sensing and tuning

Introduction

The objectives of the task is to design and implement a smart controller capable to

autonomously tune the BAN node antenna. The idea is based on novel idea of implementing

a circuit sensing the impedance of an antenna using dedicated electronics and an algorithm

running in a microprocessor. In this task has developed a circuit block integrated into the

SoC consisting an antenna tuner , or a capacitor bank. A characterization environment has

been investigated. An embedded software has been developed to control the circuit block

and to perform auto tuning algorithm calculation.

3.1 AIST hardware

3.1.1 Antenna impedance tuning

The antenna impedance tuning is implemented as a part of the SoC circuit. The tuner

contains a binary weighted n-bit capacitor array with a small unit capacitor. The binary

weighted capacitor array provides in theory linear control steps over capacitor range as

response of control but also at the same time a minimum size of total capacitance i.e.

occupied silicon area. On the other hand the obtained frequency tuning is not a linear

function of control which means that a higher number of control bits are required to cover

the whole tuning range. The selection of capacitors inside the array is performed by RF

switches. Furthermore a digital interface is available for the switch control.

Figure 19 Simplified schematic diagram of the tuner circuit.

An RF switch isolating the tuner from the radio has been added. This arrangement augments

the performance of the sensing block since the impedance vs. frequency response of the RF


22

SAW filter can be omitted. Moreover, the switch isolates the sensing signal generator

(oscillator) from the radio (LNA & PA).

A simplified diagram of the antenna tuning circuit is shown in Figure 19. The transmit RF

signal from the radio front-end goes through the RF SAW filter to the capacitor bank on SoC.

From the capacitor bank the transmit signal continues to the antenna radiator on PCB (or

IPD). The matching circuit is partly integrated on the same PCB platform on which the

antenna radiator locates. During the normal operation the tuner control circuit has set to a

fixed switch configuration to enable a proper antenna matching. The impedance sensing

circuit is disabled and disconnected from the tuner using an RF switch (SWSENSE). When the

impedance sensing routine is invoked at the system level (DSP, IcyFlex) the isolation switch

(SWRF) between the tuner and the RF SAW filter is switched off and the sensing circuit is

activated. The main design parameters of the tuning circuit are listed in Table 4.

Table 4 Design parameters of the tuning circuit.

Parameters Comments Values (1) Units

Shunt capacitor, CSH

Cu Unit capacitor, tuning step 0.10 pF

Cf Fixed capacitance, minimum capacitance 41* Cu pF

Cmax Maximum capacitance 56* Cu pF

n Number of control bits, 2n steps 4

Serial capacitor, CSER

Cu Unit capacitor, tuning step 0.21 pF

Cf Fixed capacitance, minimum capacitance 5* Cu pF

Cmax Maximum capacitance 20* Cu pF

n Number of control bits, 2n steps 4

Qc Capacitor Q-value 270

Ron, SW RF switch on resistance < 1 (2) Ω

L SW RF switch isolation at 2.4 GHz tbd (2) dB

(1) Typical process parameters

(2) Trade-off between values, to be redefined later


23

3.1.2 Antenna impedance sensing

An optimal antenna tuning is performed by using a novel impedance sensing method. The

method requires a direct signal access to the antenna input port and can not be used at the

same time when the radio (Tx/Rx) operates. Therefore the main control for the sensing and

tuning is done at system level by IcyFlex (or externally) which invokes the sensing/tuning

algorithm when necessary. When the device environment does not change or changes only

slowly over the time (small proximity effect) the antenna sensing/tuning system is activated

infrequently after the boot-up of the system.

A simplified schematic diagram of the antenna impedance sensing circuit is shown in Figure

20. The system level control for the circuit block is performed by a digital interface

(registers) which is controlled at the system level. In the default condition the sensing block

is in idle state i.e. disabling all activity except setting control for the capacitance bank

switches. When the sensing or tuning operation is requested at the system level the sensing

circuit is activated and the frequency shift of the antenna tuning is analysed and the proper

configuration for the tuner is determined. After the sensing sequence has been finished a

status of the tuning is delivered to the system level: tuning succeeded / failed. In the first

version of the ASIC the digital interface will be also utilized to characterize the circuit

operation and to test different algorithm versions for an optimal tuning.

The antenna impedance sensing will be performed by using a controllable oscillator circuit

which is highly sensible to loading effect. The operational point of the oscillator is controlled

digitally. The output response is obtained from a frequency counter and it is analysed by a

specific algorithm. The acquired response vs. control is analysed and recorded at first in the

nominal condition, when no unwanted proximity effects are present. In case the sensing is

activated the impedance sensing/analysis routine is repeated and a deviation from the

nominal condition is calculated. The calculated difference is used to re-adjust the tuning

circuit for a proper impedance match.

Figure 20 Simplified schematic diagram of the impedance sensing block.


24

Table 5 Design parameters of the sensing circuit.

Parameters Symbols Comments Min Typ Max Units

Supply voltage VDDx 1.2 V 10 % 1.08 1.2 1.32 V

Operating temperature TA -40 27 85 °C

Total active current consumption IAIST_DD 1 mA

Total standby current consumption IAIST_STB < 1 nA

3.1.3 AIST realisation

AIST-block was integrated as a part of the WiserBAN SoC. The circuit blocks for sensing and

tuning were packed together in order to make the whole SoC construction easier. This

arrangement leads to a suboptimal system level layout, since the tuner was not possible to

place at vicinity of the radio port of the SoC.

Figure 21 The layout of the AIST-block (size 280 µm x 120 µm).

The layout of the AIST-block is shown in Figure 21. In addition to shunt and series tuner

capacitor banks and the oscillator providing the sensing operating a few auxiliary were

required. A dedicated logic was implemented to interface HAL-layer, construct low level

control signals and provide accurate timing for the frequency counter timing. A sub-

threshold MOS-transistor based voltage reference generator, a digital-to-analog converter

and a buffer amplifier were designed to provide a proper control for the sensing oscillator

circuit.

More sophisticated control operations are done at system level. The control of the AIST-

block is performed by using the HAL-layer of the IcyFlex microprocessor. In the HAL-layer

there are specific ANACTRL-registers that can be directly accessed, which enables a directly

control over sense and tuning functions. The input ANACTRL-registers can be used to fetch in

commands and parameters which triggers AIST-block into specified operational mode. In the

simplest case when only a capacitor bank settings need to be altered corresponding

ANACTRL-register value should be written by IcyFlex. The sense operation is more complex

and requires a set of commands to be fetched into AIST-block, and a dedicated software is

required. The low control of the AIST-block is illustrated in the document ‘AIST low level


25

control specification’, where the ANACTRL-registers, control commands and a proposed

tuning algorithm are specified.

3.2 AIST characterization

3.2.1 Characterization platform

The antenna impedance tuning could be in theory performed as an 'open-loop' control

system, where no (measured) feedback signal is available, and tuning would be applied

based on fixed tuning setting. In general, this is not usually possible because the component

parameter tolerances etc. make an accurate tuning impractical. Moreover, here one target is

to provide a real-time feature which requires an acquired feedback signal.

The proposed idea in the WiserBAN demonstrators is to tune the antenna impedance

against a baseline impedance measurement. The fingerprint of the antenna impedance

(control vs frequency curve) is highly sensitive to all devices and parasitics coupled into

sense node of the AIST. In order to develop a proper tuning algorithm or form a look-up

table tuning data will demand empirical results.

The functionality characterization of the AIST-block has been done utilizing MPW1/MPW2

characterization boards. These boards provide an USB-link between PC and a

communicational chip providing proper control signals for the WiserBAN SoC. A simplified

block diagram of the proposed strategy to characterize the AIST block is shown in Figure 22.

A system level denotes an external control over IcyFlex like console running in PC-machine.

In the system level test vectors to characterize functionality are generated and response

data will be acquired and post-processed. Initial method to manipulate test vectors is a

script-based (perl) interface. Test vectors are fetched into IcyFlex via suitable data link, e.g.,

direct access to memory space using JTAG-interface. In the IcyFlex is running a simple code

interpreting register calls and performing specified control and characterization operations.

The control and data transfer between IcyFlex core and the AIST-block are performed using

a hardware abstraction layer (HAL) or dedicated registers (ANACTRL). Register are used to

set antenna tuner to certain tuning position and applying a proper control for the antenna

impedance sense circuitry. Also the acquired impedance sense raw data are delivered for

post-processing using the ANACTRL-register. Finally AIST-block is responsible to set the tuner

and control switches in a proper position corresponding to required operation condition.


26

Figure 22 A block diagram of the AIST characterization setup.

The automatic antenna tuning algorithm will be implemented at demonstrators using

software running in the IcyFlex core integrated into the WiserBAN SoC. In the proposed

strategy the tuning operation will be implemented as a function call(s) from the main

program running in the WiserBan demonstrator. This strategy adds flexibility upon different

user cases and platforms where the demand for the tunability varies. For instance some

applications might require only one calibration/tuning phase at assemble line where some

cases frequent tunability over proximity effects might be mandatory. Required functionality

for AIST-block control software layer can be summaries as:

System level control - Console at PC - script/GUI control JTAG - control vector generation - data acquisition/post-processing

//main program

//definitions

// declarations

Infinite_loop

maintain_jtag_io {}

memory_access{}

inteprept_commands{}

.

aist_ant_tun_auto {}

loop end

HAL-layer

IcyFlex

JTAG/USB

PC

IcyFlex

Memoryspace - global variables

- access ANACTRL

AIST-block

ANACTRL_IN3,4

ANACTRL_OUT14,15

Memory/

register

access

aist_ctr aist_data

Tuner Sense

Radio/antenna hardware

Register access

Antenna/Radio

Physical RF-interface


27

- Function calls to provide access to IcyFlex registers

- Function call for sensing and tuning algorithm

- Sample main program (for characterization)

- Interface layer between IcyFlex and system level

3.2.2 Experimental results

The results shown below are acquired from MPW2-evaluation board. This setup limits the

tuning performance and results are considered as a functionality test of the AIST. It should

be emphasized that this setup cannot be used to any radio transmission, but only to

preliminary characterize AIST-block and speed-up software development. When a real

WiserBAN-demonstrator is available there results (algorithm and software) can be hopefully

applied with small modifications.

The main problem of characterization the AIST-block is the need of an antenna connected in

the radio port. The sensing block does not operate properly when the impedance seen in

port differs considerably from the design values. More precisely the sensing block does

operate at high impedance range but experimental data is needed to fine tune algorithm. A

major drawback of the characterization boards is a lack of a possibility to attach a miniature

antenna within the project. In the MPW1-board there was added a SMA-port to connect an

antenna, but it was found that an antenna providing set features can not be connected using

a cable without a major loss of performance. In the MPW2-board the external antenna was

replaced with a microstrip structure providing an electrical impedance emulating VTT's

miniature antenna (see Figure 23). The impedance of the microstrip and tuner circuit can be

measured using the radio port in SoC (see Figure 19). It should be noted that the microstrip

resonance frequency, emulating antenna matched frequency band, is lower than used the

the WiserBAN radio. However, it was considered that lower frequency is still useful to

characterize the basic functionality of the circuit blocks and characterize the autonomous

tuning performance.

Figure 23 (a) MPW2-evaluation board with a ‘microstrip antenna’. (b) a model of the microstrip

antenna with added capacitor tuner circuit.

An equivalent model of the designed ‘microstrip antenna’ electric is shown in Figure 23. The

simulated response (abs[S11] ) of the microstrip antenna for series (Cse) and shunt capacitor

(Csh) bank settings are shown in Figure 24. The measured responses (see Figure 25) show

the tuner and antenna arrangement can be controlled with IcyFlex so there are no system


28

level issues to control AIST-tuner. The measured response differs considerably from the

simulated in respect of the resonance frequency and losses at higher frequencies. When this

issue was investigated in more detail, it was found that MPW2 –board material and

dimensions differs from the design values, which causes deviation of the electrical response.

However, for testing of the AIST-block functionality this can be accepted since the block is

based of differential measurements between operational conditions where the absolute

values are not critical.

Figure 24 Simulated tuning response of the microstrip antenna.

Figure 25 Measured tuning response of the microstrip antenna.

The sense function and autonomous tuning of the AIST-block has been characterized as

follows. Based on the scattering parameters with the tuner, the sense block controls the

parameters and setting for the AIST-block were determined. Initial testing was performed

via using a low level commands i.e. direct write in to ANACTRL-register using JTAG-link at PC.

The low level commands were generated by a perl-script, which enables a fast and easy

debugging of functionality, but lacks the real-time speed because each command has to be

transferred from PC to IcyFlex.

The frequency response of the microstrip antenna coupled was acquired for altered

conditions while sweeping over the (whole) AIST-block sensing range. This set of frequency


29

responses can be expressed in a 3D mesh indicating a ‘fingerprint’ of the antenna versus

tuner settings. In Figure 26 is shown a fingerprint mesh of condition where all series tuner

positions. In the mesh plot one can detect the impedance vs frequency sensitive areas,

which will be used as a guide to adjust the autonomous tuner algorithm. With the MPW2-

board it is not possible to obtain accurate information about the final demonstrator

frequency characteristics and the current version of the auto-tuner is based on the results

already achieved.

Figure 26 A measured microstrip antenna fingerprint for different series tuner positions (Cser). The

selected operational point for autotuning is shown with a red arc.

The auto-tuning feature has been characterized using an operation point found at slightly

higher frequencies than a sharp ‘edge’ area due to resonance frequency seen by the AIST-

block. This region is beneficial for auto tuning, because the measured response is

moderately continuous in respect of the series tuner position as can be seen in Figure 27. In

addition to tuner position settings the microstrip antenna is highly sensitive to an improvised

proximity effect or touching with a fingertip. This feature was used to make a real-time auto

tune demonstrator with following algorithm:

1. Tuner position was set to ‘middle point’

2. A baseline frequency was measured and recorded

3. An auto-tune mode is enable to make continuous loop

a. Frequency measured again

b. If frequency deviates from recorded value new tuner values are

calculated

c. Radio port was enabled to record scattering parameters with the

network analyzer

It was found that even with a sub-optimal setup auto tuning feature was possible to

demonstrate. When a finger was moved at vicinity of the antenna an upward frequency shift

at the network was observed. At the same time at AIST-block control console indicates a


30

frequency deviation of baseline and a new tuner values were calculated and updated. A low

operating speed of the control strategy made a real-time tuning inconvenient, but this can

be easily achieved by having the tuning program running in IcyFlex core. The auto tuning

software was later coded by SignalGenerix and results are shown in next chapter.

Figure 27 A measured microstrip antenna fingerprint for different series tuner positions (Cser) at

fixed operational point shown in Figure 26.

3.3 AIST software

3.3.1 Introduction

The objectives of this task is to design and implement a smart self-tuning antenna-to-radio

interface which consists of impedance sensing, control unit with associated software, drivers

and impedance tuning element (on-chip CMOS capacitor bank) for the BAN node

demonstrators developed in WP5.

The task was executed in stages

(i) The initial application was written using Perl Script running from A PC (VTT).

(ii) Initial antenna characterization was carried out by changing antenna tuning

parameters manually (VTT).

(iii) An application and associated libraries were created to perform a parametric

sweep of all the antenna tuning parameters and to plot the results.

(iv) Two different auto-tuning applications/libraries were developed, one using a

‘Brute Force’ method where the values for the tuning parameters are stepped

through every combination and the values that provide the most successful

outcome are selected, the other uses a iterative tuning approach that converges

on the tuning value to give minimum error.


31

3.3.2 Principle of Operation

With reference to Figure 19, the antenna is tuned by automatically adjusting the values of

the shunt capacitor CSH and the series capacitor CSER to achieve the best antenna matching.

Because the antenna impedance can vary continuously, depending on its position on or

inside the body and the dielectric properties of the surrounding material, the network must

be retuned periodically to maintain optimum matching and hence maximal antenna

performance.

In order to determine that the matching network is tuned optimally, a sensing circuit is

implemented as shown in Figure 20. This uses a local oscillator feeding the antenna port,

tuned by varying VREF, and an integrating ADC that measures the resonance frequency at

the antenna port and hence can determine that the performance of the matching circuit is

within the target limits.

3.3.3 AIST Tuning Algorithms

Description

The requirement of the AIST tuning function is:

(a) To be able to measure the resonance frequency of the counter/integrator output.

(b) To adjust the series and/or shunt capacitor in the antenna matching circuit so that

the resonance frequency is as close as possible to the pre-defined target frequency,

within limits.

(c) To provide status information that the tuning was a success, or failure if the target

cannot be met within the range of the tuning capacitors or the target frequency is

outside the pre-defined tolerance.

There are two approaches:

(a) Using a ‘Brute Force’ tuning method which sets the series and shunt capacitors to

every possible combination, each time taking a measurement, and selecting the best

result. This is useful for initialising the values in a static environment as it gives the

optimum result, but is considered to be too slow in response to a rapidly changing

environment, due to the number of iterations. The flowchart for this approach is

shown in Figure 28.

(b) Using a ‘Gradient’ tuning method, this adjusts only the series capacitor, the shunt

capacitor being set to a constant value found previously. The adjustment is made in

small increments or decrements, each time taking a measurement, to converge

towards the target result. This method is more rapid as it is possible to control the

number of tuning steps, down to just one step; hence the operation could be fitted

into an allocated processor timeslot and the matching network will converge

towards the target over several executions. The flowchart is given in Figure 29.

Both approaches use a common algorithm for measuring the resonance frequency. The

radio is temporarily disconnected from the tuner during measurement and a frequency


32

counter/integrator is used to measure the resonance frequency. The flowchart for the

sensing algorithm is given in Figure 30.

Figure 28 Flowchart – Brute-Force tuning algorithm.


33

Figure 29 Flowchart – Gradient tuning algorithm.


34

Figure 30 Flowchart – AIST Sensing Function.


35

Embedded Software Implementation - Overview

The AIST libraries/functions and associated test applications are shown in the tree diagram

in Figure 31.

Figure 31 Summary of Embedded Software Libraries and Applications – AIST.

Antenna Control

This standalone application allows the AIST parameters to be monitored and controlled from a PC connected to the WiserBAN module or development board via the UART, using commands entered from a terminal program. This allows the AIST parameters to be individually selected and tuned manually for initial testing and calibration. An example of the terminal interaction is shown in Figure 32.


36

Figure 32 AIST Manual Tuning Application - Terminal Interaction Example.

AIST Characterisation

This application uses the AIST library to initialise the AIST registers and then performs a

parametric sweep of the RF frequency and series and parallel tuning capacitors to obtain the

resonance response of the tuning circuit for various antenna types and scenarios. The

associated Matlab application plots and stores the result, as shown in Figure 33. For each

run, sixteen 3-dimensional plots are generated, one for each setting of the shunt capacitor.

Figure 33 AIST Characterisation Application – Generated Result Example.

AIST Auto-tune

This test application uses the AIST library to initialise the AIST registers to their pre-defined

values (found during the characterisation phase) and then implements either the ‘Brute

Force’ or ‘Gradient’ tuning function to perform auto-tuning on the matching circuit

connected to the antenna, depending on the selection in the make file. An example of the


37

terminal display is shown in Figure 34 for the Brute Force algorithm; and in Figure 35 for the

Gradient algorithm. The auto-tuning occurs on a timed interrupt.

Figure 34 AIST Auto-tuning Test Application, Bruce Force Method – Dummy Antenna.

Figure 35 AIST Auto-tuning Test Application, Gradient Method – Dummy Antenna.


38

4 Miniature frequency agile antenna

4.1 Reminder of the antenna design

4.1.1 Design and performances

A compact antenna has been designed to be integrated within a new generation of

miniature hearing aid electronic device [1]. The maximum volume available for the antenna

including its ground plane has to be very small: the maximum dimension is equal to λ0/25, λ0

being the wavelength at the operating frequency of 2.45 GHz. To reach these size

specifications, the antenna structure is based on a monopolar wire-patch antenna which

provides dipolar-like radiation characteristics [2]. The designed antenna is represented in

Figure 36. As a classical monopolar wire-patch antenna, it is composed by two metallizations

etched on each face of a dielectric substrate. The lower metallic plate acts as ground and the

upper metallic plate constitutes the antenna top hat. This kind of antenna is fed by a coaxial

probe which is connected to the top hat through the ground plane and the dielectric

substrate. The ground wire acts as a short-circuit to the capacitance of the antenna

constituted by the top hat above the ground plane and allows achieving a new low-

frequency parallel resonance. The resonance frequency is smaller than the classical antenna

fundamental cavity mode. It is primarily set by the size of the top hat, the height of the

antenna, the permittivity of the substrate and the ground wire diameter.

The main antenna parameters to adjust the antenna impedance matching to 50Ω are:

The ground wire radius. The smaller the radius is, the higher the maximum of the

input impedance real part is.

The radius of the feeding probe. The higher the radius is, the lower the input

impedance imaginary part is.

The ground wire – feeding probe separation. The Q-factor is increasing when the

length between the ground wire and the feeding probe core is increasing.

As presented in [3], [4] the use of a closed slot into the antenna top hat involves a significant

reduction of the resonant frequency. Indeed, the introduction of a slot in the hat of the

antenna changes the equivalent capacity of the antenna short-circuited hat by increasing its

value. The longer the electrical length of the slot is, the lower the resonant frequency is. As

the antenna dimensions are limited to λ0/25 at 2.45 GHz, the electrical length of the slot is

maximized by folding on the allowed surface on top hat. This slot has a 0.4mm-width.

Moreover, to decrease the resonant frequency by an even higher degree, a discrete

capacitor is loading the slot [4]. The designed antenna structure is depicted in Figure 36. The

characteristics of the used dielectric substrate (RO4003, chosen for its moderate permittivity

and low loss) are εr=3.55 and tan(δ)=0.0027. The overall dimensions of the antenna are 5

mm x 5 mm by 2 mm high.


39

Figure 36 Miniaturized monopole wire patch antenna.

Antenna impedance is directly matched to 50 Ohms. The very small dimensions of the

antenna involve a reduced operating bandwidth. The discrete capacitor loading the slot can

be adjusted according to the desired resonant frequency and impedance matching band. It is

for example possible to cover the entire frequency band going from 2.4 GHz to 2.483 GHz for

capacitance values between 0.63 pF and 0.74 pF. The simulated input impedances and

reflection coefficients for different capacitance values are plotted in Figure 37. Moreover,

Figure 37 (b) shows the antenna efficiency for each matching band.

(a)

(b)

Figure 37 Input impedances (a), |S11| parameter and total efficiency (b) according different

capacitance values.

5mm

5mm=λ0/25

2mm= λ0/62.5

Capacitor

Feeding probeDielectric substrateRO4003 Ground wire

Capacitor

0,2mm

0,45mm0,4mm

1,1mm

1mm

0,5mm

0,3mm1mm

2,4mm0,3mm

2.35 2.4 2.45 2.5-10

0

10

20

30

40

50

60

70

80

90

100

Frequency (GHz)

- R

e(Z

in)

( )

--

Im(Z

in)

( )

C=0.63pF

C=0.66pF

C=0.7pF

C=0.74pF

2.35 2.4 2.45 2.5

-35

-30

-25

-20

-15

-10

-5

0

Frequency (GHz)

|S11| (d

B)

2.35 2.4 2.45 2.50

0.01

0.02

0.03

0.04

0.05

To

tal eff

icie

ncy (

x100)

(%)C=0.63pF

C=0.64pF

C=0.65pF

C=0.66pF

C=0.67pF

C=0.68pF

C=0.69pF

C=0.7pF

C=0.71pF

C=0.72pF

C=0.73pF

C=0.74pF


40

For sake of briefness, the antenna measurements in next section will be shown only for one

antenna configuration, i.e. one capacitance value.

4.1.2 Considerations on antenna miniaturization

Given the antenna dimension and performance limitations in terms of bandwidth and

efficiency, here we carry out an analysis with respect to fundamental limits of electrically

small antennas [5]-[8]. Several studies deal with the quality factor Q: authors in [9] have

been concluded that the impedance bandwidth roughly equals 1/Q. Recently, Best [10] and

Yaghjian [11] have adjusted this first approximation. Indeed, they defined the approximate

formula for the fractional matched Voltage Standing Wave Ratio (VSWR) bandwidth,

FBWv(ω0) as:

s

s

Z

RFBWv

1.

)(

)(2)(

000

000

(1)

where )( 00 Z is the first derivative of the antenna impedance, 0R is the real part of the

antenna impedance, and s the VSWR.

Finally, they derived the relationship between FBWν(ω0) and Q(ω0) through the

maximum allowable VSWR as:

s

s

FBWQ

v

1.

)(

1)(

0

0

(2)

The minimum Q value attainable by an infinitesimal electric dipole, or similarly by the

azimuthally symmetric TM10 spherical mode, has been investigated thoroughly. This

minimum, deriving from small antenna analysis [5] is:

))(1()(

)(2123

2

kaka

kaQHansen

(3)

where a is the minimum radius of the sphere enclosing the antenna and k is the wave

number (k=2π/λ).

Another Q physical limitation using spherical modes has been demonstrated by Collin et al.

[7]. They have shown that for the first spherical mode:

kakaQCollin

1

)(

13 (4)

In [10] it has also shown that the lower bound on Q is depending on the expense of

efficiency as shown the equation below:

kakaQlb

1

)(

13

(5)

Thus, to compare antenna performances with the physical limitations previously shown, the

antenna efficiency has to be taken into account. Therefore, here we consider the antenna

quality factor normalized with respect to the antenna efficiency.


41

Considering the previous antenna results, the antenna size and performances can be

compared with physical limitations. Figure 38 shows the sphere circumscribing the antenna.

Its radius a corresponding to the distance between the antenna center and its apexes equals

to 3.674 mm (i.e. λ0/34 at 2.4 GHz).

Figure 38 Sphere circumscribing the antenna.

The ratio Q/η is plotted versus the ka values in the Figure 39 for three relevant cases:

The Hansen limit of (3).

The Collin limit of (4).

The considered antenna results from (2).

It should be noted that the efficiency here considered is the one of the simulated antenna

when it is matched.

Figure 39 Q/η versus the ka values for three cases.

From these results we can see that the considered antenna is very close to the physical

limits. It should be noticed that Hansen and Collin limits are equivalent in the case of a very

small antenna (ka


42

Figure 40 η/Q versus the (ka)3 values for the considered antenna.

4.2 Antenna Impedance characterization

As described in the deliverable D3.2, measurement of such an antenna using classical

invasive methods is a challenging issue given its small dimensions. The reduced size of the

antenna ground plane implies that antenna performances are disturbed by parasitic RF cable

effects, which is needed for antenna measurements.

Compact antennas for mobile handsets, which use their “large” ground plane to improve

their performances, can be correctly measured using coaxial cable suitably protected and

positioned [14]. However, the feeding cable becomes problematic for a miniature antenna

over a limited ground plane, and can hardly prevented. Indeed, placing the cable in the

reactive zone of the antenna can modify the near field distribution, and consequently

perturbs both its radiation and impedance characteristics, which differ from those of the

isolated antenna in free space.

Some methods have been proposed in literature to reduce the feeding cable effects during

the measurement process. For low frequencies antennas, i.e. working up to 400 MHz,

ferrites can be integrated on the feeding cable in order to absorb unwanted radiation

[15][16]. As a consequence because of this frequency limitation this technique cannot be

applied for our antennas.

In this section firstly, all issues raised by the use of a feeding coaxial cable in simulation are

presented. Then, a new methodology to counter the cable issue for impedance

measurement process is proposed. Indeed, correctly simulate an infinite cable connected to

the miniature antenna allows a consistent comparison with measurement. These results will

be used to develop a method which will permit to extract the real antenna impedance from

the measured ones with an intrusive infinite coaxial cable.

4.2.1 Cable effect on impedance characteristics

4.2.1.1 Finite length cable

This subsection investigates the introduction of a coaxial cable with a finite length using 3D

electromagnetic simulation tool.

6 6.5 7 7.5 8

x 10-3

5

5.5

6

6.5

7

7.5

8x 10

-3

(ka)3

/ Q

linear fittingy = x


43

Figure 41 shows the surface current distribution on a finite coaxial cable for two different

lengths. The electrical field amplitude is equally plotted for the long cable configuration.

Both surface current and electrical field cartographies show that there is a standing wave on

the cable.

(a) (b)

Figure 41 Surface current on the cable for two cable lengths (a) and E field for L=10 cm (b).

This result suggests that such a standing wave can differently disturb both antenna

impedance and radiation properties depending on cable length. This is also confirmed by the

variation of the input impedance according to the cable length (Figure 42).

Indeed, Figure 42 shows that the input impedance is modified depending on the cable length

Lcable, and clearly different from the single isolated antenna input impedance. We note a

decrease of the quality factor of the antenna with a long cable suggesting an expansion of

the antenna. The |S11| parameters are therefore different and the antenna is even

mismatched for coaxial cable cases.

(a)

L=3cm

L=10cm

2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.80

10

20

30

40

50

60

70

80

Frequency (GHz)

---

Re(Z

in)

( )

-

- -

Im

(Z

in)

( )

L

cable = 3 cm

Lcable

= 10 cm

Single antenna


44

(b)

Figure 42 Input impedances (a) and |S11| parameter (b) for three cable lengths (Lcable).

Radiation properties are also affected. Figure 43 compares the directivity pattern for the

single antenna case with the one obtained with feeding coaxial cable of 10 cm length.

(a) (b)

Figure 43 Directivity patterns of the single antenna (a) and with the finite feeding cable (b).

Thus, the radiating element is no longer the single antenna and has to be replaced by the

single antenna and the cable (see Figure 44). The radiation is then similar to that of an off-

centered dipole with an equivalent length greater than the wavelength [17].

Figure 44 {Antenna + cable} radiating structure.

4.2.1.2 Infinite length cable

In practice, the antenna is measured in anechoic chamber with a measurement cable several

meters long, which disappears in RF absorbers. This cable configuration is similar to that of

an “infinitely” long cable supporting propagating currents. Similar cable configurations have

2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8-20

-15

-10

-5

0

Frequency (GHz)

|S11| (d

B)

Lcable

= 3 cm

Lcable

= 10 cm

Single antenna

Radiating structure: {Antenna + cable}


45

been investigated using Finite-Integration Technique (FIT) electromagnetic simulation tool.

Absorbing boundary conditions (Perfectly Matched Layers (PMLs) in our case) are applied

directly at the extremity of the cable to simulate an infinite cable. This condition avoids the

appearance of a standing wave on the cable in simulation, as shown in Figure 45.

(a) (b)

Figure 45 Surface current on the cable (a) and E field (b) with PMLs boundary conditions.

Figure 45 shows that the propagating current along the cable is absorbed by the PML layers.

Consequently, there is no standing wave due to absence of significant reflected wave at the

cable extremity. To validate the exact PMLs behavior in the simulation it’s been verified that

the antenna input impedance is unchanged considering different cable lengths when PMLs

conditions are integrated.

In this case, the influence of the measuring cable is stabilized and we can properly compare

measured and simulated impedances. However, it is important to note that the input

impedance of the {antenna + infinite coaxial cable} structure is different from the one of the

single antenna case.

4.2.1.3 Quarter wave length stub (bazooka cable)

At frequencies where ferrite is not efficient, a quarter wavelength sleeve can be used to

reduce cable effect [18]-[20]. However, quarter wavelength sleeves can be used only on 10%

of bandwidth, which make them not suitable for wideband antennas characterization.

Alternatively dual band baluns can be designed for multiple frequency antennas [21], [22].

As first attempt to control the antenna surrounding and thus to properly compare

measurement and simulation, a quarter wave length stub (bazooka cable) has been

integrated on the cable [1]. To avoid the stub being in the antenna reactive area, the L

distance (Figure 46) between the antenna and the stub has been optimized.

PMLs boundary conditionsPMLs boundary conditions


46

(a) (b)

Figure 46 Radiated {antenna + stub} structure (a) and surface current on the cable with the stub.

A prototype of the {antenna + stub} structure was manufactured, by employing a

0.64pF capacitance value on antenna hat, and integrating a quarter wave length stub on the

RF feed cable (see Figure 47).

Figure 47 Prototype of the {antenna + stub} structure.

Measured {antenna + stub} structure results were compared with the simulated ones of the

same structure. Impedance performances (input impedance and |S11| parameter) present a

very good agreement between simulation and measurement as shown in Figure 48.

Radiated structure: {antenna + stub}

L’

Reactive areaReactive area

L


47

(a)

(b)

Figure 48 Comparison between simulated and measured input impedance (a) and |S11|parameter

(b) for the {antenna + stub} structure.

Nevertheless, even if the currents are strongly reduced on the cable (below the stub), the

use of bazooka cable is not straightforward when located in reactive field zone of electrically

small antenna. The impedance is affected by the presence of such element, and the results

are different from the one of the isolated antenna. Thus in small antenna designs, such as

the one here presented, it would be better to compare simulation and measurement

without introducing additional potential radiating elements other than the cable.

The next section presents a new methodology to take into account the cable effect, by

precisely characterizing its influence, and to recover the intrinsic properties of the isolated

antenna.

2.35 2.4 2.45 2.5 2.55 2.60

10

20

30

40

50

60

70

80

90

100

Frequency (GHz)

Zin

(

)

Measured Re(Zin

) Antenna+Stub

Simulated Re(Zin

) Antenna+Stub

Measured Im(Zin

) Antenna+Stub

Simulated Im(Zin

) Antenna+Stub

2.35 2.4 2.45 2.5 2.55 2.6-10

-9

-8

-7

-6

-5

-4

-3

-2

-1

0

Frequency (GHz)

|S11|

(dB

)

Measured S

11 Antenna+Stub

Simulated S11

Antenna+Stub


48

4.2.2 Impedance measurement methodology and results

To compare measurement and simulation, a prototype of the {antenna + infinite coaxial

cable} structure has been manufactured by employing a 0.64pF capacitance value for the

discrete capacitor on the antenna hat (see Figure 49).

Figure 49 {antenna + infinite coaxial cable} prototype.

The 0.505 mm diameter coaxial cable probe is directly soldered to the antenna top hat. The

outer conductor of the coaxial is welded to the antenna ground plane. The measurement

methodology to obtain consistent comparison between measurement and simulation is

explained in this subsection.

Firstly, measured {antenna + infinite coaxial cable} structure results were compared with the

simulated ones for the same configuration (with PMLs). As shown in Figure 50, there is a

very good agreement between measured and simulated impedance results (the reference

plane is the antenna ground plane) and reflection coefficient. It proves that the

measurement context is correctly considered both in experiment and in the 3D

electromagnetic simulation tool.

(a)

2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.80

10

20

30

40

50

60

Frequency (GHz)

Zin

(

)

Measured real part

Measured imaginary part

Simulated real part

Simulated imaginary part


49

(b)

Figure 50 Comparison between simulated and measured input impedance (a) and |S11| parameter

(b) for the {antenna + infinite coaxial cable} structure.

Since the measurement context is properly taken into account with 3D electromagnetic

simulation tools, we can use the simulator results to characterize the measurement cable

alteration and consequently to extract the antenna impedance alone. The interaction of the

measuring cable with miniature antenna depends on both elements’ position, orientation

and size. We propose to model the effect of the cable through a complex transfer function,

Himp(f), defined as the ratio between the antenna impedance Zca including an infinitely long

measuring cable and the isolated antenna impedance only Za. This transfer function is

specific to the antenna associated to the studied cable configuration (Figure 51).

(a)

2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9-8

-7

-6

-5

-4

-3

-2

-1

0

Frequency (GHz)

|S11| (d

B)

Measured |S11

|

Simulated |S11

|

Himp

H-1impZa Zca


50

(b)

Figure 51 Illustration of the impedance transfer function definition (a) and the infinite cable impact

on the input impedance (b).

)(

)()(

fZ

fZfH

a

ca

imp (7)

This transfer function (7) can be determined from the simulated single antenna impedance

and {antenna + infinite coaxial cable} results, the latter being in agreement with

measurements. Thus, by using the inverse transfer function with measurement results, we

are able to retrieve the properties of the antenna prototype without the disturbances

introduced by the measuring cable.

The comparison between simulated and measured single antenna results obtained by the

method presented above is presented in Figure 52.

(a)

2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.80

10

20

30

40

50

60

70

80

Frequency (GHz)

Zin

(

)

Himp

H-1imp

Without cable: real partWithout cable: imaginary part

Infinite cable: real partInfinite cable: imaginary part

2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.90

10

20

30

40

50

60

70

80

Frequency (GHz)

Zin

(

)

Measured real part

Measured imaginary part

Simulated real part

Simulated imaginary part


51

(b)

Figure 52 Comparison between simulated and measured input impedance (a) and |S11| parameter

(b) for the single antenna structure.

The excellent agreement between measured and simulated impedance results validates the

antenna design as well as the proposed impedance extraction methodology. The accuracy of

these results is directly related to two critical aspects of the technique: i) the correct

extraction of the transfer function obtained by simulating the infinite cable effect on the

miniature antenna; ii) the good agreement between measurement and simulation results of

the perturbed antenna configuration (this one also providing a good validation of the first

one). As consequence of the linearity of the transfer function, a bias in the estimation of

Himp(f), or in the measurements with cable, leads to inaccurate impedance results.

Nevertheless, even in presence of a small bias, the methodology provides more reliable

characterization of the isolated antenna, than the one obtained by neglecting the cable

effects.

4.3 Antenna radiation characterization

For the single antenna structure case, radiation and directivity patterns present

omnidirectional properties in the azimuth plane. The 3D directivity pattern at 2.47 GHz,

corresponding to the resonant frequency of the antenna loaded by the 0.64 pF capacitor, is

presented in Figure 53.

Figure 53 Simulated 3D directivity pattern at 2.47 GHz.

Radiation null of the dipolar pattern is tilted with respect to z-direction in the yOz

plane because of the off-centered feeding. The maximum realized gain for the single

antenna in the air is -18dBi.

2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9-25

-20

-15

-10

-5

0

Frequency (GHz)

|S11| (d

B)

Measured |S11

|

Simulated |S11

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