rlect · 2019. 7. 18. · 330 12 scsi 5.0m 3.5- 240k £999 425 12 scsi 5.0m 3.5 240k £1179 520 12...

THE INTERNATIONAL ELECTRONICS MAGAZINE

OCTOBER 1991 UK 1 90

RLECT

Digital function generatorFourterminal networks

Computer-controlleweather station

50 MHz 8 -bit DAC

Audio spectrumshift techniques

AM Receiver

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DENTALFUNCOON

GENERATOR

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FLOPPY DRIVES5.25': 360KB £54; I .2MB £63

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14' 1024X768 .28mm £2251024X768 .28mm £595

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SYSTEM CASES 4- PSUFlip top compact table top, 200W £ 85Low profile table top. 200W £ 99Stylish compact table top, 200W £ 99Mini -Tower table top, 200W £109Wide case (Full AT) desk top, 220W £125Midi -Tower (Full AT) table top, 230W £149Std Tower (Full AT) floor base 250 W £175

FLOPPY ADAPTORSAll cards are 8 - bit for PC /ATSupport 360K/720K /1.2M /1.44M:1 or 2 drives, intem. A / B £19

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RU. NO 8KB 225K 8 - bit £42WM YES 2KB 440K 16 -bit £58RU. YES 8KB 690K 16 -bit £69RLL YES 32KB 1 I M 16 -bit £89IDE NO depends ch drive 8 - bit £22IDE YES depends on drive 16 - bit C. 18

IDE YES plus 2 -serial & parallel 16 - bit £27ESDI YES 8KB 770K 16 -bit £90ES01 YES 32KB 1.IM 16 - bit £144

ESDI YES 64KB i 2.0M 16 - bit £189

9:31 -various 8 & 161:it for DOS.C6 /2. XENIX UND: E89 to E300

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MONITOR ADAPTORS & CONTROLLERSVGA adaptors all 16 - bit and alldo 640 x 480 x256 colours:256K8 800 x 600 x16 co::Jr £42512KB 1024 x 768 xI6 colour £791M8 1024 x 768 x256 £991MB D-4000 card compatb!e

OrcKd Predesigner 11 £139

TTL Graphics adaptor / printer portIBM / Hercu es £18; CGA emu aton £22p us NTSC / PAL ouu: £27T1GA graphic controller cards withAUTOCAD, WIndows3, etc drivers:16 - bit 34010 1MB video RrW, £44932 - bit EISA 34020 4MB video £1185

BELL 80X86 MOTHERBOARDScpu caK Lvm,20

RAMNAAx

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V20 12 5 I MB - 8/-/- £31

V30 10 6.6 IMB - 8/-/- £39

286 12 16 4MB - I/6/- £66

286 16 21 4MB 1/6/- £77

286 25 32 8MB - 1/6/- £139

386S 16 21 8MB - 1/5/- £179

386S 20 26 8MB - 1/5/- £219

386 20 27 I 6MB 2/6/- £299

386 25 34 (6MB 2/6/- £349

386 25 40 16MB 64 3/5/- £449

386 33 50 16MB 64 2/5/- £529

386 40 62 32MB 64 1/7/- £599

486 25 90 16MB OPT 2/5/- E I 149

486 33 117 32MB 64 -/8/- £1449

486 40 144 32MB 64 -/8/- £1649

486 25 90 32MB 128 -/2/6 £1899

486 33 120 32MB 128 -/2/6 E2 1 99

HARD DISC DRIVESFOWATTFJ

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TYPE1,,F

BUFFERWAX1X / RX j SIZEbytes / sec .. tires

21 40 1

1

MFM 625K 3.5' - £12932 40 1 RLL 937K 3.5' - £14949 28 RLL 937K 3.5' - £18965 28 RLL 937K 5.25- 1 - £22445 25 IDE 7.4M I 3.5' 1 32K £15990 19 IDE 7.4M 3.5" 1 32K £279

135 19 IDE 7.4M 3.5' 1 32K £399180 19 IDE 7.4M 3.5- ! 32K £489115 19 ESDI 1.2M 3.5' ' £359150 16 ESDI 1.2M j 5.25' i - £499350 14 1 ESDI 1.87 5.25' 1

- EB80660 16 I ESDI 1.87 5.25' I - £1190105 14 SCSI 4.7M 3.5' 64K £379210 14 SCSI 4.7M 3.5' 64K £679330 12 SCSI 5.0M 3.5- 240K £999425 12 SCSI 5.0M 3.5 240K £1179520 12 SCSI 5.OM I 3.5' 1240K £1349670 16 SCSI 4.0M 5.25' 64K £1399

1070 14 SCSI 4.0M 5.25' 256K £17991400 14 SCSI 5.OM 5.25' 256K £2299

ELEKTOR ELECTRONICS OCTOBER 1991

n5-177/67TOLtii 'Lli-liir_tik,A CONTENTS October 1991

Volume 17E! Number 193-ji ' i L W it 'sr_fi / _.T.

Front coverFunction generators areintended to produceseveral differentwaveforms and are,therefore, very flexibleinstruments. Most of themgenerate a sine,rectangular and triangularwaveform over afrequency range of up to1-50 MHz. Some alsohave the capability ofproducing pulsewaveforms. Generally, theperformance of a functiongenerator is inferior tothat of dedicatedinstruments, but thedigital design of whichthe first of three parts ispresented in this issue hasthe capability of accuratefrequency setting coupledwith very low distortionof the sine wave output.

In next month's issue

Class A amplifierDissipation limiterFunction generator [2]Experimental quadri-form ferrite antenna24 -bit full -colour videodigitizerFour -terminal networks- Part 2Computer -aidedelectronics design

Copyright _ 1991 Elektuur BV

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11 Scientific travels in real and virtual worlds

APPLICATION NOTES

59 Switch -mode voltage regulators LM2575/LM2577National Semiconductor

AUDIO & HI-FI

56 The digital compact cassettebased on an original article by P. van Willenswaard

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Upgrade for MCS-BASIC-52p.14

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COMPUTERS & MICROPROCESSORS

14 Upgrade for MCS BASIC -52 V1.1 - Part 1 (of 2)by Dusan Mudric and Zoran Stojsavljevic

DESIGN IDEAS

16 Audio spectrum shift techniquesby C. White Halfoat

GENERAL INTEREST

28 PROJECT: Computer -controlled weather station:Part 2: Electronic hygrometerby J. Ruffell

46 Four -terminal networks - Part 1 (of 2)by Steve Knight

INTERMEDIATE PROJECT

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23 Build the Opticalock - Part 2 (fmal)by Michael Swarizendruber

RADIO. TELEVISION & COMMUNICATIONS

-19 Fiber optics - Part 2 (final)by Joseph J. Carr

52 PROJECT: AM broadcast receiverby R. Shankar

SCIENCE & TECHNOLOGY

42 A review of coding theoryby Brian P. McArdle

TEST & MEASUREMENT

26 PROJECT: 50 MHz 8 -bit DACby R. Shankar

32 Measurement techniques -Part 8by F.P. Zantis

34 PROJECT: Digital function generator - Part 1 (of 3)by T. Giffard

MISCELLANEOUS INFORMATION

Electronics scene 12; Events 13; New books 13;Readers services 63; Switchboard 64;Terms of business 64; Index of advertisers 74.

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COM LETE P KITS FOR ALL -ELEKTORELEKTOR ELAWRONICS PROJECTSaftBefore you start buying components for the latestproject, please contact us.We can supply completekits of components (excluding PCB and enclosure)for all of the projects featured in this and otherissues.

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SCIENTIFIC TRAVELS IN REAL AND VIRTUAL WORLDSAN exciting new v. orld is being built in

Britain that appears, in some respects,to be a fantasy, straight from the pages of ascience fiction novel, but it is utterly unlikethe worlds envisaged by classic SF writerssuch as H.G. Wells and Jules Verne-it is theworld of Virtual Reality.

Virtual Reality (VR) is a fast-growingbranch of Information Technology (IT), oneof Britain's strongest suits in today's com-petitive research market. VR is entirely com-puter -generated and is as like the real worldas computer graphics can make it.

In its simplest form, VR is experiencedby one person wearing a pilot -style helmetthat seals the senses off from the outsideworld. A typical VR helmet contains twoVDU screens, filling the visual field with acomputer -simulated landscape fed via ca-bles from a powerful, specially -program-med computer. The computer reacts to headmovements by updating the visual displayinside the helmet so the wearer feels as if heor she is surveying an actual landscape.

In some sophisticated VR systems, theparticipant also wears a special glove that re-lays hand movements to the computer. Inthese systems, a hand depicted in computergraphics also appears in the visual displayinside the helmet.

Thus, if you reach out to touch a virtualobject in the virtual reality environment, yousee your virtual hand mimic the movementsof your real one. Some gloves are alsoequipped to give the wearer an impressionof touch and weight so that objects in thesimulated world can be apparently caughtand held, as if they had physical substance.

Serious applicationIn more complicated systems, two or morepeople can interact and, because the virtualworld is entirely computer -generated, dis-

tance is no object. Delegates at a conferencein the United States recently watched VDUscreens in amazement as two people in dif-ferent cities shook hands in VR via linkedcomputers and, of course, the indispensablehelmet and gloves.

Two British companies are among thefront-runners in this strange new technol-ogy. One, Advanced Robotics Research ofSalford is developing remote -controlled ve-hicles that can be manoeuvred in VR by a

Helmet -mounted display system asused in 'virtual reality' experiments at

Advanced Robotics Research Ltd.

driver sitting somewhere else.This has the serious practical aim of pi-

loting vehicles into hostile environments,such as the cores of nuclear reactors or sub-merged wrecks, without exposing a driver toobvious danger. He or she will view a virtualenvironment fed from sensors on the vehicleitself.

The other company, W Industries ofLeicester, is exploring VR's potential for en-tertainment. It has launched a machine of -

fering a variety of simulated experiences thatcan be enjoyed in a purpose-built seatequipped with two joysticks and a helmet(called a Visette) wired for sound.

One of the offered experiences is that ofpiloting a Harrier jump -jet on a fighter mis-sion involving attack by enemy aircraft,mid-air refuelling, and a hair-raising landingon the deck of an aircraft carrier.

Improving computersDesigners will soon be able to experiencetheir designs at simulated first hand. Archi-tects will enter VR to walk around theirbuilding on the other side of the world at theplanning stage. In the United States, sur-geons are already experimenting with safevirtual worlds in which they can practise vir-tual operations on virtual patients, beforetrying their hand at the real thing.

Meanwhile, British computer scientistsare trying to make computers more power-ful. A team at Essex University has begunlooking for ways to change the basic logicalstructure of computers, which has changedsurprisingly little in the past 25 years.

Professor S.H. Lavington believes that,although silicon chip technology is slowlymaking computers faster and cheaper, manycomputers and their languages are becominginadequate for solving many of the problemsnow confronting science and industry. Theinadequacies, he says, relate not only toshortcomings in performance, as measured,for instance, by speed of execution, but tothe time and effort taken to produce error -free programs.

Solutions being investigated includevarious forms of parallel processing inwhich information can be fed into linkedcomputers simultaneously, allowing them todivide processing tasks between them forgreater speed and accuracy.

Produced and published by ELEKTOR or +44 580 200 657 (International ) GERMANY NETHERLANDSELECTRONICS (Publishing) Telefax: (0580) 200616 (National) Elektor Verlag GmbH Elektuur BV

or +44 580 200 616 (International) Siisterfeld StraBe 25 Peter Treckpoelstraat 2-IEditor Publisher: Len Seymour 5100 AACHEN 6191 VK BEEKTechnical Editor: J. Bulling European Offices: Editor: EJ.A. Krempelsauer Editor: P.E.L. KersemakersEditorial Offices: Postbus 75 GREECE PORTUGALDozen House 6190 AB BEEK Elektor EPE Ferreira & Bento Lda.Broomhill Road The Netherlands Kariskaki 14 R.D. Estefini, 32-1«,LONDON SW18 41Q Telephone: +31 46 38 94 44 16673 Voul a - ATHENA 1000 LISBOAEngland Telex: 56617 (elekt n1) Editor: E. Xanthoulis Editor: Jeremias SequeiraTelephone: 081-877 1688 (National t Fax: +31 46 37 01 61 HUNGARY SPAINor +44 81877 1688 (International ) Managing Director: MALI. Landman Elektor Elektionikai folyorat Resistor Electronica AplicadaTelex: 917003 t LPC G) 1015 Budapest Calle Maudes 15 Endo C.Fax: 081-874 9153 (National Distribution: Batthydny u. 13 28003 MADRIDor +44 81874 9153 (international 1 SEYMOUR Editor: Lakatos Andras Editor: Agustin Gonzales Buelta

1270 London Road LNDIA SWEDENAdvertising: PRB Limited LONDON SW16 4DH Elektor Electronics PVT Ltd Electronic Press AB3 Wolseley Terrace Chhotani Building Box 5505CHELTENHAM GL50 ITH Printed in the Netherlands by NDB, 52C, Proctor Road, Grant Road (El 14105 HUDDINGETelephone: (0242)510760 Zoeicreoude BOMBAY 400 007 Editor: Bill CedrumFax: (0242) 226626 Editor: C.R. Chandarana USA & CANADA

ISRAEL Elektor Electronics USASubscriptions: Overseas editions: Elektorcal P 0 Box 876World Wide Subscription Service Ltd. FRANCE PO Box 41096 PETERBOROUGH NH 03458-0876Unit 4, Gibbs Reed Farm Elektor sari TEL AVIV 61410 Publisher: Edward T. DellPashley Road Les Trois Tillculs Publisher: M. AvraharnTICEHURST TNS 7HE B.P. 59. 59850 NIEPPETelephone: (0580) 200657 (National ) Editors: D.R.S. Meyer.G.C.P. Raecirrsdorf


ELECTRONICS SCENE

CONSUMER ELECTRONICSMARKETS IN 1990Overall, for most consumer electronics prod-ucts, 1990 maintained the downturn thatbegan in the last few months of 1988 as in-terest rates rose and continued throughout1989. Television deliveries in total were22% down on their peak in 1988, especiallyin small screen sizes; larger sets benefitedfrom a core, non -postponable, distress -re-placement demand. Video cassette recordersthrough most of the year were barely af-fected owing to falling prices, improving prod-uct specification and a strong element offirst-time buying. Sales of camcorders andCD players continued to expand, buoyed byfirst-time demand for these innovative prod-ucts (albeit at reduced rates of growth com-pared with last year). In general, sales of audioproducts were weak.

As NICAM transmissions spread (cover-ing around 75% of the population by the endof the year), sales of NICAM-equipped TVsand VCRs continued to grow strongly, pro-viding an overall lift to large -screen TVs es-pecially.

During the last quarter, VCR demand ap-peared to begin to weaken, while a flat year-on-yearperformance by camcorders reflectedthe absence of discounted secondary -brandmodels this year rather than static demand forthe product.

INFRA -RED LIGHT BARRIER KITSA DIY infra -red light barrier can be con-structed easily with two new kits: the K2549transmitter and the K2550 receiver. Theseunits may be placed up to five metres apart.

The PULSAR digital logic simulator from Number One Systems, priced at only £195,is the answer to many a digital design engineer's prayers. Until now, the restrictedcomponent libraries, complexity and high cost of the majority of available digitallogic simulator programs put them out of reach of most engineers. Further informa-tion from Number One Systems, Harding Way, Somersham Road, St Ives, Huntingdon,Cambs PE17 4WR; Telephone (0480) 61778; Fax (0480) 494 042.

The kits are available by mail order fromElectronic & Computer Workshop Ltd, Unit1, Cromwell Centre, Stepfield, Witham,Essex CM8 3TH; Telephone (0376) 517 413.The price is £54.00, incl. VAT and p&p.

STEREO SOUND ON BBC TVWork on equipping 10 main transmitters and402 dependent relay stations of the BBC to

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White Technology's new Model WS-XM xX module is a low -power 64-Mbit CMOSSRAM in a compact 120-pin hermetically sealed flatpack that can be user -configuredas 8 Mbit by 8 bits, 4 Mbit by 16 bits, or 2 Mbit by 32 bits. The module will also op-erate in three modes depending on which signals are used. Additional informationfrom White Technology. Inc., 4246E. Wood St. Phoenix, Arizona 85040; (602) 437 -

broadcast BBC -1 and BBC -2 programmesin stereo has now been completed and trans-missions have started.

Initially, nearly three quarters of the Britishpopulation will be able to receive TV withstereo sound, but BBC Television says it iscommitted to extending the service through-out the country, and over the next four yearsfour additional transmitters are to be mod-ernized to increase stereo coverage to 84 percent of the population.

The Near Instantaneously CompandedAudio Multiplex (NICAM) system was ac-cepted in 1986 as the UK standard, and, in1987, it was also recommended by the EuropeanBroadcasting Union (EBU) as the digitalstandard for terrestrial TV stereo broadcastsin Europe. Already, over 100 different typesof TV receiver and 30 video recorders fittedwith NICAM decoders are on sale in theUnited Kingdom.BBC Engineering, White City, 201 WoodLane, London W12 7TS.

WORLD LEAD IN FIBRE OPTICSGEC -Marconi has demonstrated a new op-toelectronics telecommunications system, de-veloped under the European Commisions'sResearch in Advanced Communication s forEurope (RACE).

The demonstration, the first of its kind inEurope, is claimed to represent an advancein the application and engineering of opto-electronics technology that gives Europe atechnical lead over the USA and Japan.GEC -Marconi Materials Technology Ltd,Caswell, Towcester, Northamptonshire.


DIG1TALSYSTEMSREFERENCE BOOKEdited by B. Holdsworth andG.R. MartinISBN 0 7506 1008 5Price £120.00 (hard cover)This monumental reference book provides acomprehensive coverage of the field of dig-ital systems in a concise and authoritativeform. Concise it may be, but this reviewer,and many of his engineering acquaintances,could not fmd anything missing that mightbe required during a normal day's work inthe laboratory, college or (technical writer's)office.

The work is divided into five parts:Fundamentals; Devices for digital systems;System design and techniques; System de-velopment; and Applications. Each part con-tains several self-contained chapters, eachof which has been written by an acknowl-edged expert from industry or academia.

As an up-to-date, authoritative and com-prehensive reference work, this book shouldmeet the needs of everyone working in thedigital field, including engineers, academicsand scientists.Published by Butterworth -Heinemann, thebook may be ordered fromReed Book Services Ltd,PO Box 5,Rushden NN10 9YX.

USING TELEVISION AND VIDEOIN BUSINESSby Andrew CroftsISBN 1 85251 099 4166 pagesPrice £14.95 ($24.95) (hard cover)Technological developments are turning thetelevision screen into the perfect mediumfor two-way communication. At the sametime, changes in international laws are open-ing up opportunities for sponsored televi-sion and for video publishing projects thatcan provide companies with new opportuni-

NEW BOOKS

ties to sell and promote themselves and theirproducts to new audiences.

Before long, television screens will be asintegrated into daily business life as telephonesand computers are now; companies that knowhow to use them effectively will gain a sig-nificant competitive edge.

Thisbook takes thereader through the vari-ous different ways in which television andvideo can be used and misused within thebusi-ness world, and attempts to predict what thefuture will bring.The Michael Hyde Partnership44 High StreetBalsham, Cambridge CBI 6EP

AUDIO ANTHOLOGY VOLUME FOURWhen audio was youngcompiled by C.G. McProudISBN 0 9624191 9 2Price £10.95 ($16.95)This book is a reprint of a collection by thefounder/editor of Audio Engineering, and wasfirstpublished in 1958. Articles were selectedfrom the pages of what had then becomeAudio magazine. It is the fourth of the sixvolumes in the series.

Apart from transistors amplifiers, the bookalso covers a number of valve circuits. Thisis especially apposite today when vacuumtube technology is being rediscovered and of-fered by a host of manufacturers in what istermed 'the high end' of audio.

Also included are 'How to plan your hi-fi system', 12amplifiers and pre -amplifiers,introduction to solid-state techniques, six loud-speaker designs, and some designs for tapeamplifiers for stereo tape playback: a verynew technology in those days.

This fourth collection reflects the matur-

IEE AND IEEIE PROGRAMME

2-4 Oct-Electronic security systems.11 Oct-What's new in the 16th edition? (For

college lecturers).28-31 Oct-Eighth International Conference

on Automotive Electronics.

Information on these events organized bythe lEE or the IEEIE may be obtained fromthe IEE, Savoy Place, London WC2R OBL,Telephone 071 240 1871 or from the IEEIE,Savoy Hill House, Savoy Hill, London WC2ROBS, Telephone 071 836 3357.

During October, Frost & Sullivan will con-duct seminars on a variety of topics in thefield of electronics, computers and com-munications. Further information from Frost& Sullivan, Sullivan House, 4 GrosvenorGardens, London SW1W ODH, Telephone071 730 3438.

EVENTS

Total Solutions, the control exhibition to beheld at the National Exhibition Centre,Birmingham from 29 to 31 October, willinclude a PC Workshop in which visitorswill be able to gain 'hands on' experienceof the latest developments in programmablecontroller systems.More information from Evan SteadmanCommunications Group, The Hub, EmsonClose, Saffron Walden, Essex CB1O 1HL,Telephone (0799) 26699.

The IBTS'91-International Audio, Video,Broadcasting and Telecommunications Show-and MeM'91-International Market forAudio and Video Programs and Services-will be held in Milan from 17 to 21 October.

ing of home sound systems, the appearanceof stereo and the beginnings of an evolutiontowards solid-state devices. It provides aclear and authoritative picture of audio as acraft in the late 1950s.Audio Amateur Publications305 Union StreetPost Office Box 576Peterborough NH 03458 - 0576, USA

SERVICING AUDIO ANDHI-FI EQUIPMENTby Nick BeerISBN 0 7506 0076 4210 pages - well illustratedPrice £25.00 (hard cover)Written especially for service technicians andengineers, this book is intended for use as abench -side companion and guide. Its purposeis to ease and speed up the processes of faultdiagnosis, repair and testing of all classes ofhome audio equipment receivers, amplifiers,recorders and playback machines.

It examines the mechanics and electron-ics of domestic audio in a down-to-earth andpractical way, concentrating on what goeswrong, how to track down problems andhow to solve them. Based on hard-won ex-perience on the bench and in the field, thisbook is the next best thing to having a time -served engineer sitting alongside at thebench:advising, guiding, cautioning, pointing thereader in the right direction.

Contents range from the simplest AMradio to the intricacies of CD and DAT sys-tems. Along the way such diverse subjectsas servos and speakers, dial cords and dirtyheads, motors and microprocessors, turnta-bles and transistors are examined, togetherwith the techniques and test equipment neededto sort them out and set them up.Butterworth -HeinemannLinacre House, Jordan HillOxford OX2 8DP

I B TS, 20149 Milano, ViaDomenichino. 11:Telephone (02) 481 5541.

The Computers in Manufacturing Showwill be held at the National Exhibition Centre,Birmingham, from 8 to 11 October.More information from Independent ExhibitionsLtd, Weybourne House, 2 London Street,Chertsey KT16 8AA, Phone (0932) 564 455.

Within the framework of TELECOM 91, the6th World Telecommunication Forum willbe held in Geneva from 10 to 15 October.The forum, highly technical and scientific,yet universal in character, is now recognizedas the summit for the interchange of ideas inthe fast evolving scene of world telecom-munications.More information from Forum Secretariat,ITU, Place des Nations, CH -1211 Geneva 20.


09

V II

2A117 FOR3AS =52

ir'oivi 1)The 8052AH-BASIC from Intel is a versatilemicrocontroller with a powerful BASICinterpreter lurking in its on -boardmask -programmed ROM. The authors,having worked with this IC for some time,discovered certain flaws in the BASICinterpreter, and set out to produce abetter, faster version that can be run fromEPROM.

by Dusan Mudric and ZoranStojsavljevic

TO be able to make changes to the MCS-1. 52 BASIC interpreter in the 8052AH-

BASIC microcontroller, it is necessary to firstunload it from the IC. This is done basicallyas described in an earlier article on the MCSBASIC -52 interpreter, Ref. 1. The result ofreading the 8-KByte ROM is a file in Intel -Hex format that contains the machine codeof the MCS BASIC -52 interpreter (version1.1).

MCS BASIC -52 (Ref. 3), extracted fromthe 8052AH-BASIC V1.1 microcontroller,was dicassembled and texts, tables and con-stants were extracted in order to produce anassembler version of the interpreter. The sizeof this assembler file was approximately4,000 lines. Studying the program, we foundthat the operation of the interpreter could beimproved by rewriting certain lines of as-sembler code. Subsequently, a number of al-gorithms were developed and substitutedfor the ones originally implemented by Intel.Furthermore, errors found in a number ofroutines were corrected.

Floating point nucleusOne of the routines in the BASIC interpreterfound to contain programming errors is thefloating-point arithmetic nucleus. The errorscan be demonstrated by running two smallprograms:

10 a=.10000001E3020 b=.99999993E2930 ?a-b

The result, 2.74E22, is erroneous, and shouldbe 1.7E22. Similarly,

10 a=.10000001E3020 b=.99999997E2930 ?a-b

ADDR CODE INSTRUCTION

19F2H 752A00 NOV 2AH,100H ; +0001

19F5H 71C8 ACALL 1BC8H

I9F7H 7F04 NOV R7,I04H ; +004T

19F9H 792E NOV R1,12EH ; +0461

19FBH 749E NOV AMEH ; +1581

19FDR C3 CLR C

19FEH 9C SUBB

I9FFH D4 DA

1A00H CC XCH

1A01H 7001 M.1A03H FC MOV

1A04H 845000 CJNE

IA07H 302318 JHB

1A0AH 83 CPL

IAOBH 5119 ACALL

1A0DH 5008 JNC

IAOFH 052A INC

A,R4

A

A,R4

1A04H ; $ 03H

R4,A

A,150H,1A07H ; +080T ; $ + 03H

23H,1A22H ; $ + 1BH

C

1A19H

1A17H ; $ + OAH

2AH

910128-12

Fig. 1. Original floating-point nucleus in the ACS BASIC -52 interpreter.

produces 1_34E22 instead of 1.3E22.The disassembly listing of the original

floating point nucleus developed by Intel isgiven in Fig. 1, and the version developed bythe authors in Fig. 2. When implemented inthe BASIC interpreter, the nucleus shown inFig. 2 produces the correct answers to theabove subtractions.

Other corrections

Further improvements were made to thehex -to -BCD conversion routine, both in re-gard of efficient programming and speed.For example, two approaches are possiblefor extracting BCD digits a, b, c, d, and e in

xyzwH = aD*10000D + bD*1000D + cD*100D+ dD*10D + eD*1D

These possibilities are:1. successive extraction of BCD digits

starting with the most significant digit, a;2. successive extraction of BCD digits

starting with the least significant digit, e.

If the original version of the hex -to -BCD con-verter is studied, it is seen that the first pro-cedure is employed. The DPTR is used as a'weighted register', and the procedure isbased on finding a suitably weighted sub-traction number from a variable value.


IIUPGRADE FOR MCS BASIC -52 V1.1 (Part 1)

ADDR COST INSTRUCTION

19F2 7521111 5699 nov FP 159-1,919 ; preparation of eg:al erne/lents

19F5 713 5798 ACALL SHIFT RIGHT

19f7 7fI4 5711 4CV R7,1LEN_SYIE

19f9 792E 5792 1518 R1,1FPISB

5713

5794 ;FPEFPEFFEFFEFPEPEREFPEFPEREPEFFUPEFPEFPEREFfEFPEREFPEFPEREFfEffE

5715

5796 ; FLOATING POINT MRS, FOUND BY 0.81DRIC AND 1.STOJEAVLIEVIC

571?

f"?' ; NOV A,19E4 ;ERRS; WISER 91

573;

:711 ; VALUE IN R4 63ST BE COPLEMENTED WITH 191D f99A41, IT. HOST BE TrE

5711 ; FIRST CONPLEIENT

5712

5713 ; C..R

5714 ; SUSS AM5715 ; DA A

5716 ; Its

5717 ; 3RD 4+3

5713 NOV 84,4

5719

5729 ; ERROR HVBER 92

5721

5722 ; WITH SURSTRACTION, AFTER REDUCERS BOTH TIE MINEND MD THE

5723 SUBTRAHDE TO THE SAX EIMGENTS, MEN R4 (I t, IT IS OBVIOUS

5724 THAT CIE ALWAYS HAS TO RAKE A WARNING FRO!! THE FIRST HIGHER

5725 ; 105IT73I CF TIE HINVEND, NOT AS IT IS STATED BY THE DRISINAL

5726 ; INEFE IT IS RAM ONLY WHEN R4 514

5727

5728 ; UNE 4,1514,1+3 ; deal uith rounding

5729 1113 FFISER,FP SlIBB ; test for subtraction iatention to carry)

¶73$

573: ;FFSFPEFFEFFEREFPEFFEFFEITIFPFJPEPEFFEREFFEFPUPEfFEFFEFBEFFEREFPEFFE

5732

5733 ; PROPER ROUNDING, 11EVELOPU BY Lk-1MM

5731

MB 749A 5735 NOV AMAX191D C3 5735

:::E 9C 5737 SUB 1,44

:A 5738 DA A

:Aia CC 5739 IC4 4,84

Ikil 39231E 5749 343 FASEER,FPSOD

1414 54913 5741 CUT A,15f4,1+6

1417 B1 5742 13 1,1,1

IAIB it

:41; 8!

5743 ; cantina loraal cede

5744

:AM 33 5745 Eft C

:An 5:19 5746 AC4 477 7H,4

:AC 5112 5747 at 7r,

:6!E 1524 5748

_":_73S

INE77 .7,

910128-13

Fig. 2. Assembly listing of the improved floating-point nucleus.

OLD:

10801090

OF A505 6D

0212

0619

9FA3

12

1205 73OE A5

7B 00A3 B9

79OD

0700

A3 II40 22

IAA3

12EO

19F0 7F OA 75 24 00 71 C8 7F 04 79 2E 74 9E C3 9C D41400 CC 70 01 FC B4 50 00 30 23 18 B3 51 19 50 08 05

IFOO E4 38 FB 22 84 90 27 10 Fl 21 90_03 E8 Fl 21 90IFIO 00 64 Fl 21 90 00 OA Fl 21 90 00 01 Fl 21 60 201F20 22 78 FF OE CA B5 83 00 CA 40 12 C8 95 82 C8 CA1F30 95 83 CA 50 EE C8 25 82 C8 CA 35 83 CA 4E 60 801F40 74 30 2E 88 AO F3 09 B9 00 01 OB 22 DI A7 A2 36

NEW:

10801090

OE AS12 19

02A3

0600

9F00

12

000500

7300

7600

00B9

79OD

0700

A3 12 05 6D40 22 A3 EC,

19F0 7F OA 75 2A 00 71 CS 7F 04 79 2F. 74 9A C3 9C 041400 CC.30 23 1E B4 50 03 00 00 00 B3 51 19 50 08 05

1F00 E4 3B FB 22 7D 00 EA 75 FO OA 84 FA E8 54 FO 451E10 FO C4 75 FO OA 84 C4 FE £8 54 OF C4 45 FO C4 751E20 FO OA 84 4E F8 E5 FO 24 30 OD CO EO EA 48 70.1161F30 DO EO 8B AO F3 09 B9 00 01 OB DD F4 22 00 00 00

' 00 00 00 00 00 00 00 00 00 00 00 00 DI 47 A2 36

910128 -11

The second procedure is based on suc-cessive division of the variable value by 10,where, with each division, one BCD digit ap-pears as the remainder.

By simplifying the conversion to suc-cessive division -by -10, the original code of72 bytes is reduced to 57 bytes. The im-proved hex -to -BCD converter no longer re-quires the DPTR contents to be stored,returned and incremented after the conver-sion is finished. As a result, it is faster thanthe original converter implemented by Intel.Table 1 shows a few 'bench -mark' ex-amples.

Table 1. Hex -to -decimal conver-sion time

P (hex) P (dec) t-, -a (ms) t (ms)

0000 0000 141 56

0009 0009 276 56

4000 16384 507 268

EA5F 59999 820 268

Action!If you are interested in the improvementsmade to the MCS BASIC -52 interpreter asdescribed here, unload the interpreter from a8052AH-BASIC, modify it, put it back into anEPROM and run it either with an 80C32, oras a turnkey -EPROM with an 8052AH-BASIC. All this is pretty straightforward anddescribed at length in Refs. 1 and 2. First,copy the interpreter into an EPROM. Next,use the OLD function of the EPROM pro-grammer on the BASIC computer (Refs. 4, 5)to make sure that an exact copy is available.Then load the contents of the EPROM into anEPROM programmer, and use the edit func-tion to make the changes indicated in Fig. 3.Finally, program a new EPROM with themodified BASIC interpreter. The EPROMused may be a 27C64, a 27C128 or a 27C256.Alternatively, you may want to use an EE -PROM Type 2864A.

References:1. "CMOS replacement for 8052AH-BASIC",Elektor Electronics January 1990.2. "ROM -copy for 8052 BASIC computer",3. "MCS BASIC -52 Users Manual", IntelCorp., 1986.4. "BASIC computer", Elektor ElectronicsNovember 1987.5. "8032/8052 Single -board computer",Elektor Electronics May 1991.® MCS BASIC -52 is a registered trademarkof Intel Corp.

Fig. 3. Overview of address locations to bemodified in the 8-KByte interpreter. Remem-ber. you can not overwrite data just like thatin an EPROM -a 0 will remain a 0 unless youerase the EPROM using ultra -violet light. Themodified version of the interpreter must.therefore. be loaded in a blank EPROM.


16

DESIGN SEAS

The contents of this article are based solely on information obtained from the authorand do not imply practical experience by Elektor Electronics.

AUDIO SPECTRUM SHIFT TECHNIQUES

Although spectrum shiftand spectrum inversionhave been known for quitesome time as a means ofprotecting voice linksmainly over the telephone,little has been written onthe design and operationof the electronics involved.This article aims at offeringyou a design base and twopractical circuits to startexperimenting with aninteresting form ofanalogue audio scramblingand descrambling whichmay be used to, say, personalize a voice link (cassette mail, telephone, wirelessbabysitter, etc.). Remarkably, you will not be able to tell the difference between theencoder and the decoder until you compare the two units and spot a couple ofcomponents with different values.

by C. White Halfoat

(-NNE may wonder why analogue audioV scrambling is still used when digital en-coding systems are relatively simple to im-plement whilst offering a good degree ofsecurity. The reason is simple: digitizedaudio data requires a bandwidth which isgreater than that of the analogue source sig-nal. Thus, where a limited bandwidth isavailable to convey an audio signal - say,3 kHz for voice signals, and 15 kHz formusic signals - digital encoding is simplyout of the question, and different meansmust be sought to protect the information asit is conveyed from the transmitter to thelicensed receiver.

Encoding principleAn all -analogue encoding system is de-scribed that renders an audio signal totallyunintelligible by shifting the entire spectrum

between 50 Hz and about 10 kHz by 1 to2 kHz within the existing channel. When theshift is 1 kHz, a source signal of 50 Hz isshifted to 1,050 Hz, while a source signal of10 kHz is shifted to 11 kHz. In this system,the frequency range below 1,050 Hz is, inprinciple, empty after encoding. The shiftoperation is shown diagrammatically inFig. la. Figure lb shows an alternative ver-sion, where the audio spectrum is addition-ally mirrored around a certain frequency.Both encoding systems yield a new audiosignal that is virtually unintelligible, withspeech pauses and stress on utterances (i.e.,volume variations) as practically the onlyrecognizable features (in fact, we have heardsuch scrambled signals described as thesound of "a lot of squabbling chipmunks").

Each of the two encoding systems de-scribed by Figs. la and lb has its advantagesand disadvantages. The second system(Fig. lb) is simple to realize both at the trans-mitter and the receiver. However, when ap-plied to an FM (frequency modulation)

Fig. 1. Principle of spectrum shift (upperdrawing) and spectrum inversion (lowerdrawing). Both techniques are used toscramble audio signals conveyed over air,via cables or magnetic media.


AUDIO SPECTRUM SHIFT TECHNIQUES 1111

communication link, the signal-to-noise(SIN) ratio at the receiver side suffers be-cause the shifted audio signal contains morehigh -frequency components than the orig-inal (source-) signal. By contrast, when thefirst encoding system (Fig. la) is applied, thebulk of the intelligence is carried by compo-nents in the lower part of the spectrum, sothat SIN degrading will not occur (in fact,there is psychoacoustic evidence that mostinformation in speech and music is con-tained in the frequency range from 300 Hz to1,000 Hz).

The system whose spectrum is shown inFig. la yields almost unintelligible outputsignals, and is not easy to decode. The de-coding process required to restore the signalto its original spectrum can be subdividedinto a number of operations as illustrated inFig. 2. As shown in Fig. 2a, the first functionof the decoder is to limit the bandwidth ofthe encoded signal, i.e., to fit this into a fre-quency range from 500 Hz to 10 kHz. The fil-ter slope at the high end of the spectrumprevents intermodulation products beinggenerated in the decoding process, while thehigh-pass filter prevents whistles between1 kHz and 2 kHz as a result of low -frequencycomponents that are possibly added to thechannel.

The block diagram of the decoder (Fig. 3)shows that the input signal is taken througha low-pass filter before it is amplified and ap-plied to a high-pass filter. A switch con-nected ahead of the output buffer allows theuser to select between encoded and non -en-coded signals. When the switch is set to 'non -encoded', the input signal is fed direct to theoutput buffer, i.e., it is not amplified. This isdone to ensure that the signal level of non -encoded signals is not changed by the

decoder. Thus, the amplifier merely serves tocompensate losses introduced by the phasefilters used for encoding or decoding theinput signal.

Frequency transforms:Hilbert & Fourier

Shifting an audio spectrum can be achievedin three ways, which are familiar from radiocommunication techniques used for genera-ting and demodulating SSB (single -side -band) signals. The first two systems arebased on narrow -band filters. These arewidely applied, and do not present problemswith audio signals whose frequency range,for radio communication purposes, extendsfrom 300 Hz to 3,000 Hz only.

Transmitting music via a communicationlink based on spectrum shifting poses moreproblems than speech because a much largerfrequency spectrum must be conveyed with-out distortion. It is for this, and other, reason-s, that the present encoding system is basedon the so-called 'third method', a term fam-iliar to most licensed radio amateurs andradio engineers (Ref. 3).

So how does the encoder work? First, theaudio signal is split into two components.Both have the same amplitude, and containthe source signal. The only difference be-tween these components is the phase shiftbetween the frequencies they consist of. Thisphase shift is 900, and one of the componentsis called the Hilbert transform of the other(the two signals are called a Hilbert pair). AHilbert transform is a frequency shiftingoperation that unfortunately exists in theoryonly. In practice, however, a number of sys-tems have been developed that yield dose

Fig. 2. Multiplication with two phase -lockedcarriers, filtering and sideband suppressionare the main functions into which the decod-ing process can be subdivided.

approximations of such an ideal transform.One of these systems is applied in the pres-ent encoder.

Fig. 3. Block diagram of the frequency shift encoder/decoder. The FC H L input selects one of two system clocks from which the amountof frequency shift is derived. Note that the encoder/decoder has a bypass function. which is controlled via the S. 0 1 input.


18 AUDIO AND HI-FI

The Hilbert pair is required to suppressone of the sidebands that is generated whenthe filtered input signal is shifted ('trans-formed') to a higher frequency. The multiplyoperation carried out after shifting the phaseof the input signal over +45° and -45° givesrise to two double-sideband (DSB) signals. Inthe decoder, these signals appear around a24 -kHz component - the actual frequencydepends on the shift used during the encod-ing, and will be discussed further on.

As shown in Fig. 2b, one of the sidebandsdisappears when the two DSB signals areadded. The mathematics underlying thisoperation are shown inset in Fig. 4, a draw-ing reproduced from an earlier article in thismagazine on frequency transformation(Ref. 1). The operations shown in Fig. 4 de-scribe largely what happens in the decoder.The equations go to show that there is noway to avoid a Hilbert transform functionwhen one of the sidebands is to be selectedwithout resorting to (very complex) filters.As shown in Figs. 2b and 2c, the upper side -band is removed in the present design.

Multipliers or double -balanced mixerscan take a number of practical shapes. Al-though their design is mostly plain sailing,they do have to be balanced with the aid ofadjustments, and may have insufficient li-nearity for some applications. Good resultsat a very small outlay, and without the needof adjustments, can be achieved with elec-tronic analogue bilateral switches, e.g., thosecontained in the 4053 CMOS multiplexer IC.These switches are particularly well suited touse in the decoder where they enable highsuppression of the 24 kHz component to beachieved. When this suppression is insuffi-dent, a 1.5 -kHz signal (whistle) would read-

ily appear in the decoded spectrum, corre-sponding to the amount of frequency shiftapplied. The main disadvantage of the anal-ogue switches is that they generate a fre-quency spectrum with spurious componentscentred around odd -numbered harmonics ofthe 24 -kHz switching signal. It will be re-called that switching is the same as multi-plying with a rectangular signal. Thus, interms of the spectrum, switching can be de-scribed as a multiplication with the Fourierseries

sin OW- V3sin(30)0+1frasin (5o.11)-1/2sin...

The sidebands of the third harmonic, whichare visible in Fig. 2c, are removed with theaid of a low-pass filter designed for a steeproll -off at 30 kHz. This is done to prevent thesidebands causing interference in the mixingoperation that restores the original fre-quency band.

The actual decoding process is relativelysimple. As shown in Fig. 2d, the encoded sig-nal is subjected to a second multiply oper-ation, this time with a frequency of 22.5 kHz.The equations in Fig. 4 once more prove thatthe decoding operation works and yields thedesired output signal.

Finally, the decoded signal is takenthrough a low-pass filter to remove spuriouscomponents generated by, among others,the switching mixers.

Which frequencies?The frequency shift of 1.5 kHz used in thedecoder is achieved by first shifting the spec-trum to 24 kHz. Next, the spectrum is shifteddown again with the aid of a signal of

225 kHz, or the difference between 24 kHzand 1.5 kHz. This restores the spectrum tothe frequency range it has at the input of theencoder (which may be thousands of milesaway, or just round the corner). In otherwords, we have decoded the scrambled sig-naL Note that it is also possible to shift thespectrum 'direct' by 1.5 kHz, but not whenanalogue switches are used as mixers.

Even though a special kind of frequencysynthesis had to be implemented to ensurethe required stability of the frequency trans-form, the 'third method' is used in thedecoder for reasons already mentioned. Forexample, a stability of /5 Hz is equal to0.33% at 1,500 Hz. However, the 1,500 Hzsignal is not generated directly, but indi-rectly as the difference between 24 kHz and22.5 kHz. At 24 kHz, an error of Hz isequal to 0.021%, while at 22.5 kHz it equals0.022%. When these two switching frequen-des are generated independently, the accu-racy of each of them must be 0.01% to ensurethat a total error of±5 Hz is not exceeded. Itis readily seen that 0.01% is a pretty tough re-quirement as regards stability, particularlywhen the relevant oscillators are to be de-signed as free -running types to enable a freechoice of the shift frequency. However, it canbe shown that if these two frequencies havefixed phase relation, the common (central)oscillator from which they are derived isgood enough if it has a relative deviationequal to the maximum permissible error atabout 1,500 Hz, which is 0.33%. In the pres-ent decoder, this is realized with the aid of acentral oscillator in conjunction with twodigital dividers. Basically, the first divider isset for a divisor of 15, the second fora divisorof 16. The frequency of the central dock is

SPECTRUM SHIFT DECODER:

sin ((ost) sin (ok0 = 0.5 cos (o1,-, -(1)s) - 0.5 cos l(1),-; uls)cos (ost) cos (cwt) = 0.5 cos (co` - ms) -i- 0.5 cos (toc os)

cos (u, - (as)

cos (ok - ws) sin (0k2) = 0.5 sin (w. cocz - cos) + 0.5 sin (toe - ur +0k)substituting, and after the low-pass filter.

cos (coc - cos) sin (cik2) = 0.5 sin [ 2n4fs - 1500) I

(1k = 24 kHz (x 2n)ok2 = 22.5 kHz (x 21T)cos = encoded audiofs = shifted audio signal

910105 -14

Fig. 4. The 'Feedback killer' (Ref. 1) is a practical realization of a Hilbert transform.



U

Uf2

0

CI 192-I

C2

MI MNI- C1 d

Ua0

1029

18

EMU

R7 RBO EM

1009

la f

C7

10

Sat

C3 C6

fl

RIM

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C30

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03 -PO OS 2

IC6 05

as 107 28FIEF

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C31 12

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023983 Hz0232588 Hz

R11

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R12 RIIWE NM

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CS

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03628 Hz031626 Hz

R1S

[MIRI6

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IC3a

0356 Hz07331 Hz

819

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s3o11 Hz067857 Hz

P25

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09140 Hz015728 Hz R31

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IC3dIC3c1 12EMI

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01382 Hz02139 Hz

R33

1C4

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935

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C12 C13same A A

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I924

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IC4d

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C16

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P2 02

P3 034516

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U12

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IC5c

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1945 R47

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Ur2

90/320

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C36CU C34 C37

22,259

474

259110

Pos.* Encoder Decoder

02 tea 320p

C3 In MACl 4709 10

C7 297 IPA

CS fed 25% 2.7 23%

CIO 244 24% 649 2.51C11 175 23% 217 23%

CIS 3309 5409

C20 I. 1411

C21 3309 3524

0

IC2a

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U12

0

C27 C2I

10114 1014

0,

tt 002. 11 ttIC5

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0 0 0 0 0 0 016164646*


na Ica 5% 383 5.

87 138 SS 303 5%

R9 7039 SS

R12 2515 I% 2003 1%

8I3 mil 1% 3979 5%

R13 432 1% 1231 1%

R17 QM 511 39A 5%A20 11:00 IS 144 I%

821 102 IS 104 I%

R74 7115 IS WS 11


10 240 1%

1130 84C0 5% 1300 51

R33 1031 1% 4315 I%

RII 101 3% 2115 ISR43 1344 5% QS 5%

844 ISA 5% 14 514

R45 132 5% 154 ISR46 2711 5% 221 51832 271 3% 204 51

C29 3,3

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IC1. IC3, IC4 - 71.084

IC2. IC5 = 4053

IC8 - 4013

*we me A. ZS% pOrgre-,

910105 - 15

Fig. 5. Circuit diagram of the spectrum shift encoder decoder.


AUDIO AND HI-FI

Fig. 6a. Track layout (mirror image) of the single -sided printed circuit board designed for the spectrum shift encoder/decoder.

calculated as follows. Assuming a spectrumshift of 1,500 Hz, then:

f/15-f/16=1,50016fsc / 240 - 15 f,5, / 240 = 1,500

.-.1;./240= 1,500fosc = 360 kHz.

Since the 24 -kHz frequency must be avail-able as four signals with a phase shift of 90°,i.e., 0°, 90°,180° and 270°, the actual oscilla-tor frequency is four times higher than calcu-lated, i.e., 1.44 MHz, while the divisors of 15and 16 change to 60 and 64 respectively. Thisis illustrated in Fig. 3. The 24 -kHz signal is

Fig.

fed in quadrature to two identical mixers(4053). After adding the results of the multi-plication, the upper sideband is compen-sated. Via a 30 -kHz low-pass filter, the signalis fed to the third multiplier where it is trans-formed to the original frequency band.

The encoderSo far, we have not discussed the encoder,and this has a very good reason: it is virtuallyidentical to the decoder! In fact, the same cir-cuit is used, which allows the encoder andthe decoder to be built on identical printed -circuit boards. The only difference between

. Completed printed circuit board. A decoder is shown here.

the two units is the value of 29 passive com-ponents, of which a number determine thefrequency response of the phase filter builtaround two opamps, IC3 and IC4 (see the cir-cuit diagram in Fig. 5). These componentsdefine a pass -band of 1 kHz to 10 kHz in thedecoder, and a pass -band of 120 Hz to 7 kHzin the encoder. The accuracy of the phase fil-ter is determined by the accuracy of thepassive components used. The performanceof the filter depends on the number of sec-tions it consists of, and the ratio between thehighest and the lowest input frequency. Thisratio is about 10 in the decoder, which, basedon a phase filter with three sections, achievesan SIN ratio of about 60 dB. Also based onthree filter sections, but designed to handle amax/min frequency ratio of about 58(7,000 Hz/120 Hz), the encoder has a worseSIN performance of about 40 dB, which isnot bad given the simplicity of the design. Inany case, the results are perfectly acceptablefor speech.

The shift of +15 kHz in the encoder (in-stead of -1.5 kHz in the decoder) is achievedby suppressing the lower sideband insteadof the upper sideband (see also Fig. 2b).Since the spectrum in Fig. 2b then extendsfrom 24 kHz to about 31 kHz, the roll -off fre-quency of the 30 -kHz low-pass filter must beshifted up to about 36 kHz. In addition, theinput and output filters must be adapted tothe spectrum to be passed. All the changesnecessary to effect this, i.e., to turn thedecoder into an encoder (or vice versa), aresummarized in the components table youfind inset in Fig. 5.



Fig. 6b. Component mounting plan.

COMPONENTS LIST

Resistors: 1 330pF 2.5% polystyrene C30 ENCODER DECODER COMPONENT1 33k11 R1 1 180pF 2.5% polystyrene C31 VALUES5 22k0 R2:R3;1=15:R6;R58 1 68pF ceramic 0324 2k.02 R911:150:R53;R60 1 22&F 25V radial C33 Pos. Encoder Decoder1 2200 R10 2 47uF 25V radial C34,C35 C2 1nF2 820pF16 10K0 140 R11;R24:R14:R15; C3 1nF 680pF

R18;R19:R22:R23; Inductors: C6 470pF 1nFR27;R28:R31;R32: 2 7A1S assembly (Neosid): L1;L2 C7 2nF7 1nF8R35:R36:R37;R25 for winding details see text C9 6nF8 2.5% 2nF7 2.6%

3 1001(0 R38:R65:R66 2 47mH radial choke: L3;L4 010 24nF 2.5% 6nF8 2.5%2 4k.(270 R39:R40 Toko 181LY473 C11 1nF5 2.5% 2nF7 2.5%3 41(07 R41:R54:R61 C19 330pF 5609F4 2kS-27 R42:R47;R51;R57 Semiconductors: C20 1nF 1nF82 4700 R48:R49 1 1N4148 D1 C21 330pF 560pF2 271c0 R55:R56 3 TL084 IC1;103:1042 3k03 R62:1:163 2 4053 IC2;1C5 R4 10 ic0 5% 3k031 100 R64 1 HEF4060' 106 R7 10k0 5% 3 k0.3 5%1 471(0 R59 1 4516 IC7 R8 100k0 5% 33k0 5%1 lk.S1 1167 1 4013 IC8 R12 25k051% 2k11491%1 50k52 preset H P1 3 BC54713 Ti :T2:T3 R13 2k52371% 39005%

1 BC5578 T4 R16 431:02 1% 121(01 1%Capacitors: R17 6801(0 5°,:. 390k0 5%13 100nF C1:C4:C5;C15; ' do not substitute by CD4060, HCF4060 or R20 100k0 10k0 1%

C16;C17:C18; MC14060 R21 16k02 1% 10k0 1%C27:C28:C36-C39 R26 71(015 1% 5k049 1%

1 1nF5 2.5% polystyrene C8 Miscellaneous: R29 43k1221% 24k523141 2nF7 2.5% polystyrene C12

1 pr;nted-circuit board 910105 R30 680k0 5% 820k.0 5%1 6nF8 2.5% polystyrene C13 R33 100k0. 1% 47k1)5 1%1 1uF 25V radial C14

R34 6k08 5°/..; 21K05 1%1 1nF2 C22 R43 6k0.8 5% 5k06 5%1 3nF3 C23 R44 18k52 5% 15k0. 5%1 390pF ceramic C25 R45 18k0 15k05%1 1 nF5 C24 R46 27k0 5% 22k0 5%1 OnF2 C26 R52 27ki2 5% 22'612 5%1 4uF7 25V axial C29


p

III AUDIO AND HI-FI

Fig. 8. The rear panel of the Teko 222 enclosure has on it the frequency select switch (FCH L). the scrambling on off switch (S. 0 1). the input and output sockets. a DC adaptor socket,and a 7812 voltage regulator with decoupling capacitors.

The circuit in detailThe following description applies to the en-coder as well as to the decoder, which areelectrically identical circuits. Where necess-ary, a distinction will be made.

The audio signal applied to the input ofthe circuit is fed to a low-pass filter builtaround ICId and ICIc. The roll -off frequencyis set to 10 kHz in the decoder, and to6.6 kHz in the encoder. The previously men-tioned high-pass filter is formed by capaci-tors CI, C4 and Cs and resistors RI, R4 and R7.The roll -off frequency is set to 500 Hz in thedecoder, and to 150 Hz in the encoder. Apartfrom functioning in the active filter, ICIc am-plifies the input signal about 10 times.

The phase filter is formed by the circuitaround opamps IC3 and IC4. The indicatedcomponent values must be maintained to en-sure the correct operation of the filters. Thefrequency -determining capacitors are typesfrom the E12 series with a tolerance of 2.5%or better. Here, polystyrene types are used incombination with 1% resistors from the E96series. The theoretical roll -off frequencies ofevery filter section are indicated in the circuitdiagram. The equation

fc =1 / RC

applies for the roll -off frequency, and maybeused to match capacitors and resistors if youcan not secure the ones given in the circuitdiagram and the parts list (consult Ref. 2).The resistors must not be made smaller thanabout 2142, or larger than about 200 Ica Forinstance, if you can not get hold of a 24nF2.5% capacitor (Cut in the encoder), look atthe pole of the filter section, which is 356 Hz.Verify this using the above equation andgiven that R=116.2 k.C2 (R20 -1-R21). Next, sub-stitute Cto by, say, 27 nF 2.5%. The total re-sistance of R2o+R21 works out at 103.2 kQ.Since resistors are available in far morevalues than capacitors, this value is easier tocreate than 24 nF.

The multiplication with the 24 -kHz sig-nal is effected by electronic switches ICsband IC5c. Two 1% resistors, R39 and R40, addthe product signals. Transistor Tt functionsas an impedance transformer to ensure thatlow-pass filter L3 -L4 -C19 -C.20 -C21 is termi-nated correctly. Next, the filtered signal isfed to a second multiplier built around T2, T3and IC2b. This circuit supplies the decodedsignal in the decoder, and the encoded signalin the encoder. A low-pass filter fitted at theinput of opamp ICIa suppresses the un-wanted high -frequency products, and pro-vides the drive signal for the output buffer,ICH,. An electronic switch ahead of ICIballows you to disable the decoder/encoderby applying +12 V to the so/1 (scramblingon/off) terminaL

The central clock oscillator is contained inIC6, an HEF4060 CMOS oscillator/divider,which is used with an inductor, Lt or L2, in-stead of the rather more usual R -C combina-tion, to determine the oscillator frequency.Properly constructed, the L -C oscillatormeets the 0.33% accuracy requirement. Notethat capacitors C30 and C31 must be typeswith a low temperature co -efficient (NPO- orCOG -class devices; these are ceramic capaci-tors with a black band at the top side of thebody). Alternatively, fit 2.5% tolerance poly-styrene types, which are more readily avail-able. The presence of two oscillatorinductors enables you to switch between twodifferent frequency shifts. The selection is ef-fected via switch IC2c, which is controlled byapplying +12 V or 0 V to the FC H/L (fre-quency control high/low) input of theboard. This is also useful when a singlescrambled source makes use of two encod-ing frequencies.

The HEF4060 combines the function ofcentral oscillator and divide -by -64 prescalerto supply the 22.5 -kHz switching signal. Thedivide -by -60 function is realized by a 4516,ICS, and 4013, ICs. The former is wired todivide by 15, the latter to divide by 4. The Qand Q outputs of the bistables in the 4013

supply the control signals for multipliersICsb and IC5c.

Construction and adjustmentAs already mentioned, the printed -circuitboard (Fig. 6) can be used to build thedecoder as well as the encoder. The construc-tion itself is straightforward, and not ex-pected to cause problems. Note that there isone wire link on the board, and that a fairnumber of parts are fitted vertically. Makesure you select the right component valuesfrom the parts list when building up thedecoder and/or the encoder. The value cod-ing of the 1% E96 -series resistors may causeconfusion if you have never used these de-vices before - when in doubt, use your digi-tal multimeter to establish the value.

The inductors, Li and L2, consist of100 turns of 0.1 -mm dia. (5WG40) enamelledcopper wire on Type 7A1S formers from Ne-osid. After winding the inductor and sol-dering the wires to the appropriate pins(look on the component overlay!), fit the fer-rite cup, and secure it with a few drops ofwax or glue. This must be done to ensure thestability of the oscillator. Next, fit the screen-ing can and the core, check the continuity ofthe inductor at the base pins, and mount theassembly on the PCB. Solder all the pins, in-cluding those of the screening can, quickly toprevent overheating. The self-inductance ofLt and L2 is about 100 µH.

The central oscillator may not workproperly if you do not use an HEF4060. TheHEF4060 is a Philips Components product,and a LOCMOS version of the ubiquitousCMOS 4060. We tested a few 'ordinary'4060s of different makes in the circuit, andfound that none of these achieved the relia-bility and stability of the HEF4060.

Apply 12 V to the decoder, and an en-coded input signal. Connect an amplifier tothe output. Adjust PI for an input amplitudeof about 200 mVpp. Leave the tic H/L and snitpins unconnected, and adjust Lt until youmeasure 1.44 MHz at pin 9 of the HEF4060(IC6). Remember, the central dock frequencydetermines the shift, which you are free toset between 1 kHz and 2 kHz. A shift of 1,500Hz is obtained with a 1.44 MHz dock. Takethe FC H/L pin logic high (+12 V), and adjustL2 similarly. Listen to the output signal, andfine-tune the inductor(s) for best results.Your ears are the best test instruments forthis purpose.

The adjustment of the encoder is identical- the only difference in the set up is that theoutput is connected to a transmitter, a cas-sette recorder, etc., instead of to an amplifierfor direct sound reproduction.

References:1. "Feedback killer", Elektor Electronics Fe-bruary 1990.2. "Wideband phase -shift networks", in Elec-tronic Filter Design Handbook (1981), by Art-hur 13. Williams, McGraw-Hill Publishers.3. "HF transmitters", in Radio CommunicationHandbook, published by the Radio Society ofGreat Britain (RSGB).


INTERMEDIATE PROJECTA series of projects for the not -so -experienced constructor. Although each article

will describe in detail the operation, use, construction and, where relevant, theunderlying theory of the project, constructors will, none the less, require an

elementary knowledge of electronic engineering. Each project in the series will bebased on inexpensive and commonly available parts

BUIIL D 0 PICAby Michael Swartzendruber

TP

U,,LOW

3.5-4V d.o. Fil3H

* Radioshack Type 276.142

IS detector

key under test

IR emitterarray

channel undertest (cn)

all otherchancels ofi

910091 -11- 14

9 V d.c.

Fig. 7. Auxiliary circuit for testing the key assembly.

("..°-"i°normally closed Mop

normally cpen Icop

break In the alarm loop added c,nr.es;:zn

addition ofrelay 'branch' added connection

- N° relay coil driven- by alarm trigger

com

z1 70co I relay coil driven

r=1-71 by alarm Ulna

91C.91 -II - 15

9

Fig. 8. How to use relays as trigger switches.

PAS 2

Test procedures for thekey assembly1) Construct the circuit shown in Fig. 7.2) Enable one of the emitter circuits at a time

by leaving all DIP switches off, except thechannel currently on test: that channel'sswitch should be turned on.

3) Aim the emitter on test at the test circuit.4) Look at the voltage or logic level of test

point A.5) If the emitter passes test, disable this chan-

nel and proceed to the next channel untilall have been tested.

6) Trouble -shoot as necessary. If all emittersfail, double-check the battery connections,or check the device polarity for correctinstallation on to the board, or check thedevice with a junction tester.

Test procedures for thedetector/amplifier array1) After the ascPmbly is completed and passes

all visual inspections and voltage tests,install the two transistor arrays and con-ned the assembly to a 5-V d.c. supply.

2) The assembly is tested with the aid of thekey assembly. Enable all emitters on thekey by setting all DIP switches to 'on.

3) Locate the test node for the first amplifierchannel and place the probe of the volt-meter or logic probe at this node.

4) Aim the key at the completed detectorarray (these two assemblies should havebeen constructed so that the key arraymates with the detector array) and enablethe key by dosing the push-button switch.

5) The voltage or logic level at the node ofthe channel on test should swing to logiclevel 1 or about 4 V and should remain atthis level as long as the key is aimed atthe detector. When the key is removed,the logic or voltage level should return tozero.

6) Proceed to the next channel of the arrayand repeat the test. Trouble -shoot as nec-essary.

7) If all detectors fail the test, check the wiringof the supply, all polarized devices for

EI.EKTOR ELECTRONICS OCTOBER 1991

24 INTERMEDIATE PROJECT

Fig. 9. Mounting method of the optional bypass capacitors con- Fig. 10. View of the Opticalock's main logic board and the 'key-nected across each photodetector of the infra -red detector hole' connected by a ribbon cable.array should they be required to remove fluorescent interfer-ence.

Fig. 11. Two views of the Opticalock's 'key'. The one on the right shows a detail of the method used to mount the infra -red arrayat a right angle to the board. A 20 -pin, 0.1" -0.1" connector is soldered to a right-angle header assembly and the array is insertedinto the socket.

Fig. 12. General view of the key and keyhole. Fig. 13. General view of the entire Opticalock system.


111BUILD THE OPTICALOCK - PART 2

correct placement with particular atten-tion to the infra -red detector array.

Test points for the infra -red detector/am-plifier can be found at the following pointsof the circuit.

Channel IC Pin1 1 122 33 1 54 105 2 126 2 3

7 2 5

8 2 10

Interfacing the logic driverboardThe relay driver outputs of the logic drivercircuit can be used in a number of ways.Some simple examples are given to aid anycustom applications.

Example I. How to use the lock relay to drive asolenoid dead bolt.

A) Solenoid coil can be driven by 24 V d.c. at250 mA. Directly power a solenoid coil withPwrDrv-1 by connecting the solenoid coil leadsto 'load +' and 'load -' pads next to IC8,which becomes actuated by the lock drivercircuitry.

B) Load is high voltage or high current load.Use 'load +' and 'load -' pads adjacent toIC8 or IC7 to actuate a 24 V d.c. coil relay ortriac. Use the switch contacts or device for

load isolation and control.

&ample 2. How to use the relays as trigger switches.

To interface the alarm relay into an existingalarm loop, use the appropriate relay contactsand insert them into the loop-see Fig. 8.

For instance, to hook the Opticalock's alarmrelay into a parallel, normally closed loop, con-ned the the normally dosed and common con-tacts of the relay into the loop and simplyadd the relay contacts as another parallelbranch.

To interface the relay into a series (normallyopen in most cases), break the loop at someconvenient point and splice the relay intothe series by using the normally open andcommon contacts.

Example 3. How to use the unlock relay to pro-vide an alarm enableldisann switch.

To use this technique, a method must be de-veloped to hold a voltage level high or lowwithout having to excite the relay continu-ously. One very easy way of doing this is touse the normally open and common contactsof a relay to trigger a flip-flop (bistable)-see Fig. 9.

Programming and initialsystem testingSet the combination of the DIPswitch on thekey to match the combination of the DlPswitchon the main logic board. Align the key withkeyhole and press the push-button key. TheLED indicating that the lock is 'unlocked'should light.

relay

logicboard

Ccmdebounce

91C:91 - II - 1E

monostable

JK t. Sflip-flop(bistable)

Change the setting of oneof the DIPswitches and pressthe push-button on the keythree times, whereupon thealarm drive should becomeactuated. If you wish, youmay elect touse a logicprobeon the Q outputs of the JKbistables (flip-flops) whileperforming the three ille-gal -entry attempts to moni-tor the counting action ofthis chip.These tests should give apretty good idea as to howthe system works. SwitchS3 may be used to disablethe 'unlocked' output al-together to prevent the cir-cuit from providing an un-locksignal even when some-one has the IR key with thecorrect combination.

Finishing touchesandinstallation

Now the electronic part ofthe combination lock has

Fig. 14. How to use the unlock relay to provide an alarm been completed, we needenable/disarm switch. to look at some installation

ideas to make the project a useful additionto an existing security system or to help for-tify some accessible doorway by providing apick -proof lock system.

To augment a security system, the contactsof the alarm relay can be interfaced easily withan existing alarm loop, or its contacts can beused as a switch for an alarm horn circuit.

The lock relay may be used to drive asolenoid -driven lock system such as an elec-tric car door lock or it may simply be connectedto any solenoid (modified for use as a dead -bolt) with a spring return latch assembly.With these approaches, the lock will alwaysbe in a locked mode unless the relay is actu-ated. The lock relay may also be tied to anexisting alarm as the arm/disarm switchusing the contacts of the relay to toggle abistable (flop a flip-flop).

A word of caution: you may void anywarranty of a professionally installed alarmsystemby attempting to connect the opticalockto it. Consult your professional installer if youare not sure or have any questions regardingconnection of the device to this type of alarm.

Experienced computer enthusiasts maywant to consider the followingcomputer/op-ficalock interface possibilities. The lock driveroutput and the alarm driver outputs may beused to drive a bistable and this flip-flop canbe monitored by a monitoring control system.In addition, the 8 -position DIP switch on themain logic board can be driven by a paralleloutput port on a control system.

As for security, the detector array canbe located at any remote location in rela-tion to the rest of the system. If the dis-tance between the detector and logic boardexceeds more than a couple of feet (abouta metre), additional pull-up and bypass cir-cuitry may be required at each end of theconnecting cable.

The detector should be well -armouredagainst vandals and attempted intrusions.However, its destruction will not give ac-cess to any unauthorized persons; bear inmind, though, that access may also be de-nied then to authorized persons. So youmay consider a rather beefy metal plateassembly to guard it.

The lock disable switch should be locatedinside at a convenient position.

The alarm reset switch should be situ-ated in a semi -secure area that is easy toget to should it have to be used by thosewho should know its whereabouts.

Should you opt to use a solenoid -drivendead -bolt, or indeed any electric lock, thatmay be located at a remote position also.

The system can easily be made to workin many different situations. You may wishto experiment with it on some inner -struc-ture door, such as a closet, or your labora-tory door, to find out the finer points of apossible application.

The opticalock makes an innovative useof opto-electronics, and with the numerousbenefits of this type of lock system, it wouldnot be surprising to see more designs likethis very soon. Who knows, perhaps youhave just built the key of the future.


26

50 MHz 8 -BIT DAC

The digital -to -analogue converter described here is fast, based ondiscrete components, and comparable - in terms of performance -

to pretty expensive integrated circuits.

by R. Shankar

3V75

20

110V

0_Z

0_11.

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03374/574

CPI 04

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R15

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C31 R33

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111111

R30

E12:1R32

R29

15

12

15

04

R28

13

R27

16

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826

12R25

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1 I70

1V25 C)

C2 =11P

10/1

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015

/117 86

014 1313 012 DII 010

X XR5 R4 A R3 A

MB EEO EB7171 I= EMIR14 R13

Dt...D16 = ND4981.76

R12 R117' R/0C41

At.

Ti...T9 = BC560C

09

R2 111

3705

R9

Eli13

R24

T10

BC548

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R23

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01111%

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son

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5

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BC548BC560C

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101 0

LM336 910076.11

Fig. 1. This fast digital -to -analogue converter is based on discrete components rather than the latest in IC technology.

converter can directly drive a 7542load at a full scale voltage of 0510 V, and

a settling time of less than 10 ns to 1 LSB. Thetemperature coefficient of the output voltageis 0.005%/K, which is significantly betterthan that of many integrated DACs.

The circuit is based on the conventionalR -2R ladder network, with current switchingaccomplished by Schottky diodes and ahigh-speed Advanced Schottky (ALS) TTLlatch. Resistors RI to R14 form the ladder net-work, the output impedance of which is75 Q. This enables the DAC to drive a loadvia a 75-0 coax cable.

Transistors Ti to Is form the currentsources that supply 6.67 mA each to the lad-der network via D9 to D16. When any outputof ICI goes low, the resulting current isdiverted to ICI via Di to Ds. This arrange-ment requires the outputs of ICI to go nega-tive with respect to ground. Therefore, ICI isoperated with a dual supply of -1.25 V and

+3.75 V. To make the inputs TTL compatible,a level translator like the one shown in Fig. 2may be used at each input. The input loading('fan -in) of the translator is equal to that of3 standard TTL inputs. The circuitry aroundT9, Tio, IC2 and IC3 is necessary to obtain alow temperature coefficient.

A suggested power supply circuit isshown in Fig. 3. The current consumption ofthe DAC is of the order of 170 mA.

Construction hintsAlthough no PCB design is available for thisproject, the construction is fairly simple evenon a prototyping board. Di to rti 6 can be anySchottky diode that meets the following spe-cifications: ID 10 mA; I/BR V; V60.5 Vat 10 mA.

The connections of the ladder resistors,the Schottky diodes, ICI and C2 should bekept as short as possible. A large ground

Fig. 2. TTL-compatible input level transla-tor.


50 MHz 8 -BIT DAC

Fig. 3. Regulated power supply for the DAC.

Popular modem IC set upgraded toinclude FAX

Fax transmit capability can be addedeasily and at nominal expense to stand-alone modems or for such applications asPCs, laptops and portable computers,using Silicon Systems' newSSI 73D2417 modem IC set.

The new chip set adds fax capabilityas an upgrade to its popularSSI 73D2407 2,400 baud modem chipset which offers a turn -key 'AT' com-mand interpreter with MNP classes 2-5for error control and data compression.

The SSI 73D2417 combines V.27terfax transmit capability at 4,800/2,400baudrates with 2,400 baud multimode(V.22bis/V.22/V.21, Bell 212A/103)data communication and `AT' commandcontroller.

N EW PRODUCTS

The SSI 73D2417 is said to providean easy way to enhance conventional2,400 baud modem designs by addingMNP5 and basic fax features without sig-nificantly increasing cost. As an exten-sion of Silicon Systems' proven73D2404 modem IC set, it gives users areliable basis on which to develop PC faxdesigns with a minimum of design effort.

The SSI 73D2417 modem IC set iscomprised of four ICs. The 73M214 is ananalogue processor that performs the fil-tering, timing adjustment, level detec-tion and modulation functions. The73D215 is the receiver digital signal pro-cessor. The 73D218 is a command pro-cessor that provides supervisory controland command interpretation. TheSSI 73D225 is a ROM that provides stor-age for integral control software. Thefour devices are available in both DIPand surface -mount packages.Silicon Systems 14351 Myford Road Tustin CA 92680. Telephone (714)731-7110 ext. 3575.Silicon Systems International Wood-peckers The Common West Chilt-ington Pulborough RH2O 2PL.Telephone: 07983 2331. Fax: 079832117.

plane should be provided, and the earthedends of Rs-R14,Ill and C2, should meet at onepoint.

Without component matching, the li-nearity is of the order of 2 LSB. However, bycarefully matching the resistors with the aidof a digital multimeter, a linearity of betterthan 0.5 LSB can be achieved (tip: use theclosely matched resistors towards the MSB).Similarly, the transistors can be matched bymeasuring the resistance of the base -emitterjunction, and noting the values. This meas-urement should be carried out with the col-lector and the base connected via the testprobe. Additionally, Ti to T9 may bemounted on a common heat -sink for closethermal tracking.

TN RO analysis and dish aimingsoftware

Greg Grissoms' TVRO System Ana-lysis and Antenna Aiming Computer Pro-grams for Satellite Technicians is asoftware package aimed at satellite TVenthusiasts, professionals as well asTVRO dealers. It can also be a usefuladjunct in demonstrating how changingparameters such as dish size or LNBnoise figure affect picture quality.

The upgraded version of the program,version 1.1, includes a system configurescreen that allows a number of factorsthat refine the program operation to beset. These include the cutoff angle abovethe horizon below which satellites cannot be `seen', noise weighted factor, pre -emphasis factor, peak -to -peak conver-sion factor and video noise bandwidthfactor. The last four factors differ be-tween NTSC, PAL and SECAM broad-casts. The new version of the programoffers a user -edited database of satellitepositions.

The users manual and software, oneither a 5.25 or 3.5 -inch diskette is avail-able at £35.00 plus £2.00 P&P fromBaylin Publications 24 River Gar-dens Purley Reading RG8 8BX.Telephone (0734) 414 468.Baylin Publications 1905 Mariposa Boulder CO 80302. Telephone: (303)939-8720. Fax: (303)449-4551.


28

COWSTATIONPART 2: ELECiitoiliC

The electronic hygrometer,or relative humidity (RH)sensor, discussed here isremarkably accurate, andfunctions automatically aspart of the PC -controlledweather station we set out todescribe earlier this year.

by J. Ruffell

AMONG the basic measurements carriedout daily by amateur meteorologists are

those of air temperature, air pressure andrelative humidity. In the first instalment ofthis series, Ref. 1, we described an electronicindoor/outdoor thermometer that works inconjunction with an advanced I/O card forPCs and compatibles (Ref. 2), and dedicatedcontrol software. This month we propose toextend the function of this PC -based systemwith a relative humidity (RH) sensor. Sen-sors to capture other meteorological datasuch as air pressure, wind speed and winddirection, will be described in future instal-ments.

The present circuit measures the relativehumidity of ambient air with the aid of aspecial sensor manufactured by PhilipsComponents. The sensor is basically a non -polarized variable capacitor whose capacit-ance is a function of the ambient airhumidity. Unfortunately, the capacitancedoes not change linearly with humidity, butthat can be corrected fairly easily with theaid of the computer, which is at the heart ofour weather station. The computer alsogreatly facilitates the calibration of the sen-sor.

Relative humidityRelative humidity, RH (or U in physics), isthe ratio of the pressure of ambient air, e, tothat of saturated air over a plane liquid watersurface at the same temperature, e', and isnormally expressed as a percentage, that is,RH = 100e/e'. The relative humidity dependsstrongly on air temperature. A relative hu-midity of 0% indicates a total absence ofwater particles in the air. In the evening andat night, RH values can rise to 80-90%, whilein dense fog the RH is virtually 100%. Typi-cal daytime RH values are between 60% and70% in sunny weather.

Professional meteorologists measure RH

PUTM-CONTROLLF0 WFATHER

HYGROMETER

\ACwith a wet and dry bulb hygrometer. Thisconsists of two mercury -in -glass ther-mometers mounted side by side. The drybulb thermometer shows the air temperaturein the usual way, while the other has its bulbkept moist continually (by means of a smallreservoir mounted underneath it) and ex-posed to an adequate draught. The rate atwhich the air takes moisture from the reser-voir depends on how far it is being saturatedwith water vapour. If the air is already satu-rated, it takes up no more and the two ther-mometers read the same (that is, RH=100%).The lower the humidity of the ambient air,the more it tends to take up moisture, and thelower the reading of the wet bulb ther-mometer compared with that of the dry one.The difference in temperature gives a highlyreliable measure of relative humidity, whichis read from tables.

Another, less accurate, method tomeasure the RH makes us of a hair. Thisbased on the fact that the length of a strainedhair depends on the humidity of the ambientair. This inexpensive way of measuring theRH is applied mainly in simple hygrometersof the type hung up in the living room.

RH measurement: theelectronic wayUnlike most electronic hygrometers, whichare based on a hygristor (a componentwhose electrical resistance varies with hu-midity), the une described here is based on aPhilips Type H1 sensor whose capacitancechanges with humidity. The capacitance ofthis device varies between about 110 pF and145 pF for RH values of between 10% and90%. The RH-dependent capacitance is usedto vary the frequency of an oscillator basedon the well-known 555 timer IC - see thecircuit diagram in Fig. 1. The RH sensor andthe external components around the 555 givethe oscillator a nominal frequency rangefrom 43.6 kHz to 33.1 kHz. Since the capacit-ance of the sensor has a maximum tolerance

MAIN SPECIFICATIONS

Electronic hygrometerRange: Cr/. to 95%Channels: 2 (indoor and out -

Recording:

Measurement:Measured values:

Indication:

Software:

door)continuous (interval= 10 minutes)once every minutecurrent, day max-imum, day minimumpercentage RH;low, normal, highmemory -residentdata logger, cali-

bration utilities andfull -colour graphdisplay programs

System requirementsIBM PC XT, AT. 386, or compatible640 KByte RAMEGA or VGA display adapterDOS 3.3C or later

of 15%, the lower frequency may drop to28 kHz, while the upper frequency may riseto about 52 kHz. Capacitance -to -frequency(C -to -F) conversion is applied here to allowthe measured RH value to be carried over aconsiderable distance via a cable.

The software that runs on the PC takescare of the linearization of the sensor charac-teristic. The method developed for this pur-pose is remarkable because it reduces thenumber of adjustments in the hygrometer toone! By making use of the 6 -digit frequencymeter function provided by the measure-ment card for PCs, the electronic hygrometerachieves a resolution of 1%.

Two H1 sensors and associated interfacesare used since it is usually required tomeasure the indoor and outdoor RH valuessimultaneously. The two frequencies pro-duced by the sensor interfaces are applied tothe F6 (RH indoor) and F7 (RH outdoor) in-puts of the PC measurement card. These twoconnections are located on connector K6.


LillELECTRONIC HYGROMETER

Fig. 1. Circuit diagram of the capacitance -to -frequency converter.

The two sensors may be connected to thecomputer via ordinary wire, e.g., a 4 -wiretelephone cable. Cable lengths up to 15 m(50 ft) should not present problems. Each in-terface requires a supply voltage of 5 V,which is conveniently taken from the stabi-lized +5-V supply rail brought out on con-nector K6 of the PC measurement card.Decoupling capacitors 0 and C2, and reser-voir capacitor C3 compensate the impedanceof the (long) supply wires, which wouldotherwise degrade the stability of the C -to -Fconverter. With Cr, C2 and C3 fitted, themeasured value, at RH=50%, varies no morethan 0.004% per mV of variation in the sup-

_ ply voltage.In some cases, it is not desirable that the

sensor interfaces are powered by the PC. Thealternative is to build a small, regulated 5-Vsupply based on, for example, an 7805, andfit this in the sensor enclosure. The stabilityof the C -to -F converter is best when the volt-age regulator is located as dose as possible tothe sensor interface board. The enclosureused to house the sensor should offer plentyof space for the regulator and the usual de -coupling capacitors at its input and output.

ConstructionFigure 2 shows the printed circuit board onwhich the sensor is mounted. The construc-tion of the interface is downright simple, andmerits no further discussion. The PCB fitsexactly in a Type E406 enclosure from Bopla.Other enclosures may be used, but may re-quire some drilling and filing to fit the PCB.

Drill two 1.5 -mm holes in the cover of theendosure to allow the connecting terminalsof the H1 sensor to pass. Extend the H1 ter-minals with 5 -cm long wires before securingthe device to the cover with a few drops oftwo -component glue. The two wires are sol-dered to the PCB holes marked Rhcl. Drill ahole in the side panel of the case to allow thecable to the PC to pass. This hole should bejust large enough for the cable to pass. Fit astrain relief on the cable at the inside of theenclosure.

The interface must be protected againstmoisture. This is best achieved by applying alittle silicone compound on the edges of theenclosure before this is dosed. Also use thecompound to seal the holes for the sensorwires and the cable to the PC. In both cases,apply the compound at the inside of the en-closure. Before the enclosure is dosed, theholes in the bottom half must be sealed withcompound. Like the thermometer module,the sensor is temporarily connected to the PCvia a small piece of stripboard.

Software in actionThe floppy disk supplied for this project,ESS 1561, contains the software required forthe adjustment of the sensor, the measure-ment routines, and the automatic loggingroutines. In addition, it contains the first up-date (version 1.1) of the control software forthe thermometer module. The new softwareis marked by an extended version of thememory -resident data logger. It integratestemperature and RH measurements, andprovides links for the wind speed and winddirection modules sensors to be discussed infuture instalments of this article.

The name of the background programhas been changed from TLOGGER intoXLOGGER. The update takes into accountthat XLOGGER is the only program that ad-dresses the PC measurement card - i.e.,XLOGGER now performs measurementscontinuously, and adds measured values tothe log file every 10 minutes. The programsfor the graphics display of temperature andrelative humidity, TEMP and HUM respec-tively, translate the data contained in the logfile into graphs on the PC screen, and at thesame time open a RAM data path to XLOG-GER. This path allows them to receive thecurrent measurement results, i.e., maximum,minimum and current temperature, andmaximum, minimum and current relativehumidity. This data transfer avoids complexsynchroniiation protocols between XLOG-GER and other programs, and prevents thehardware on the PC measurement cardbeing addressed simultaneously by two syn-chronous routines, which would cause thePC to crash.

The communication between XLOGGERand the calibration utilities, HADJUST andTADJUST, is similar to that described above.XLOGGER performs all its functions in thebackground, i.e., as a memory -residentutility. This allows you to use the PC as be-fore for word processing or drawing. Therelative humidity is measured once everyminute, and the temperature once every15 minutes. XLOGGER can be actuated bytyping XLOGGER /I from the commandlevel. The program takes about 50 KBytes ofthe base memory in the PC.

.LOG and .CFG filesUsers of TLOGGER should note that log filesproduced by this program (Tyymmdd.LOG)are not compatible with the new log filesproduced by XLOGGER (Xyymmdd.LOG).The old files may still be viewed with version1.0 of TEMP, however, provided XLOGGERis removed from memory. This is achievedby typing XLOGGER /U.

The new Xyymmdd.LOG files contain

COMPONENTS LIST

Resistors:2 1 00;(0 1% R1 R3

Capacitors:100nF C1

1 22nF ceramic1 10uF 16V C3

Semiconductors:1 TLC555 IC1

Miscellaneous:1 H1 humidity sensor Rhc1

{Philips Components 2322 691 90001)1 ABS enclosure, size approx.

65x50.'30 mm: e.g., Bopla EG4061 Printed circuit board 900124-21 Control software on disk ESS 1561

Fig. 2. Single -sided printed -circuit board for the relative humidity sensor.


30 GENERAL INTEREST

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900124 - 4 - 13

Fig. 4. Screen layout of HADJUST (left) and HUM (right).

voltages, frequencies and binary codes only.The calibration utilities create transfer files,HTRANS.CFG and TTRANS.CFG, thatallow these quantities to be converted intorelative humidity and temperature, respec-tively. TTRANS.CFG contains two linearfunctions, while HTRANS.CFG is used tostore the arithmetically determined frequen-cies that correspond to a relative humidity of0%. The status windows of TADJUST andHADJUST show the content of these transferfiles. It should be noted that the transfer filesmust be saved before the sensors are recali-brated, which may be necessary after sometime. Without the saved transfer files, it isimpossible to convert the existing log files

into the correct values.The RAM channel conveys frequencies,

voltages and codes only. The calibrationpoints are recorded in the TCALI andHCALI configuration files.

The main program, XLOGGER.EXE, usesa configuration file, XLOGGER.CFG, fromwhich it reads a number of initial par-ameters. XLOGGER.CFG is a text file whichmay be opened and modified with anyASCII -compatible word processor. The firsttext line defines the path and the directorywhere the log files are stored. The path is alsoused by TEMP and HUM, and may have tobe modified depending on your require-ments. The next four lines in XLOG-

XLOGGER - background to a complex and powerfulprogram

AiD conversionXLOGGER controls the A -D converter on board the PC measurement card in an interestingway. Each measurement result is based on an average of 500 measurements. The start -of -

conversion command for the ADC is generated by the system clock of the PC. Unfortunately,the USER-S1C-TIMER-TICK interrupt can not be used without modifications because it wouldresult in excessive wait times for 500 or so conversions. This interrupt is executed at a rateof only 18.2 Hz, which is so slow that it would cause the total conversion time to rise to27.5 seconds. XLOGGER reduces the conversion time to 5 seconds by reprogramming theprescaler in the system clock. When XLOGGER is installed, the standard interrupt BIOSroutine $08, which controls, among others, the real-time clock. and is started at the end ofthe SIC handler, is replaced by a new routine that takes care of the data acquisitionoperations. This interrupt -$08 routine calls the standard BIOS routine (and with it interrupt -

$1C) every 55 ms on average, so that the real-time clock continues to function normally.

Pop-up handler and frequency meterXLOGGER uses interrupt-S1C to pulse the PBO line when an error condition occurs. Interrupt -

$1C also starts a so-called pop-up handler every 15 seconds. The pop-up routine reads theresults of the interrupt -controlled frequency measurement, starts the next measurement witha new channel number, and adds the captured values to the log file every ten minutes. Themain advantage of the pop-up handler is that DOS operations such as disk I/O may continueto run asynchronously. By virtue of the pop-up handler, the fact that DOS is not 're-entrant'is no stumbling block any more in the development of interrupt routines and memory residentprograms. When the pop-up handler is run, the PBO line is briefly actuated.

Install UninstallWhen XLOGGER is started. the presence of a valid parameter, the il (install) or rU (uninstall)switch, is checked. The program terminates with an error message when the parameter ismissing or incorrect. When the .1 option is used. XLOGGER is installed only when it is notalready present in the memory. The uninstall option, U. works only if no other memory -resi-dent programs were loaded after XLOGGER. Note that DOS PRINT is a memory residentprogram!

GER.CFG are the labels, or names, assignedto the sensors. These labels are used by TAD -JUST, HADJUST, TEMP and HUM, and maybe changed if necessary.

One line of text has been added toADCF.CFG - this will be reverted to below.

InterruptsXLOGGER can not be used without thehardware on the PC measurement card.Since the hygrometer makes use of the fre-quency meter on the measurement card, oneof the jumpers JP2-JP7 must be fitted to en-able the interrupts to be handled correctly.First, determine which interrupt line is stillfree in your PC (in most cases, this is IRQ2).Fit the respective jumper on the X -row. Next,check that jumper JP8 is fitted in position E.The interrupt line used by XLOGGER isfound in the last line of the configuration fileADCF.CFG, and must be the same as the lineselected by the jumper. Make sure that this isthe case by opening ADCFG.CFG with yourword processor, or typing TYPE ADCF.CFG,and looking at the interrupt number. Ifnecessary, change the number to match thehardware interrupt selection.

Software: features andapplicationsHADJUSTThis is the calibration program for the hy-grometer function. The layout of the screen,and the use of pull -down menus are similarto those of TADJUST, which may already befamiliar from the earlier article on the ther-mometer module. Adjusting the RH sensoris much less complicated than the tempera-ture sensor because it involves only one set-ting for each sensor. This is achieved byvirtue of purposely written calibration rou-tines contained in the HADJUST program.The principle of linearization is explainedseparately elsewhere in this article. Beforethe calibration is started, the two sensors areplaced in an environment of which therelative humidity is known. It is best to use a


ELECTRONIC HYGROMETER 131

calibrated hygrometer for this purpose, andyou may need to go to your local weathercentre to ask for assistance.

Install XLOGGGER, and run the HAD -JUST utility. The menu appears, and twowindows, one containing the calibrationstatus, the other the currently measuredvalues. When the program is run for the firsttime, the calibration window shows a de-fault status, which is simply a rough esti-mate of the expected calibration data. Theother window displays the frequencies pro-duced by the two sensors. From the installa-tion of XLOGGER, it takes about one minutebefore the two frequencies are measured forthe first time. Until then, the frequency vari-ables remain at nought. The CurrentlyMeasured Values (CMV) window alsoshows the relative humidity values derivedfrom the frequencies and the calibrationstatus.

When the first frequency measurement iscomplete, select the 'inside' option from themenu, after which the 'EDIT H' window isopened. Press the return key, and enter theactual humidity (the value read from the ref-

erence hygrometer) at the cursor location.You will notice that the calibration statusand the calculated relative humidity areautomatically updated. The other sensor(option 'outside') is calibrated likewise.

Leave the program via the 'quit' option,and subsequently select 'update' to save thetransfer and calibration files. Do not select'abandon' at this stage, as this will terminatethe program without storing the previouslyobtained calibration data.

HUMHUM (for humidity) handles the graphicspresentation of currently measured as wellas previously saved values. The screenlayout of HUM is basically the same as thatof TEMP. HUM displays two analogue hy-grometers with an RH percentage scale inthe left-hand bottom corner of the display. Itshould be noted that RH measurements arecarried out only when XLOGGER has beeninstalled. The current RH value is alsoshown numerically on the screen, togetherwith the minimum and maximum valuesmeasured during the last 24 hours. The anal-

ogue RH meters have scale areas markedlow' (for RH values smaller than 45%), 'nor-mal' (for RH values between 45% and 65%),and 'high' (for RH values greater than 65%).

The control menu appears in the top left-hand corner of the screen, while the auto -ranging 24 -hour graphs take up most of theright-hand side of the screen. Below thegraphs you find the name of the log file andthe minimum and maximum values in this.

Finally, function keys F3 and F4 allowyou to request the RH value logged at everyfull hour. The other function keys serve to se-lect certain program options: a graphics ornon -graphics graticule; inside RH or outsideRH graph or both; producing hardcopy(Epson FX-85 mode only); loading anotherlog file; and leaving the program.

References:1. "Computer -controlled weather station -Part 1: indoor/outdoor thermometer", Elek-tor Electronics March 1991.2. "Multifunction measurement card forPCs", Elektor Electronics January and Fe-bruary 1991.

immmommillo- HUMIDITY SENSOR CALIBRATION - WITH A SINGLE ADJUSTMENT 411Ilimmomm

The Type H1 humidity sensor from PhilipsComponents is basically a non -polarized ca-pacitor whose capacitance depends on thehumidity of the air around it. The dielectricconsists of a non-conductive foil with a thingold layer at both sides. The two gold layersform the electrodes in this sensor. The dielec-tric constant. E, of the toil is a function of theatmospheric humidity. The absolute capacit-ance of the capacitor is calculated fromEq. (1). In the equation. r is the relative hu-midity, A the surface of the electrodes. and dthe thickness of the foil. Unfortunately, thecharacteristic of the sensor is non-linear (seethe curves below). This means that convertinga measured capacitance (or a frequencywhich is inversely related to the capacitance)into a relative humidity value is not so simple.A further complicating factor is that the realcharacteristics of the two sensors you havelie somewhere between the Cmaz andcurves. This spread is caused by productiontolerances on the thickness of the dielectric.The problem with the spread can be solved bybasing all calculations on the capacitance fac-tor. calculated from Eq. (2). instead of onthe absolute capacitance. Substituting Eq. (1)

Table 1. RH - K correlation

r K(%l

10 2.135

20 4.265

30 6.463

40 8.865

50 11.433

60 14.130

70 16.918

80 19.767

90 23.238

in Eq. (2) proves that IC(, is independent ofthe thickness of the dielectric. Eq. (3) showsthat the dielectric constant, e(r. is the singlefactor that determines the actual capacitance.This means that 1.(,) is independent of thespread on the sensors, so that it can be usedas a repeatable starting point. Table 1 showsa few discrete values of Kir:. Remember.is expressed as a percentage.In effect, the oscillator frequency is measuredinstead of the sensor capacitance. The fre-quency is an inverse function of the capacit-ance. In Eq. (4), parameters a and b areoscillator -dependent constants which haveno further significance here. CombiningEq. (1) with Eq. (4) yields Eq. (5). Finally.substitution of Eq. (5) in Eq. (3) yields theoutput frequency formula, Eq. (7).

CalculationsAssuming that fo is known. Eq. (6) enablesus to calculate the capacitance factor on thebasis of a measured frequency fir. Therelative humidity (RH) associated with thevalue of t(,) may then be deduced. by inter-polation. from Table 1.The frequency at 0% RH, is also deter-

mined arithmetically. This is done by perfor-ming a frequency measurement at one,known, RH value. The capacitance factor fol-lows from Table 1 via inverse interpolation.Next, the measured frequency and the resultof the interpolation are entered into Eq. t7).The result is a frequency, fo:. expressed inHertz.

E(r) x AC(r) -

Cir) - (i)Ko)

C / 100

[F] Eq. (1).

Il Eq. (2).

Substituting Eq. (1) in Eq. (2) yields:

Ku-) - Eir) E(u) X 100E(r)

r7ci Eq. (3).

The oscillator frequency is a function of thehumidity as expressed by

fir) = b x1I-Izt Eq. (4).


a x dE(r)=Ax6xf(r) [IrndEq.(5).


KW= ( 1-irl)

X100- f

[%1 Eq. (6).

After calibration. Eq. (6) can re rewritten as

-K(r)/ 100 [Hz] Eq. (7).


mrAsuk LT QJIn - 8

Vi{easurements in digit.

WOSUREMENTS in digital circuits falltwo categories: (1) voltage, current

and resistance; and (2) data. The first cate-gory presents no problems other than thosealready discussed in previous instalments ofthis article. The measurement of data andassociated circuits will be described in thispart in the series.

Measurementtechniques

for digital circuits


or investigating theesign of the circuit


or investigating thedata flow

- oscilloscope. Ingle analyser

900113-V111-

Fig.1. Measurements in digital circuits fallinto two categories.

Techniques for examiningcircuit designTypical of digital circuit designs is the useof densely populated printed -circuit boards,whose tracks, owing to space limitation, areextremely thin. This makes faultfinding par-ticularly difficult and it is, therefore, advis-able to carry out a thorough inspection ofthe board before any work on it is begun.For this, an ohmmeter, multimeter, oscillo-scope with component tester or a simplecontinuity tester is perfectly adequate.

Once the board has been populated, amultimeter is useful only for checking thevarious supply voltages. A component testeris also very useful: this quickly shows whethera component is all right ordefect. Voltage lev-els and edges of digital signals can be checkedwith an oscilloscope.

Data pulses can cause problems with thetriggering of an oscilloscope, however. Someoscilloscopes are, therefore, provided withspecial facilities, such as pre -trigger or hold -off. The pre -trigger function allows it to ex-amine the trigger pulse just prior to its firstmesial (see p. 41). In other words, the signalbehaviour before triggering is displayed.The trigger pulse is then not shown at theleft-hand of the screen, but in the centre, or

by F.P. Zantis

even at the right-hand.The hold -off function allows the contin-

uous lengthening of the delay between twotime -base sweep voltages as shown in Fig. 2.Pulses or other spurious signals occurring dur-ing the hold -off time cannot influence the trig-gering, which is particularly useful in the caseof aperiodic pulse bursts of constant ampli-tude. The oscilloscope can be triggered ondifferent points of the waveform.

A digital storage oscilloscope is particu-larly useful for measuring an aperiodic sig-nal. Note, however, that in moderately pricedmodels short spurious pulses or glitches maynot all be identified The scanning rate andstorage capacity limit the use of these oscil-loscopes.

Examining the data flowAs is well-known, logic signals in digital

engineering can only be 1 (high) or 0 (low).Errors caused by insufficient pulse voltagelevels or resulting from too long a first tran-sition duration of the pulse can be tracedwith standard test methods.

A typical problem encountered in mea-surements in digital circuits is that the ex -

s

amination of only one signal usually doesnot yield enough information about the func-tioning of the circuit. Generally, two or, incomplex circuits, even more signals mustbe examined simultaneously.

For analysis of two signals, a dual -beamoscilloscope is required (not a dual -tracetype, because that cannot capture two fast tran-sient events). Examination of the data busof an 8 -bit microprocessor system requires an8 -channel oscilloscope. Even that is not suf-ficient if a control signal such as 'write' or`read' must be triggered.

Some, rather expensive, oscilloscopespermit the simultaneous display of four sig-nals-see Fig. 3. A dual -beam oscilloscopemay be converted with a multi -channel adap-tor to make relatively simple, qualitativetest on a number of channels possible. Themajor disadvantage of such adaptors is theirlow chopping rate, which allows examinationof relatively slow signals only.

In contrast to oscilloscopes, which mea-sure voltage as a function of time, logic anal-ysers are designed primarily for examiningthe data flow in complex digital circuits andmicroprocessor systems.

A logic analyser is suitable for testing hard -

Signal

voltage at thehorizontal (a) plates

Signal

voltage at thehorizontal (a) plates

altered hold -off time-.

Fig. 2. Illustrating the hold -off function. At the top. this facility is not operating. Theoscilloscope is then triggered when the electron beam is in its output position and aleading edge appears in the signal. The waveform in bold print is then shown on thescreen: the picture runs across the screen. Analysis is not really possible. In thelower half, the hold -off time is altered in a manner that ensures that triggering alwaysoccurs at identical leading edges, so that the picture stands still. It is then possibleto identify any glitches or other short, spurious pulses clearly.


MEASUREMENT TECHNIQUES - PART 8 ID

A

aRrEw6.- 7

Fig. 3. Display of four logic signals on a 4 -channel oscilloscope.

I

logic analyser

prole probe

logic generator

clockdriver

0

4.3

0N

driver

I

I

I

toao.cr

cr

unit t..7.= :es;(e.g., fr.;:,77',3ry)

900113,1111- 13

Fig. 4. Typical application of a combination of a logic analyser and a logic generator.

desk data In printer mode options100 ilHZ

18 X CH: 1 -1 1' J1-11-1-1-n-iFir H: 7I zo tin ,-F-

1 X CH: 3-1 1 9Hz '--1-.n.n.n.n.r

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0 4 0 cannel 1-16 0 1 01 read << i >>1 < >

ware as well as software; it measures data,that is, a pulse is considered as a bit with avalue of 1 or 0, and not as a rectangularwaveform with a certain amplitude. At thetime of writing, logic analysers with up to72 input channels are commercially available.

A combination of a logic analyser and alogic generator, which provides sample bitsthat can be fed to the circuit under test, isespecially useful. It allows the analysis of cir-cuit sections even when these are driven byexternal sources.

A typical application of such a combina-tion is shown in Fig. 4, in which the unit ontest is controlled via a data bus and an ad-dress bus. The logic generator provides theaddresses, data input and clock, while theanalyser records the test data.

The most important part of a logic analy-ser is the control computer, whose softwareserves to operate the analyser and to anal-yse and display the test results. In that pro-cess, it is necessary for the relevant infor-mation to be extracted from a vast mass ofdata. In the simplest case, it is only neces-sary for a given sample bit to be triggered.More complex systems permit triggeringeven when, for example, the operating con-ditions are normally set by one or a numberof data buses.

Differences between various logic anal-ysers, reflected in their price, are manifestedparticularly in the possibilities of data ex-traction and the way this is carried out.

The data display on the screen may takevarious forms. Apart from the usual timingdiagram as shown in Fig. 5, which showsthe behaviour of signals as a function oftime, it is also possible for the status or thedisassembler to be shown.

The status diagram shows the input chan-nels in a circuit section that were determinedpreviously by a configuration menu.

The disassembler diagram is particularlyuseful for an analysis of the interaction be-tween hardware and software. The recordeddata show the converted machine code ofthe computer system on test in mnemonicform. This process is called disassembly.

Every logic analyser is provided withspecial probes that facilitate proper connec-tion of the many inputs to the unit undertest. It is, of course, essential that fast pulsebursts be transmitted only via short linesthat introduce no or negligible losses.

The probes contain drivers that amplifythe signal prior to transmission to the ap-propriate analyser input.

Reference:

`Logic analyser' (5 parts), ElektorElectronics,January, February, April, June, July 1991.

Fig. 5. Timing diagram displayed on a logic analyser.


IIGITA N CT o\ k Ai 0 IQ Mrby 'f. Giffard

The outstanding properties of this function generator are itsaccurate frequency setting and very low distortion of the

sine -wave output. It is, therefore, eminently suitable for use asan a.f. generator for test and design.

many function generators pro-1...Jrefover the past 15 years or so, theone described here is NOT based on theubiquitous XR2206, but on discrete compo-nents. Although this does not necessarily makeit easier to construct, digital ICs are fairlyinexpensive and a digital circuit has fewercalibration points.

Basic designThe blodc diagram in Fig. 1. shows that the de-sign is based on four printed-thruit boards, in-dicated by the dashed lines. The heart of thefrequency -synthesis board at the top left isthe phase -locked loop (PLL). A phase com-parator compares the output frequency of acrystal oscillator with that of a voltage -con-trolled oscillator (VCO). Any phase differ-ence between the two signals causes the com-parator to generate a voltage that is used to syn-chronize the VCO with the reference oscilla-tor.

The wanted output frequency of the gen-erator is determined by the divisor of a di-vider in the PLL, which is set by means of akeyboard on the front panel. The divider alsoprovides the information for the four -digitLED display.

The VCO and the display share a com-mon power supply, although the supply volt-age to the VCO is 6.6 V, whereas that to theremainder of the circuit is 6.0 V.

The divider provides a rectangular sig-nal, which is directly proportional to the out-put frequency of the generator, to a digital -to -analogue (D -A) converter. This converterconsists of a shift register and an appropri-ate resistance network.

Each period of the wanted sine -wave sig-nal is built up of 32 clock pulses providedby the divider. This means that the clock sig-nal for the D -A conversion must lie between32 Hz and 3.2 MHz if the generator is to pro-vide an output frequency of 1 Hz to 100 kHz.

Since, in addition to the high precision ofthe wave shaping, the D -A converter is fol-lowed by a phase -locked filter, the distor-tion factor is even smaller than that of a Wien -bridge type sine -wave generator commonlyused for a.f. design and test measurements.Usually, function generators produce a sinewave that is converted into rectangular andtriangular signals of the same frequency: thistechnique does not result in low distortion fac-tors.

FRONitOVERPRU,,'PCT

Fig. 1. General view of the digital function generator.

Frequency

Output voltage

Offset

Technical data

Sine wave Rectangular

1 Hz -99.99 kHz

1 Vrms. e.m.f.

0.9-1 Vrms into 5011

±2.5 V (U0=0 V)

±4.75 V (U0=1 Vrms)

1 Hz -99.99 kHz

0-12 V e.m.f.

0-6 Vpp in to 500

±-12 V

Output impedance 6000 500

Frequency stability Three digits better than shown on display

Frequency resolution 0.01% of full-scale reading

Third -harmonic

distortion 0.03% typical

Slew rate 130 Wps

Triangular

10 Hz -99.99 kHz

0-12 V e.m.f.

0-6 Vpp into 500

-±12 V

500

ELEE.FOR ELECTRONICS OCTOBER 1991

DIGITAL FUNCTION GENERATOR - PART 1 Ea

oscillalorphasem.p.g.. vC0

PLL presettabie

shiftregister

! _ _

R -

networklockedfilter

17/7/7/70 0 0 E

Rt

l k-Jr' UNLOCK

Al1PLITUDE

sine wave!

AUP IJTUDE

1111_

MA

Fig. 2. Block diagram of the digital function generator.

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After the amplitude has been set, the sine -wave signal is fed to the output socket via abuffer amplifier.

For the other two wave forms, the shift reg-ister in the sine -wave section is used 'as a di-vider to ensure that the frequency remains thesame as that of the sine -wave signal. The sig-nal is fed to the rectangular/triangular out-put socket via a voltage amplifier, change -overselector and power amplifier. At the same time,it is used to drive an operational transcon-ductance amplifier (OTA) which convertsthe rectangular signal into a triangular one.

Divisors

An expanded, more detailed block diagramof the PLL frequency synthesis section isgiven in Fig. 5. Apart from the crystal oscil-lator, the VCO and the presettable divider, thesection contains several more dividers and

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Fig. 5. Block diagram of the phase -locked loop (PLL) section.


36 TEST & MEASUREMENT

Resistors:R1 = 2.2 kf2R2.R14,R15.R17.R19 =1 Mf2R3 = 10 S2R4 = 470R5 = 390 flR6,R7,R10,R20,R22 = 100 ItS2R8.R9,R13.R18,R21.R23 = 10 k12R11.R12.R16,R61-R65,R67.

R68 = 47 ki2R30 -R60 = 47011R66.R69 = 1 kf2

Capacitors:Cl = 100 pFC2 = 45 pF trimmerC3,C51-053 = 100 nF ceramicC4 = 47 pF, 10 V, tantalumC5 = 10 ILF. 40 V. bipolar, radialC6.C44 = 100 !IF, 10 V. radialC7,C8 = 47 nFC9.C10.C12,C14,C15,C17,

C18 = 100 nFC11 = 470 nFC13,C16 = 3.3 nFC19-C37,C45 = 100 nF, ceramicC38-C41.C50 = 47 nF, ceramicC42 = 1000µF, 25 V, radialC43 = 10 pF, 10 V, radial

Semiconductors:D1 = BB 212D2-D6.D10 = 1N4148D7 = 1N4001D8,D9,D11 = LED, 3 mmB1 = B80C1500LD1-LD4 = HD 11310T1 -T4 = BC 547 BT5 = BC 557 B101 = 74 HC 4060IC2 = 74 HC 40103IC3 = 4013IC4 = 74 HC 4046IC5 = 74 HCU 04IC6 = 74 HC 4040107 = 74 HC 151IC8 = 40591C9,1C10 = 74 HC 4518IC11 = 4067IC12 = 74 HC 40171C13-IC16 = 74 HC 696IC17 = 74 HC 682IC18,1C20 = 4093IC19 = 4001IC21 = 7806IC22-1025 = 74 HC 4543

Miscellaneous:X1 = quartz crystal 5.12 MHzS1,S2,S4,S6,S7 = push -to -make

switchS3 = double change -over switchS5 = SPST switchTr1 = mains transformer,

secondary 9 V, 500 mAPCB 910077-1PCB910077-2Front panel foil 910077-FKl. K3 = 34 -way D -type plug for

PCB mountingK2 = 14 -way D -type plug

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counters.The heart of the section is IC4, which con-

tains the phase comparator and a VCO thatis not used here. The comparator comparesthe two input signals, Cin (the reference fre-quency) and Sin (the signal frequency). Its out-put goes high when the phase of the signalfrequency leads that of the oscillator fre-quency, and low in the opposite situation(see also Fig. 3). For this purpose, it uses thefirst transitions, not the duty factor, of thesignals. After both transitions, irrespectiveof in which order, have been received, the out-put of the comparator has a high impedanceuntil the next first transition arrives. The du-ration of the low' and 'high' periods is, there-fore, directly proportional to the phase shift.

The low-pass loop filter at the output ofIC4 determines the properties of the loop asregards capture range, bandwidth, transientresponse and stability. After passing throughthis filter, the output voltage of the com-parator controls the VCO

The state when the two signals are in per-fect synchronization, that is, are locked, isindicated by the LED at the LOCK output ofIC4 going out.

The reference frequency of 160 Hz is ob-tained by dividing the 5.12 MHz oscillatorfrequency by 32 000.

The required frequency range of 32 Hzto 3.2 MHz is, however, not so easily ob-tained, because a large bandwidth leads to along transient response and poorstability. Thatis why the range, outside the loop, is splitinto five by decadic divider IC9-IC10: thewanted range is selected by multiplexer IC11.This means that the PLL is only active whenthe highest sub -range, that is, 320 kHz-32MHz,is selected.

The highest sub -range is itself further di-vided into four, so that the frequency of theVCO must be a multiple of two, which makesthe PLL very stable. Depending on the setfrequency,IC_6-IC7divides the VCO frequencyby one, two, four or eight. The combined di-visor of IC3b and ICg, which counts down froma preset number (max. 9 999) to 1 000, andIC6-1C7 lies, therefore, between 16 000 and32 000. Since the frequency of the signal fedby IC31i into IC4 must always be 160 Hz (ref-erence!), the frequency range of the VCO is2.5-5.12 MHz.

The setting of the divisor within the PLL,and thus the frequency within one of thesub -ranges, is effected by four cascaded up -down counters, IC13-IC16. These counteis havebeen set to a predetermined start value byjump leads. The start value may be overrid-den at any time by pressing the `up' or 'down'push-button key.

The set divisor is strobed by comparatorIC17, whose output is active as long as the ratiois lower than 3 200. This is of importance forthe locked filter in the analogue section.

The signal for the analogue section is takenfrom junction IC7-1C8 (pins 5 and 1 respec-tively). The frequency of this signal lies be-


Fig. 6. Circuit diagram of the digital sec-tion of the function generator.


AO

tween 320 kHz and 3.2 MHz.The signal is applied to divider IC9-4C10,

whose outputs have frequencies of 100,10-1,10-2, 10-3 and 10-4 times that of the input.

The sub -range is selected by multiplexerICB, whose setting is determined by a 0-4counter.

The analogue section thus receives a sig-nal whose frequency lies between 32 Hz and3.2 MHz and information, from DO -D4 of IC12,

as to which sub -range has been selected.The display board is fed with data repre-

senting the frequency setting via bus linesQ0 -Q15, D0 -D4. It is also connected to the'unlock' pin (PP) of IC4.

Circuit descriptionDigital sectionThe reference oscillator is formed by fourteen -

stage shift register IC1. The 5.12 MHz dockis divided internally by 27. The resulting40 kHz signals is taken from pin 6 and fed toa synchronous, binary down -counter, IC2. Astart count, related to the divisor (here :125)can be set at inputs PO -P7. From that start,the IC counts to zero, resets itself via pin 15and at the same time triggers flip-flop) IC3a.

The bistable functions as a binary scaler,which provides a pure signal with short tran-

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DIGITAL FUNCTION GENERATOR - PART 1 39

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sition durations as required by IC4. The loopfilter consists of R4 -R8 -C9 -Co.

The VCO consists of inverter IC5d, whichfunctions as an amplifier, and an LC circuitthat is tuned by dual capacitance diode DI.The control voltage for that diode is derivedfrom potential divider R6-117. The VCO sig-nal is taken from across IC5c.

The setting of divider IC.6-IC8 is rather com-plex. The fundamental VCO frequency ex-ists at pin 15 of IC7; scaled by two at pin 9 ofIC6; divided by four at pin 7 of IC6; and di-vided by eight at pin 6 of IC6. Which of thesesignals is passed on depends on the threebits at inputs A -C of multiplexer IC7.

The output of IC7 (pin 5) is fed to IC9-and thence to the analogue section

and is also used as the clock for programmablen -divider IC8. This chip consists of four cas-caded four -bit counters, which operate asdecade scalers. The decade range of the threethat determine the thousands, hundreds andtens is determined by appropriate bits ateach of the groups of inputs J13 -J16, J9 -J12and J5 -J8 respectively. That of the fourth de-pends on the level at inputs KA, KB and KC.This counter determines the units of the over-all divisor. If, for instance, n is to be 1507,J13 -J16 must be set to 1000; J9 -J12 to 1010;J5 -J8 to 0000; and J1 -J4 to 1110.

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Fig. 9. Circuit diagram of the analogue (display) section of the digital function generator.



The first three are also applied to IC7 insuch a manner that when the divisor lies be-tween 1 000 and 1 999, the VCO frequencydivided by eight (pin 4 of IC7) is internallycoupled to the output (pin 5) of IC7. Whenthe divisor is between 2 000 and 5 999, pin 3(VCO frequency divided by four) is con-nected to pin 5; when it lies between 6 000and 7 999, pins 1-2 are connected to pin 5;and when it is between 8 000 and 9 999, pin15 (VCO frequency) is coupled to pin 5. Theoverall divisor can, therefore, be set between8 000 and 16 000.

The signal is applied to IC9-IC10, whereit is divided by powers of 10 (101-104) as de-termined by the last transition of the clock.The four resulting signals, as well as the di-rect output of IC7, are applied to pin 9 ofmultiplexer IC11. This makes it possible tocover five decadic ranges with the same over-all divisor, that is, 32-320 Hz, 320-3200 Hz,3.2-32 kHz, 32-320 kHz and 0.32-3.2 MHz.

The multiplexer connects one of inputsX0 -X4 or X8 to pin 1 (COM) and thence, viaK2, to the analogue section.

The multiplexer is controlled by DECADEswitch S7. Combination R10 -C9 forms an anti -bounce network. When S7 is pressed, a pulseis generated by Schmitt trigger IC20c andused to clock Johnson counter ICp. Successiveoutputs of IC12 go high at each clock input.

To avoid elaborate settings of the divisorby means of binary coded decimal switchesand to make presetting possible, UP andDOWN switches, S2 and Si respectively, areused in combination with counters IC13-1C16.These circuits function as synchronous decadescalers with presettable output latches. Figure6 shows that these circuits consist of threestages: the counter/divider proper; a mem-ory (4 -bit register); and a three -state output.The circuits are cascaded in such a mannerthat IC13 processes the lowest decade, andIC16 the highest. Preset inputs A -D on eachof the scalers are connected to ground or thepositive supply line by wire bridges or jumpleads, except pin 3 of IC16, which is con-nected permanently to +6 V to ensure thatthe start-up value is always 1 000. It is, ofcourse, possible to set this value to a highernumber.

The counters may operate in either of twomodes, determined by RECALL/U-D switchS3. If their pin 11 is high, the output bufferwrites the content of the register, which inits turn stores the data of the counters whena first transition appears at pin 9 (RCLK). Thistransition is generated by switch S6 (STORE).In practice, this means that, when S3 is set toRECALL, S6 must be pressed briefly to ensure

that the counter status is stored and passedto the output of the IC. When S3 is set toU/D, pressing S6 causes the counter statusto be written into the register when the ICsreceive thedodcsignal at their pin2. Dependingon the level at pin 1, the counters count upor down at every first transition.

The correct timing of the above process isensured by the networks associated with

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FREQUENCY

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Fig. 11. The foil for the front panel is available through our Readers' services.


DIGITAL FUNCTION GENERATOR - PART 1 41

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switches Si and S2.When Si is pressed, a rectangular pulse is

generated by IC18,, and applied directly tothe U/D input of IC13-1C16. The frequencyof the pulse is determined by C12 and C13.Theclock is delayed slightly by network R18-Ci8and inverted by IC18c-ICtsd-IC19b-ICI9d-IC20b and then applied to pins 2. This ar-

rangement ensures that the level at U/D isalways low before the first transition arrives.

When S2 is pressed, a pulse is generatedby ICuit, which ensures that pins 1 remain highso that the ICs can count upwards.

Switch S3 enables selection of a low orfast dock. The dock speed can be altered bychanging the values of C13 and C16 (fast: S3

open) or C12 and C15 (slow: S3 dosed).Although the clock is connected to all four

pins 2, the ICs do not count identically, be-cause the RCO output of IC13 enables theENT input of IC14 only when in counter po-sition nine the next first transition arrives, thatis, at every tenth input pulse. The same hap-pens between IC14 and IC15 and betweenIC15 and 1C16.

Gates IC19a-b-d, 'Cm and a (discrete) ORgate disable the counters when position 1000or 9999 is reached. For instance, when the10 000th pulse is received, the RCO outputof IC16 is applied to NAND gate ICI8b so thatthe clock is disabled. The same happens whenthe four highest bits, Q12 -Q15, are low si-multaneously (divisor <1000) and the out-put of the wired -OR gate, D2 -D5, is also low.

The counters can be reset to the start valueby pressing the DEFAULT switch, $4. NetworkIC213a-C11-R15 generates the required pulse,which is applied to the LOAD input.

Analogue (display) sectionThe display section (circuit diagram in Fig. 9is constructed on a separate board, shown inFig. 11 (see also Fig. 10). All inputs to the sec-tion are via plug and socket (K1).

The Q -bus is split into four. Each of the fourgroups is coupled to a BCD -to -seven -bit con-verter, IC22-1C25, which drive the four displaysegments. The decimal points and the Hz/kHzLEDs are controlled via the D0 -D4 lines.

Diode Du lights when the PLL is not locked.

The second instalment of this three-part articlewillbe published in our November 1991 issue.

Anatomy of a practical pulse

First transition(previously:leading edge)

Pulse start line

1

1

4-

1 I i

First transition duration I

Overshoot

Pulse stop line

1

1

I kr\ 1

Last transition duration/11 (previously: decay or fall time)

1

(previously:risetime) I

Pulse duration (width)

Anatomy of a practical pulse

Top line

Last transition(previously:trailing edge)

- Base line

91E2'31-11

TECHNICAL literature is sprinkled withvague, misleading and ambiguous terms

applied to a pulse. What does the term 'pos-itive edge' mean when applied to a negativepulse? Is it the 'positive -going' edge, thatis, in this case, the last transition, previouslycalled 'decay' or 'fall' time, or is it the 'firsttransition', previously 'leading edge'?

Another common error is to assume thatthe 'pulse duration (or width)' is the dura-tion between the first and last proximals,whereas, in fact, it is the duration betweenthe pulse start and stop lines.

It should also be noted that the term 'dutycycle' should not be used in connection withpulses; the proper term for the ratio of thepulse duration to the pulse repetition periodof a periodic pulse train is 'duty factor'.

Fortunately, there are national and inter-national standards (among them British Standards

and IEC) that normalize the terms describ-ing properties of a pulse waveform: the mostimportant are shown in the drawing.


42

A [review of coding theoryby Brian P. McArdle

1. IntroductionThe general area of Coding Theory for telecommunications andcomputer applications is reviewed to provide a simple introductionto the subject. For further information, the reader should consult thebooks in the reference section.

There is no formal definition of a code. Essentially, messages arerepresented in some form more easily transmitted than normal writ-ten language. In this article, a code is a digital electronic signal thatrepresents a message symbol, such as a letter or number. For exam-ple, a teleprinter code would have to have a signal for every possi-ble symbol (26 letters, 10 numerals and other symbols) and signalsfor every operation (that is, space, carriage return and line feed con-trols). Figure 1 shows the arrangement.

SourceSource Alphabet rap

Channel

EncoderEncoding Operation E

DecoderDeccairg Operation E

Receiver

not distinguish between upper and lower case letters. The decodedmessages may be printed in upper case letters only. Thus, apartfrom small variations that should not affect normal understanding,the encoding operation, irrespective of its complexity or purpose,must be exactly and uniquely reversible.

A more formal mathematical definition is that the set of codedsymbols f Kai] must be uniquely partitioned (that is, can be di-vided into subsets that do not overlap) such that each partition canbe associated uniquely with a source symbol. Figure 2 shows the ar-rangement.

The remainder of this article attempts to explain the meaning ofE in different applications, such as error -detection -correction andencryption. It is always assumed that the encoding operation is uniquelyreversible unless otherwise stated. Another important assumption isthat a memory -less source is involved. The probabilities Pr(ay)s arethe probabilities of the general occurrence of these symbols in nor-mal language. In reality, letters occur in groups (digraphs and di-graphs). i before e, except before c is a well-known expression.Consequently, the probability of occurrence of a particular letter couldbe influenced by preceding letters. It is also assumed that a mem-ory -less source is being considered unless otherwise stated. Theanalysis is mostly confined to digital coding except for Section 6,which deals with coding for analogue signalling.

2. Different codes910102-11

Codes can be analysed from many different viewpoints, but engineersare generally concerned only with two main categories.

Fig. 1. Encoding/decoding operation.

An encoding operation E turns a message symbol ai into codedform for transmission over a channel. The set (ai) is the source al-phabet and Pr(apl is the set of probabilities associated with this al-phabet: Pr(di) is the probability that ai occurs. In normal language,this is the probability of occurrence of letters. The word 'channel'has a general meaning. It could be a cable, radio link or storage mediumwhere the receiver is retrieving the messages at some later time.Obviously, the receiver must be able to apply a decoding operationE -t. Hence, the principal requirement for a satisfactory code is thatthe coded symbols be uniquely decodable. In mathematical terms,E[ai] cannot represent more than one symbol. E[ai] cannot equal E[ai]unless ai and di are effectively the same symbol. For example, E might

91017:2.12

(a) Fixed length codesEvery character of such a code is represented by a block of bits withevery block having the same length. A typical example would be acomputer code, such as ASCII or EBCDIC. Both of these use blocksof eight bits. Thus, there is a total of 28 or 256 difference blocks.Any two blocks of the same code would have to differ in at leastone bit. The blocks need not be symbols (letters, numbers, punctu-ation marks) but can be controls (carriage return, line feed, etc.).

(b) Variable length codesConsider an alphabet of five symbols (a, b, c, d, a}. In (a), thiswould require a code of three bits per block or symbol with 23-5 = 3redundant blocks. However, if the following arrangement of threeblocks of two bits per block and two blocks of three bits per blockis used,

a=00,b= 01,c= 10,d= 110,e= 111,

the average length of a message would be reduced. Blocks still havea specific length, but it is no longer the same fixed length. The basicrequirement for unique decoding must still be maintained. For ex-ample, the bit sequence 011100011110 is easily decoded to bdaecwith no errors. This must apply for all combinations of the symbols.An important quantity is the average length L of a block given by

L = ri [Eq. 1]

Fig. 2. Partition of the set of coded symbols.


A REVIEW OF CODING THEORY 43

where pi is the probability of occurrence of block type j with ti bits.Ideally, this should be as small as possible to minimize the totalnumber of bits per message. To ensure this requirement, the largeprobabilities would be paired with the smaller blocks. Morse codeis another example where the common letter e is a dot, but z is twodashes followed by two dots.

This particular example has a special significance in addition tovariable length blocks. If it is rewritten in the form of a diagram asbelow, it seems to have a tree -type structure with different branches.

10 = C

je;,-700 = a 01 =b

110 = d

111=e

message seems instinctively correct. However, this method of mea-suring information has no connection with an actual signalling sys-tem. The entropy for A=fai I is given by

H( A) = -1Pr ( ai)logPr (a.) [Eq. 3]

and is a measure of the average information. An alternative expla-nation, which has become more common in recent years, is that it isa measure of the uncertainty in the information. H(A)=0 means thatPr(a)=1 and Pr(ai)=0 for all other messages. Consequently, thereis no doubt about the message. The maximum value occurs whenthe probabilities are the same and all messages are equally likely. Ifthere are n possible messages, H(A) is between 0 and log(n). The di-mension is information bits per symbol.

To apply information theory to coding theory, consider Fig. 3 wherethere is a noisy channel between sender A and receiver B. The jointentropy is given by the equation

H(A,B)=H(A)+H(B/A) [Eq. 4]

910102-16 where H(B/A) is known as the conditional entropy or equivocation.This in turn is defined as

Each branch is terminated by a symbol. The branches join togetherat nodal points which do not in themselves represent symbols. Thisarrangement indicates an instantaneous code. This means that thedecoding operation does not require a 'memory', that is, it does notrefer to blocks before or after any block that is being decoded. Inthe decoding of bdaec, it was not necessary to test the 3rd bit beforedeciding that the first two bits, 01, represented b. This property re-mained true for the full operation and for all decoding operations ir-respective of the combinations of symbols. (This should not be con-fused with a memory -less source, defined earlier, where there is norelationship between the occurrence of different symbols.). In aninstantaneous code, no block can be a prefix or suffix for anotherblock. Huffman codes, which are too involved to be considered in asimple overview, come into this category. However, it must be em-phasized that any collection of blocks of varying lengths does notmake an instantaneous code. There is a specific requirement givenby the Kraft inequality

lz)<_I

j 2 jto form such a code. Further analysis is outside the scope of thispaper and the reader is referred to the Reference Section for furtherstudy.

3. Information theoryInformation theory has steadily increased its profile over the pastfew years and it is no longer possible to study telecommunications,especially coding, without touching on it somewhere. At first glance,the ideas behind it can appear too general and abstract for simple,direct applications. The fundamental fact is that the basic conceptsof entropy, equivocation and channel capacity come from informa-tion theory, which, in turn, has influenced coding theory, and re-quire some explanation.

There is a fundamental difference between an electronic signal andits value as information. In sound broadcasting, un unmodulatedcarrier would not convey any programme content to a listener.Therefore, there is a need to be able to quantify the value of a sig-nal as information. In the 1920s, Hartley put forward the idea thatthe logarithmic function could be used as a measure of information.This was one of the landmarks in information theory. If two messages,ai and a, are independent,

log f Pr(ai) and Pr(ai) }=log Pr(ai) I +log I Pr(ai) I [Eq. 2]

and the base 2 is normally used. Remember that log' is not a linearfunction. The idea that the information contents of two independentmessages is simply the sum of the information of each separate

H(BI A)= IH(B/ ai) Pr ( aj). [Eq.5]

In non -mathematical terms, H(B IA) is a measure of the informa-tion loss in transmission. The channel capacity is given by

C(A,B)= maximum H(A)-H(A/B). [Eq. 6]

This appears correct because the limit on the information conveyedover a channel is determined by the original uncertainty of that in-formation (before reception) reduced by the uncertainty after recep-tion. Essential capacity is limited only by noise and the Hartley -Shannon law sets an upper limit of Wlog( I+SIN), where W is the in-formation bandwidth, S is the signal power and N is the noise power.For technical reasons, present-day systems operate well below thislimit. The reader is referred to the Reference Section for further study.

Sender A ill Receiver B

910102.13

Fig. 3. Communications channel.

From the point of coding and electronic engineering, Eq. 6 canbe simplified for normal use. Consider the case for a binary channelwhere A and B represent the input and output respectively. In gen-eral, the probability of a '0' or isl /2, which gives H(A)=log(2)=1.(The entropy of the source alphabet could be computed from the prob-ability of occurrence of the various symbols, but it is the channelthat is now under consideration.). If p is the probability of an errorwhere a '0' is received as a 1', or vice versa, the channel condi-tional probabilities are

A B 0 1

0 (1-P) p1 p (1-p)

From Eq. 5:

H(A,B)=Pr(0)[-(1-p)log( 1-p)-plog(p)1+Pr( 1 )[-( 1-p)log(1-p)-plog(p)] [Eq. 7]

which gives a new expression for the channel capacity:


44 SCIENCE & TECHNOLOGY

Bit:

C(A,B)= 1 +plog(p)+ (1-p)log(1-p) [Eq. 8]

in bits per symbol. This is the usual expression in most textbooks ontelecommunications. If the signalling rate is R symbols per second,the right-hand side is multiplied by R to give bits per second. Thus,information theory can be useful in the analysis of codes. The entirearea has become extensive and has been treated only superficiallyhere.

4. Error detection and correctionError detection and correction is one of the main applications ofcoding theory and paralleled its development. Figure 3 showed theproblems with errors where a '0' can be received as a 1' or viceversa. The use of the word 'receiver' is general in that it could rep-resent a storage medium, and so on. It suffices to say that data iscorrupted, which limits its value upon reproduction. Section 3demonstrated that channel capacity is limited only by noise. To re-duce the effects of errors, and therefore noise, extra bits are addedto a block of data bits to create new and larger blocks, which in turnallow errors to be identified.

Consider the (7,4) Hamming code as follows:

Position: 7 6 5 4 3 2 1 c4=(d7+d6+d5) mod 2

d7 d6 d5 C4 d3 c2 c1 c2=(d7+d6+d3) mod 2

c1=(d74-(15-Ed3) mod 2

There are three check bits in positions 1, 2 and 4 which havebeen derived from the data bits in the other four positions. The codeis linear in the sense that the check bits are linear combinations ofthe data bits and the encoding operation is simply the application ofthe three linear equations. Since every block will have a total ofseven bits without exception, the code is in the fixed length cate-gory. The position of the check bits within the block is very signif-icant. A receiver generates the check bits from the received databits and applies the decoding rule in Appendix 1. For example, if d3has been altered, c1 and c2 will not be validated and so on. The ar-rangement to check five data bits is given in Appendix 2. In boththese examples, the set of coded blocks is such that the minimumvariation between any two blocks in the same set is three bits. Thisis known as the Hamming distance. The reader is referred to Reference2 for a more detailed explanation. The main poinuo-note is that themethod identifies only one error per block. In general, r check bitshave (2r-1) possible combinations and thus r bits in a total size of nbits must satisfy the condition (2r-1)?n in orderio_identify andtherefore correct one error. To correct two or more errors per block,a code with a larger Hamming distance and more complicated ar-rangement would be needed.

Cyclic codes are the most commonly used for error detection andcorrection. These are also of the fixed length variety. For a block oftotal size n, the check bits are produced by a generator polynomialwhich is a factor of (xn+l). A typical example is the specificationMPT 1317 for the transmission of data over radio links. The formatis as follows:

DATABit 64 63 62 ....17

CHECK I PARITY16 15 . . . .2 1

with a block size of 64 bits. However, the first bit is for parity andis generated by the other 63 bits. The 15 check bits are generatedfrom the 48 data bits using the generator polynomial

gt0=x154.x141-x13-1-xlli-x4-Ex24.1 [Eq- 9]

which is a factor of (x63+1). Refer to Appendix 3 for an exact break-down. The data bits are the coefficients of the terms x62 down to .05inclusive. Some books write the data bits on the right-hand side ofthe format, but this is not important provided they represent thehigh power terms of the polynomial. The polynomial consisting ofonly the data bits is divided by g(x). The remainder is then added

back to produce a new polynomial such that g(x) is now a factor ofthe new polynomial. Since the check bits are essentially the originalremainder, they represent the terms x14 down to xo. Then a parity bitis added in order to detect odd numbers of I s and the full 64 -bitblock should have even parity. Refer to Appendix 4 for the genera-tion of a parity bit. The overall result is that the code can identifyand correct up to four errors per block. This is a considerable im-provement on the (7,4) Hamming code, but the operation is muchmore involved and the block size nine times larger. To check for er-rors, the receiver divides the polynomial by g(x) and there shouldbe no remainder..

Another example is the POCSAG code for paging, which usesthe format

DATA CHECK I PARITYBit 32 31 30 . . . .12 11 . . . . 2 1

and the generator polynomial

g (x) =x10+x9+x8+x6-1-x5+x3+1 [Eq. 10]

which is a factor of (x31+1). Refer to Appendix 5 for an exact break-down. The overall method is the same with the 21 data bits gener-ating the 10 check bits to produce a 3I -bit block plus an extra par-ity bit.

5. EncryptionIn encryption, the E operation, defined in Section 1, represents a se-crecy operation and is usually written as EK in most textbooks. Theparameter K is known as the key and its purpose is to vary the op-eration. This is in complete contrast to error -detection -correctionwhere exactly the same operation is performed on all blocks with-out exception. The importance of K is that it is generally the part ofEK that is kept secret. In a publicly known algorithm, such as theData Encryption Standard (DES), the complete algorithm is known.A user chooses a key from the set of possible keys {K} and encryptsthe data. Thus, only encrypted data appears on the channel of Fig. 1.The data can be recovered by the inverse or decryption operation EK-Iwhich also requires the correct key. If the key in use is kept secretand only known to authorized receivers, the data is kept secret.Obviously, { K} must be sufficiently large to prevent an unautho-rized user from trying each key in turn. There are a number of otherrequirements that are outside the scope of this paper.

There are three main methods of encryption.

(a) Stream encryptionIn Fig. 4, each bit of the data is added modulo 2 using an XOR logicoperation. A sequence of key bits is produced by the key generatorsuch that each data bit is encrypted by its own particular key bit.

910102-14

Fig. 4. Stream encryption.

The authorized receiver must know the method of key generation toreproduce the exact same sequence. The inverse operation is simplyto apply the key sequence in the correct order to the sequence of en-crypted bits. It would be too complicated to discuss the varioustechniques of key generation, but the most common method usesshift registers to generate a pseudo -random binary sequence. Generally,part of this process must be kept secret, such as the number of stagesand feedback arrangements. The current proposals to provide en-


A REVIEW OF CODING THEORY 45

cryption facilities on the cellular system GSM or for digital short-range radio DSRR are believed to use a form of stream encryption.However, the information is confidential and it is very likely thatthe exact method will not be made public.

(b) Block encryptionBlock encryption-see Fig. 5-differs from stream encryption in thata block is encrypted as a single unit. The most widely known methodis the US Data Encryption Standard, which uses blocks of 64 bitsfor input and output. The algorithm was published in 1977 and theexact method is for public information. The actual key is a 56 -bitblock, so that the number of possible keys is 256. In operation, theauthorized receiver would know the particular key in use and applythe inverse or decryption algorithm. Controversy has always sur-rounded the key size and recent articles have suggested methods foran improved DES.

Input BlockBlock

OPej,31II:m Output BlockEk

ik

Key

9101D2 - 15

Fig. 5. Block encryption.

Jfhe main advantage over (a) is that a satisfactory block opera -on creates interdependence between the bits of a block. If one bit

of an input block is varied, a number of bits in the output block arealtered. However, it is generally much slower than stream encryp-tion and cannot be used for high-speed telecommunications appli-cations.

(c) Public key encryptionPublic key encryption differs from the methods in (a) and (b) in thatpart of the key is made public, whence its name. The main require-ment is that the part which must remain secret should not be easilydeducible from the public part. A typical example is the RSA methodintroduced in 1978. Each user publishes two numbers, N and e. N isvery large, of the order of 80 digits, and the product of two primes,P and Q, while e and d satisfy the equation

1=ed mod (P -1)(Q-1). [Eq. 11]

Only e is made public; d, P and Q remain secret. If user A wishes toforward the message 'a' to user B, A looks up the parameters N ande for B and transmits

b= ae mod N.

User B recovers the original message from

a=ba mod N;

processing was in use. However, codes are also used in analogueelectronics, but their application is rather limited. For example, inthe PMR service, CTCSS (continuous tone controlled signallingsystem) has been around for a number of years. During transmis-sion, an encoder generates a specific audio tone that modulates theradio frequency carrier. This tone is continuous for the duration ofa message. In the absence of a CTCSS signal, the decoder at the re-ceiving end is deactuated.

Another example is tone selective calling, such as EEA andZVEI. In these methods, a sequence of five tones is used to form anaddress for a receiver. Both EEA and ZVEI have a total of 12 pos-sible tones. Each possible address consists of a set of five, whichactuates the receiver from the point of the user. However, despite theseexamples, coding has remained almost exclusively digital and the cel-lular GSM standard actually prohibits the use of tones.

On the secrecy side, there are voice scramblers that use fre-quency inversion, but increasingly the trend has been to digitize speech(for instance, ADPCM-Appendix 6) and to apply the techniquesof Section 5. Coding in analogue signal processing is very restrictedand need not be considered seriously.

7. References1 Information Theory and its Engineering Applications

D.A. Bell; Pitman (1968)

(2) Coding and Information TheoryRichard W. Hamming; Prentice -Hall (1980)

(3) Information and CodingJ.A. Llewllyn; Chartwell-Bratt (1987)

(4) A First Course in Coding TheoryRaymond Hill; Oxford University Press (1986)

(5) Codes and CryptographyDominic Welsh; Oxford University Press (1988)

(6) Coding for Digital RecordingJohn Watkinson; Focal Press (1990)

Appendix 1

The receiver re -calculates the check bits and validates them againstthe received values.

ci and c, are not validated = d3 is incorrectc1 and c4 are not validated = d5 is incorrectc, and c4 are not validated = d6 is incorrectc1, c2 and c4 are not validated d7 is incorrect

[Eq. 12] The sum of the indices of the check bits indicate the location ofthe erroneous bit. The correction process replaces a '0' by a '1' orvice versa. The principal difficulty is that two errors can cause acorrect bit to be changed.

[Eq. 13]

since d is one of the secret parameters, this cannot be done by anyother user. From a secrecy point of view, an unauthorized userwould have to factor N into P and Q to calculate d. Thus, as long asN is sufficiently large, this is impractical and the method is secure.There are other methods, such as the Merkle-Hellman-KnapsackMethod, but they all follow the same principle of a public and a pri-vate key. Equations 12 and 13 are the equivalents of the encodingand decoding operations.

Appendix 2The Hamming code for five data bits is d9 c8 d7 d6 d5 c4 d3 c, elandrequires four check bits with the same procedure as in the (7,4)code. r check bits can test up to 2r-1 locations. For r=3, this givesn=23-1, which leaves four data bits. For r=4, there is a block sizen=15 which allows for 11 data bits and four check bits in the order:

d15 d14 d13 d12 d11 d10 d9 c8 d7 d6 d5 c4 d3 c2 c1.

6. Coding for analogue signallingIn the preceding four sections, it was assumed that digital signal


Appendix 3For a 63 -bit block, the factors of the modulus are:

46 SCIENCE & TECHNOLOGY

(x63+1)=(x+1)(x2+x+1)(x3+x+1)(x3-Er2+1)(x6+x+1)(x6+x3+1)(x6+x4+.r2+x+1)(x6+x4+x3+x+1)(x6+x5+1)(.16-Fx5+x2+x+1)(x6+x5+_r3+x2+1)(x6+x5+x4+x+1)(x6+x5+.r4+x2+1).

Each of these factors is irreducible in that it cannot be factoredfurther while keeping real coefficients that are 0 or 1. In MPT 1317,the generator polynomial has the following factors:

x 15+.1-14+x13+x I I -Fx4+x2+1 =(x3+x2+1)(x6+x+1)(x6+.14+x2+x+1)

and is a factor of (x63+1) as per the mathematical conditions.

Appendix 4

as per the table. In Appendix 3, the generator polynomial is not afactor of (x64+1), but of (x63+1). The 64th parity bit has the genera-tor polynomial (x+1).

Appendix 5For the POCSAG, the modulus has the following factors

(x31+1)=(x+1)(x5+x41)(x5+x3+1)(x5+x3+x2+x+1)(x5+x4+x2+x+ 1 )(x5+.0+x3+x+1 )(x5+x4+x3+x2+ 1)

and the generator polynomial consists of the factors

Consider the generation of a parity bit in the following table x10+x9+.178+x6+.0-Fx3+1=(x5+x2+1)(x5+.14+.1-3+x2+1)

d3 0 0 1 1 and thus divides (x3I+1).d2 0 1 0 1

cl 0 1 1 0 Appendix 6In mathematical terms, the appropriate generator polynomial, g(x),divides the polynomial (d3x2-Ed2x+1). Let

g(x).(x+1) and d3x2-Fd2x+c1=(wix+w2)(x+1)

Equating terms on each side gives

w =d3 w1+w2=d2w2=w14-d2=d3+112=c1

ADPCM (adaptive differential pulse code modulation) was ac-cepted by the CCITT in 1984 for encoding speech. Each sampledspeech signal is originally encoded into a 12 -bit block. This is com-pared with 16 quantizing levels and the nearest level chosen. Thismeans that a block of 12 bits can now be replaced by a block of onlyfour bits. The actual mechanism is quite complicated and is consid-erably different from PCM. The final result is a transmission rate of12 kbit/sec.

FOUR -TERMINAL NETWORKS - PART 1

Getting to grips with attenuators

by Steve Knight, BSc

Most electronic equipment has an attenuator or attenuators of onesort or another. The object of these is to reduce to manageable

levels a signal we have elsewhere worked like mad to build up. Theusual objective is to turn out more than we want, then cut it down tothe size we do want. This might sound like an easy option but, likemany other things electronic, what we want and what we get aren'talways identical. So, apart from the intrusion of Sod's Law which

reigns universally, I hope that what follows will cast a ray of light onthe often neglected subject of attenuator systems.

might define transmission networksin general terms as circuits that have

two input terminals and two output termi-nals which are introduced between a gener-ator and its load impedance. These networks,which are referred to as four -terminal net-works, have properties that depend on thework they have to do in the transmissionsystem; in the case of an attenuator, whichis our main concern, it must enable us to ob-tain as output some desired fraction of theinput which is entirely independent of the sig-nal frequency. Clearly, such an attenuator sys-tem must be built up from purely resistive Fig. 1. Elementary attenuator.

1

elements, since reactive components will leadto frequency discrimination over all or cer-tain parts of the band. Further, existing im-pedance conditions in the system into whichthe attenuator is introduced must not be dis-turbed.

The most elementary attenuator of all isthe potential divider network that usually turnsup in the form of a variable resistor. Figure 1shows the system; our input goes between ter-minals 1 and 2 and we get the output fromterminals 3 and 4. This is a four -terminalnetwork, two of whose terminals are com-moned, and we obtain an output that can be


FOUR -TERMINAL NETWORKS - PART 1 47

a fraction of the input lying between the lim-its of 0 and 1. This is, no doubt, quite a sat-isfactory arrangement for turning the vol-ume up and down on a radio receiver, but itfails dismally if we harbour ambitions formaking precise quantitative measurements.Outside of the simplest applications to whichsuch an attenuator might be put, the disad-vantages are not hard to find.

The output has to feed into some kind ofload impedance; assume for a moment thatthis is resistive. What the input terminals 'see',therefore, is not a constant resistance, butone that is made up of a non-linear combi-nation of two resistances in parallel. Hence,the generator loading is variable and the po-tential difference-p.d.-at the input termi-nals is itself changing as the attenuator isoperated. Not really the sort of thing wewant if we are concerned with exactly howmuch attenuation we are getting and how itmight affect the characteristics of the trans-mission path. What do we require from anattenuator if it is to do a worthwhile job? Well,we want it to introduce any needed degreeof attenuation, but at the same time we wantthe input and output resistances to be suchthat the impedance conditions existing inthe circuit are not upset in any way: if Iwant to insert an attenuator into a 300 SIline, the attenuator impedance must also be300 CI.

Let us see how well-defined characteris-tics can be applied to attenuator networks sothat we can get the quantitative results wewant for our own particular requirements.

Basic characteristicsFour -terminal networks may come in two for-mats: symmetrical, in which we can inter-change the input and output terminals with-out affecting in any way the electrical char-acteristics of the circuit; and asymmetrical,in which this last condition does not hold. Thesimple attenuator of Fig. 1 is clearly asym-metrical. Each of the two formats may bebalanced or unbalanced, definitions to whichwe shall revert later on.

Symmetrical systems have two funda-mental characteristics that are essential to ourunderstanding of their function in life: thecharacteristic impedance, symbolized Zo inthe complex case, and the attenuation con-stant a. For purely resistive networks, wecan talk about characteristic resistance, Ro,and attenuation factor N.

Fig. 2. Defining the characteristic resistance of a network.

Characteristic resistanceLet us use our imagination for a moment.Suppose we have a network made up of aninfinite number of identical repetitive sec-tions as in Fig. 2a. Each section contains anumber of resistances, but how these are ac-tually arranged is not important at this stage.Suffice it to say that the resistance measuredat the input terminals, 1 and 2, will have acertain magnitude that will depend only onthe nature and circuit arrangement of the in-dividual sections. Suppose this resistance tobe R..

Now, it might appear that we are gettinginto deep water by this approach: as the net-work is infinitely long, what chance do wehave of calculating Ro for a specific casewhere the contents of only one (or of a finitenumber) of the sections is known? Fortunately,there is a way out of the impasse by a rela-tively simple dodge: suppose we remove thefirst section of the infinite chain as in Fig. 2b.The input resistance of the remainder of thearray as measured at terminals 5 and 6 willbe the same as that measured originally atterminals 1 and 2, because the infinite na-ture of the assembly is unaffected by re-moving the first section (or any finite num-ber of sections, come to that); so we could stillmeasure R. The input resistance of the sec-tion we have removed, however, is very un-likely to be R0, but we can make it so in oneof two ways: by replacing the rest of the in-finite chain (not a very practical way, to besure), or by putting across terminals 3 and 4a single resistance of a value equal to R0.

Now, this second method gives us theclue we want: no matter how many finitesections we remove from the infinite chain,if we terminate them with a resistor of value

R0, the input resistance will also be equal toR, since these sections are effectively ter-minated by Ro when reconnected to the restof the infinite array. With this proper valueof termination, that is, the characteristic re-sistance of each section, therefore, the inputconditions are such that anything connectedto the input terminals cannot distinguish be-tween an infinite network or a finite net-work so terminated.

Hence, we can define the characteristic re-sistance, Ro, of an attenuator network: anysymmetrical netvork terminated in R, willhave an input resistance also equal to Ro. Andthis, of course, must also be true when work-ing from output to input.

Attenuation factorWhat about the attenuation constant, or theattenuation factor as it is more generallyknown for resistive networks? Well, thissimply tells us the loss sustained in eachsection of a network. This loss may be ex-pressed as a fraction of the input or as a re-duction in decibels (dB); if the output is one-half the input, for example, the voltage (orcurrent) attenuation is 6 dB.

Each section of an attenuator networkwill attenuate at identical rates, but the ac-tual amount of attenuation is different as weproceed along the chain. Suppose, for instance,that we start off with unit input and lose halfof this input in each section. The output ofsection 1 will be lb and this becomes the inputto section 2. Here again, half the input islost, so the output of section 2 will be 1/2x1/2

or 1/4. After the following section, the out-put will be I/4x1/2=1/8. Hence, the amountof loss differs for each section because the

Fig. 3. The forms of the T -section and the 7: -sectionattenuators.


Fig. 4. Extreme termination conditions enable Ro to becalculated.

SCIENCE & TECHNOLOGY

magnitude remaining to be attenuated at anystage is becoming progressively smaller.The rare of loss, however, remains constant.

For n sections, each reducing the inputby some fraction p, the output will be pn; ifthe attenuation is expressed in dB, n sec-tions, each having a loss of p dB, will havean overall loss of lip dB.

Finding R0

Let us get down to the job of calculating thecharacteristic resistance of an attenuatorsection from a knowledge of its componentparts.

The forms taken by individual sectionsof the general attenuator array arc T-sec-tions or rc-sections; these are illustrated re-spectively in Fig. 3a and Fig. 3b. The T-sec-tion, as its configuration clearly implies, ismade up of a divided series -arm and a cen-tral shunt -arm. Used between equal impedances,the section is symmetrical when the series -arm contains two equal resistances. Theit-section, again as its form implies, con-sists of a single series -arm and two shunt -arms. This section is symmetrical when theshunt -arms are equal resistances. Both sec-tions, though symmetrical in the way de-scribed, are unbalanced in the sense that theseries -arm members are on one side of thesection 'through' wires. It is essential not toget 'symmetry' confused with balance whentalking about attenuator sections.

Working on the T-section forconvenience(the rr-section will lead us to exactly the sameend -result), suppose we terminate the sectionat points 3 and 4 with the extreme condi-tions of, first, an open -circuit, and secondly,a short-circuit. Figures 4a and 4b show thesecases. Clearly, in the first case, the input re-sistance seen at terminals I and 2 will be

Roc=R 1+R7,

and in the second case, it will be

Rsc=Ri+RiR21(R1+R7).

Now, between these open -circuited andshort-circuited conditions at the termina-tion, there can be an infinite range of resis-tance values; as the termination changesthrough this range, the input resistance willchange also. It seems reasonable, therefore,that there will be some value of the termi-nating resistance that will make the inputresistance also equal to this value. This value

RI

C5 -1=-1-1=0-IR2 riRo

,c5

910103- 1 E

Fig. 5. A correctly terminated section.

must be, in accordance with our earlier stateddefinition, the characteristic resistance ofthe attenuator section. So, from Fig. 5 we have

R0=R1+1?-4Ri+R.)1(Ril-R71-R0).

Solving this for Ro, we get

Ro=4(Ri2+2RIR2).

This enables us to find the characteristicresistance of a section from a knowledge ofthe resistor values making up the section.

We can now go one step further and findRo without necessarily knowing the valuesof the elements used in a section. All weneed to know is the input resistance (whichcan easily be measured) when the outputterminals are either open -circuited or short-circuited. For, looking back a few lines, wehave the product R0(.1? expressible as

Rock,=(R i+R2)[Ri+R 1R2/(R 1+R,)],

and multiplying this out, we get

RaRsc=R 12+2R IR,.

The right-hand side of this is R02, so that

Ro=4(Rock,),

which provides us with a very neat way ofcalculating the characteristic resistance of anysection.

Cascaded sectionsAlthough a single section will operate suc-cessfully as an attenuator, it is usual to havea number of sections in cascade or tandemso that a range of attenuation is provided.Once the R. of a particular section has beenfound, another section may be added to it with-out affecting the overall characteristic resis-tance. For, if in Fig. 6a section A is terminated

correctly by Ro, an identical section connectedin place of 1?, will in turn correctly termi-nate A, since section A will be unable todistinguish between the presence of sectionB or the presence of a single terminating re-sistor equal in value to 1?.

It is plain that no matter how many suchsections are wired in cascade, the input re-sistance will remain at R0. What will changeas we progress along the chain is the totalattenuation: each section will introduce thesame attenuation, but the desired overall at-tenuation can be achieved by using the re-quired number of sections.

Finding the attenuationKnowing the characteristic resistance of a cir-cuit into which an attenuator is to be in-serted, the problem of design boils down tofinding suitable values for R1 and R2, givenRo and the required attenuation.

A desired value of R, can be obtainedwith numerous combinations of Ri and R7;looking at Fig. 7, for instance, both sectionsshown have a characteristic resistance of 30(check on this for yourself!), but the net-work on the right will provide a greater de-gree of attenuation than the one on the left.

As we have already mentioned, the at-tenuation may be expressed as a fraction,that is, as a ratio of output voltage (110 toinput voltage (Ui). Expressed in decibels,

attenuation=20log(L/i/U0).

If the ratio of the input power, Pi, and theoutput power, P, is taken,

attenuation=l0log(Pi/Po) [dB]

=20logq(Pi/P0) [dB],

whencet/i/Uo=11(Pi/P0)=N ,the attenuationfactor. Notice that with this notation the at-tenuation is expressed as a whole number, nota proper fraction. This often makes calcula-tions easier.

In next month's concluding part of thisarticle, we shall see how the attenuation canbe provided, and how practical attenuatorscan be built to suit a variety of occasions.

Fig. 6. How sections can be cascaded. Fig. 7. Networks with identical R, but differentattenuations.


Fiber Optics -Part 2Last month we discussed the basic theory behind optical fiber

systems. It was learned that fiber optics are the light equivalent ofmicrowave waveguides, and that they are capable of immense

bandwidth communications. In this month's final instalment we willlook at fiber communications and some of the circuits that can be

used with experimenter grade optical fiber kits.

Losses in optical fibersystemsDecibel notation can be used for optical fiber sys-tems and refers to the gain or loss of each stage orcomponent in the system. The dBkm scale usesdecibels of gain or loss relative to the attenuationof a standard optical fiber section over a one ki-lometre (1 km) length. Alternatively, either dBmi(dB loss relative to attenuation over 1 mile) ordBl (a normalized unit length) can be used.

The light power at the output end of a opticalfiber (P0) is reduced compared with the inputlight power (PO because of losses in the system.As in many natural systems, light loss in the fibermaterial tends to be decaying exponentially(Fig. la), so obeys an equation of the form:

Fio=pine(-AiL)

Where:A is the length of the optical fiber being con-sidered;L is the unit length, i.e., the length for whiche-^11-=

There are several mechanisms for loss in fiber op-tics systems. Some of these are inherent in anylight -based system, while others are a function ofthe design of the specific system being con-sidered.

Defect lossesFigure lb shows several possible sources of lossowing to defects in the fiber itself. First, in uncladfibers, surface defects (nicks or scratches) thatbreech the integrity of the surface will allow lightto escape. In other words, not all of the light ispropagated along the fiber. Second (also in un-clad fibers), grease, oil or other contaminants onthe surface of the fiber may form an area with anindex of refraction different from what is ex-pected and cause the light direction to change.And if the contaminant has an index of refractionsimilar to glass, then it may act as if it were glassand cause loss of light to the outside world. Fi-nally, there is always the possibility of inclusions,i.e., objects, specks or voids in the material mak-ing up the optical fiber. Inclusions can affect bothclad and unclad fibers. When light hits the inclu-sion it tends to scatter in all directions, causing aloss. Some of the light rays scattered from the in-

by Joseph J. Carr

a

0.75

0.632

050

0.37

025

0.1350125

b

LOST LIGHT

SCRATCHEDSURFACE

.0"

GREASE, OIL, etc.

1L

SCATTEREDLIGHT

o

u

IO

-1.25

-2

-4.3

-6

-g

LENGTH ( L )

9100512-11- Ha

INCLUSION

LOST LIGHT -A

_1C:32 -II. 11b

Fig. 1. (a) Exponential decay of signal strength along an optical fiber; (b) mechanisms ofloss in fibers.

clusion may recombine either destructively orconstructively with the main ray, but most do not.

Inverse square law lossesIn all light systems there is the possibility oflosses caused by spreading (divergence) of thebeam. If you take a flashlight and point it at awall, and measure the illuminance per unit area atthe wall at a distance of, say, one meter, and then

back off to twice the distance (two meters) andthen measure again, you will find that the illumin-ance has dropped to one-fourth. In other words,the illuminance per unit area is inversely propor-tional to the square of the distance (1/d2).

Transmission lossesThese losses are caused by light that is caught inthe cladding material of clad optical fibers. This


50 RADIO, TELEVISION AND COMMUNICATIONS

Fig. 2. (a) Air coupling creates reflection losses: (b) matching liquid coupling improves thejunction.

light is either lost to the outside, or is trapped inthe cladding layer and is thus not available forpropagation in the core.

Absorption lossesThis form of loss is caused by the nature of thecore material, and is inversely proportional to thetransparency of the material. In addition, in somematerials absorption losses are not uniformacross the light spectrum, but are thought to bewavelength sensitive.

Coupling lossesAnother form of loss is caused by coupling sys-tems. All couplings (of which more is said later)have loss associated with them. Several differentlosses of this sort are identified.

Mismatched fiber diametersThis form of loss is caused by coupling a largediameter fiber (DL) to a small diameter fiber (Ds);i.e., the larger diameter fiber transmits to the

lesser diameter fiber. As a ratio, this loss is ex-pressed by:

loss= -10 log(Ds)D

(dB) (21L

Numerical aperture (NA) coupling lossesAnother form of coupling loss occurs when thenumerical apertures of the two fibers are mis-matched. If NAR is the numerical aperture of thereceiving fiber, and NAT that of the transmittingfiber, then the loss is expressed as:

NARloss= -10 log (-NAT) (dB) 131

Fresnel reflection lossesThese losses occur at the interface between theoptical fiber and air-see Fig. 2a-and arecaused by the large change of refractive index atthe glass -air barrier. There are actually two lossesto consider. First is the loss caused by internal re -

DATAFIBER FIBER-110.OPTIC OPTIC

DATA OUTTRANSMITTER RECEIVER

DATA IN

DATA A

b

TRANSMITTER A

RECEIVER B

DATA B

FIBEROPTICUNK

Y COUPLER

DATA

{Y COUPLER

910092 - - 12 a

IK012-111- t3s

DATA B

TRANSMATTER B

RECEIVER A

DATA A

flection from the inner surface of the interface,while the second is caused by reflection from theopposite surface across the air gap in the coup-ling. Typically, the internal reflection loss is ofthe order of 4 per cent, while the external reflec-tion is about 8 per cent.

Any kind of reflection in an optical trans-mission system may be compared to reflectionsin a radio transmission line (standing -wave ratio).Studying standing waves and related subjects inbooks on RF systems can yield some under-standing of these problems. The amount of reflec-tion in coupled optical systems uses similararithmetic:

r_ipi -132)2

Pi 1-P2141

Where:r is the coefficient of reflection;pi is the coefficient of reflection for the receiv-ing material;pl is the coefficient of reflection for the trans-mitting material.

Mismatched reflection coefficients are analogousto a mismatch of impedances in a radio trans-mission system. Whereas the cure in a trans-mission line system is to use an impedancematching device, in fiber optics, a coupler thatmatches the 'optical impedances', that is, thecoefficients of reflection, is used. Figure 2bshows a coupling between the two ends of fibers(lenses may or may not be used depending on thesystem). The gel or liquid used to seal the coup-lings must have a coefficient of reflection similarto that of the fiber. The reflection losses arethereby reduced or even eliminated.

Optical fibercommunications systemsA communications system requires an informa-tion signal source (e.g., voice, music, digital data,or an analogue voltage representing a physicalparameter), a transmitter, a propagation media (inthis case optical fibers), a receiver preamplifier, areceiver, and an output. In addition, the transmit-ter may include any of several different forms ofencoder or modulator, and the receiver may con-tain a decoder or demodulator.

Figure 3 shows two main forms of communi-cations link. The simplex system is shown inFig. 3a. In this system a single transmitter sendslight (information) over the path in only one di-rection to a receiver set at the other end. The re-ceiver can not reply or otherwise send data backthe other way. The simplex system requires onlya single transmitter and a single receiver moduleper channel.

A duplex system (Fig. 3b) is able to simulta-neously send data in both directions, allowingboth send and receive capability at both ends. Theduplex system requires a receiver and a transmit-ter module at both ends, plus two-way beam split-ting Y -couplers at each end.

There is also a half -duplex system known incommunications, but this is of little interest here.A half -duplex system can transmit in both direc-tions, but not at the same time.

Fig. 3. (a) Simplex communications system: (b) duplex communications system.


OPEN - COLLECTORINVERTER S ,1

* RI U°1":"LED( al slccn t

Fig. 4. (a) Digital driver circuit; (b) anal-ogue driver circuit; (c) d.c. offset bias isneeded to prevent distortion of the analoguesignal.

Receiver amplifier andtransmitter driver circuitsBefore the fiber optics system is useful for com-munications, a means must be provided for con-vening electrical (analogue or digital) signalsinto light beams. Also necessary is a means forconverting the light beams from the optical fibersback into electrical signals. These jobs are doneby driver and receiver preamplifier circuits re-spectively.

Figure 4 shows two possible driver circuits.Both circuits use light emitting diodes (LEDs) asthe light source. The circuit in Fig. 4a is useful fordigital data communications. The signals arecharacterized by on/off (high/low or I/O) states inwhich the light emitting diode is either on or off,indicating which of the two possible binary digitsis required at the moment.

The driver circuit consists of an open -collec-tor digital inverter. These devices obey a verysimple rule: if the input (A) is high, the output (B)is low; if the input (A) is low, the output (B) ishigh. Thus, when the input data signal is high,

point B is low, so the cathode of the LED isgrounded. The LED turns on and sends a lightbeam down the optical fiber line. But when theinput data line is low, point B is high, so the LEDis ungrounded (and therefore turned off) - nolight enters the fiber. The resistor (RI) is used tolimit the current flowing in the LED to a safevalue. Its resistance is found from Ohm's law andthe maximum allowable LED current:

(U+)-0.7121 =

imax[5]

An analogue driver circuit suitable for voice andinstrumentation signals is shown in Fig. 4b. Thiscircuit is based on an operational amplifier. Thereare two aspects to this circuit: the signal path andthe d.c. offset bias. The latter is needed in orderto place the output voltage at a point where theLED lights at about one-half of its maximum bril-liance when the input voltage, Mb is zero. Thatway, negative signals will reduce the LED bright-ness, but will not turn it off (see Fig. 4c). In otherwords, biasing avoids clipping of the negativepeaks. If the expected signals are monopolar, U1is set to barely turn on the LED when the inputsignal is zero.

The signal Uin sees an inverting follower witha gain of RF/Rin, so the total output voltage (ac-counting for the d.c. bias) is:

rin +1))

[6]

Because the network R2/PI is a resistor voltagedivider, the value of U1 will vary from 0 volts toa maximum of:

(U+)PtUi =

Therefore, we may conclude that Uo(nax) is:

[7]

RF) rp.)(4+,)[8]Uc(max)- Rin + R, +Pt Rin

Three different receiver preamplifier circuits areshown in Fig. 5: analogue versions are shown inFigs. 5a and 5b, while a digital version is shownin Fig. 5c. The analogue versions of the receiverpreamplifiers are based on operational amplifiers.Both analogue receiver preamplifiers use aphotodiode as the sensor. These p -n or p-i-n junc-tion diodes produce an output current, 4,, that isproportional to the illumination on the diodejunction.

The version shown in Fig. 5a is based on theinverting follower circuit. The diode is connectedso that its non -inverting input is grounded, i.e., atzero volts potential, and the diode current is ap-plied to the inverting input. The feedback current(IF) exactly balances the diode current, so the out-put voltage will be:

U0 = RP

Fiber Optics - Part 2

Fig. 5. (a) Inverting follower receiver; (b)non -inverting follower receiver; (c) digital re-ceiver based on Schmitt trigger.

U0=1 RLRF

+ 1Rin

[10]

Both analogue circuits will respond to digital cir-cuits, but they are not at their best for that type ofsignal. Digital signals will have to be recon-structed because of sloppiness caused by disper-sion. A better circuit is that of Fig. 5c. In thiscircuit, the sensor is a phototransistor connectedin the common emitter configuration. When lightfalls on the base region, the transistor conducts,causing its collector to be at a potential only a fewtenths of a volt above ground potential. Conver-sely, when no light falls on the base, the collectorof the transistor is at a potential close to U+, thepower supply potential.

The clean-up action occurs in the followingstage: a Schmitt trigger. The output of such a de-vice will snap high when the input voltage ex-ceeds a certain minimum threshold, and remainshigh until the input voltage drops below anotherthreshold (these thresholds are not equal). Thus,the output of the Schmitt trigger is a clean digitalsignal, while the sensed signal is a lot moresloppy.

[91Conclusion

The non -inverting follower version shown inFig. 5b uses the diode current to produce a volt-age drop (U1) across a load resistance, RL. Theoutput voltage for this circuit is:

Fibre optics can be used by experimenters andsmall users: even Radio Shack (Tandy) has fiberoptics kits available. Other suppliers can be foundby perusing the ads of this magazine. EdmundScientific (USA) has a particularly nice selection.


AM BROADCAST RECEIVER

In these days of stereo FM, medium -wave AM reception is oftenviewed with contempt, and the quality of a sophisticated hi-fi stereotuner is solely judged by the specifications of the FM section, withthe AM section taken for granted. No wonder that the AM reception

of these hi-fi tuners is no better than that of any cheap portableradio.

article describes a top -of -the -rangeAM receiver with the following features:High quality audio output with a -6 dBbandwidth of 12 Hz to 4.5 kHz, and atotal harmonic distortion of only 0.5%.High input signal range of nearly 60 dBwith virtually no change in the audio out-put level.

A phase -locked loop (PLL) which keepsthe intermediate frequency (IF) at exactly455 kHz even when the receiver isslightly mistuned to a station. Since asmall difference between the tuning of

by R. Shankar

the antenna and that of the oscillator sec-tions is inevitable in any receiver, this re-laxation can be considered as a kind offine tuning. Also, oscillator drift has noeffect on the IF.

An extremely steep IF cut-off charac-teristic thanks to a communications -grade ceramic filter.

Principle of operationBasically a superheterodyne type, the pro-posed receiver is considerably more sophis-

ticated than a good many hi-fi stereo tuners.A few salient differences are listed below.

AFCAs mentioned before, the local oscillator fre-quency is adjusted by a PLL so that the IF isheld at exactly 455 kHz. This AFC (automat-ic frequency control) function may be turnedoff by a switch, in which case the local oscil-lator runs free, functioning just as in mostother receivers. A certain level of carrier hasto be present for the AFC to function. Other-wise, it is automatically switched off to en-

CA3046

.1111 111L.11L-11ammociem

Si LEVEL HI

RI

EMI

C2

20p

Ll

C3

15p

Cl - 20-20517

LO BF256A

R3

.7.

C4

lOn

520.-1610 kHz

0EMI

T5...T9 = IC1 = CA3046T10 = BC548B

L2. L3 = YRCS12374AC

C5

10n

CEO

1R12

R5R7

:On

T2 VO 2x 73

BF494

mincio

BF C6710n

494 --1

R6

R8

TIO

15

220n

L2

R13 R16

0r

T4

OD

R18 R20

C11 R19

220n

RIO R14

fl C8

10n

R9

T6

1710n

O

Rloa11

770

R17

C12NI=Ton

UAGC

V?1N4148 UL

DI

O

T8

C13

10n

R21

12V

C14

ion

R22 R24

711

R23 BF495

R26

R25 115

R15

20n

H C)+50. A...2.5rnA

T50

O

910093 - 11

L3

Fig. 1. Front end of the AM broadcast receiver.


111AM BROADCAST RECEIVER

Fig. 2.Basic schematic and response of the of the automatic gain control (AGC).

able tuning to a different station.

AGCMost AM receivers use a single transistor forthe AGC (automatic gain control) function.By contrast, this receiver uses an elaboratetwo -stage AGC circuit to obtain a level con-trol range of nearly 60 dB without overload-ing or excessive noise.

Active detectorWhereas most AM receivers use a singlepoint -contact diode for the detector, the

present design is based on an active detectorstage, which has a distortion of less than0.5% rather than the more usual 2% intro-duced by a diode.

Design backgroundThe total circuit, excluding the power sup-ply, uses 6 standard ICs and 15 transistors.Two ceramic filters are used. At first, it wasdecided to have only two IF stages, but in theend four were found necessary. The reasonis as follows: in order to keep the audio out-

put as flat as possible up to 4.5 kHz, the Qfactor of each IF stage has to be reduced to amaximum of 25 by loading it. Inevitably, re-ducing the Q reduces the gain. Further gainreduction is caused by the insertion loss ofthe ceramic filter. So, in order to obtain ahigh input sensitivity (27 µV for 20 dB sig-nal-to-noise ratio), four IF stages were foundnecessary.

For 4.5 kHz sidebands, the relative gainroll -off of the IF stages alone (including theceramic filter) amounts to about 4 dB. Theroll -off of the antenna tuned circuit is an -

R29

C16

lOn

R27

6V

R30

C1810n

R31

n

-

kLR28

T12

L4

1==IFlt

BF495

1-(6V5)

C19

C20

10n

R34 R36

113

R38

BF459R35 R37

139

R40 R45

R46

2N3906R42

100LS C21

MEI

43 V C2510n

C23 C24

n 10n

R48 1R52

116

I 11

R44T15

erita... IF

C22

70n

N3904

R31 R47

12V

0

D3

BC548B

C27

10n

TI7

D5 BC558B

4x1N4148

R51

RIDC26

7nt. NIB

UAGC R56

R55

C30Fl 1 - CFG455FISFD12 EMI I

4_7L4 = YRCS1109BAC

LS = THCS11100AC T19C31 R54

1-71_2

BC550C

C28

70n

R53

14.

12V

AUDIOOUT

*C29 )2p2 -

T18

BC558B

810093 -13

Fig. 3. IF amplifier and demodulator.


54 RADIO AND TELEVISION

other 2 dB. This means that the total audioattenuation at 4.5 kHz is 6 dB, which is con-sidered good for an AM receiver.

An important point must be made aboutthe IFTs (intermediate frequency transfor-mers) used in this circuit. These are basically468 -kHz types with built-in 180-pF capaci-tors. Though these IFTs can be tuned to455 kHz, the stability suffers. For this reason,every IFT is fitted with an external 10-pF ca-pacitor across the primary winding. Notethat these 10-pF capacitors are not shown inthe circuit diagram. The IFTs have not beenused in the order suggested by the manufac-turer, Toko. The reason that these particular468 -kHz types were selected is that therewere no good 455 -kHz types available fromthe author's supplier, Cirkit. Since the signallevels and impedances in the IF amplifierhave been painstakingly ascertained, no at-tempts should be made to use types of IFTsother than the ones indicated, or the purposeof this project is defeated.

As regards distortion in amplifiers basedon bipolar transistors, odd -order harmonicdistortion is negligible for base -emitter volt-age swings of less than 5 mV. At 3 mV, forexample, the third -order harmonic distor-tion is only 0.06%. The second -order har-monic distortion, however, is nearly 3%.Fortunately, even -order products do notmatter in the IF stages as they fall outside thepass -band. The upshot is that the simplesingle -ended common -emitter stage is per-fectly acceptable. Not so in the mixer stage,however, where even -order intermodula-tion products are troublesome when the re-ceiver is tuned to about 910 kHz (twice theIF), because they coincide with the image fre-quency. As the second -order products areconsiderable even for small input swings, adifferential amplifier should be used for themixer.

Contrary to popular belief, bipolar tran-sistors are better suited to AGC circuits thanFETs or MOSFETs. There are two reasons forthis. First, the linearity of a FET or MOSFETdeteriorates rapidly when its transconduct-ance is made smaller than a tenth of its maxi-mum value. Interestingly, precisely thatcondition would arise when a strong signalis received. Second, the exponential charac-teristic of a bipolar transistor results in anAGC loop gain which is independent of thecarrier strength. For a FET, it can vary by a

factor of 100, giving rise to wildly varyingloop response times and lower cut-off fre-quencies.

The circuitInput stageFigure 1 shows the input stage of the re-ceiver, which comprises the mixer, the firstIF amplifier, and the AGC. The antennatuned circuit comprises Li, Cia, C2 and C3,with the ferrite rod, Li, tapped at a fourthfrom earth. When the signal strength is high,a resistor can be switched in series with thetuned circuit for two purposes. First, it in-creases the bandwidth so that the high -fre-quency response is improved. Second, it

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Fig. 4. Local oscillator.

decreases the signal level, and prevents dis-tortion.

Transistor Ti is a buffer that prevents themixer stage from loading the tuned circuit,especially at the higher end of the band. Tioperates at fd4ds., which results in maxi-mum gain and linearity.

The mixer stage is formed by T2, T3 and

T4, with the local oscillator signal injectedinto their emitters. The mixer stage also func-tions as a part of the AGC. At low input le-vels, T4 conducts fully so that the maximumpossible gain of the mixer stage is achieved.The operation of the AGC now dependsmainly on the current through Tn. When theinput signal is fairly strong, T3 steals some of

UL

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Fig. 5. Phase -locked loop.


AEI BROADCASTBROADCAST RECEIVER

27pV...20mV 55_V 2.7mV 2.1mV 6,11V 0.21V

910093. 16

Fig. 6. Overview of crucial signal levels. quality factors and transconductance requirements at various points in the receiver.

the current through T4, so that the signal atthe base is limited to about 3 mV. At stillhigher input levels, T4 starts to conduct lesswhile Tit has a flat gain. Figure 2 shows asimplified model of the AGC stage. Here,current sources have been used instead oftransistors. Currents h and 12 are 1 mA and50 µA respectively. It can be shown that thetotal gain varies purely exponentially withthe control voltage.

The two -stage AGC ensures that the firstIF stage can not be overloaded by strong sig-nals. This means that the signal handling ca-pability of the receiver is increased by afactor equal to the mixer stage gain.

Transistor To) is included because thebase currents of T5 and T7 are comparable tothe collector current of T6. Compensation isachieved with Cs and C7 which also ensurean AC path from the bases of T5 and T7 toground. To improve stability, resistors Rs,R14 and R17 provide RF isolation for 10. The100-52 resistors likewise prevent spuriousVHF oscillations of the transistors.

IF amplifier and AM detectorThe circuit diagram of the second, third andfourth IF stage, and the active detector, isshown in Fig. 3. The input and output of theceramic filter are matched to 1.5 kn.

The active AM detector is formed by T14-117. 114 and T15 form a voltage -to -currentconverter with the collectors of the transis-tors at a relatively high impedance. A push-pull rectifier based on D2 -D3 is used. Therectified RF current that flows through 133develops a voltage across R50, which issmoothed by C26. The circuit around TI6, T17,D4 and Ds provides a quiescent current ofabout 1 µA through D2 and D3 to keep thevoltage swing at their junction to a mini-mum. The detected audio signal across R50 isfiltered by R51 and 0s, and buffered by 118before it is taken to the output of the receiver(to reduce traces of IF to an absolute mini-mum, one more ladder section consisting ofa 47042 resistor and a I 0-nF capacitor can beadded before Tis).

The AGC feedback is also taken from thispoint. The instantaneous carrier level is inte-grated by T19, and fed to the base of Ts. Ex-cluding T19, the gain of the AGC from thebase of TS to the base of Tis is Vb./ V1, orabout 23. The lower cut-off frequency of the

audio signal is 23/ (2nRs4C3o), or 20 Hz. Asmentioned earlier, this cut-off frequency isvirtually independent of the carrier strength.

Local oscillatorFigure 4 shows the local oscillator based on asingle CA3046 transistor array. The oscilla-tor is basically a Hartley type, although aportion of the current through 1.6a is also fedto L6b by T23. This causes the effective induct-ance of L6 to change by a small amount. Theratio of the currents through T23 and T24varies between 3:1 and 1:3, corresponding toVc values of 4 V and 8 V respectively. Thetotal frequency variation achieved is about25%. At the lowest oscillator frequency,975 kHz, the variation is about 25 kHz -more than enough to correct tracking devia-tions between the antenna and oscillator sec-tions. When the AFC is disabled, Vc is heldat half the supply voltage to minimize fre-quency drift. In this case, TB and T24 conductequally.

T22 is the positive feedback transistorwhose d.c. setting determines the strength ofthe oscillations. Teo and T21 are used to con-trol the amplitude to about 1 mA(0.6/R64) byreducing the d.c. voltage at the base of T22when Do and 137 start conducting.

For optimum tracking of the antenna andoscillator sections, the parallel combinationof C36 and C37 should be equal to 0.978 timesthe total antenna tuning capacitance at thelowest frequency. This works out at 285 pF.Capacitors C36 and 07 should be 1% types.When C2 and 09, the trimmers associatedwith the tuning capacitor, are properly ad-justed, a worst -case deviation of smaller than5.5 kHz is possible.

PLL sectionThe zero crossings of the IF output signal areshaped by Ni and N2 and fed to one phasecomparator input of the 4046 PLL. The otherinput is supplied with the reference fre-quency of 455 kHz from a ceramic resonatoroscillator based on N3. The output of thephase comparator is fed to a ladder filter toremove traces of the product IF/4. When theAFC is switched on, the 'A' inputs of IC4a

and IC4c are made logic 0, provided a suffi-cient carrier strength exists (for this, the cur-rent through In has to be smaller than1 mA). The output of the PLL is fed to the

loop filter formed by R81, R82 and C44. TheAGC time constant is about 0.05 s while thedamping factor is about unity.

If Vc is between 4 V and 8 V when theAFC is switched on, Ds lights to indicate thatthe PLL is locked. If the AFC is off, or if thecarrier is too weak, or if Vc exceeds its range,Ds is off. When the AFC is not on, IC4a selectsthe resistive divider, and Vc becomes 6 V. Atthe same time, IC3 is disabled.

Signal levelsFigure 6 shows the signal levels at variouspoints in the RF-IF chain. For the IFTs with190 pF total capacitance, the parallel loadingcorresponding to a Q factor of 25 is 46 kfl.Since the IFTs have an inherent Q of 90, ad-ditional loading is necessary to reduce the Qfactor to this value. With the transistor typesused, the base -emitter swings of all stages,except the mixer, have been kept in the 2 to3 mV range as this is the best compromisebetween noise and distortion. Note, how-ever, that even the mixer can handle up to13 mV of RF, and still produce less than 1%distortion. At higher signal levels, RI may beswitched on.


'IL{L T OMPACT C

WILL IT SUCCC`vill?

based on an original article by P. van Willenswaard

Just when consumers were getting reconciled with the idea thatdigital audio tape (DAT) is virtually dead and started to buy

more CDs, the two DAT protagonists, Sony and Philips,decided to confuse the market afresh by each announcing

another new system: the Mini Disc (MD) and the Digital CompactCassette (DCC) respectively. We briefly explain the reasons

behind these new systems and have a closer look at thedigital compact cassette technology.

Wthe demise of the digital audio tape(rAT)system-lcilled by the music in-

dustry which believed (probably rightly)that a system that enables the making of per-fect copies from copies would cost themthousands of millions of dollars-audio equip-ment manufacturers began to reformulate theirdevelopment programmes with the aim of get-ting a larger slice of tomorrow's market.

They found that cassettes are bought pri-marily for portable or mobile equipment; CDsand LPs almost exclusively for use at home.Moreover, three times as many cassettes(recorded and blank) as CDs and long-play-ing records (LPs) are sold world-wide. Becauseof their higher prices, CDs and LPs bring ina much larger revenue, of course. See Fig 1.This schism is further illustrated by consid-ering the apportioning of tape decks over.he various types of audio equipment. About12 per cent are used in hi -ft separates or in-tegrated systems; the remainder, that is, al-most 90 per cent, are used in car radio/cas-sette players, walkman equipment and otherportable sets-see Fig. 2.

Annual sales of compactcassettes amount to a

staggering 2,600 million

The realization that the bulk of the demandfor audio equipment is for portable/mobileunits form the basis of Sony's and Philips'development programs.

Sony, still smarting from the collapse ofthe DAT market on which they had pinnedmuch hope for the future, decided to getaway from cassettes for the time being andhave another look at their tremendously suc-cessful CD system (CDs today account forabout half the £13,000 million -$22,000

- world market for recorded music).Knowing that CDs cannot compete with

the cassette in the popular mobile/portable

markets, Sony recently unveiled Mini Disc(MD). The MD is considerably smaller thanthe Compact Disc, but it has the tremendousadvantage over its big brother that it is record-able and erasable. Moreover, the system isimmune to shocks and jolts, which means thatit can be used on the move, that is, in portableand mobile (car) units, with impunity.

Philips has gone back to its earlier inven-tion, the compact cassette, but now in digi-tal format: the Digital Compact Cassette(DCC) system. Since a large part of the 2,600million cassettes sold every year are used infairly inexpensive equipment, Philips rea-soned that, to stand a chance of cornering asizeable part of that market, they had to offersomething that works with relative inexpen-sive tape decks and tape. Even better if thenew system would also accept standard com-pact cassettes.

Both systems are promised (threatened?)to be on the market in late 1992. The suc-cess, or otherwise, of the new systems is inthe hands of confused consumers and a cau-tious music industry which will insist that bothsystems are fitted with appropriate circuitrythat will prevent copying from copies. It isstill their, and many others', opinion thathome taping is nothing but theft.

DCC requirementsRecording 16 -bit audio samples at a rate of44 100 times a second in two channels re-quires 1.4 Mb/s; if space for encoding anderror correction is included, that figure shouldbe doubled. This kind of high -density record-ing is impossible on cheap tape. Ways have,therefore, to be found to make recordingwith fewer data possible.

At the same time, the choice of inexpen-sive, slightly modified compact cassette tapedecks means a considerable reduction in de-velopment cost and time. The most frequentlyheard complaint about the current cassettesystem is the varying quality of sound re-production: there is loss of high frequencies

1

RV 1

Percentages

World-wide 1988

147)1(11

Blank cassettes

World-wide 1990

V .11 fri)

Blank cassettes (MI%) 91XE.:

Cassette players in

Portables (42%) 910043-12


THE DIGITALDIGITAL COMPACT CASSETTE

Fig. 3. Division of the tape into nine digital tracks;at the end of the tape. head and direction reverseand the lower half is written to or read from. Thehead also has two slits for the playing of analoguerecordings.

IN 1

A /D-PASC

D /A

OUT

L

O R

S UB BANDFLTERS

SIMIAN°ENCODER/DECODER

ERROR CORRECTIONENCODER/DECODER

I

CHANNELMOD & DEMOD

HEAD

Fig. 4. Block schematic of a DCC player.

owing to misaligned ordirty heads;the sound quality is proportionalto the quality of the tape trans-port (causing fading at high fre-quencies and wow and flutter)and to minor damage of the tape.

A digital recording, if ade-quate error correction is pro-vided, is fundamentally immuneto such imperfections. A digitalcompact cassette would, there-fore, in many instances give anappreciable improvement in soundreproduction, which would makeit immediately attractive to largenumbers of potential users. Thispresupposes, of course, that pro-duction costs are low enough tomake the sales price acceptableat the lower end of the market.This is quite feasible, because thetape decks of the DCC playerare not very different from thoseof current cassette players, whilethe cost of the electronics wouldbecome low enough if manu-facturing quantities were large.There are an estimated 200 mil-lion cassette players in the world;even replacement of 5-10 percentof these would ensure success.

Reverting to the question ofaudio quality: how is good qual-ity to be ensured if the audio sig-nal is stored digitally in a muchsmaller space than 1.4 Mb/s?The tape width of a compact cas-sette is 3.78 mm. The vertical tol-erance of transporting the tapeacross the head is about 0.05 mm.This means that the DCC re-mains within that tolerance ifhalf the tape width is divided intonine tracks, each 185 pm wide,of which 70 pm is used for read-ing (play -back) and the remain-der for writing (recording)-seeFig. 3. Eight of the nine tracksare allocated to audio; the ninth,to system information.It has been found that at the stan-

dard tape speed of cassette play-ers (4.76 cm/s), about 100 kb/sper track can be saved reliably.There is thus about 800 kb/s spacefor the audio signal; this has beenstandardized at 768 kb/s. Halfof that, however, is used for theReed -Solomon interleaving (inaid of effective error correction),synchronization bits, and a 10:8modulation mode, which meansthat only 384 kb/s remain forthe audio information proper.That is about a quarter of theearlier established 1.4 Mb/s; inother words, at a sampling fre-quency of 44.1 kHz, there wouldonly be space for a four -bit am-plitude description per channel.That is, of course, quite unac-ceptable. Philips has, therefore,

developed precision adaptive sub -band cod-ing-PASC.

Precision adaptive sub -bandcodingPASC is not, as one might suppose, a highlydeveloped form of data compression as usedin computers or data links. Whereas pulse -code modulation (PCM) systems give a totaland perfect (within the constraints of 16 -bitlinear encoding) digital representation ofthe sound that reaches a microphone, PASCuses a perceptive approach: it takes humanhearing as reference. PASC analyses the sig-nal and computes what may be audible andwhat not, and stores only what is in a veryefficient notation. Subsequently, the result-ing digital information is reorganized in sucha manner that the available space is reallo-cated optimally.

Only what may be audible isstored and encoded

The way a signal is processed in DCC isshown schematically in Fig. 4. First, the a.f.range is divided into 32 sub -bands: the rea-son for this will become clear later. The di-vision is linear, not logarithmic as in humanhearing: each sub -band is, therefore, about600 Hz wide. In digital signal processing(DSP), it has been possible for some time todivide the a.f. band into sub -ranges that canlater be recombined without any errors ordeterioration as far as frequency and phaseare concerned. Philips is confident of beingable to put this entire process on to one chipin the near future.

Threshold and maskingof hearingThe next step occurs in every sub -band. Theminimum intensity a sound requires in orderto be heard depends on its frequency-seeFig. 5. This means that in each sub -band athreshold has to be determined; any infor-mation below that limit is ignored.

Another aspect is masking: it is well knownthat there are circumstances in which a strongsignal will mask, that is, make inaudible, aweaker signal at a frequency not far removedfrom its own-see Fig. 6. When someonewhispers something to you in a quiet street,you can understand it perfectly. If, however,just then a noisy lorry passes by, you have tobe a lip-reader to understand.

Another form of masking occurs in time.If a weak sound is preceded a few millisec-onds earlier by a strong sound, we cannot hearthe weak sound. Even the opposite can hap-pen: a weak sound emitted a few millisecondsbefore a strong one may, in certain circum-stances, mask the strong sound. It all de-pends on the difference in intensity and theseparation in time.

The PASC processor analyses the signalin each of the sub -bands and constantly adapts

I_ I_ E KTOR ELECTRONICS OCTOBER 1991

58 AUDIO & HI-FI

the various thresholds to the actual signal.The thresholds are, therefore, not static, butfollow the signal content. This results inhigher efficiency, because only the part ofthe signal above the threshold, i.e., that whichis audible, is stored and encoded.

Mantissa and indexThe amplitude of the part of the signal thatremains above the thresholds must be quan-tized. Whereas compact disc technologyuses 16 -bit linear quantization, PASC uses afloating-point technique. This works in a man-ner similar to the presentation of a largenumber on a pocket calculator: with a man-tissa and an index (exponent). The mantissaof most pocket calculators is normally eightdigits long and represents the value of thenumber. This is followed by the exponent,normally two digits (positive or negative),which weights the numbers.

In PASC, the mantissa is 15 bits long, butthe processor may decide to work with ashorter one if not all the 15 bits appear nec-essary. This may happen, for instance, whenthe signal is just above the noise level or thresh-old of hearing. The mantissa thus determinesthe resolution of the system, while the indexshifts the area over which the mantissa op-erates up and down the dynamic range.

The index is not linear over the entire dy-namic range, but reflects the non-linear ac-tion of the ear. After all, the ear does not dif-ferentiate between sounds at 118 dB and 120dB: both are too loud. At the other end ofthe range, however, 2 dB steps would be muchtoo coarse, because the ear is highly sensi-tive to subtle differences when listening toweak sounds. Defusing the amplitude in thisflexible manner gives a further increase inefficiency compared with the linear encod-ing used in PCM.

If PASC fulfils its earlypromise, it will mean a

revolution in digital audio

The maximum length of the mantissa of15 bits is reflected in DCC by what Philipscalls the (theoretical) THD+N figure of -92 dB,which is 1 bit, or 6 dB, above the -98 dB ofCD technology. But that technology is notflexible, so that the THD+N figure is alwaysreferred to the same absolute level: a -20 dBsignal will, therefore, have its THD+N fig-ure only 78 dB lower. Because of the float-ing-point amplitude description, in DCC theTHD+N distance of 92 dB will, however, bemaintained until the noise threshold of thesystem is reached. Philips states a dynamicrange of -108 dB, which is 10 dB better thanthat of CD.

All internal calculations are carried outwith 24 -bit wide words. The associated noiselevel is -146 dB, but Philips claims that theresolution made possible by the mantissaand index system will handsomely exceedthat level. The real problems arise not in this

part of the system, but in the ana-logue -to -digital (A -D) converter thatneeds to precede the PASC. Thecrux of the matter is that even thebest state-of-the-art converters can-not provide more than 18-19 bitsof reliable a.f. information.

ReallocationOne of the nicer properties of musicis that the waveform varies con-stantly. This means that not all avail-able space in the 32 sub -bands is inuse all the time: that in a number ofthem will, in fact, be empty (abovethe threshold, that is). A certainspace is reserved in the ultimatecode that is written on to the tapefor each of the sub -bands. A sub -band that is temporarily partially,or wholly, empty would not be veryefficient. When the recording codeis being composed, PASC moves in-formation from full sub -bands to par-tially, or wholly, empty ones (andprovides it with an address code toensure that during play -back thesignal is reconstructed correctly).The process of moving the infor-mation is called reallocation.

The gain in efficiency resultingfrom this method of signal pro-cessing is surprising: from 16 bitsper channel for linear PCM to an av-erage 4 bits per channel for PASC.This means a gain of a factor 4 indata content. PASC does not workwith two channels, but with a stereosignal (the correct encoding of stereosignals proved to be quite difficult; stabilitygave particular problems). Just as there isno sharp division between left-hand andright-hand, there is none between one sam-ple of the signal and the next: PASC workswith small groups of samples. On average,the code for each stereo sample containseight bits of space that are written on to thetape. These are not bits of waveform: the 8 -bit code generated by PASC is much moreintelligent than that. The 8 -bit words shouldbe considered as pieces of a continuouslydeveloping jigsaw puzzle, which are resolvedby the decoder during playback.

The decoder need not be high-tech, be-cause the intelligence is stored in the codeitself, since this contains all necessary keysforencoding and reallocation. In other words,the keys have become part of the 8 -bit codes.

Storing the keys in ROM near the de-coder would seriously limit the system (per-haps even make it unworkable), becauseeach sound is different from others and thusrequires its own unique PASC code. Thekeys in the 8 -bit code control the operationof the decoder: what is the reallocation in-formation; which mantissa and index wereused; and so on. The code is the brain: thedecoder only carries out orders.

The decoder is, therefore, suitable notonly for working with DCC codes, but withall codes within the same family, such as Digital

Fig. 5. Characteristic to indicate the threshold ofhearing. both as regards loudness L (vertical scale)and frequency F (horizontal scale). Sounds belowthe curve can be omitted. Sound A is just percep-tible.

Fig. 6. Strong signal B raises the threshold ofhearing locally, as it were, so that sound A is nolonger audible: B masks A.

Audio Broadcasting (DAB) codes (in theframework of the European Eureka project,Philips partakes in DAB experiments), or evensimple 2 -bit codes of limited audio quality.Moreover, the code may change from one toanother instantly because the decoder doesnot have to switch over.

Breakthrough in digital audioIf PASC fulfils its promise, it will mean arevolution in digital audio. Storage spacefor the large amount of data generated bydigital audio has always been a problem.Until now, storage with simple means, alonger playback time, a substantially highersampling rate than the current 16 bits ofamplitude, and so on, were not possible inpractice. It may also well be, because a sig-nal after being processed in PASC is tech-nically less complex, that the analogue elec-tronics in the play -back chain can becomeless complex.

As far as the sound quality of DCC is con-cerned, it is too early to make a definitivejudgement. However, first impressions gainedduring a recent demonstration of the sys-tem to a number of international journal-ists were good. When amplifiers of reason-able (not high -end) quality were used, therewas no perceptible difference between aCD and a DCC.


1-6)//NLI" j N_ N LjEiS

The contents of this article are based on information obtained frommanufacturers in the electrical and electronics industry and do notimply practical experience by Elektor Electronics or its consultants.

SWITCH-IVIODE VOLTAGE REGULATMS11/12575/LM2577(National Semiconductor)

TIIErecently introduced LM2575 andLM2577 switch -mode voltage regulators

in the Simple Switcher series from NationalSemiconductor have a wide voltage range,and are remarkably easy to get going withonly a handful of external components. Thedevices in the LM2575 series are step-downregulators, and those in the LM2577 seriesstep-up regulators. Offering an efficiency ofover 80%, these ICs are a good alternative tothe ubiquitous 1-A regulators in the 78)o.series.

The new step-up and step-down conver-ters are based on switch -mode voltage regu-lation, which gives them a higher efficiency(70-90%) than linear voltage regulators (30-60%). By virtue of the on -chip power stages,the external component count remains low- basically, only a choke, a power diode anda few decoupling capacitors are required.The adjustable regulators in addition requirea voltage divider that determines the outputvoltage. All devices in the LM2575 and theLM2577 series are available in a 5 -pin TO220case (LM257xT), and a 4 -pin TO3 case(LM257xK). The case of these devices is al-ways connected to ground. Fixed voltage aswell as variable voltage devices are available- see Table 1. The LM1575 and LM1577should be used in heavy-duty applicationswhere an extended temperature range (Ti upto 150°C) is required. The main technicaldata of the regulator family are summarizedin Table 2.

Since the switch -mode regulators arebased on a fixed oscillator frequency with avariable duty factor, they do not require theusual minimum load current to operate cor-rectly. The ICs feature an internal thermaloverload protection which is actuated atTff125 °C, and an output current limitingcircuit. A further advantage of these new ICsis that they are inexpensive and uncritical ofthe choke type used.

Step-down regulator LM2575The operating principle of the so-called buck(step-down) regulator is illustrated in Fig. 1.Switches Si and Sz close in alternate fashionat a certain rate. When Si is dosed, the self-inductance of the charge choke, L, causes a

Table 1. Overview of devices in LM2575 LM2577 Simple Switcher family

TypeStep-down

Step-up

Output voltage%,/

12 V

15Vadjustable

12V15Vadjustable

10220 case T03 case0125751-5.0 LM2575K-5.0LM2575T-12 0.12575K-12LM2575T-15 LM2575K-15LM2575T-ADJ LM2575K-ADJ

LM2577T-12 LM2577K-120.42577T-15 0.12577K-15LM2577T-ADJ 0.12577K-ADJ

slowly rising current to flow. This currentcauses energy to be stored in the form of amagnetic field in the core of the choke. Whenthe switches toggle (Si is opened, and Sz isclosed), the self-inductance first causes a cur-rent IL to be fed through L. As a result, themagnetic energy stored in the choke, andwith it IL, decreases gradually. This processis repeated as Si and Sz dose and open alter-nately. Capacitor C smooths the output cur-rent, /our, because it is charged as /L. rises,and discharged by the load as IL drops again.

lour

of 1T ZT 3T R Si' ST Ti ST 91MOMS- f

Fig. 1. Basic operation of the U.12575step-down regulator.

By controlling the mark -space ratio of indi-vidual pulses, /our can be set such that thedesired output voltage Uour appears acrossthe load. The drawing in Fig. 1 illustrates thisprocess when the output current drops.Switch Si is actually a power transistor con-trolled by a rectangular wave generator(Fig. lb). The other switch, S2, is formed by adiode that enables IL to continue flowingwhen the power transistor is switched off. Toprevent losses as a result of the relativelyhigh currents, the diode must have a fastswitching characteristic. This requirement ismet by the use of a Schottky power diode.

The internal circuit of the step-downregulator Type LM2575 is shown in Fig. 2.The power transistor is an n -p -n type thatswitches the input voltage, UN, to the outputpin marked OUT. The transistor is driven by

LM2575

o u.1 A switching transistor

thermallimiting

ErVIXT

Um

error amplifie

= 1.23 V

driver

52 kHzoscillator

current

liMitirig

min

ilX!!. I'

Fig. 2. Internal schematic diagram of the012575.


60 APLICATION NOTES

an oscillator with a fixed output frequency of52 kHz. The pulsewidth of the oscillator out-put signal is controlled by an error amplifier,which compares the voltage supplied by apotential divider connected across the out-put with a fixed voltage supplied by an inter-nal 1.23-V reference.

Apart from the protection circuits for thepower transistor (junction temperature; out-put overload), there is a reset generator,which is also used to switch off the outputvoltage via the TTL-compatible ON/OFFinput of the regulator.

A typical application circuit of theLM2575 is shown in Fig. 3. Here, the adju-stable device Type LM2575T-ADJ is used.The output voltage is set with the aid of anexternal circuit. When a fixed -voltage deviceis used in this circuit, pin 4 must be con-nected to the output. A 3-A Schottky diodeType 1N5821 is used as the previously men-tioned power diode. Components R2, D2 andC5 supply a voltage of +5 V for the ON/OFFcontrol input, whose function may be testedwith the aid of jumper J1. When the ON/OFFfunction is not required, the 5-V supply andthe jumper may be omitted, and pin 5 con-nected permanently to ground. As indicatedin Fig. 3, the circuit must have a single ear-thing point. This arrangement prevents volt-age fluctuations as a result of current peaksupsetting the control loop, and should betaken into account when a PCB is designedfor the supply. Spurious pulses are sup-pressed by electrolytic capacitors C2 and C3at the input and output of the regulator. In

Table 2. Lf/125751LM2577 main technical data

Parameter

Input voltage

Output voltage

Output currentCurrent limit (Mo.)Switch saturation voltageQuiescent currentReference voltageInput current at VFBOscillator frequencyEfficiency

Operating temperatureProtections

Special features

UOUT

/OUT

USAT

/a

UF B

IFB

tog°

LM2575 LM2577

3.5 - 35 V1.23 - 30 V(always < VIN)

A

2.2 A0.9 V

10 mA

1.23 V (±2%)50 nA 100 nA52 kHz (±10%)

80%

-400=C < Ti < +1250'Cinternal current limitthermal shutdownON/OFF input Soft -start

undervoltage detector

3.5 - 40 Vmax. 60 V(always > \fit()2 A (depends on VOJT)4.3 A

0.5 V

some cases, the a.c. resistance of C3 may betoo high, when the capacitor is best replacedby a number of capacitors in parallel with thesame total capacitance (small electrolytic ca-pacitors have a lower parasitic inductancethan large ones). The value of C3 must al-ways be greater than 330 µF. The datasheetadvises against the use of tantalum capaci-tors, which cause instability of the controlloop. When the output voltage is higher than25 V, C3 must be replaced by a type with a

*see textUIN

UOUT <UIN1.23...25V

1AIC17...35V LM2575T-ADJ (max.30V)

UIN OUT500RH

R25TVOFF GND UFB P1

5 410k

+1.23V

C I C22D2= * cl J 1

* i ridRI CO C4

in= min100n 2205

40V__..700n f3

5V61r= orb

1 -2 =ON2-3= OFF 1N5821 47070n

25V

910018 -13

Fig. 3. Application circuit with the LM2575.

Table 3. Test data for application circuit in Fig. 3

12 V to 5 V

20 V to 12 V

/our (A) POUT (W) /IN (A) PIN (W) Efficiency Ripple (my)0.1 0.5 0.07 0.84 60%0.3 1.5 0.2 2.4 63%0.6 3.0 0.35 4.2 71% 1.2

0.1 1.2 0.08 1.6 75% 2

0.3 3.6 0.2 4.0 90% 2

0.6 7.2 0.4 8.0 90% 2.21.0 12.0 0.7 14.0 86% 2.5

working voltage of 40 V or more. Capacitors0 and C4 form a virtual short-circuit for thefast pulse edges, and so secure the necessaryRF suppression.

The output voltage, Um', of the regulatoris simple to calculate. Potential divider Pt-Rtreduces the desired output voltage to 1.23 V.The equation for the current, I, allows us todetermine the values of Pt and Rt:

/ = 1.23 V /12.1 = (Uour - 1.23 V) / Pt

This allows the value of Pt to be determinedwhen RI is known:

= (Uour- 1.23 V) /1.23 V

Pt= R1 (Uour / 1.23 V - 1)

The value of PI should lie between 1 kf2 and10 kf2. Evidently, the operating principle ofthe regulator does not allow an output volt-age that is higher than the input voltage. Theresults of a test on the circuit of Fig. 3 at twodifferent settings are given in Table 3. Par-ticular attention was paid to the behaviour ofthe LM2575 when the input voltage was in-creased slowly. The result: the set outputvoltage was maintained exactly (by contrast,some other regulators produce spurious out-put voltages higher than the set value whenthe input voltage drops below a certain mini-mum value). The measured values show anefficiency of between 60% and 90%, and anoutput voltage ripple of a few millivolts(measured in mV,,, with an a.c. millivolt -meter).

Step-up regulator LM2577Before discussing the operation and applica-tion of the LM2577, it may be useful to reca-pitulate the principle of the step-up or boostvoltage regulator. Figure 4 shows the basics.It is seen that the charge inductor, L, is con-nected ahead of switches St and S2 (compareFig. 1) which are dosed alternately. When S2


SWITCH -MODE VOLTAGE REGULATORS LM2575/LM2577 61

L St lour

Up

/0

I min,..43UTI,01

To

la ....

o -... - _ _

o

,,,,

.

CUT

COI IT 2T 3T 4T 57 ST ST ST SI"

9%018 - II

Fig. 4. Basic operation of the Lf.12577step-up regulator.

is dosed, the choke is connected to the fullinput voltage, Um. Because of the self-in-ductance, the current through L can only riselinearly, i.e., not pulse -like. As in the step-down regulator, magnetic energy is stored inthe core of the choke. As soon as S2 is opened,and Si is closed, the self-inductance the

IC1

Lt.12577T-ADJIN S'e.ITCH

Fig. 6. Application circuit with the Lf.12577.

choke forces the current IL to keep flowing inthe same direction. The voltage developedby the self-inductance is polarized such thatit is added to the input voltage. Hence, theoutput voltage is higher than the input volt-age. Capacitor C smooths the output current,Lour. In the LM2577 (Fig. 4b), switch S2 isformed by a power transistor, and Si by afast switching diode. The diode is reversebiased when the transistor conducts. Thevoltage and current waveforms that occur inthe circuit when the output current is re-duced are shown in Fig. 4c.

Figure 5 gives the block schematic diag-ram of the LM2577. Like the LM2575, theLM2577 contains a 52 -kHz pulse -widthmodulated oscillator driven by an error am-plifier which obtains its control informationfrom the voltage at the VFB input. In addi-tion to thermal and current overload protec-tion circuits, the LM2577 features a soft -startgenerator that prevents a high rush -in cur-rent through choke L. Another preventivefunction is that provided by the undervolt-age limit circuit. When the input voltagedrops below 2.9 V (typ.), the power stage isautomatically switched off to prevent it con-ducting almost continuously. When theundervoltage detector is actuated, the out-put voltage drops to virtually the input volt-age.

An application circuit of a step-up volt-age converter based on the LM2577 is givenin Fig. 6. Here, the adjustable device Type

Table 4. Test data for application circuit in Fig. 6

lOUT (A) POUT (W) lit4 (A) PIN (W) Efficiency Ripple (mV)5 V to 12 V 0.1 1.2 0.3 1.5 80%

0.2 2.4 0.6 3.0 80% 8

0.3 3.6 0.9 4.5 80% 15

12 V to 28 V 0.1 2.8 0.3 3.6 77% 0

0.2 5.6 0.6 7.2 77%

Ci.3 8.4 0.9 10.8 77% 15

U42577

Inet.:11 Swill:nonOen,t.,,s7Itage 1,1 tr:

I Te

Son 52 Stiroscillator Slider

Z 2.. .v; ,7

tiia°®

t211

CulferalIrnornor- snotAcr

"r 1.23V

91]Ct5-15

5. Internal schematic diagram of theU.12577.

LA,42577T-ADJ is used. The 1N5821 and the500-RH choke are familiar from the step-down regulator (Fig. 3), as are the centralground point and the possible problemswith a single electrolytic capacitor in posi-tion C3.

Since the same reference voltage of 1.23 Vis used, the calculation of the output voltageis identical to that shown above for theLM2575. It should be noted, though, that thecircuit in Fig. 6 can not supply output volt-ages lower than the input voltage. The maxi-mum output current depends on theproperties of choke Li, the ratio Uou-r/LInv,and the maximum permissible currentthrough the power transistor in the LM2577(4.3 A typ.). The datasheet provides the fol-lowing information on Jour and Uour:

(Jour < 60 VUST < 10 UN/our < 2.1 A ULN UOUT

Also, when the output voltage is higher than30 V, the 1N5821 should be replaced by atype with a higher rated voltage.

The main results of tests carried out onthe circuit of Fig. 6 are listed in Table 4.Clearly, the efficiency rises as the differencebetween the input and the output voltagebecomes smaller. The ripple on the outputvoltage is higher than with the step-downregulator, but a mere 10 mVru or so will notbe a problem for most applications. Whenexperimenting with the LM2577, make surethat the voltage divider is always connected.Without a properly dimensioned and dosedcontrol loop, the IC may be destroyed easily.

Source:Datasheets LM1575-ADJ / LM2575-ADJ -Simple Snitcher Step -Down Regulator. Na-tional Semiconductor 1990.Datasheets LM1577-ADJ 1 LM2577-ADI -Simple Snitcher Step -Up Regulator. NationalSemiconductor 1990.


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Computer -controlled 113 10.00 1.75Teletext decoder

Plotter driver (B. Lewetz) 117 5.75 1.00Fax interface for IBM PCs 119 7.00 1.23RAM extension for BBC -B 123 5.00 0.88EPROM simulator 129 5.75 1.00RS -232 spliner 1411 5.75 1.00Centronics ADC/DAC 1421 5.75 1.00Transistor characteristic 1431 6.50 1.14plotting (Atari ST b.sw)

ROTA -copy for BASIC 1441 6.50 1.14computerMultifunction measurement 1461 6.50 1.14card for PC58751 programmer 1471 6.50 1.14PT100 thermometer 1481 6.50 1.14Logic analyser: IBM soft- 1491 16.50 2.89ware on disk. incl. GALLogic analyser Atari soft- 1501 16.50 2.89ware on risk. incl. GALPlotter driver (D. Sijtsma) 1541 9.50 1.66

111 10.00 1.75

PROJECT

All11111111111PC -controlled weath,- 1551station -PC -controlled weather 1561station - 2I/O interface for Atari 1571

Tek/Intel file converter 1581

EVW video digitizer 1591Timecode interface 1611RTC for Atari ST 1621

6.50 1.14

6.50 1.14

6.506.509.506.50

'50

1.141_141.661.14

14

PRINTEWRCUET BOARDS

APRIL 1991L ogic analyser ;3..:- control board

MIDI programmechangerSurf generator 9061188 -bit I/O for Atari 9100056-m band transvener 910010Wattmeter- meter board 910011-1- display board 910011-2

Moving -coil (MC) 910016preamplifierAM -FM receiver 910021Dimmer for halogen lights- transmitter 910032-1- receiver 910032-2

PC -controlled semi-conductor tester

MAY 19918032/8052 Computer 910042Battery tester 906056Laser (1)Moving -magnet (MM) 900111preamplifierUniversal I/O interface 910046for IBM PCs

Virmieband active rod 910023antenna

900094-5 15.75 2.76900t38 5.75 1.00

Not available10.50 1.849.75 1.71

5.50 0.963.50 0.619.00 1.58

Not available

3.50 0.613.75 0.66

ELV project

10.25 1.793.50 0.61

ELV project5.75 1.01

9.25 1.62

Not available

JUNE 1991Universal battery charger 900134 8.00 1.40Logic analyser - 4- power supply board 900094-7 7.50 1.31- Atari interface 900094-6 10.75 1.88- IBM interlace 900094-1 12.25 2.14

Digital phase meter 910045-1/2/3 22.25 3.89(set of 3 PCBs)

Video ADC/DACLight transceiverVariable AC PSU

910009 Not availableUPBS-1 1.95 0.34900104 5.25 0.92

if,:10JECT No. Pr ice VAT(t) (El

Light switch vs. TV IR r/c 910048RTC for Atari ST 910006Stepper motor board -1 910054-1- PC bisection card

4.75 0.835.25 0.92

24.75 4.33

JULY/AUGUST 1991Modern LED clock 896139 Not availableMultifunction tr0 for PC5 910029 20.75 3.63

WW video digitizer 910053 19.25 3.37

8088 SBC PCB available from authorStepper motor board - 2 910054-2 24.25 4.23- power driver boardLaser - 3 ELV project

Electronic reversing cct 904098 Not availablefor model trainsLED voltmeter 914005 4.75 0.83

Wien bridge 914007 3.50 0.61Windscreen wipe`wash 914009 Not availableRemote temperature 914011 Not availablemodel for MAW

Car battery monitor 914014 Not availableAngled bus extension 914030 10.25 1.79card for PC51Mbit adaptor for EPROM 914035 Not availableprogrammerField strength meter 914041 Not availableS -meter for SW receivers 914050 Not availableSync separator 914077 3.75 0.66Video enhancement for 914051 Not availableArchimedes

September 1591Peak indicator for loud-speakersTimecode interface for stick control- main board 910055 20.75

- display board 87291.9a 3.50

ELV project

3.630.61

Asymm-symm converter 910072 4.75 0.83Opticalock 910091 Not available

October 1991PC -controlled weather 900124-2 3.25 0.57Station (2)

Digital function generator main board 910077-1 18.50 324- display board 910077-2 10.75 1.88

Audio spectrum shift 910105 8.75 1.53encoder/decoder

ELEKTOR ELECTRONICS OCTOBER 1991dlist of all PCBs, software liadods and front panets available thin:nigh the Readers Serviees is

ished in the March: May, September aro De:en-tar Issues el E:.-)Vic1 :5

64

SWITCHBOARDSwitchboard allows all PRIVATE READERSof Elektor Electronics one FREE advertise-ment of up to 108 characters. includingspaces, commas, numerals, etc.. per month.

Write the advertisement. which MUST re-late to electronics, in the coupon on thispage; it MUST INCLUDE a private telephonenumber or name and address: post officeboxes are NOT acceptable.

Elektor Electronics (Publishing) can notaccept responsibility for any correspondenceor transaction as a result of a free advertise-ment or of any inaccuracy in the text of suchan advertisement.

Advertisements will be placed in the orderin which they are received.

Elektor Electronics (Publishing) reservethe right to refuse advertisements without giv-ing reasons or without returning them.

WANTED. Lecson stereo circuit board layoutdiagram or non -working API, AMRL, IFER.Phone Cheltenham 242 185.

WANTED. Circuit and construction details onmoving message displays. Write to S. Syed,Flat B, 9 Shelgate Road, London SW11 1BD.

HELP. Get my EE Sat TV receiver working. Willpay all costs. Write to D. Weaver, 14A, QueensHeights, 16 Siu Wo Road, Shatin, Hong Kong.

CORRESPONDENCE. Electrical engineer withbasic practical knowledge would like tocorrespond with others anywhere. Please writeto Peter F. Murphy, 52 Ardkill Road, Ardmore,Londonderry, Northern Ireland BT47 3RH

SALE of components, software, books, hard-ware. PCBs. Send 30p A4 s.a.e. toV. Williamson. 21 Duchess Drive, Newmarket.Suffolk CB8 8AG

FOR SALE. Dartron dual -beam oscilloscope£100: selection of bakelite valve radios from £20or offers for lot. Phone Andy Mills on(0737) 552 693.

FOR SALE. Disk drives, dual 8", 500 k, casedwith PSU £30; 5114', 40T SS, cased with PSU£10; 3112, 720 k, £10. Phone 081 977 8305.

Send this coupon to: Elektor Electronics(Publishing), Down House, BroomhillRoad, LONDON SW18 4JQ.

Block capitals please - one character to each box

ELEKTOR ELECTRONICS 10/91

Name and address MUST be given

TERMS OF BUSINESSHOW TO PAYUnless you have an approved credit account with us, allorders must be accompanied by the full payment, includ-ing VAT(where applicable)and postage and packing charges

as stated overleaf in section 1.

Payment may be made by cheque, postal order, VISA,

ACCESS, MASTERCARD or EUROCARD (when pay-ing by credit card, the order must go to the cardholder'saddress). Do not send cash through the mail. Chequesand postal orders should be crossed and made payable to

ELEKTOR ELECTRONICS.Payment may also be made by direct transfer from a

private or business Giro account to our Giro AccountNo.331523801 by completi ng and sending to the National

Giro Centre, in a National Giro postage paid envelope, aNational Giro transfer/deposit form. Do not send Giro trans-

fers direct to us, as this will delay your order.If you live outside the UK, payment may also be

made by Bankers' sterling draft drawn on a London clear-

ing bank, Eurocheque, or US or Canadian dollar cheque.If you pay by Bankers' sterling draft, make clear to

the issuing bank that your full name and address MUSTbe communicated to the London clearing bank.

Our bankers are National Westminster Bank PLC,100 High Street, BRENTFORD TN\ S 8 kY. England.Our account number is 4285 0134

Eurocheques should be made out in pounds sterlingand have the holder's guarantee card number written onthe back. US and Canadian dollar cheques, as well asEurocheques, not made out in sterling are accepted at theexchange rate prevailing at the time your order is re-ceived: these cheques should be increased by the equiv-

alent of £5.00 to cover the bank's negotiating fee.

DELIVERYAlthough every effort will be made to dispatch your orderwithin 2-3 weeks from receipt of your instructions, wecan not guarantee this time scale for all orders.

RETURNSFaulty goods or goods sent in error may be returned forreplacement or correction, but not before obtaining ourconsent. All goods returned should be packed securely ina padded bag or box, enclosing a covering letter statingthe dispatch note number. If the goods are returned because

of a mistake on our pan, we will refund the return postage.

Goods returned for refund must be in resaleable condi-tion and will be subject to a 10% handling charge with aminimum charge of £2.50.

DAMAGED GOODSClaims fordamaged goods must be received at our London

office within 10 days (UK); 14 days (Europe) or 21 days(all other countries) from the date on our "RecordedDelivery" slip.

CANCELLED ORDERSAll cancelled orders will be subject to a 10% handlingcharge with a minimum charge of £2.50.

PATENTSPatent protection may exist in respect of circuits, de-vices, components, and so on, described in our books ormagazines. Elektor Electronics (Publishing) do not ac-

cept responsibility or liability for failing to identify suchpatent or other protection.

COPYRIGHTAll drawings, photographs, articles, printed -circuit boards,

EPROM, and cassettes published in our books or magazines

(other than in third -party advertisements) are copyrightand may not be reproduced or transmitted in any form orby any means, including photocopying and recording, inwhole or in part, without the prior permission of ElektorElectronics (Publishing) in writing. Such written permis-sion must also be obtained before any part of these pub-

lications is stored in a retrieval system of any nature.Nothwithstanding the above, printed -circuit boards

may be produced forprivate and personal use without prior

permission.

LIMITATION OF LIABILITYElektor Electronics (Publishing) shall not be liable incontract, tort, orotherwise, for any loss ordamage sufferedby the purchaser whatsoever or howsoever arising out of,

or in connexion with, the supply of goods or services byElektor Electronics (Publishing)other than to supply goodsas described or, at the option of Elektor Electronics(Publishing), to refund the purchaser any money paid inrespect of the goods.

LAWAny question relating to the supply of goods and servicesby Elektor Electronics (Publishing) shall be determinedin all respects by the laws of England.


Please mention ELEKTOR ELECTRONICS when contacting advertisers

[AUTONA LTDUK s leading module manufacturer since 1972

51K POPPY ROADPRINCES RISBOROUGHBUCKS HP17 9DBTEL: (084 44) 6326 FAX: (034 .14)7102Add VAT -E150 on all orders. Export add

Callus byappn.ntment

only

HST

MODULES AND EQUIPMENTFULLY BUILT AND TESTED For Projects & Applications in: COMPREHENSISVE INSTRUCTIONS

*AUDIO * SECURITY * INDUSTRIAL * DIGITAL VOLTMETERS** SECURITY **

i MINIATURE PASSIVE INFRA -RED SENSOR-RP33Swiichable Dual range, detects intruders up to 6 or 12 metres

This advanced sensor operates by detecting the bodyheat of an intruder moving within the detection field.

Slow ambient changes such as radiators, etc. areignored. Easily installed in a room or hallway. Providing

reliable operation from a 12V supply, it is ideal for usewith the CA 1382 or equivalent high quality control unit.

Size 80x 50x 40mm Supplied with full instructions..u ntity discounts start at 3 units

DIGITAL ULTRASONIC DETECTOR-US 5063Crystal controlled movement detection module operating

at 50kHz with an effective range up to 20ft Suitable foroperation in household or vehicle security systems. 12V

operation and built-in timing makes it suitable for awide range of applications.

Easily Installed

Onhi

£44.95

ADVANCED CONTROL UNIT-CA 1382

Automatic Loop Test & Switch On * Automatic SirenRe -Set * Audible Entry/Exit Warning Buzzer * Two

Separate Loop Inputs -24-hr Circuits *Easily Installed, Full Instructions Supplied.

This advanced control panel provides effective andreliable control for all security installations, yet its

operation is sheer simplicity for all members of thefamily, and is supplied with two keys. Housed in a steel

case with an attractive moulded front panel, it compareswith units costing twice the pricy;

LOW-COST CONTROL UNIT-CA 1250This tried and tested control unit provides the finest

value for money in control systems, with manythousands protecting houses all over the country.A suitable steel enclosure is available separately.

The unit offers the following features: Built-in electronicsiren, drives two loudspeakers incorporating exit &entry delays * Anti -tamper and panic facility * Screwconnector for ease of installation, etc. etc.

FULL RANGE OF SECURITY ACCESSORIES STOCKED PROVIDINGEVERYTHING YOU NEED TO PROTECT YOUR HOME

DIGITAL VOLTMETER MODULESDVM 456 HIGH PERFORMANCE ;94ensirt),*

31/2 DIGIT PANEL METER Input ImpedanceSupply VoltageDimensions

DVM 356 VERSATILE3 -DIGIT PANEL METER

...:Atio 11140

1-999VWithin at± 1 digit10041 ohm

8V -12V955x 55x 1 lmm

This exciting new module provides alarge, bright digital read-out with an

accuracy within 0.1%. It incorporates abuilt-in regulator which allows it to be

used from an unregulated supply ofbetween 8V -12V. Full over -load

protection is included and the unit issupplied with a mounting bezel and

filter, together with full applicationinstructions showing how to extend its

range and measure resistance, currenand temperature

DTIO TEMPERATURE MEASUREMENT MODULEcs:rnp!e.tr.z_:;:n ett.-,:0we module whith when

onstructed prondes a linear output at lOrnV(Cover the temperature range -10 to -100C.

£3.95 VATPS209 DUAL POWER SUPPLY

Fully built mains pmter supp:iC .rtputs of up to 250mA each. s'z -se h :

erther DM modules and other egg; 4-nent£6.65 + VAT

The DVM 356 is a low-cost module offering3 -digit performance with an FSD of -1- 999mV and-9 9mV. Supplied with acomprehensive Data Sheet.No bezel available

ATTRACTIVE DISCOUNTSAVAILABLE FOR QUANTITY USERS

** AUDIO **AL 12580-125W POWER AMPLIFIERA rugged, high powered module that is ideal foruse in discos & P.A. Systems where powers of upto 125W, 4 ohms are required. The heavy dutyoutput transistors ensure stable and reliableperformance. It is currently supplied to a largenumber of equipment manufacturers wherereliability and performance are the mainconsiderations, whilst for others its low price is themajor factor. Operating from a supply voltage of a40-80V into loads from 4-16 ohms.

AL 5070-ULTRA LOW DISTORTION50W AMPLIFIERProvides sound reproduction of the highest quality withdistortion levels below 0.02%. this module offerssuperlative performance in all types of audio equipment_Full over -load protection is incorporated ensuringreliability of the highest order. Supplied with its ownheat sink, it opertes from a 40V -65V supply rail intoloads of 8-16 ohms.

£17.19VAT

AL 2550-COMPACT LOW-COST 25W AMPLIFIEROne of our most popular audio modules with tens ofthousands installed. Ideal for domestic applicationswhere low distortion and compact size arethe prime requirements. Used with supplyrails of 20V -50V into loads of 8-15 ohms.

£6.55VAT

AL 1030-RUGGED lOW AMPLIFIERThis low cost unit provides a powerful 10W Output itideal for all medium power applications requiring q gistreproduction with rugged performance. Repre-senting excellent value for money it operatesfrom a supply of 18V -30V into loads 018-16 ohms.

MM 100-BUDGET 3 -INPUT MIXERWith a host of features including 3 individual level controls, a master volume andseparate bass and treble control, it provides for inputs for microphone.magnetic pick-up and tape, or second pick -u (selectable), and yet costsconsiderably less than competitive units.This module is ideal for discos and publicaddress units and operates from 45V -70V.

MG 100GAs MI1100 gitar +1 microphoneinpit intended ; _ 'Jr anckber appEcatixo.

50Fr INFRA -RED BEAM -1R1470The 1111470 consists of a separate transmitterand receiver providing a beam of up to 50ftwhich, when interrupted, operates a relay in the receiver whichin turn may be used to control external equipment The systemrequires only 65mA from a 12V supply. Size: (each unit) 82 x 52 x 57mm

TIMER SWITCH & POWER SUPPLY-DP3570Tne 0P3570 consists of an adjustable timer switch and 12Vstabilised power supply designed to provide switching of loadsup to 4A at 240V A.C. for a preset time between10 secs and 6 mins, the timed period being initia-ted by the normally open or normally closed inputs

£17.49

GENERAL PURPOSE ULTRASONIC AMR.MOVEMENT DETECTOR US4012This module uses ultrasonic techniques to detect movement atdistances up to 5 metres with an operating range of 60'. Supplyvoltage 10-14V (12mA). Size: 147 x 52.5 x 15mm.

STABILISED SUPPLY & SWITCHING UNIT-PS1265The PS1265 provides stabilised 12V output for current levels up to 700rnA. Addition-

ally it incorporates a highimpedance input for switchingloads up to 1kW at 240Vwithout timing

£12.95


rlect · 2019. 7. 18. · 330 12 scsi 5.0m 3.5- 240k £999 425 12 scsi 5.0m 3.5 240k £1179 520 12...

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