Optics Communications 283 (2010) 396–399

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Optics Communications

journal homepage: www.elsevier .com/ locate/optcom

Photonic instantaneous measurement of microwave frequency using fiberBragg grating

Ze Li, Bo Yang, Hao Chi *, Xianmin Zhang, Shilie Zheng, Xiaofeng JinDepartment of Information and Electronic Engineering, Zhejiang University, Hangzhou 310027, China

a r t i c l e i n f o a b s t r a c t

Article history:Received 6 July 2009Received in revised form 16 October 2009Accepted 16 October 2009

Keywords:Microwave photonicsMicrowave frequency measurementFiber Bragg grating (FBG)

0030-4018/$ - see front matter � 2009 Elsevier B.V. Adoi:10.1016/j.optcom.2009.10.036

* Corresponding author. Tel.: +86 571 87176827.E-mail address: chihao@zju.edu.cn (H. Chi).

A photonic approach to realizing instantaneous measurement of microwave frequency based on opticalmonitoring using a fiber Bragg grating (FBG) is proposed and demonstrated. In the approach, a frequency-unknown microwave signal is modulated on an optical carrier in a Mach–Zehnder modulator biased atthe minimum transmission point. After detecting the transmission and reflection optical powers at theoutput of the FBG, the microwave frequency can be determined according to the value of transmis-sion-to-reflection power ratio, due to the fixed relationship between the microwave frequency and thepower ratio. A proof-of-concept experiment has been performed, which demonstrates that a measure-ment resolution of ±0.08 GHz over a 10 GHz measurement bandwidth is achieved. The measurement per-formance in terms of resolution is better than previously reported results.

� 2009 Elsevier B.V. All rights reserved.

1. Introduction

Due to the advantages of wide bandwidth, light weight, low lossand immunity to electromagnetic interference (EMI), the genera-tion, distribution, and processing of microwave signals using pho-tonic methods have attracted great interest for more than onedecade [1–3]. In modern radar systems and other military applica-tions, it is required for a microwave receiver to estimate the fre-quency of a received unknown microwave signal over a largebandwidth. The conventional electronic techniques for the micro-wave frequency measurement are thought to be slow, bulky, lim-ited in bandwidth and vulnerable to EMI. Recently, photonicapproaches for microwave frequency measurement have beenintensively investigated [4–13].

Several photonic scanning receivers and channelizers for fre-quency measurement were proposed for the microwave frequencymeasurement [4–7]. A Fabry-Pérot (F–P) etalon-based temporalscanning receiver system was proposed in [4]. A high-resolutionfree-space optical diffraction grating-based channelizer was dem-onstrated in [5]. In [6], an array of ultra-narrow phase-shifted grat-ings was employed as wideband microwave channelized receiver.It was demonstrated that microwave channelizer could also berealized using an integrated optical Bragg-grating F–P togetherwith an integrated hybrid Fresnel lens system [7]. Recently, a num-

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ber of photonic techniques for instantaneous microwave frequencymeasurement have been proposed and demonstrated [8–13]. In[8–12], the microwave frequency was estimated according to themapping relationship between the frequency and the ratio of twodispersion-induced RF power-fading functions. The RF power fad-ing can be easily realized in an intensity-modulated or phase-mod-ulated dispersive radio over fiber (RoF) link. The limitation wasthat the expensive high-speed photodiodes (PD) should be em-ployed in the approach based on RF power monitoring and com-parison. Another approach was based on optical powermonitoring and comparison [13], which can be realized using alow-frequency PD. The use of low-frequency PD reduces the sys-tem cost. However, the approach in [13] needs two laser sourcesand the employed Sagnac-loop optical filter in the approach hasstability problem, which inevitably leads to a relatively low mea-surement resolution.

In this paper, we propose and demonstrate a novel approach tothe instantaneous measurement of microwave frequency with ahigher resolution based on optical power monitoring, in which awell-packaged athermal fiber Bragg grating (FBG) is employed.Thanks to the complementary nature of the FBG’s transmissionand reflection spectrum, only one laser source is needed and afixed relationship between the microwave frequency and the opti-cal power ratio can be obtained. Since the optical power, not the RFpower, is measured, low cost PD with low-frequency response canbe employed in the approach. In addition, the achieved measure-ment resolution is much higher than previous approach due tothe use of a well-packaged athermal FBG. It is experimentally dem-onstrated that measurement error less than ±0.08 GHz over abandwidth larger than 10 GHz is realized.

Z. Li et al. / Optics Communications 283 (2010) 396–399 397

2. Operation principle

The schematic diagram of the proposed system is shown inFig. 1. A light wave from a laser diode (LD) is sent to an MZMthrough a polarization controller (PC). A frequency-unknownmicrowave signal is applied to the MZM via the RF-in port. TheMZM is biased at the minimum transmission point in order to sup-press the optical carrier. The modulated optical signal is sent to anFBG via an optical circulator. The wavelength of the LD is alignedwith the center wavelength of the FBG. The transmitted and the re-flected signals are monitored by two low-frequency PDs. The dccurrents from the two PDs, which are proportional to the transmit-ted and reflected average optical powers, are sent to a processingunit to acquire their power ratio. RF frequency can be estimatedbased on the power ratio value.

Let us denote the output optical power of the LD as P0. As thepower of the carrier-suppressed signal is equal to the power ofthe two first-order sidebands, the transmission and reflection pow-ers can be expressed as

PT ¼ P0J21ðbÞTðfmÞ ¼ P0J2

1ðbÞ½1� RðfmÞ� ð1ÞPR ¼ P0J2

1ðbÞRðfmÞ ð2Þ

where b = pVm/Vp is the modulation index, Vm is the amplitude volt-age of the microwave signal, Vp is the half-wave voltage of theMZM, J1(�) denotes the first-order Bessel function of the first kind,fm is the microwave frequency to be measured, and T(fm) andR(fm) are respectively the transmission response and the reflectionresponse of the FBG. According to (1) and (2), we then obtain thepower ratio of the transmitted and reflected signals

r ¼ PT

PR¼ TðfmÞ

RðfmÞ¼ 1� RðfmÞ

RðfmÞ: ð3Þ

Notice that the power ratio r is independent of the modulationindex or the input RF power. In [13], the transmission and reflec-tion responses of the filter were all sinusoidal. In fact, it can alsobe any other symmetric function. To demonstrate clearly the prin-

Fig. 1. Schematic diagram of the proposed system for microwave frequencymeasurement.

Fig. 2. Simulation results of the Gaussian-shaped FBG example. (a) Dependence of the dNormalized frequency response of the FBG.

ciple of the frequency measurement, we give a simulation result asshown in Fig. 2. In the simulation, we assume the reflection spec-trum of the applied grating is Gaussian-shaped. Fig. 2a gives thedependence of the transmission power PT, the reflection power PR

and the power ratio r on the microwave frequency fm based on(1)–(3); and the assumed reflection and transmission spectra areshown in Fig. 2b. As can be seen, for a given FBG spectrum, themicrowave frequency can be deduced from the power ratio valueunambiguously.

3. Experimental results

A proof-of-concept experiment with the setup as in Fig. 1 hasbeen implemented. A well-packaged athermal FBG with its centerwavelength at 1549.17 nm and bandwidth �0.18 nm is applied.The transmission and reflection spectra of the FBG are shown inFig. 3b. The wavelength of the tunable laser source (ILX lightwave)is tuned to match the center wavelength of the FBG, with the out-put power fixed at 16 dBm. The lightwave is modulated by a micro-wave signal from a signal generator (Agilent E8267C) at a MZM(EOSPACE) which is biased at the minimum transmission point.The optical powers are recorded by an optical power meter (Agi-lent 81635A). Both the signal generator and the power meter arecontrolled by a computer through a GPIB interface.

Firstly, we record four groups of optical power data and use theaverage value to obtain the power ratio function. This can beunderstood as the calibration process. The RF power is set as8 dBm. This average power ratio function is stored in a look-up ta-ble which will be used to determine the microwave frequency inthe following experiment. Fig. 3a gives the averaged transmis-sion/reflection optical power vs. microwave frequency and the cal-ibrated power ratio function.

We then use the calibrated power ratio function to measure themicrowave frequencies. Two groups of data at the RF power 8 and4 dBm are recorded. The experimental results are shown in Fig. 4.The measured frequencies vs. input RF frequencies are given inFig. 4a and the corresponding measurement errors are presentedin Fig. 4b (‘‘8 dBm-1” and ‘‘4 dBm”). It is shown that the measure-ment error less than 0.07 GHz over a bandwidth from 1 to 10 GHzis achieved. If a smaller measurement bandwidth is considered, forexample, 6–10 GHz, the measurement resolution can be improvedto ±0.05 GHz. Note that the frequency measurement performancerealized here in terms of measurement resolution is much betterthan previously published results as in [13] due to the applicationof stable filter. In order to observe the stability of the frequencymeasurement system, we test the system after 2 h. The resultsare shown as the data 8 dBm-2 in Fig. 4a and b. It is shown that

etected optical power and the power ratio on the input microwave frequency. (b)

0

0.05

0.1

0.15

0 2 4 6 8 100

3

6

9

12

PR

PT

Frequency (GHz)

(a)

Opt

ical

Pow

er (

mW

)

Pow

er r

atio

1549 1549.1 1549.2 1549.3Wavelength (nm)

Opt

ical

Pow

er (

dBm

)

Reflection

Transmission

(b)

-62

-58

-54

-50

-46

Fig. 3. (a) The averaged transmission and reflection optical power and the calibrated power ratio function. (b) The transmission and reflection spectra of the applied FBG.

-0.1

-0.05

0

0.05

0.1

0 2 4 6 8 10 12

8dBm-1

RF power

4dBm

8dBm-2

Frequency (GHz)

(b)

Mea

sure

men

t E

rror

(G

Hz)

0

2

4

6

8

10

12

0 2 4 6 8 10 12

Mea

sure

d F

requ

ency

(G

Hz)

8dBm-1

RF power

4dBm

8dBm-2

(a)

Frequency (GHz)

Fig. 4. (a) Estimated frequencies based on measured power ratio function. (b) Measurement errors for the microwave signals at different power levels and different time.

398 Z. Li et al. / Optics Communications 283 (2010) 396–399

the measurement resolution and bandwidth are almost kept un-changed as the measurement errors at 2 h later can still maintainas ±0.08 GHz. From Fig. 4b, we can see that if the measurementis performed immediately after the calibration and the input RFpower is kept unchanged, the error will be kept in a small range,say, ±0.05 GHz (‘‘8 dBm-1”). But if the measurement condition ischanged, for example, the input RF power is changed to 4 dBm,or there is wavelength drift in the FBG (‘‘8 dBm-2”), the errors willbe larger. Since the measurement errors in low frequencies are lar-ger than those in high frequencies in the data of ‘‘8 dBm-2”, we canconclude that the influence of the wavelength drift is smaller in thehigh frequencies than in the low frequencies. It can be explained bythe relatively flat power ratio curve in the low frequencies asshown in Fig. 3a. Note that the resolution realized here is accept-able in practical applications since an estimation of the microwavefrequency realized by this type of receiver can effectively reducethe complexity in other specialized receivers.

4. Discussions

There are three main error sources which influence the mea-surement resolution in the proposed system. The first comes fromthe wavelength drift of the FBG and LD. The second is the incom-plete carrier suppression and the bias drift of the MZM. The lastis the generation of higher order harmonics in the modulation pro-cess. The last two factors determine the sensitivity and dynamicrange of the system.

According to our previous measurements and records, the resid-ual wavelength drift of the applied FBG is around 1 pm/�C. As we

know, an FBG without temperature compensation is sensitive toenvironmental temperature change. The temperature compensa-tion technique using material with a negative thermal expansioncoefficient can largely reduce the temperature sensitivity of theFBG [14].

We have analyzed the measurement error induced by the wave-length drift of the applied FBG by computer simulation in whichthe reflection and transmission spectra shown in Fig. 3b are ap-plied. The simulation results are shown in Fig. 5. It is evident thatthe influence of wavelength drift in the high frequencies is smallerthan that in the low frequencies, which agrees well with Fig. 4b. Ifthe temperature change is large in applications, say �50 �C to+85 �C, a temperature-control mechanism should be introducedto stabilize the temperature of the FBG according to the practicalrequirement. For example, if a ±0.2 GHz error tolerance is required,the temperature change of the FBG should be controlled within±8 �C. The case of the LD wavelength drift induced measurementerror is similar to the case of the FBG wavelength drift. For high-quality LD, there is a thermoelectric cooling module for tempera-ture control. Usually, the temperature change of an LD can be con-trolled within ±0.5 �C and the wavelength drift is typically within1–2 pm in an environment with the temperature change from�40 to 70 �C. Therefore, it can be estimated that the measurementerror due to the wavelength drift of the LD is within ±0.02 GHz.

Next, we analyze the dynamic range of the measurement sys-tem. Since the modulation nonlinearity in an MZM leads to thegeneration of higher order harmonics which results in the mea-surement error, the maximum RF power that can be measured un-der a given measurement error tolerance depends on the higherorder harmonics generated in the modulation process. On the

3 4 5 6 7 8 9-0.2

-0.1

0

0.1

0.2

0.3

0.4

RF Frequency (GHz)

Mea

sure

men

t Err

or (

GH

z)

Wavelength Drift

12 pm

8 pm

4 pm

Fig. 5. Frequency measurement error induced by the wavelength drift of the FBG.

605550454035302510

15

20

25

30

35

40

45

Extinction Ratio (dB)

Dyn

amic

Ran

ge (

dB)

Error Tolerance

± 0.2GHz

± 0.1GHz

Fig. 6. Dynamic range of the measurement system.

Z. Li et al. / Optics Communications 283 (2010) 396–399 399

other hand, the minimum RF power that can be measured dependson the optical noise and the incomplete carrier suppression. In thesystem, the optical signal-to-noise ratio (OSNR) is determined bythe relative intensity noise (RIN) of the LD since there is no opticalamplification. A typical RIN is around �165 dB/Hz. The system’sOSNR is around 65 dB (in a 10 GHz optical bandwidth), which ismuch larger than the extinction ratio of a typical MZM (40 dB).Therefore, the residual carrier power is much larger than the opti-cal noise. In other words, the minimum measurable RF power un-der a given measurement error tolerance is mainly limited by theincomplete carrier suppression of the MZM.

Here, we quantitatively analyze the dynamic range of the mea-surement system by computer simulation. In the analysis, it is as-sumed that the half-wave voltage of the MZM is 6 V. According tothe reflection and transmission spectra of the applied FBG, we cancalculate that the maximum measureable input RF power inducedby the modulation nonlinearity is 20.2 dBm under a ±0.2 GHz errortolerance and 18.9 dBm under a ±0.1 GHz error tolerance. Then, wecalculate the minimum input RF power under the error tolerance of±0.2 GHz and ±0.1 GHz in the case of different carrier suppressionwith the extinction ratio from 60 to 25 dB. The simulated dynamicrange performance is given in Fig. 6. It is seen that in the case of60 dB carrier suppression and ±0.2 GHz error tolerance, the dy-namic range is around 43 dB. We find that a poorer carrier suppres-sion would lead to a smaller dynamic range, and a larger error

tolerance would lead to a higher dynamic range. For a typicalMZM with 40 dB extinction ratio, the dynamic range is around25 dB in the case of ±0.2 GHz error tolerance, which is not adequatefor a high-performance receiver in practical applications. Note thatthe MZM with over 60 dB extinction ratio is commercially avail-able today, which can effectively improve the dynamic range. Inaddition, a wideband low noise microwave preamplifier can beconsidered to largely improve the dynamic range [4].

As the bias drift of the MZM would lead to a poorer carrier sup-pression and thus a larger measurement error or a smaller dynamicrange, we have developed a single chip microcomputer-based cir-cuit with feedback mechanism to keep the MZM working at theminimum power transmission point.

In practice, pulse-modulated RF and multiple frequencies mea-surement are usually required. In [15], simultaneous microwavefrequencies measurement has been realized based on frequency-to-time mapping method in optical domain. Since our system isdesigned for the instantaneous frequency measurement whereonly a single frequency is measured in a short time, it is not appro-priate for the pulse-modulated RF or multiple frequencies mea-surement. New mechanisms should be included in the system todeal with the pulsed RF signal or multiple frequenciesmeasurement.

5. Conclusion

An approach to realizing instantaneous measurement of micro-wave frequency based on optical power monitoring and compari-son was proposed and experimentally demonstrated. The keycomponent in the system was a well-packaged athermal FBG,which leads to a higher measurement resolution. A proof-of-con-cept experiment has been implemented, which shows a measure-ment resolution better than 0.08 GHz over a 10 GHz bandwidth.The error sources and dynamic range of the system are analyzed.The realized measurement results in terms of measurement reso-lution are better than previously published results. The proposedapproach shows a good prospect in military applications.

Acknowledgements

This work was supported by the National Natural Science Foun-dation of China (Nos. 60871011 and 60801003) and the ZhejiangProvincial Natural Science Foundation of China (No. Y1080184).X. Zhang was supported by the Program for New Century ExcellentTalents in University (No. NCET-05-518).

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