logic gates 1 - ntut.edu.twdkao/chap03.pdf · 2020. 12. 4. · static complementary logic gate...
TRANSCRIPT
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VLSI Design : Chapter 5-1 1
Schedule
11. 11/27/20 Chapter 2 (CMP, Design Rule )
12. 12/04/20 Chapter 3 (Logic Gates, Noise Margin)
13. 12/11/20 Chapter 3 (Power, fan-out and loading, timing )
14. 12/18/20 Quiz 2, Chapter 4 (Simulation, Cross Talk)
15. 12/25/20 Chapter 4 (ATPG & DFT)
16. 01/01/21元旦放假
17. 01/08/21 Final Examination
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VLSI Design : Chapter 5-1 2
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VLSI Design : Chapter 5-1 3
Logic Level (Gate Level)
MOS Level
Layout
Cross-section
Process
Equation
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VLSI Design: Chapter 3-1 4
Chapter 3: Logic Gates
Combinational logic functions
Static complementary logic gate structures
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VLSI Design: Chapter 3-1 5
Combinational logic expressions
Combinational logic: function value (outputs)
is a combination of inputs (and inputs only).
A logic gate implements a particular logic
function.
Both specification (logic equations) and
implementation (logic gate networks) are
written in Boolean logic.
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VLSI Design: Chapter 3-1 6
Gate design
Why designing gates for logic functions is
non-trivial:
may not have logic gates in the library for all
logic expressions;
a logic expression may map into gates that
consume more area, delay, or power.
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VLSI Design: Chapter 3-1 7
Boolean algebra terminology
Function:
f = a’b + ab’
a is a variable; ab’ is a term.
A function is irredundant if no term can be
removed without changing its truth value.
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VLSI Design: Chapter 3-1 8
Static complementary gates
Complementary: have complementary pullup
(p-type) and pulldown (n-type) networks.
Static: do not rely on stored charge.
Simple, effective, reliable; hence ubiquitous.
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VLSI Design: Chapter 3-1 9
Examples (1)
f = (a+b) * (a+b’)
. = a*a + a *b’ + b * a + b * b’
= a + a *b’ + b * a
= a + a (b’ + b )
= a + a
= a
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VLSI Design: Chapter 3-1 10
Examples (2 & 3)
f = a*d + a *e + b * d + b * e+ c*d + c *e
= (a+b+c) * (d+e)
f = a*c*e + a*d*e + b*c*e + b*d*e +
a*c*f + a*d*f + b*c*f + b*d*f
= (a*c + a*d + b*c + b* d) * (e+f)
= (a+b) * (c+d) * (e+f)
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VLSI Design: Chapter 3-1 11
Examples (4)
f = a + b*a’ + a’ *b’
. = a + a’ * (b + b’)
= a + a’
= 1
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VLSI Design: Chapter 3-1 12
Completeness
A set of functions f1, f2, ... is complete iff every Boolean function can be generated by a combination of the functions.
NAND is a complete set; NOR is a complete set; {AND, OR} is not complete.
Transmission gates are not complete.
If your set of logic gates is not complete, you can’t design arbitrary logic.
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VLSI Design: Chapter 3-1 13
Static complementary gate
structure
Pullup and pulldown networks:
pullup
network
pulldown
network
VDD
VSS
outinputs
p trs
n trs
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VLSI Design: Chapter 3-1 14
Inverter
a out
0
1
1
0
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VLSI Design: Chapter 3-1 15
Inverter Layout
VDD
GND
NMOS (2/.24 = 8/1)
PMOS (4/.24 = 16/1)
metal2
metal1polysilicon
InOut
metal1-poly via
metal2-metal1 via
metal1-diff via
pdiff
ndiff
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VLSI Design: Chapter 3-1 16
Inverter
VDD
Rn
Vout = GND
Vin = V DD
VDD
Rp
Vout = Vdd
Vin = 0
VOL = 0
VOH = VDD
VM = f(Rn, Rp)
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VLSI Design: Chapter 3-1 17
0
0.5
1
1.5
2
2.5
0 0.5 1 1.5 2 2.5VDS (V)
X 10-4
VGS = 1.0V
VGS = 1.5V
VGS = 2.0V
VGS = 2.5V
NMOS transistor, 0.25um, Ld = 0.25um, W/L = 1.5, VDD = 2.5V, VT = 0.4V
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VLSI Design: Chapter 3-1 18
Inverter Layout
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VLSI Design: Chapter 3-1 19
NAND gate
a b out
0 0 1
0 1 1
1 0 1
1 1 0
* 串+ 並
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VLSI Design: Chapter 3-1 20
NOR gate
a b out
0 0 1
0 1 0
1 0 0
1 1 0
* 串+ 並
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VLSI Design: Chapter 3-1 21
Layout of a NOR Gate
P-diffN-diffPolyM1N-wellP-subVia
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VLSI Design: Chapter 3-1 22
NOR & NAND
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VLSI Design: Chapter 3-1 23
AOI/OAI gates
AOI = and/or/inverter; OAI = or/and/inverter.
Implement larger functions.
Pullup and pulldown networks are compact:
smaller area, higher speed than
NAND/NOR network equivalents.
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VLSI Design: Chapter 3-1 24
AOI example
out = [ab+c]’:
symbol circuit
and
or
invert
* 串+ 並
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VLSI Design: Chapter 3-1 25
Pullup/pulldown dual network
Pullup and pulldown networks are duals.
To design one gate, first design one network,
then compute dual to get other network.
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VLSI Design: Chapter 3-1 26
O = A • B • C • D
* 串+ 並
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VLSI Design: Chapter 3-1 27
4-input NAND
DCBA
D
C
B
A CL
C3
C2
C1
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VLSI Design: Chapter 3-1 28
O = D + A • (B + C)
Try this one
* 串+ 並
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VLSI Design: Chapter 3-1 29
Try this one
O = D + A • (B + C)
D
A
B C
D
A
B
C
O
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VLSI Design: Chapter 3-1 30
Try again
z = c • (a + b)
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VLSI Design: Chapter 3-1 31
Try again
c
c
z
a b
a
bz = c • (a + b)
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VLSI Design: Chapter 3-1 32
Gate to Logic K-Map
Truth table Karnaugh Map
X Y Output
0 0 1
0 1 0
1 0 0
1 1 0
Y
0 1
X 1 xy’ xy
0 x’y’ x’y1
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VLSI Design: Chapter 3-1 33
X Y Z Output
0 0 0 0
0 0 1 1
0 1 0 0
0 1 1 0
1 0 0 0
1 0 1 1
1 1 0 1
1 1 1 1
K-Map
xyz’ + xyz + x’y’z + xy’z
= (xyz’ + xyz) + (x’y’z + xy’z) + (xy’z + xyz)
= (xy (z’ + z)) + (y’z (x’ + x)) + (xz (y’ + y))
= (xy 1) + (y’z 1) + (xz 1)
= xy + y’z + xz
YZ
00 01 11 10
X 1 1 1 1
0 1
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VLSI Design: Chapter 3-1 34
X Y Z Output
0 0 0 0
0 0 1 1
0 1 0 0
0 1 1 0
1 0 0 0
1 0 1 1
1 1 0 1
1 1 1 1
K-Map
xyz’ + xyz + x’y’z + xy’z
= (xyz’ + xyz) + (x’y’z + xy’z) + (xy’z + xyz)
= (xy (z’ + z)) + (y’z (x’ + x)) + (xz (y’ + y))
= (xy 1) + (y’z 1) + (xz 1)
= xy + y’z + xz= xy + y’z
YZ
00 01 11 10
X 1 1 1 1
0 1
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VLSI Design: Chapter 3-1 35
W X Y Z Output
0 0 0 0 1
0 0 0 1 0
0 0 1 0 0
0 0 1 1 X
0 1 0 0 1
0 1 0 1 0
0 1 1 0 0
0 1 1 1 X
1 0 0 0 1
1 0 0 1 1
1 0 1 0 X
1 0 1 1 1
1 1 0 0 1
1 1 0 1 X
1 1 1 0 1
1 1 1 1 1
K-Map (1)
Statements…… (not equation)
= w + y’z’
YZ
00 01 11 10
WX 00 1 X 0
01 1 0 X
11 1 X 1 1
10 1 1 1 X
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VLSI Design: Chapter 3-1 36
W X Y Z Output
0 0 0 0 1
0 0 0 1 0
0 0 1 0 1
0 0 1 1 X
0 1 0 0 1
0 1 0 1 1
0 1 1 0 0
0 1 1 1 X
1 0 0 0 1
1 0 0 1 0
1 0 1 0 X
1 0 1 1 0
1 1 0 0 1
1 1 0 1 X
1 1 1 0 1
1 1 1 1 1
K-Map(2)
Statements…… (not equation)
= xz + wx+ x’z’ +y’z’
= (x + z’) + wx + y’z’
YZ
00 01 11 10
WX 00 1 X 1
01 1 1 X
11 1 X 1 1
10 1 X
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VLSI Design: Chapter 3-1 37
W X Y Z Output
0 0 0 0 1
0 0 0 1 0
0 0 1 0 1
0 0 1 1 X
0 1 0 0 1
0 1 0 1 1
0 1 1 0 0
0 1 1 1 X
1 0 0 0 1
1 0 0 1 0
1 0 1 0 X
1 0 1 1 0
1 1 0 0 1
1 1 0 1 X
1 1 1 0 1
1 1 1 1 1
K-Map(3)
Statements…… (not equation)
=
= x y’ + wx + x’z’
YZ
00 01 11 10
WX 00 1 X 1
01 1 1 X
11 1 X 1 1
10 1 X
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VLSI Design: Chapter 3-1 38
Chapter 3: Logic Gate
Electrical properties of static combinational
gates.
Effects of parasitics on gate.
Driving large loads.
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VLSI Design: Chapter 3-1 39
Transfer characteristics
Transfer curve shows static input/output
relationship—hold input voltage, measure
output voltage.
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VLSI Design: Chapter 3-1 40
O = (D + A) • (B + C)
Try this one
* 串+ 並
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VLSI Design: Chapter 3-1 41
Inverter transfer curve
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VLSI Design: Chapter 3-1 42
Logic thresholds
Choose threshold voltages between points where slope of transfer curve = -1.
Inverter has a high gain between VIL and VIH
points, low gain at outer regions of transfer curve.
Note that logic 0 and 1 regions are not equal sized—in this case, high pullup resistance leads to smaller logic 1 range.
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VLSI Design: Chapter 3-1 43
Noise margin
Noise margin = voltage difference between
output of one gate and input of next. Noise
must exceed noise margin to make second
gate produce wrong output.
In static gates, voltages are VOH = VDD and
VOL = VSS, so noise margins are VDD-VIH
and VIL-VSS.
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VLSI Design: Chapter 3-1 44
Logic levels
Solid logic 0/1 defined by VSS/VDD.
Inner bounds of logic values VL/VH are not
directly determined by circuit properties, as
in some other logic families.
logic 1
logic 0
VDD
VSS
VH
VL
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VLSI Design: Chapter 3-1 45
Inverter transfer curve
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VLSI Design: Chapter 3-1 46
Schedule
12. 12/04/20 Chapter 3 (Logic Gates, Noise Margin)
13. 12/11/20 Chapter 3 (Power, fan-out and loading, timing )
14. 12/18/20 Quiz 2, Chapter 4 (Simulation, Cross Talk)
15. 12/25/20 Chapter 4 (ATPG & DFT)
16. 01/01/21元旦放假
17. 01/08/21 Final Examination
18. 01/15/21 Exam review and FinFet
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VLSI Design: Chapter 3-1 47
Logic Level (Gate Level)
MOS Level
Layout
Cross-section
Process
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VLSI Design: Chapter 3-1 48
Power Domains
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VLSI Design: Chapter 3-1 49
Logic level shifter
Levels at output of one gate must be sufficient
to drive next gate.
Vdd1 Vdd2
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VLSI Design: Chapter 3-1 50
ShifterVdd1
Vdd2
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VLSI Design: Chapter 3-1 51
Power consumption analysis
Total Power = Pstatic + Pdynamic
Pdynamic = Pswitching+ Pshort
Most of power consumption comes from
switching behavior.
Static power dissipation comes from leakage
currents.
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VLSI Design: Chapter 3-1 52
Dynamic power consumption (1)
Switching Power = IV
= (Q/t) V = (CV/t) V
= C V2 f
Dynamic power consumption is independent of the
p, n transistors’ size
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VLSI Design: Chapter 3-1 53
Dynamic power consumption (2)
Short Circuit Power = tsc VDD I f
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VLSI Design: Chapter 3-1 54
Static power consumption
Short Circuit Power = Isleakage VDD
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VLSI Design: Chapter 3-1 55
Power Breakout List
250nm
Switching: 75%
Short: 20%
Leakage: 5%
90nm
Switching: 32%
Short: 10%
Leakage: 58%
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VLSI Design: Chapter 3-1 56
Leakage Increased
0.16 0.17 0.18 0.19 0.20
30
40
50
60
70
80
90L
eakag
e C
urr
en
t (p
A)
Drawn Gate Length (um)
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VLSI Design: Chapter 3-1 57
Delay
Assume ideal input (step), RC load.
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VLSI Design: Chapter 3-1 58
Delay assumptions
Assume that only one transistor group (p or n)
is on at a time. This gives two cases:
rise time, p on, n off;
fall time, p off, n on.
Assume resistor model for transistor.
Ignores saturation region,
but results are acceptable!!
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VLSI Design: Chapter 3-1 59
Inverter delay circuit
Load is resistor + capacitor, driver is resistor.
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VLSI Design: Chapter 3-1 60
Inverter delay
Vout(t) = VDD(1-e-t /(Rn+RL) CL)
t1 = 0.9 VDD; t2 = 0.1 VDD; tf = t2 - t1 ;
tf = 2.2 (Rn + RL) CL
For pullup time, use p transistor(s) resistance.
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VLSI Design: Chapter 3-1 61
Quality of RC approximation
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VLSI Design: Chapter 3-1 62
Parasitics and performance
b
a
c
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VLSI Design: Chapter 3-1 63
Driving large loads
Sometimes, large loads must be driven:
off-chip;
long wires on-chip;
large fanout number.
source
sink
sink
sinkFanout = 3
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VLSI Design: Chapter 3-1 64
Linear model
2 4 6 8 10 12 14 16
tpNOR2
t p(p
sec)
eff. fan-out
All gates
have the
same drive
current.
tpNAND2
tpINV
Slope is a
function of
“driving
strength”
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VLSI Design: Chapter 3-1 65
Effect of parasitics
Resistance slows down static gates, may cause
function failure.
Increase transistor’s size to increase driving,
but also increase input capacitance which
reduces input slope.
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VLSI Design: Chapter 3-1 66
Parasitics and performance
VDDVDD
VinVout
M1
M2
M3
M4Cdb2
Cdb1
Cgd12
Cw
Cg4
Cg3
Vout2
Fanout
Interconnect
VoutVin
CL
SimplifiedModel
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VLSI Design: Chapter 3-1 67
Chapter 3: Logic Gates
Delay
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VLSI Design: Chapter 3-1 68
Buffer Sizing
Sizing up the driver transistors only pushes
back the problem—large driver presents
larger capacitance to earlier stage.
Use a series of buffers (inverters)
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VLSI Design: Chapter 3-1 69
Buffer Sizing 2
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VLSI Design: Chapter 3-1 70
Buffer sizing
Use a chain of inverters, each stage has
transistors larger than previous stage.
Optimal number of stage nopt = ln (Cbig/Cg).
nopt must be an even number (for inverter)
Driver sizes increased exponentially.
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VLSI Design: Chapter 3-1 71
Buffer sizing
Narrower Width
=
Lower current through channel
Length
Width
GATE
W
L
L
Width (W)
Wider Width
=
Higher current through channel
GATE
Length
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VLSI Design: Chapter 3-1 72
Buffer sizing (N-MOS)
“1X” NMOS (W/L = 6)
GND
OUT
L = 0.25 um
W = 1.5 um
IN
0.25 um
GND
3 um OUT
IN
“2X” NMOS (W/L = 12)
1.5 um
GND
0.25 um
OUT
IN
“2X” NMOS (W/L = 6 + 6)
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VLSI Design: Chapter 3-1 73
Wire delay
Wires have parasitic resistance, capacitance.
Parasitics start to dominate in deep-submicron
wires. (70 ~ 80 % of delay comes from
wires in deep submicron)
Distributed RC introduces time of flight along
wire into gate-to-gate delay.
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VLSI Design: Chapter 3-1 74
Wire Delay-2
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VLSI Design: Chapter 3-1 75
Wire Delay in Deep Sub-Micron
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VLSI Design: Chapter 3-1 76
RC distribution line (L-model)
Assumes that dominant capacitive coupling is
to ground, inductance can be ignored.
Elemental values are ri, ci.
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VLSI Design: Chapter 3-1 77
Elmore delay & RC trees
Elmore defined delay through linear network
as the first moment of the network impulse
response.
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VLSI Design: Chapter 3-1 78
Different Models
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VLSI Design: Chapter 3-1 79
p-model, 6 p-model
…+
-
Vin
Cload
+
-
Vout
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VLSI Design: Chapter 3-1 80
Delay Calculation
Ti = C1R1 + C2R1 + C3(R1+R3) + C4(R1+R3) + Ci(R1+R3+Ri)
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VLSI Design: Chapter 3-1 81
RC Elmore delay
Can be computed as sum of sections.
Resistor ri must charge all downstream capacitors.
From tf = 2.2 (Rp + RL) CL
Where both R and C proportional to the wire length,
tf proportional to the wire length square.
Delay grows as square of wire length.
Minimizing rc product minimizes growth of delay
with increasing wire length.
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VLSI Design: Chapter 3-1 82
Buffer Insertion
If the length is 10 unit, then the tf will be 100
times than a unit length delay
Cload
+
-
Vout
+
-
Vin
Cload
+
-
Vout
+
-
Vin
… …10R * 10C = 100 RC
10*(RC)+ 9*Tb
RC
Tb
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VLSI Design: Chapter 3-1 83
Example of a Timing Report
Startpoint: pt100_core/main_dp/src2_oprand_ff/DFFR_31_14/out_reg
(rising edge-triggered flip-flop clocked by clk)
Endpoint: cp15/dr_index_ff/DFF_2_0/out_reg
(rising edge-triggered flip-flop clocked by clk)
Path Group: clk Path Type: max Des/Clust/Port
Wire Load Model Library
------------------------------------------------
pt110_logic_test_1 G200K fs90a_b
Point Incr Path
--------------------------------------------------------------------------
clock clk (rise edge) 0.00 0.00
clock network delay (ideal) 0.00 0.00
pt100_core/main_dp/src2_oprand_ff/DFFR_31_14/out_reg/CK (QDFZP)0.00 #
0.00 r
pt100_core/main_dp/src2_oprand_ff/DFFR_31_14/out_reg/Q (QDFZP)0.76
0.76 f
pt100_core/main_dp/src2_oprand[14] (main_dp_test_1) 0.00 0.76 f
pt100_core/U26/O (BUF4) 0.32 1.08 f
pt100_core/arith_unit/src2_oprand[14] (arith_unit_test_1)0.00 1.08 f
pt100_core/arith_unit/U459/O (INV4) 0.19 1.28 r
pt100_core/arith_unit/arith_adder/add_360/B[14]
(arith_unit_DW01_add_33_2_test_1) 0.00 1.28 r
pt100_core/arith_unit/arith_adder/add_360/U6/O (OR2) 0.32 1.60 r
pt100_core/arith_unit/arith_adder/add_360/U218/O (ND4T)0.36 1.96 f
pt100_core/arith_unit/arith_adder/add_360/U408/O (INV4)0.18 2.15 r
pt100_core/arith_unit/arith_adder/add_360/U464/O (ND2F)0.21 2.36 f
pt100_core/arith_unit/arith_adder/add_360/U419/O (INV4)0.11 2.47 r
pt100_core/arith_unit/arith_adder/add_360/U280/O
(AOI12)0.19 2.66 f
…….
…….
………..
cp15/U1229/O (OA12P) 0.29 6.64 r
cp15/U1399/O (ND2P) 0.10 6.75 f
cp15/U903/O (INV2) 0.06 6.80 r
cp15/dr_index_ff/DFF_2_0/out_reg/RB (DFZCRBN) 0.00 6.80
r
data arrival time 6.80
clock clk (rise edge) 6.00 6.00
clock network delay (ideal) 0.00 6.00
clock uncertainty -1.00 5.00
cp15/dr_index_ff/DFF_2_0/out_reg/CK (DFZCRBN) 0.00 5.00
r
library setup time -1.07 3.93
data required time 3.93
--------------------------------------------------------------------------
data required time 3.93
data arrival time -6.80
--------------------------------------------------------------------------
slack (VIOLATED) -2.88
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VLSI Design: Chapter 3-1 84
Buffer Tree
rootleaves
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VLSI Design: Chapter 3-1 85
Max Transition/Cap
1x
2x1x
1x
1x
Maximum Transition
Rule ViolationMaximum Transition
Rule Met
Upsized Driver or Added Buffers
Aft
er O
pti
miz
atio
n
Bef
ore
Op
tim
izat
ion
46
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VLSI Design: Chapter 3-1 86
Placement and wire capacitance
unbalanced load
more balanced
dvr
g1
g2
g3
g4
dvr
g1
g2
g3
g4
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VLSI Design: Chapter 3-1 87
Wire sizing
Wire length is determined by layout
architecture, but we can choose wire width
to minimize delay.
Wire width can vary with distance from driver
to adjust the resistance which drives
downstream capacitance.
Source sink
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VLSI Design: Chapter 3-1 88
Tapering of wiring trees
Different branches of tree can be set to
different lengths to optimize delay.
Optimal tapering improves delay by about
8%.source
sink 1
sink 2
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VLSI Design: Chapter 3-1 89
Speed up the circuit (1)
1. Buffer sizing
2. Buffer insertion, buffer tree
3. Bring critical signal closer to sink
4. Circuit replication
5. Cycle stealing, multi-cycle (memories, …)
6. Re-route
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VLSI Design: Chapter 3-1 90
Speed up the circuit (2)
7. Re-placement (new floorplan, Chapter 7)
8. Logic re-write, Logic change (CKT change, re-synthesis, CKT replication, change libraries/cells/IPs…)
9. Wire sizing
10. Process retarget
11. Architecture change, algorithm change
12. Change spec…… and the last one…
Pray….
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VLSI Design: Chapter 3-1 91
Home works assignment
Chapter 3:
3-4, 3-10, 3-11