linear technologlinear technology - analog deviceslinear technology magazine • august 1998 3...

LINEAR TECHNOLOGYLINEAR TECHNOLOGYLINEAR TECHNOLOGYAUGUST 1998 VOLUME VIII NUMBER 3

, LTC and LT are registered trademarks of Linear Technology Corporation. Adaptive Power, Burst Mode, C-Load,FilterCAD, Hot Swap, Linear View, Micropower SwitcherCAD, No RSENSE, SwitcherCAD and UltraFast are trademarks ofLinear Technology Corporation. Other product names may be trademarks of the companies that manufacture the products.

The LTC1625 CurrentMode DC/DC ControllerEliminates theSense Resistor by ChristopherB. UmmingerIntroductionPower supply designers have a newtool in their quest for ever higherefficiencies. In the past, when design-ing a step-down DC/DC converter,one had to choose between the highefficiency of voltage mode control andthe many benefits of current modecontrol. Although voltage mode con-trol offers high efficiency and a simpletopology, it is difficult to compensate,has poor rejection of input-voltagetransients and does not inherentlylimit output current under fault con-ditions, such as an output shortcircuit. Current mode control over-comes these problems by adding acontrol loop to regulate the inductorcurrent in addition to the output volt-age. Unfortunately, a sense resistor isrequired to measure this current,which adds cost and complexity whilereducing converter efficiency. How-ever, with the new LTC1625 NoRSENSE™ controller, one can enjoy allof the benefits of current mode con-trol without the penalties of using asense resistor.

The LTC1625 is a step-down DC/DC switching regulator controller thatincorporates popular features fromthe LTC1435 and related parts. It iscapable of a wide range of operationwith inputs from 3.7V to 36V. Fixedoutput voltages of 5V and 3.3V can beselected or an external resistive dividercan be used to obtain output voltages

from 1.19V up to nearly the full inputvoltage. The controller provides syn-chronous drive for N-channel powerMOSFETs and retains the advantageof low dropout operation typicallyassociated with P-channel MOSFETs.Burst Mode™ operation maintainsefficiency at low load currents, butcan be overridden to assist second-ary-winding regulation by forcingcontinuous operation. In addition toeliminating the sense resistor, theLTC1625 further reduces the exter-nal parts count by incorporating theoscillator timing capacitor. The oscil-lator frequency can be set to 150kHz,225kHz, or can be injection locked toany frequency between these points.

Current Mode Controlwithout a Sense ResistorHow does one implement currentmode control without a current senseresistor? The answer is to make thestandard power MOSFET switches dodouble duty as current senseelements. By measuring the drain-to-source voltage of a MOSFET operatingin the triode region, one can use theRDS(ON) of the device as a senseresistance. Although conceptuallysimple, this is tricky to implement inpractice because inductor currentinformation can only be obtained whena MOSFET is turned on. Figure 1

continued on page 3

IN THIS ISSUE…COVER ARTICLEThe LTC®1625 Current Mode DC/DCController Eliminates theSense Resistor ............................... 1Christopher B. Umminger

Issue Highlights ............................ 2

LTC in the News… ......................... 2

DESIGN FEATURESLTC1627 Monolithic SynchronousStep-Down Regulator MaximizesSingle or Dual Li-Ion Battery Life..................................................... 5

Jaime Tseng

Clock-Tunable, High Accuracy,Quad 2nd Order, Analog FilterBuilding Blocks ............................. 8Philip Karantzalis

New DC/DC Controller Enables HighStep-Down Ratios ........................ 11Greg Dittmer

VID Voltage Programmer forIntel Mobile Processors ................ 13Peter Guan

Fixed Frequency, 500kHz, 4.5A Step-Down Converter in an SO-8Operates from a 5V Input ............ 16Karl Edwards

48 Volt Hot Swap™ Controller for Negative Voltages ................. 19Henry Yun and Robert Reay

Hot Swapping the PCI Bus Usingthe LTC1643 ............................... 22Robert Reay

Eliminate Pipeline Headaches withNew 12-Bit 3Msps SAR ADC ......... 25Dave Thomas and William C. Rempfer

DESIGN INFORMATIONComponent and MeasurementAdvances Ensure 16-Bit DACSettling Time (Part One) .............. 30Jim Williams

New 16-Bit Bipolar Output DAC inNarrow SO-16 Package ................ 33Hassan Malik

DESIGN IDEAS ....................... 34–37(complete list on page 34)

New Device Cameos ..................... 37

Design Tools ................................ 39

Sales Offices ............................... 40

Linear Technology Magazine • August 19982

EDITOR’S PAGE

LTC in the News…Issue HighlightsThe lead article this month fea-

tures the new LTC1625 No RSENSEDC/DC controller. The LTC1625 is astep-down DC/DC switching regula-tor controller that incorporatespopular features from the LTC1435and related parts. The LTC1625 pro-vides all of the benefits of currentmode control without the penalties ofusing a sense resistor. How does oneimplement current mode control with-out a current sense resistor? Theanswer is to make the standard powerMOSFET switches do double duty ascurrent sense elements.

Several other new power controlproducts are also introduced in thisissue. The LTC1627 is a new additionto a growing family of power manage-ment products optimized for Li-Ionbatteries. Li-Ion batteries, with theirhigh energy density, are becomingthe chemistry of choice for manyhandheld products. The LTC1627monolithic, current mode synchro-nous buck regulator was specificallydesigned to meet these demands. Withits operating supply range of 2.65V to8.5V, it can operate from one or twoLi-Ion batteries as well as 3- to 6-cellNiCd and NiMH battery packs.

LTC1435A is an improved versionof the LTC1435 synchronous DC/DCcontroller. It has all the outstandingfeatures of the LTC1435, with areduced minimum on-time of 300nsor less, and improved noise immunityat low output voltages. With theseimprovements, high performance atoutput voltages down to 1.3V can beachieved with operating frequenciesin excess of 250kHz from input sup-ply voltages above 22V.

To accommodate the precise, yetvaried voltage requirements of thenext generation of microprocessors,LTC introduces the LTC1706-19 VIDvoltage programmer. This device is aprecision, digitally programmableresistive divider that enables an entirefamily of LTC’s DC/DC converterswith onboard 1.19V references to pro-duce output voltages from 1.30V to2.00V in 0.05V steps.

The LT®1506 is a 500kHz mono-lithic buck mode switching regulator,functionally identical to the LT1374but optimized for lower input voltageapplications. Its high 4.5A switch rat-ing makes this device suitable for useas the primary regulator in small tomedium power systems.

In the data conversion arena, wedebut the LTC1412, a new 12-bit3Msps ADC that brings new levels ofperformance and ease of use to highspeed ADC applications. By raisingthe speed of the successive approxi-mation (SAR) method to 3Msps, iteliminates the many drawbacks ofpipelined and subranging ADCs inthat speed range. It is the first clean,simple to use alternative to pipelinedADCs for applications up to 3Msps.

Also in this issue, we premier sev-eral new members of the LTC1068family of universal switched capaci-tor filters. Each product contains fourmatched, low noise, high accuracy2nd order switched capacitor filtersections. High precision, high perfor-mance, quad 2nd order, dual 4thorder or single 8th order filters can bedesigned with an LTC1068 familyproduct.

Rounding out the Design Featuressection, we have four new Hot Swap™controllers. The LTC1640H andLTC1640L provide a simple, flexiblesolution to –48V hot swapping. Thechips allow a board to be safelyinserted into or removed from a livebackplane with a supply voltage from–10V to –80V. The “H” and “L” partsare intended for use with moduleswith active high and active low enableinputs, respectively. The LTC1643 isa Hot Swap controller for PC busapplication. The LTC1643H isdesigned for motherboard applica-tions, whereas the LTC1643L isdesigned for CompactPCI™ applica-tions where the chip resides on theplug-in board.

The Design Information sectionincludes part one of an article onachieving and measuring settlingtimes for 16-bit DACs and theLTC1650, a new 16-bit bipolar out-put DAC in the SO-16 package. TheDesign Ideas section includes a lownoise 33V varactor bias supply, a lownoise 5V to –5V/200mA DC/DC con-verter, a crystal oscillator with 50%duty cycle and complementaryoutputs, a circuit to provide a ground-referenced output voltage that isproportional to a measured current.We conclude with a quintet of NewDevice Cameos.

The July 13, 1998 issue of Busi-ness Week included the magazine’sannual “Business Week Global1000” rankings of the top U.S. andinternational corporations accord-ing to market capitalization. LinearTechnology made both the U.S.and global lists, coming in at num-ber 373 in the United States and768 worldwide. The Company alsomade Equities magazine’s annual“America’s Most Profitable Com-panies” issue at the end of June,ranking at number 100 among themost profitable public companiestraded on the Nasdaq.

On July 21, Linear Technologyannounced its financial results forfiscal year 1998. Said Robert H.Swanson, President and CEO, “Fis-cal 1998 was a good year for LinearTechnology as we achieved recordlevels of sales and earnings.”Specifically, the Company reportednet sales for the fourth quarter of$132,011,000 (a 27% increase overnet sales for the fourth quarter ofthe previous year), net income forthe quarter of $49,503,000 (up 32%from the previous fiscal year’sfourth quarter). For the year, theCompany ’s net sa les were$484,799,000 (up 28% comparedto the previous year), with net in-come of $180,902,000 (up 35%versus the prior year).

CompactPCI is a trademark of the PCI Industrial ComputerManufacturers’ Group.

Linear Technology Magazine • August 1998 3

DESIGN FEATURES

–

+

–

+

–

+Σ

BA×11

TA×11

TK

SW

PGND

ITH

M1

M2

IL

I

t t t

t

V SW

V T

V S ∝

IL

V B

VIN

Figure 1. The LTC1625 senses VDS across the power MOSFETs to infer inductor current.

V TUO)V(

V NI )V(

5 21 81 42 03

5.1 ✓ ✓

8.1 ✓ ✓ ✓

5.2 ✓ ✓ ✓ ✓

3.3 ✓ ✓ ✓ ✓ ✓

0.5 ✓ ✓ ✓ ✓

0.21 ✓ ✓ ✓

Table 1. Some common input and output voltage combinations available

with fOSC = 150kHz

LTC1625, continued from page 1

0

I O(M

AX) (

A)

10

DI_1068_02a. EPS

10050RDS(ON) (mΩ)

0

5

Si4420DY

Si4410DY

Si4412DYSi9936DY

Figure 2. Maximum output current isdetermined primarily by the on-resistanceof the power MOSFETs.

illustrates how this is accomplished inthe LTC1625. The drain-to-source volt-ages of the power MOSFETs are sensedthrough the TK, SW and PGND pins.Sense amplifiers TA and BA measureand amplify these voltages only whenthe respective MOSFET is conductingcurrent. The two resulting signals (VTand VB) are summed to create a sensevoltage ramp (VS) that is proportionalto the inductor current throughoutthe entire switching cycle. A currentcomparator (I) uses the sense voltageand a current threshold voltage (ITH) tocontrol the power MOSFETs. Becausethe LTC1625 fully recovers the induc-tor current information, it behaves

identically to traditional current modecontrollers that use a sense resistor.

An important feature of currentmode control is its ability to inher-ently limit load current simply byrestricting the range of the current-threshold voltage. The maximumallowed current is then inversely pro-portional to the sense resistance. Withthe LTC1625, the power MOSFET on-resistance determines the maximumoutput current. An approximate graphof this relationship is shown in Figure2, along with data points for somepopular Siliconix MOSFETs. The on-resistance is typically guaranteed tobe below a maximum specified by themanufacturer, but it will vary sub-stantially between lots and overtemperature. Thus, a design accom-modating the maximum possibleon-resistance will have a significantlyhigher maximum output current whenthis resistance is lower than normal.To ensure reasonable current levelsin case of a fault, such as an outputshort to ground, the LTC1625 willfold back the current if the outputvoltage falls significantly.

Easing the Layout ChallengeEnsuring that switching transientsdo not interfere with the control loopis a common difficulty with high out-put-current power converters. A greatdeal of effort is often expended inboard layout and component place-ment to subdue coupling effects, suchas pulse pairing or grouping. Elimi-nating the sense resistor and itsassociated signal lines makes this jobeasier. In addition, the LTC1625 usessome internal blanking around theswitching transitions to furtherreduce the possibility of jitter or pulsepairing in the control loop. The resultis a controller that is relativelyimmune to these coupling effects andremarkably easy to lay out success-fully. On the other hand, the additionof blanking increases the minimumtime in which the controller can turnthe top MOSFET on and off. Thisplaces a limit on the minimum dutycycle but does not unduly restrict themaximum input voltage for a givenoutput voltage. Table 1 shows themany common input and output volt-age combinations that the LTC1625can accommodate.

Additional FeaturesThe LTC1625 controller is designedfor synchronous, step-down applica-tions with two N-channel powerMOSFETs. Using an N-channel MOS-FET for the topside switch is morecost-effective than using a P-channeldevice, but it requires a floating top-side driver. This driver is poweredusing an external bootstrap capaci-tor and diode. If the input voltage


DESIGN FEATURES

+

+

1�

2�

3�

4�

5�

6�

7�

8

16�

15�

14�

13�

12�

11�

10�

9

EXTVCC

SYNC

RUN/SS

FCB

ITH

SGND

VOSENSE

VPROG

VIN

TK

SW

TG

BOOST

INTVCC

BG

PGND

LTC1625+

5V

CSS 0.1µF

RC1

10k

CC1 820pF

CC2 220pF

RF 4.7Ω

CF0.1µF

CIN10µF30V× 2

M1Si4410DY

M2Si4410DY

L1 7µH

R211k1%

R110k1%

COUT100µF10V0.065Ω ESR× 3

CBO.22µF

CVCC4.7µF

VOUT2.5V/5A

VIN5V–28V

DB = CMDSH-3

D1MBRS140T3

*DB

*

Figure 3. 2.5V/5A adjustable-output supply

continued on page 18

drops close to the output voltage, theLTC1625 will begin skipping cycles,leaving the top MOSFET on longer inorder to maintain low dropout opera-tion. Once in dropout, the boostcapacitor is recharged by turning offthe top MOSFET and turning on thebottom MOSFET briefly every tenthcycle. This maintains a duty cyclegreater than 99% in dropout.

Another external part was elimi-nated by incorporating the oscillatortiming capacitor into the LTC1625. Thecontroller runs at a nominal 150kHzfrequency that can be increased 50%by taking the SYNC pin above 1.2V.Frequency-sensitive applications cansynchronize the oscillator by applyinga clock signal between 165kHz and200kHz to this pin.

Providing gate charge to the powerMOSFET switches is one of the mainsources of efficiency loss in switchingregulators. The LTC1625 includes twofeatures to minimize this loss. Nor-mally, both MOSFETs are turned onand off once per oscillator cycle. Whenthe load current drops to near zero,the gate charge required for switchingbecomes a significant fraction of thetotal input current. The LTC1625addresses this problem with BurstMode operation. As the load currentdecreases below approximately onefifth of its maximum value, both MOS-FETs are turned off for a few cycleswhile the output capacitors supportthe load. In this way, gate charge issaved at the expense of fixed frequencyoperation. For applications that requireit, fixed frequency operation can bemaintained by tying the FCB pin low toforce continuous operation at low loadcurrents. This pin can also be used toadaptively override Burst Mode opera-tion when regulating the output of asecondary winding. Another problemwith gate charge is that it is normallydrawn from the high voltage inputsupply. However, power can be savedby obtaining the gate charge from ahigh efficiency external supply(between 5V and 7V) connected to theEXTVCC pin. This could be the outputvoltage of the regulator, a boosted ver-sion of the output or some other systemsupply, if available.

Removing Output VoltageConstraints along with RSENSEThe LTC1625 can be configured for awide variety of output voltages usingthe VPROG and VOSENSE pins. For basic3.3V and 5V applications, one canuse an internal resistive divider withthe output connected directly to theVOSENSE pin. Connect the VPROG pin toINTVCC for a 5V output or to groundfor a 3.3V output. Leaving the VPROGpin open disables the internal resistivedivider and connects the error-amplifier feedback node directly tothe VOSENSE pin. With an externalresistive divider, the LTC1625 iscapable of regulating an output volt-age anywhere between the inputvoltage and the 1.19V internal refer-ence. This represents a significantimprovement over previous currentmode controllers, which were con-strained by the common mode rangeof the current sense signal lines.

Start-up and shutdown of theLTC1625 can be controlled via theRUN/SS pin. This pin is connected toan external capacitor that is chargedfrom an internal 3µA current source.The controller will be shut down ifthis pin is held below 1.4V. After thepin is released, the part stays shutdown while the capacitor is charged,permitting a controlled delay forsequencing the power supply start-up. When the voltage rises above 1.4V,the controller will start but with aclamp on the current threshold volt-

age that limits the load current toapproximately one third of its maxi-mum value. Limiting the load currentreduces the inrush current into theconverter. As the voltage increasesabove 1.4V, the clamp is graduallyraised to its normal value.

Design ExamplesFigure 3 shows the LTC1625 in anapplication supplying a 2.5V outputusing an external feedback divider.Si4410DY MOSFETs from Siliconixallow this converter to deliver up to 5Aof load current. Ripple current is 1.8A(36% of full load) and current limitoccurs around 6A. Note also that theEXTVCC pin is connected to an exter-nal 5V supply. This increases efficiencyby drawing the roughly 7mA gatecharge current from a supply lowerthan VIN.

An efficiency plot of this circuit isshown in Figure 4. An LTC1435 withidentical components in the powerpath is also plotted for comparison. Atlower output voltages such as this,the sense resistor is responsible for anincreasing share of the total powerloss. By eliminating this source ofloss, the LTC1625 is easily able todeliver an efficiency greater than 90%at high load current. The benefit ofreduced I2R loss is readily apparent atthe highest loads. The controller makesa transition to Burst Mode operationbelow around 1.1 A which keeps theefficiency high at moderate loads.


DESIGN FEATURES

IntroductionThe LTC1627 is a new addition to agrowing family of power managementproducts optimized for Li-Ion batter-ies. Li-Ion batteries, with their highenergy density, are becoming thechemistry of choice for many hand-held products. As the demand forlonger battery operating time contin-ues to increase and the operatingvoltages of submicron DSPs andmicrocontrollers decreases, moredemands are placed on DC/DC con-version. The LTC1627 monolithic,current mode synchronous buck regu-lator (Figure 1) was specificallydesigned to meet these demands.

Single and DoubleLi-Ion Cell OperationThe LTC1627, with its operating sup-ply range of 2.65V to 8.5V, can operatefrom one or two Li-Ion batteries aswell as 3- to 6-cell NiCd and NiMHbattery packs. Figure 2 shows a typi-cal discharge voltage profile of a single

LTC1627 Monolithic SynchronousStep-Down Regulator MaximizesSingle or Dual Li-Ion Battery Life

by Jaime Tseng

Li-Ion battery. As shown, a fullycharged single-cell Li-Ion batterybegins the discharge cycle around 4V(it may be slightly higher or lower,depending upon the manufacturer’scharge-voltage specifications). Dur-ing the bulk of the discharge time, thecell produces between 3.5V and 4.0V.Finally, towards the end of discharge,the cell voltage drops quickly below3V. When the voltage drops further,the discharge must be terminated toprevent damage to the battery. A pre-cision undervoltage lockout circuittrips when the LTC1627’s supply volt-age dips below 2.5V, shutting the partdown to only 5µA of supply current.

MaximizingBattery Run TimeThe LTC1627 incorporates power sav-ing Burst Mode operation and 100%duty cycle for low dropout to maxi-mize the battery operating time. InBurst Mode operation, both power

MOSFETs are turned off for increas-ing intervals as the load current drops.Along with the gate-charge savings,unused circuitry is shut down betweenburst intervals, reducing the quies-cent current to 200µA. This extendsoperating efficiencies exceeding 90%to over two decades of output loadrange (see Figure 3). As the batterydischarges, the LTC1627 smoothlyshifts from a high efficiency switch-mode DC/DC regulator to a lowdropout (100% duty cycle) switch. Inthis mode, the voltage drop betweenthe battery input and the regulatoroutput is determined by the load cur-rent, the series resistance of the

–

+

350kHzOSC

2.5VUVLO

PDR

NDR

CONTROL

SYNC

RUN/SS

VFB

0.8V

ITH

SW

VDR

VOUT

VIN

LTC1627

DISCHARGE CAPACITY (mAh)0

1.5

Li-Io

n CE

LL V

OLTA

GE (V

)

2.0

3.0

3.5

4.0

5.0

400 800

2.5

4.5

200 600 1000 1200 1400

2C

1C

0.2C

SANYO 186501300mAhTA = 25°C

OUTPUT CURRENT (A)0.01

70

EFFI

CIEN

CY (%

)

75

80

85

90

100

0.10 1.00

95

L1 = 15µHVOUT = 3.3V

VIN = 5V

Figure 1. LTC1627 block diagram

Figure 2. Typical single-cell Li-Iondischarge curve

Figure 3. Efficiency vs output load current


DESIGN FEATURES

internal P-channel power MOSFETand the inductor resistance.

The internal power MOSFETswitches provide very low resistanceeven at low supply voltages. Figure 4is a graph of switch resistance vssupply voltage for both switches. TheRDS(ON) is typically 0.5Ω at 5V andonly rises to approximately 0.65Ω at3V, for both switches. This low switchRDS(ON) ensures high efficiency switch-ing as well as low dropout DCcharacteristics at low supply voltages.

ExtendingLow Supply OperationAt low supply voltages, the LTC1627is most likely to be running at highduty cycles or in dropout, where theP-channel main switch is on continu-ously. Hence, the I2R loss is due mainlyto the RDS(ON) of the P-channel MOS-FET. When VIN is below 4.5V, theRDS(ON) of the P-channel MOSFET canbe lowered further by driving its gatebelow ground. The top P-channelMOSFET driver makes use of a floatingreturn pin, VDR, to allow biasing below

GND. A simple charge pump boot-strapped to the SW pin realizes anegative voltage at the VDR pin, asshown in Figure 5. Each time the SWnode cycles from low to high and thenfrom high to low, charge is trans-ferred from C2 to C1 producing anegative voltage at VDR equal in mag-nitude to VIN – (2 • VDIODE). In dropout,when the P-channel MOSFET isturned on continuously, a dropoutdetector counts the number of oscil-lator cycles that the P-channelMOSFET remains on and periodicallyforces a brief off period to allow C1 torecharge. When 100% duty cycle isdesired, VDR can be grounded to dis-able the dropout detector.

Constant-Frequency,Current Mode ArchitectureThe LTC1627 uses a constant-frequency, current mode step-downarchitecture that provides excellentrejection of input line and output loadtransients and also provides cycle-by-cycle current limiting. Input linetransients are rejected by the feed-forward characteristics inherent incurrent mode control. The output loadtransients are rejected by the greatererror-amplifier bandwidth afforded incurrent mode control. In currentmode, the circuit behaves as if therewere a constant current feeding theparallel combination of the outputcapacitor and output load, yieldingonly a 90° rather than a 180° phaselag. This simplifies the feedback-loop design and the circuitry aroundthe error amplifier required forstabilization.

Current mode limits the peak cur-rent cycle-by-cycle, protecting theinternal main switch and synchro-nous rectifier. In extreme cases, whenthe output is shorted to ground, thefrequency of the oscillator is reducedto one-tenth of its nominal frequencyto allow the inductor current time todecay and prevent inductor-currentrunaway. The oscillator’s frequencygradually increases back to its nomi-nal frequency when VFB rises above0.3V.

The internal oscillator is set for afixed switching frequency of 350kHz,allowing the use of small surfacemount inductors. In switching-noise-sensitive applications, the LTC1627can be externally synchronized to fre-quencies of up to 525kHz. Duringsynchronization, Burst Mode opera-tion is inhibited and pulse-skippingmode is used. In this mode, when theoutput load is very low, the currentcomparator remains tripped for morethan one cycle and forces the mainswitch to stay off for the same num-ber of cycles. Increasing the outputload slightly allows constant frequencyPWM operation to resume.

Minimal ExternalComponentsSize is extremely important in mod-ern portable electronics, so theLTC1627 is designed to work with aminimum number of external compo-nents. The loop compensation,current sense resistor and the mainand synchronous switches are inter-nal. An internal catch diode is alsoprovided across the internal synchro-nous switch, eliminating parasiticcurrents or latch-up if the externalSchottky diode is omitted. Only an

Figure 4. RDS(ON) for both switchesvs input voltage

Figure 5. Using a charge pump to bias VDR

Figure 6. Dual-output 1.8V/0.3A and 3.3V/100mA application

INPUT VOLTAGE (V)1

0

R DS(

ON) (

Ω)

0.1

0.3

0.4

0.5

7

0.9

0.2

4 10

0.6

0.7

0.8

2 3 5 6 8 9

TJ = 25˚C

MAIN AND SYNCHRONOUS SWITCH

VDR SWL1

100µFCOUT

LTC1627

VOUT

VINVIN


DESIGN FEATURES

inductor, input and output filtercapacitors and two small resistorsand capacitors are needed to con-struct a high efficiency DC/DCswitching regulator (see Figure 7).The 47pF filter capacitor connectedto the ITH pin (error-amplifier output)filters out switching noise. If the loopcompensation needs to be adjustedfor a specific application, the ITH pincan also be used for externalcompensation.

Auxiliary-Winding ControlUsing the SYNC/FCB PinBesides higher efficiency and lowerswitching noise, synchronous switch-ing provides a means of regulating asecondary flyback winding. In non-synchronous regulators, power must

be drawn from the inductor primarywinding in order to extract powerfrom auxiliary windings. But withcontinuous synchronous operation,power can be drawn from the auxil-iary windings without regard to theprimary output load.

The LTC1627, with its synchro-nous switching and attendantcircuitry, provides the means of easilyconstructing a secondary flybackregulator, as shown in Figure 6. Thisflyback regulator is regulated by thesecondary feedback resistive dividertied to the SYNC/FCB pin. This pinforces continuous operation wheneverit drops below its ground-referencedthreshold of 0.8V. Power can then bedrawn from the secondary flybackregulator whether the main output isloaded or not.

ITH

RUN/SS

VFB

GND

SYNC/FCB

VDR

VIN

SW

LTC1627

CITH47pF

CSS0.1µF

COUT†100µF6.3V

25µH*

R1249k

1%

R280.6k

1%

1

2

3

4 5

6

7

8

VIN ≤ 8.4V

CIN††22µF16V

VOUT3.3V/0.5A

SUMIDA CD54-250AVX TPSC107M006R0150AVX TPSC226M016R0375

+

+*†

††

ITH

RUN/SS

VFB

GND

SYNC/FCB

VDR

VIN

SW

LTC1627

CITH47pF

CSS0.1µF

COUT✝ 100µF6.3V

D1

C10.1µF

15µH*

R1169k

1%

R280.6k

1%

1

2

3

4 5

6

7

8

VIN2.8V–4.5V

CIN††22µF16V

C20.1µF

VOUT2.5V/0.5A

SUMIDA CD54-150ZETEX BAT54SAVX TPSC107M006R0150AVX TPSC226M016R0375

D2

BAT54S**

++

***

†††

Typical Applications

1 or 2 Li-IonStep-Down ConverterFigure 7 is a schematic diagram show-ing the LTC1627 being powered byone or two Li-Ion batteries. All thecomponents shown in this schematicare surface mount and have beenselected to minimize the board spaceand height. The output voltage is setat 3.3V, but is easily programmed toother voltages.

Single Li-IonStep-Down ConverterThe circuit in Figure 8 is intended forinput voltages below 4.5V, making itideal for single Li-Ion battery applica-tions. Diodes D1 and D2 andcapacitors C1 and C2 comprise thebootstrapped charge pump to realizea negative supply at the VDR pin, thereturn pin for the top P-channelMOSFET driver. This allows Figure8’s circuit to maintain low switchRDS(ON) all the way down to the UVLOtrip voltage.

ConclusionThe new LTC1627 monolithicsynchronous buck regulator is a ver-satile, high efficiency, DC/DCconverter that is at home in a widerange of low input voltage applica-tions. Features such as precisionUVLO and optional bootstrapped gatedrive make it particularly well suitedto single-cell Li-Ion power.

Figure 7. Dual lithium-ion to 3.3V/0.5A regulator

Figure 8. Single lithium-ion to 2.5V/0.5A regulator

forthe latest information

on LTC products, visit

www.linear-tech.com


DESIGN FEATURES

Clock-Tunable, High Accuracy,Quad 2nd Order, Analog FilterBuilding Blocks

by Philip Karantzalis

IntroductionThe LTC1068 product family consistsof four monolithic, clock-tunable filterbuilding blocks. Each productcontains four matched, low noise,high accuracy 2nd order switchedcapacitor filter sections. An externalclock tunes the center frequency ofeach 2nd order filter section. TheLTC1068 products differ only in theirclock-to-center frequency ratio. Theclock-to-center frequency ratio is setto 200:1 (LTC1068-200), 100:1(LTC1068), 50:1 (LTC1068-50) or 25:1(LTC1068-25). External resistors canmodify the clock-to-center frequencyratio. High precision, high perfor-mance, quad 2nd order, dual 4thorder or single 8th order filters can bedesigned with an LTC1068 familyproduct. Designing filters with anLTC1068 product is fully supportedby the FilterCAD™ 2.0 design soft-

ware for Windows®. The internal sam-pling rate of all the LTC1068 devicesis twice the clock frequency. Thisallows the frequency of input signalsto approach twice the clock frequencybefore aliasing occurs. Maximumclock frequency for LTC1068-200,LTC1068 and LTC1068-25 is 6MHzwith ±5V supplies; that for theLTC1068-50 is 2MHz with a single 5Vsupply. For low power filter applica-tions, the LTC1068-50 power supplycurrent is 4.5mA with a single 5Vsupply and 2.5mA with a single 3Vsupply. The LTC1068 products areavailable in a 28-pin SSOP surfacemount package. The LTC1068 (the100:1 part) is also available in a 24-pin DIP package. The following fourcircuits are typical examples ofapplication-specific filters that theLTC1068 products can realize.

LTC1068-200 UltralowFrequency Linear-PhaseLowpass FilterFigure 1 shows an LTC1068-200linear-phase 1Hz lowpass filter sche-matic and Figure 2 shows its gain andgroup delay responses. The clock fre-quency of this filter is 400 times the–3dB frequency (f–3dB or fCUTOFF). Thelarge clock-to-fCUTOFF frequency ratioof this filter is useful in ultralow fre-quency filter applications whenminimizing aliasing errors could bean important consideration. Forexample, the 1Hz lowpass filter shownin Figure 1 requires a 400Hz clockfrequency. For this filter, the inputfrequencies that can generate alias-ing errors are in a band from 795Hz to805Hz (2 × fCLK ±5 × f–3dB). For mostvery low frequency signal-processingapplications, the signal spectrum isless than 100Hz. Therefore, Figure1’s filter will process very low fre-quency signals without significantaliasing errors, since its clock fre-quency is 400Hz and the aliasinginputs are in a small band around800Hz.Windows is a registered trademark of Microsoft Corp.

INV B

HPB/NB

BPB

LPB

SB

NC

AGND

V+

NC

SA

LPA

BPA

HPA/NA

INV A

1

2

3

4

5

6

7

8

9

10

11

12

13

14

INV C

HPC/NC

BPC

LPC

SC

V–

NC

CLK

NC

SD

LPD

BPD

HPD/ND

INV D

28�

27�

26�

25�

24�

23�

22�

21�

20�

19�

18�

17�

16�

15

LTC1068-200RL2 14.3k

R22 15.4k

R32 10k

R34 10k

R24 15.4k

R43 12.4k

R23 10k

R33 12.4k

R11 14.3k

R41 15.4k

R21 12.4k

R31 10k

400kHz

VOUT

5V

VIN

RL1 23.2k

0.1µF

RB3 23.2k

RL3 23.2k

R52 5.11k

R62 9.09k

–5V

R54 5.11k

R64 9.09k0.1µF

GAIN

(dB)

10

0�

–10�

–20�

–30�

–40�

–50�

–60�

–70�

–80�

–90

0.1 10FREQUENCY (Hz)

1

1.0

0.9�

0.8

0.7�

0.6�

0.5�

0.4�

0.3�

0.2�

0.1�

0.0

GROUP DELAY (s)

GAIN

GROUPDELAY

Figure 1. Linear-phase lowpass filter: f–3dB = 1Hz = fCLK/400Figure 2. Gain and group delay responseof Figure 1’s circuit.


DESIGN FEATURES

LTC1068-50 Single 3.3VLow Power Linear-PhaseLowpass FilterFigure 3 is a schematic of an LTC1068-50-based, single 3.3V, low power,lowpass filter with linear phase. Theclock-to-fCUTOFF ratio is 50 to 1 (fCUTOFFis the –3dB frequency). Figure 4 showsthe gain and group delay response.

The flat group delay response in thefilter’s passband implies a linearphase. A linear-phase filter has atransient response with very smallovershoot that settles very rapidly. Alinear-phase lowpass filter is usefulfor processing communication sig-nals with minimum intersymbolinterference in digital communica-tions transmitters or receivers. The

INV B

HPB/NB

BPB

LPB

SB

NC

AGND

V+

NC

SA

LPA

BPA

HPA/NA

INV A

1

2

3

4

5

6

7

8

9

10

11

12

13

14

INV C

HPC/NC

BPC

LPC

SC

V–

NC

CLK

NC

SD

LPD

BPD

HPD/ND

INV D

28�

27�

26�

25�

24�

23�

22�

21�

20�

19�

18�

17�

16�

15

LTC1068-50

RH2 34k

RL2 9.09k

R22 43.2k

R32 43.2k

R42 196k

R44 34.8k

R34 14.3k

R24 16.9k

R43 48.7k

R23 10.7k

R33 12.7k

R11 22.6k

R41 22.6k

R21 20.5k

R31 10k

500kHz

VOUT

3.3V

VIN

RB1 13.3k

RA1 56.2k

0.1µF

1µF

RB3 24.9k

RL3 26.7k

GAIN

(dB)

10

0�

–10�

–20�

–30�

–40�

–50�

–60�

–70�

–80

1k 100kFREQUENCY (Hz)

10k

GAIN

GROUPDELAY

150�

140�

130�

120�

110�

100�

90�

90�

70�

60

GROUP DELAY (µs)

maximum clock frequency for thisfilter is 1MHz with a single 3.3V sup-ply and 2MHz with a single 5V supply.Typical power supply current is 3mAwith a single 3.3V supply and 4.5mAwith a single 5V supply.

LTC1068-25 SelectiveBandpass Filter is ClockTunable to 80kHzFigure 5 shows a 70kHz bandpassfilter based on the LTC1068-25 oper-ating with dual 5V power supplies.The clock-to-center frequency ratio is25 to 1. Figure 6 shows the gainresponse of Figure 5’s bandpass fil-ter. The passband of this filter extendsfrom 0.95 × fCENTER to 1.05 × fCENTER.The stopband attenuation is greaterthan 40dB at 0.8 × fCENTER and 1.15 ×fCENTER. The center frequency can beclock tuned to 80kHz with dual 5Vsupplies and to 40kHz with a single5V supply. With FilterCAD, theLTC1068-25 can be used to realizebandpass filters less selective thanthat shown in Figure 6, which can beclock tuned up to 160kHz with dual5V supplies.

INV B

HPB/NB

BPB

LPB

SB

NC

AGND

V+

NC

SA

LPA

BPA

HPA/NA

INV A

1

2

3

4

5

6

7

8

9

10

11

12

13

14

INV C

HPC/NC

BPC

LPC

SC

V–

NC

CLK

NC

SD

LPD

BPD

HPD/ND

INV D

28�

27�

26�

25�

24�

23�

22�

21�

20�

19�

18�

17�

16�

15

LTC1068-25

RL2 23.2k

RH2 11.3k

R22 4.99k

R32 107k

R34 63.4k

R44 17.4k

R24 7.5k

R43 42.3k

R23 4.99k

R33 59k

R11 29.4k

R41 20.5k

R21 4.99k

R31 24.9k

1.75MHz

VOUT

5V

VIN

RH1 28k

0.1µF

RH3 15.4k

RL3 45.3k

R52 4.99k R62 56.2k

–5V0.1µF

R544.99k

R64 10k

R61 11.3k

R51 4.99k

GAIN

(dB)

10

0�

–10�

–20�

–30�

–40�

–50�

–60�

–70�

–80�

–90

20 100FREQUENCY (kHz)

6030 40 50 70 80 90

Figure 3. Low power, single 3.3V supply, 10kHz, 8th order, linear-phase lowpass filter

Figure 5. 70kHz, 8th order, bandpass filter

Figure 4. Gain and group delayresponse of Figure 3’s filter

Figure 6. Gain response of Figure 5’s filter


DESIGN FEATURES

INV B

HPB/NB

BPB

LPB

SB

AGND

V+

SA

LPA

BPA

HPA/NA

INV A

1

2

3

4

5

6

7

8

9

10

11

12

INV C

HPC/NC

BPC

LPC

SC

V–

CLK

SD

LPD

BPD

HPD

INV D

24�

23�

22�

21�

20�

19�

18�

17

16

15

14

13�

LTC1068

RL2 12.4k

RH2 55.2k

R22 10k

R32 12.4k

R24 10k

R34 10k

R11 22.4k

R21 10k

R31 12.4k

RL1 12.4k

0.1µF

RH3 38.4k

RL3 16.5k

–5V

0.1µF

5V

R33 12.4k

R23 10k

fCLK = 128 × fOUT

*COSINE WAVE OUT

*SINE WAVE OUT

3�

4�

5�

6�

11�

12�

13�

14

1�

2�

16�

10�

7�

8�

9�

15

CD4520

1Q0�

1Q1�

1Q2�

1Q3�

2Q0�

2Q1�

2Q2�

2Q3

CLK

GND

÷2

÷4

÷8

÷16

÷32

÷64

÷128

÷256

5V

0.47µF

0.1µF

fOUT =fCLK128

PIN 16'S COSINE WAVE OUTPUT IS REFERENCED TOPIN 15'S SINE WAVE OUTPUT

*

LTC1068 Square-Wave-to-Quadrature Oscillator FilterFigure 7 shows the schematic of aLTC1068 based filter that is specifi-cally designed to produce a lowharmonic distortion sine and cosineoscillator from a CMOS-level squarewave input. The reference sine waveoutput of Figure 7’s circuit is on pin15 (BPD on the 24-pin LTC1068 pack-age) and the cosine output is on pin16 (LPD on the 24-pin LTC1068 pack-age). The output frequency of thisquadrature oscillator is the filter’sclock frequency divided by 128. Theoutput of a CMOS CD4520 divide-by-128 counter is coupled with a 0.47µFcapacitor to the input to the LTC1068filter operating with dual 5V powersupplies. The filter’s clock frequencyis the input to the CD4520 counter.

The LTC1068 filter is designed topass the fundamental frequency com-ponent of a square wave and attenuateany harmonic components higherthan the fundamental. An ideal squarewave (50% duty cycle) will have onlyodd harmonics (3rd, 5th, 7th and soon), whereas a typical practical squarewave has a duty cycle less or morethan 50% and will also have evenharmonics (2nd, 4th, 6th and so on).The filter of Figure 7 has a stopbandnotch at the 2nd and 3rd harmonicsfor a square wave input with a fre-quency equal to the filter’s clockfrequency divided by 128. The filter’ssine wave output (pin 15) is 1VRMS fora ±2.5V square wave input and hasless than 0.025% THD (total har-monic distortion) for input frequenciesup to 16kHz and less than 0.1% THD

for frequencies up to 20kHz. The co-sine output (on pin 16, referenced topin 15’s sine wave output) is 1.25VRMSfor a ±2.5V square wave input andhas less than 0.07% THD for frequen-cies up to 20kHz.

The 20kHz frequency limit is dueto the CD4520; with a 74HC typedivide-by-128 counter, sine and cosinewaves up to 40kHz can be generatedwith the LTC1068-based filter ofFigure 7.

The LTC1068Product Family: VersatileFilter Building Blocks,1Hz to 200kHzThe previous four filter examples aretypical of the great variety of filtersthat can be designed with the LTC1068product family. As a general selectionguide, the following is recommended:for low noise and low frequency filtersless than 20kHz, use the LTC1068-200; for low noise and low frequencyfilters less than 40kHz, use theLTC1068; for low power filters up to20kHz operating with single 3V to 5Vsupplies, the LTC1068-50 should bethe choice; and for filters in the fre-quency range 40kHz to 200kHz, usethe LT1068-25. The FilterCAD designsoftware will recommend the appro-priate LTC1068 device for most filterapplications. For some application-specific filter designs, the choice ofthe proper LTC1068 device may notbe obvious and the assistance of aLinear Technology applications engi-neer will be helpful.

LTC1068-BasedSemicustom FiltersFor application-specific filter require-ments of single 8th order or dual 4thorder filters, a customized version ofan LTC1068-family product can beobtained in a 16-pin SO package withinternal thin-film resistors. Clock-to-center frequency ratios higher or lowerthan 200:1, 100:1, 50:1 or 25:1 canalso be obtained. Please contact LTCMarketing for details.

Figure 7. Square-wave-to-quadrature oscillator converter

Authors can be contactedat (408) 432-1900


DESIGN FEATURES

New DC/DC Controller EnablesHigh Step-Down Ratios

by Greg DittmerImportance ofMinimum On-TimeAs processor voltage requirements arepushed lower and lower, input supplyvoltages remain high, forcing DC/DCconverters to operate at lower andlower duty cycles. Since operatingfrequencies also remain high to mini-mize noise and the size of components,the on-time of the topside switch in aconstant-frequency converter mustcontinue to decrease to regulate thelower and lower output voltages. Therequired on-time is given by TON =VOUT/(VIN • f)

Unfortunately, there is a limit tohow small this time can be. In atypical current mode DC/DC con-verter, once the main switch is turnedon at the beginning of each switchingcycle, the speed at which it can beturned off is limited by the responsetime of the current comparator, thetime required for the turn-off com-mand to propagate through the logicand output driver and the timerequired to discharge the capacitanceof the topside gate. These delays addup to a few hundred nanosecondsand constitute the minimum time thetopside switch must stay on duringeach switching cycle. If the maximumVIN and frequency are fixed, this mini-mum on-time sets a lower limit on

output voltage. If an output voltagebelow this limit is required, the onlychoice is to lower the operating fre-quency, which is usually not desirable.Figure 1 shows the on-times requiredwith VIN = 22V as a function of outputvoltage for various frequencies.

Capabilities of the LTC1435The LTC1435 high efficiency synchro-nous DC/DC controller has beenextremely popular for notebook com-puters and other battery-poweredequipment due to its low noise, con-stant-frequency operation and its dualN-channel drive for outstanding highcurrent efficiency without sacrificinglow dropout operation. However, its400ns–500ns minimum on-timerequires lower operating frequencies(


DESIGN FEATURES

is approaching TON(MIN), there may bea minimum ripple current amplitude,and hence, a maximum inductancenecessary to prevent cycle skipping.Or, expressed differently, the lowerthe inductance, the higher the maxi-mum VIN that can be achieved beforethe minimum on-time is violated andcycle skipping occurs. For mostapplications, 40% ripple not onlyreduces the minimum on-time butalso optimizes efficiency.

22V to 1.6VConverter at 250kHzFigure 4 shows the LTC1435A config-ured in an all N-channel synchronousbuck topology as a 22V to 1.6V/3Aconverter running at 250kHz. The43pF COSC capacitor sets the internal

oscillator frequency at 250kHz andthe 33mΩ sense resistor sets themaximum load current at 3A. For a22V to 1.6V converter, the on-timerequired is:

TON = 1.6/(22 × 250kHz) = 291nsCan the LTC1435A do this? At

maximum VIN the inductor ripple is

∆IL =

VOUT • (1 – VOUT/VIN)

F • L

1.6 • (1 – 1.6/22)250kHz • 4.7µH

= = 1.3A

which is 43% of the 3A maximumload. From Figure 3, 43% ripple givesa minimum on-time of 235ns, whichis well below the 291ns required by

+

+

+

M1Si44412DY

M2 Si44412DY

CIN10µF, 30V×2

L1 4.7µHDB*

RSENSE0.033Ω 35.7k

1%

102k1%

COUT100µF, 6.3V×2

D1MBRS

-140T3

4.7µF

1000pF

0.1µF

COSC 43pF

CSS0.1µF

CC330pF

CC251pF

100pF

VIN

TG

SW

INTVCC

BOOST

BG

PGND

COSC

RUN/SS

ITH

SGND

VOSENSE

13�

16�

14�

12�

15�

11�

10

1

2

3

5

6

VOUT1.60V/3A

VIN4.5V–22V

SENSE– SENSE+

7 8

LTC1435A

*DB = CMDSH-3CENTRAL(516) 435-1110

RC10k

+

+ +

M1Si4410DY

M2 Si4410DY

CIN10µF, 30V×2

L1 3.3µHDB*

RSENSE0.015Ω

COUT820µF4V×2

D1MBRS

-140T3

4.7µF

1000pF

0.22µF

COSC 43pF

CSS0.1µF

CC1000pF

CC2220pF

51pF

VIN

TG

SW

INTVCC

BOOST

BG

PGND

COSC

RUN/SS

ITH

SGND

VOSENSE

13�

16�

14�

12�

15�

11�

10

1

2

3

5

6

VOUT1.30V–2.00V/7A

VIN4.5V–22V

SENSE– SENSE+

7 8

LTC1435A

*DB = CMDSH-3CENTRAL(516) 435-1110

RC10k

RF4.7Ω

CF0.1µF

VCC

FB

VID0

VID1

VID2

VID3 GND

SENSE

LTC1706-19

FROM µP

3

5

7 8 1 2 4

6

this application, so no cycle skippingwill occur. If a 10µH inductor is used,the ripple amplitude drops to 0.6A or20% and the minimum on-timeincreases to 280ns. This does notprovide much margin below the 291nson-time required, and thus the 4.7µHinductor is a better choice.

Intel Mobile ProcessorVID Power ConverterFigure 5 shows the LTC1435A usedwith an LTC1706-19 to implement anIntel Mobile Pentium® II ProcessorVID power converter. This DC/DCconverter provides digitally selectableoutput voltages over the range of 1.3Vto 2.0V in 50mV increments at 250kHzand a 7A load current. The selectableoutput voltage is implemented byreplacing the conventional feedbackresistor network with the LTC1706-19, which provides the appropriatefeedback resistor ratios internally (seethe accompanying article in thisissue). The proper ratio is selectedwith the 4-bit digital input pins.

ConclusionThe LTC1435A retains all the out-standing features of the LTC1435,such as constant-frequency opera-tion, dual N-channel MOSFET driveand low dropout, while addingenhancements such as reducedminimum on-time and improved per-formance at low output. With theseenhancements, the LTC1435A is aperfect fit for notebook computersand battery-powered equipmentrequiring high frequency, low dutycycle DC/DC converters.Pentium is a registered trademark of Intel Corp.

Figure 4. LTC1435A 22V to 1.6V/3A converter (f = 250kHz)

Figure 5. Intel Mobile Pentium II processor VID power converter


DESIGN FEATURES

VID Voltage Programmerfor Intel Mobile Processors by Peter Guan

Microprocessor manufacturers’relentless push for higher speed andlower power dissipation, especially inareas of mobile laptop computer pro-cessors, is forcing supply voltages tothese processors to a level previouslythought impossible or impractical. Infact, the supply voltage has becomeso critical that different microproces-sors demand different yet precisesupply voltage levels in order to func-tion optimally.

To accommodate this new gene-ration of microprocessors, LTCintroduces the LTC1706-19 VID(voltage identification) voltage pro-grammer. This device is a precision,digitally programmable resistivedivider designed for use with an en-tire family of LTC’s DC/DC converterswith onboard 1.19V references. Theseconverters include the LTC1433,LTC1434, LTC1435, LTC1435A,LTC1436, L TC1438, LTC1439,LTC1538-AUX, LTC1539 andLTC1624. (Consult the factory forfuture compatible DC/DC converter

+

+ +

M1Si4410DY

M2 Si4410DY

CIN10µF, 30V×2

L1 3.3µHDB*

RSENSE0.015Ω

COUT820µF4V×2

D1MBRS

-140T3

4.7µF

1000pF

0.22µF

COSC 43pF

CSS0.1µF

CC1000pF

CC2220pF

51pF

VIN

TG

SW

INTVCC

BOOST

BG

PGND

COSC

RUN/SS

ITH

SGND

VOSENSE

13�

16�

14�

12�

15�

11�

10

1

2

3

5

6

VOUT1.30V–2.00V/7A

VIN4.5V–22V

SENSE– SENSE+

7 8

LTC1435A

*DB = CMDSH-3

RC10k

RF4.7Ω

CF0.1µF

VCC

FB

VID0

VID1

VID2

VID3 GND

SENSE

LTC1706-19

FROM µP

40k

VCC

VID0

40k

VCC

VID1

40k

VCC

VID2

40k

VCC

VID3

RFB2SWITCH-CONTROL

LOGIC

RFB113k

RFB2

SENSE

FB

SGND

VCC

Figure 1. Intel Mobile Pentium II processor VID power converter

Figure 2. LTC1706-19 block diagram


DESIGN FEATURES

products.) The LTC1706-19 is fullycompliant with the Intel mobile VIDspecifications and comes in a tinySO-8 package. Four digital pins areprovided to program output voltagesfrom 1.3V to 2.0V in 50mV steps withan accuracy of ±0.25%.

Figure 1 shows a VID-programmedDC/DC converter for an Intel mobileprocessor that uses the LTC1435Aand LTC1706-19 to deliver 7A of out-put current with a programmable VOUTof 1.3V to 2.0V from a VIN of 4.5V to22V. Simply connecting the LTC1706-19’s FB and SENSE pins to theLTC1435A’s VOSENSE and SENSE–

pins, respectively, closes the loopbetween the output voltage sense andthe feedback inputs of the LTC1435Aregulator with the appropriate resis-tive divider network, which iscontrolled by the LTC1706-19’s fourVID input pins.

Figure 2 shows a simplified blockdiagram of the LTC1706-19. A 40kresistor in series with a diode fromVCC pulls up each VID input pin.Therefore, the VID pin must begrounded or driven low to produce adigital low input, whereas a digitalhigh input can be generated by eitherfloating the VID pin or connecting it toVCC. Series diodes from VCC areincluded to prevent the inputs frombeing damaged or clamped by apotential higher than VCC. This allowsthe LTC1706-19 to be fully TTL com-patible and operational over a VIDinput voltage range that is muchhigher than VCC. When all the inputsare high, the LTC1706-19 has a typi-

cal quiescent current of 0.1µA fromVCC, because all active devices areturned off. However, due to the pull-up resistors on each of the VIDprogramming inputs, each groundedinput contributes approximately68µA, (VCC – 0.6)/40k of supply cur-rent in a 3.3V system.

The top feedback resistor in theblock diagram, RFB1, connectedbetween SENSE and FB, is a 15kresistor whose value is not modifiedby the state of the VID program inputs.The bottom feedback resistor, RFB2,however, is modified by the four VID

7�

8�

1�

2

VID0�

VID1�

VID2�

VID3

6

5

SENSE

FBGND

VCC

4

3LTC1706-19

10µF

VCC 2.7V–5.5V

+

+

1�

2�

3�

4

8�

7�

6�

5

SENSE–

ITH/RUN

VFB

GND

VIN

BOOST

TG

SW

LTC1624

470pF

6.8k

100pF

1000pF

0.1µFRSENSE0.05Ω

Si4412DY

10µH

MBRS340T3

CIN22µF35V×2

VOUT1.3V–2.0V

COUT100µF10V×2

VIN4.8V–20V

inputs. The precision of the ratio be-tween RFB2 and RFB1 results in a±0.25% output accuracy.

Table 1 shows the VID inputs andtheir corresponding output voltages.VID3 is the most significant bit (MSB)and VID0 is the least significant bit(LSB). When all four inputs are low,the LTC1706-19 sets the regulatoroutput voltage to 2.00V. Eachincreasing binary count is equivalentto decreasing the output voltage by50mV. Therefore, to obtain a 1.30Voutput, the three MSBs are left float-ing while only VID0 is grounded. Incases where all four VID inputs aretied high or left floating, such as whenno processor is present in the system,a regulated 1.25V output is gener-ated at VSENSE.

Figure 3 shows a combination ofthe LTC1624 and the LTC1706-19configured as a high efficiency step-down switching regulator with aprogrammable output of 1.3V to 2.0Vfrom an input of 4.8V to 20V. Usingonly one N-channel power MOSFET,the two SO-8 packaged LTC partsoffer an extremely versatile, efficient,compact regulated power supply.

edoC 3DIV 2DIV 1DIV 0DIV tuptuO

0000 DNG DNG DNG DNG V00.2

1000 DNG DNG DNG taolF V59.1

0100 DNG DNG taolF DNG V09.1

1100 DNG DNG taolF taolF V58.1

0010 DNG taolF DNG DNG V08.1

1010 DNG taolF DNG taolF V57.1

0110 DNG taolF taolF DNG V07.1

1110 DNG taolF taolF taolF V56.1

0001 taolF DNG DNG DNG V06.1

1001 taolF DNG DNG taolF V55.1

0101 taolF DNG taolF DNG V05.1

1101 taolF DNG taolF taolF V54.1

0011 taolF taolF DNG DNG V04.1

1011 taolF taolF DNG taolF V53.1

0111 taolF taolF taolF DNG V03.1

Table 1. VID inputs and corresponding output voltages

Figure 3. High efficiency SO-8, N-channel switching regulator with programmable output


DESIGN FEATURES

Figure 4 shows the LTC1436A-PLLand the LTC1706-19, a combinationthat yields a high efficiency low noisesynchronous step-down switchingregulator with programmable 1.3V to2V outputs and external frequencysynchronization capability.

Besides the LTC family of 1.19V-referenced DC/DC converters, theLTC1706-19 can also be used to pro-

+

+ +

M1Si4412DY

M2 Si4412DY

CIN22µF, 35V×2

L13.3µH

DB*

RSENSE0.02Ω

COUT100µF10V×2

D1MBRS

-140T3

4.7µF

1000pF

0.22µF

COSC 39pF

CSS0.1µF

CC510pF

100pF

VIN

TGL

TGS

SW

INTVCC

BOOST

BGL

PGND

COSC

RUN/SS

ITH

SGND

VOSENSE

18�

21�

19

20�

17�

22�

16�

15

2

3

4

6

8

VOUT1.30V–2.00V/5A

VIN4.5V–22V

SENSE– SENSE+

9 10

LTC1436A-PLL

*DB = CMDSH-3

RC10k

VCC

FB

VID0

VID1

VID2

VID3 GND

SENSE

LTC1706-19

FROM µP

M3IRLML2803

PLL LPF PLLIN

10k 0.1µF

EXTERNALFREQUENCY

SYNCHRONIZATION

1 24

+

7�

8�

1�

2

VID0�

VID1�

VID2�

VID3

1�

2�

3�

4

8�

7�

6�

5

SHDN

VIN

GND

FB

IPOS

INEG

GATE

COMP

6

5

SENSE

FBKGND

VCC

4

3LTC1706-19

1µF

VCC 3.3V

VIN12V

10pF7.5k

1000pF

24µF

5.1Ω

220µF

IRFZ24

3.3V

VOUT1.27V–2.03IN 50.8mV STEPS

LT1575

gram the output voltages of regula-tors with different onboard references.Figure 5 shows the LTC1706-19programming the output of theLT1575, an UltraFast™ transientresponse, low dropout regulator thatis ideal for today’s power-hungry desk-top microprocessors. However, sincethe LT1575 has a 1.21V referenceinstead of a 1.19V reference, the out-

put will range from 1.27V to 2.03V insteps of 50.8mV.

The LTC1706-19 is the ideal com-panion chip to provide precise,programmable low-voltage outputs foran entire family of LTC DC/DC con-verters. Its compact size, compatibilityand high accuracy are just the rightfeatures for today’s portable electronicequipment.

Figure 4. High efficiency, low noise, synchronous step-down switching regulator with programmable output and external synchronization

Figure 5. UltraFast transient response, low dropout regulator with adjustable output voltage


DESIGN FEATURES

Fixed Frequency, 500kHz, 4.5AStep-Down Converter in an SO-8Operates from a 5V Input by Karl EdwardsIntroductionThe LT1506 is a 500kHz monolithicbuck mode switching regulator, func-tionally identical to the LT1374 butoptimized for lower input voltage ap-plications. Its high 4.5A switch ratingmakes this device suitable for use asthe primary regulator in small tomedium power systems. The smallSO-8 footprint and input operatingrange of 4V to 15V is ideal for localonboard regulators operating from5V or 12V system supplies. The 4.5Aswitch is included on the die, alongwith the necessary oscillator, controland logic circuitry to simplify design.The part’s high switching frequencyallows a considerable reduction inthe size of external components, pro-viding a compact overall solution.

The LT1506 is available in stan-dard 7-pin DD and fused-lead SO-8packages. It maintains high efficiencyover a wide output current range bykeeping quiescent supply current to4mA and by using a supply-boostcapacitor to saturate the power switch.The topology is current mode for fasttransient response and good loop sta-bility. Full cycle-by-cycle short-circuitprotection and thermal shutdown areprovided. Both fixed 3.3V and adjust-able output voltage parts are available.

5V to 3.3V Buck ConverterThe circuit in Figure 1 is a step-downconverter suitable for use as a localregulator to supply 3.3V logic from a5V power bus. The high efficiency,shown in Figure 2, removes the needfor bulky heat sinks or separate powerdevices, allowing the circuit to beplaced in confined locations. Sincethe boost circuit only needs 3V tooperate, the boost diode can still beconnected to the output, improvingefficiency. Figure 1’s circuit showsthe shutdown pin option. If this pin is

pulled to a logic low, the output isdisabled and the part goes into shut-down mode, reducing supply currentto 20µA. An internal pull-up ensurescorrect operation when the pin is leftopen. The SYNC pin, an option for theDD package, can be used to synchro-nize the internal oscillator to a systemclock. A logic-level clock signal appliedto the SYNC pin can synchronize theswitching frequency in the range of580kHz to 1MHz.

Current SharingMultiphase SupplyThe circuit in Figure 3 uses multipleLT1506s to produce a 5V, 12A powersupply. There are several advantagesto using a multiple switcher approachcompared to a single larger switcher.The inductor size is considerablyreduced. Inductor size is proportional

to the energy that needs to be storedin the core. Three 4A inductors storeless energy (1/2Li2) than a single 12Acoil, so they are much smaller. Inaddition, synchronizing three con-verters 120° out of phase with eachother reduces input and output ripplecurrents. This reduces the ripple rat-ing, size and cost of the filtercapacitors.

Current Sharing/Split Input SuppliesCurrent sharing is accomplished byconnecting the VC pins to a commoncompensation capacitor. The outputof the error amplifier is a gm stage, soany number of devices can be con-nected together. The effective gm ofthe composite error amplifier is theproduct of the individual devices. InFigure 3, the compensation capaci-tor, C4, has been increased by 3×.Tolerances in the reference voltagescause small offset currents to flowbetween the VC pins. The overall effectis that the loop regulates the outputat a voltage somewhere between theminimum and maximum referencesof the devices used. Switch-currentmatching between devices will betypically better than 300mA over thefull current range. The negativetemperature coefficient of the VC-to-switch-current transconductanceprevents current hogging.

BOOST

LT1506-3.3

VINOUTPUT3.3V4A

INPUT5V

1506 TA01

C20.68µF

CC1.5nF

D1MBRS330T3

C1100µF, 10VSOLIDTANTALUM

C310µF TO

50µFCERAMIC

D21N914

L15µH

VSW

SENSESHDN

OPENOR

HIGH= ON GND VC

+

+EF

FICI

ENCY

(%)

0 4LOAD CURRENT (A)

21 3

90�

85�

80�

75�

70

Figure 1. 5V to 3.3V step-down converter

Figure 2. Efficiency vs load current forFigure 1’s circuit


DESIGN FEATURES

A common VC voltage forces eachLT1506 to operate at the same switchcurrent, not at the same duty cycle.Each device operates at the duty cycledefined by its input voltage. This is auseful feature in a distributed powersystem. The input voltage to eachdevice could vary due to drops across

the backplane, copper losses, con-nectors and so on. The common VCsignal ensures that loading is stillshared between the devices.

SynchronizedRipple CurrentsA ring counter generates three syn-chronization signals at 600kHz, 33%duty cycle, phased 120° apart. Thesync input will operate over a widerange of duty cycles, so no furtherpulse conditioning is needed. At fullload, each device’s input ripple cur-rent is a 4A trapezoidal wave at600kHz, as shown in Figure 4. Sum-ming these waveforms gives theeffective input ripple for the completesystem. The resultant waveform,shown at the bottom of Figure 4,remains at 4A but its frequency hasincreased to 1.8MHz. The higher fre-quency eases the requirements onthe value of input filter without the 3×increase in ripple current rating thatwould normally occur. Although onlya single input capacitor is required,practical layout restrictions usuallydictate an individual capacitor at eachdevice. Figure 5 shows the outputripple current waveforms. The result-ant 1.8MHz triangular waveform hasa maximum amplitude of 350mA atan input voltage of 10V. This issignificantly lower than would beexpected for a 12A output. Interest-ingly, at inputs of 7.6V and 15V, the

++

C468nF25V

C1, C3: MARCON THCS50E1E106ZD1: ROHM RB051L-40D2: 1N914L1: DO3316P-682

+ C3C10µF25V

C2C330nF10V

D1C

D2C

1506 F15

L1C6.8µH

+ C110µF25V

5V12AR1

5.36k1%

R24.99k1%

1.8MHz

3-BIT RINGCOUNTER

+

+ C3B10µF25V

C2B330nF10V

D1B

D2BL1B6.8µH

+

+ C3A10µF25V

INPUT6V TO 15V

C2A330nF10V

D1A

D2AL1A6.8µH

VC SYNC SW GND

LT1506-SYNC

VIN BOOST FBVC SYNC SW GND

LT1506-SYNC

VIN BOOST FBVC SYNC SW GND

LT1506-SYNC

VIN BOOST FB

TIME

CURR

ENT

TIME

CURR

ENT

TIME

CURR

ENT

TIME

CURR

ENT

PHASE 1

PHASE 2

PHASE 3

TOTAL

TIME

CURR

ENT

TIME

CURR

ENT

TIME

CURR

ENT

TIME

CURR

ENT

PHASE 1

PHASE 2

PHASE 3

TOTAL

theoretical summed output ripplecurrent cancels completely. To reduceboard space and ripple voltage, C1and C3 are ceramic capacitors. Loopcompensation capacitor C4 must beadjusted when using ceramic outputcapacitors, due to the lack of effectiveseries resistance (ESR). The typical

Figure 3. Current-sharing 5V/12A supply

Figure 4. Input current Figure 5. Output current


DESIGN FEATURES

tantalum compensation value of 1.5nFis increased to 22nF (×3) for theceramic output capacitor. If synchro-nization is not used and the internaloscillators free run, the circuit willoperate correctly, but ripple cancella-tion will not occur. Input and outputcapacitors must be ripple rated forthe individual output currents.

Redundant OperationThe circuit shown in Figure 3 is faulttolerant when operating at less than8A of output current. If one powerstage fails open circuit, the outputwill remain in regulation. The feed-back loop will compensate by raisingthe voltage on the VC pin, increasingthe switch current of the two remain-ing devices.

5V to 3.3V at 2.5Aon 0.25in2 of board space,0.125in HighIn many space-sensitive applications,the component that dominates bothboard area and overall height is theinductor. One of the factors affectinginductor value choice is maximumripple current. Using the high cur-rent switch rating of the LT1506,

higher ripple current can be toler-ated, allowing the use of small, lowvalue, high current inductors. Aceramic output capacitor also reducesboard area and improves voltageripple. Using Figure 1’s circuit withthe SO-8 LT1506 and the componentchanges in Table 1, a very small, lowprofile, step-down converter can beimplemented.

Conclusion The LT1506 is a compact, easy touse, monolithic switcher. The inter-nal 4.5A switch covers a wide range ofmedium power applications. Its inputoperating range of 4V to 15V andavailability in SO-8 or DD packagesmake it ideal for very space-efficient,local onboard DC/DC converters.

traP eulaVV /rodne

#traP

3C,1C V01,Fµ22nikoT

Z622GZ0121

CC Fn22

1L Hµ2.2adimuS2R234DC

Table 1. Component changes for a low profileversion of Figure 1’s circuit

A circuit demonstrating the wideoutput range of the LTC1625 is shownin Figure 5. This application usesSi4412DY MOSFETs to deliver a 12Voutput at up to 2.2A. Note that theSYNC pin is tied high for 225kHzoperation in order to reduce theinductor size and ripple current.

1

EFFI

CIEN

CY (%

)

100�

95�

90�

85�

80

DI_1068_02a. EPS

3LOAD CURRENT (A)

2 4 50

VIN = 20VVOUT = 2.5V

LTC1625

LTC1435 +

+

1�

2�

3�

4�

5�

6�

7�

8

16�

15�

14�

13�

12�

11�

10�

9

EXTVCC

SYNC

RUN/SS

FCB

ITH

SGND

VOSENSE

VPROG

VIN

TK

SW

TG

BOOST

INTVCC

BG

PGND

LTC1625+

CSS 0.1µF

RC1

10k

CC11nF

CC2 220pF

RF 4.7Ω

CF0.1µF

CIN22µF35V× 2

M1Si4412DY

M2Si4412DY

L1* 39µH

R235.7k1%

R13.92k1%

COUT100µF16V0.030Ω ESR

CBO.1µF

CVCC4.7µF

VOUT12V/2.2A

VIN12V–28V

L1 = SUMIDA CDRH127-390MCDB = CMDSH-3

INTVCC

D1MBRS140T3

**DB

***Figure 4. Efficiency vs load current

Figure 5. 12V/2.2A adjustable-output supply

ConclusionThe LTC1625 step-down DC/DC con-troller offers true current mode controlwithout the expense and difficulty ofusing a sense resistor. Popular fea-tures from Linear Technology’s othercontrollers, such as fixed frequencyoperation, N-channel MOSFET drive,Burst Mode operation, soft-start andoutput voltage programming make

this controller useful in a variety ofapplications. By eliminating the powerloss in the sense resistor, even higherefficiencies can be achieved than werepreviously possible, making theLTC1625 an excellent choice forDC/DC converter designs requiringthe highest performance.

LTC1625, continued from page 4



DESIGN FEATURES

48 Volt Hot Swap Controllerfor Negative VoltagesIntroductionAs supply voltages for PC boards con-tinue to drop, designers face thedifficult task of minimizing the volt-age drops through distributed powersystems. At operating voltages of 3.3Vor lower, the voltage drops acrosspower busses, connector pins andinrush control circuitry can cause asupply voltage to drop out of toler-ance. A solution to this problem is todistribute power at a high voltage,commonly 48V, and then step thevoltage down to the final desired valueon each board in the system, usingpower modules.

Most 48V power modules requirean input bypass capacitor with a typi-cal value of hundreds of microfarads.When the board is hot-plugged into alive 48V power rail, the input capaci-tor can draw huge inrush currents asit charges. The inrush current cancause permanent damage to theboard’s components and createglitches on the system power supplythat can make the system functionimproperly.

The LTC1640H and LTC1640L pro-vide a simple, flexible solution to –48Vhot-swapping problems. The chipsallow a board to be safely inserted intoor removed from a live backplane witha supply voltage from –10V to –80V.They feature programmable inrushcurrent control, programmableundervoltage and overvoltage protec-

by Henry Yunand Robert Reay

tion, a programmable electroniccircuit breaker and direct power-mod-ule-enable control.

Power Supply ControlA typical LT1640 application is shownin Figure 1. The input voltage of thepower module on a circuit board iscontrolled by gradually increasing thegate voltage of the external N-channelpass MOSFET (Q1) in the power path.R1 provides current fault detectionand R2 prevents high frequencyoscillation. Resistors R4, R5 and R6provide undervoltage and overvoltagesensing. Resistor R3 and capacitorC2 act as a feedback network to accu-rately control the inrush current. Thewaveforms are shown in Figure 2. Theinrush current can be calculated withthe following equation:

IINRUSH = (45µA × CL)/ C2where CL is the total load capacitance.

Resistor R3 helps keep Q1 off whenthe power pins first make contact.When the power pins make contact,they bounce several times. While thecontacts are bouncing, the LT1640senses an undervoltage condition andthe GATE is immediately pulled low.

Once the power pins stop bounc-ing, the GATE pin starts increasinguntil when Q1 turns on and the GATEvoltage is held constant by the feed-back network of R3 and C2. When the

DRAIN voltage has finished increas-ing, the voltage on the GATE pin thenrises to its final value.

Electronic Circuit BreakerThe LT1640 features an electroniccircuit-breaker function that protectsagainst short circuits or excessivesupply currents. By placing a sense

+

VEE

VDD

LT1640L

SENSE

C133nF25V

C30.1µF100V

C4100µF100V

C5100µF16V

Q1IRF530

R210Ω5%

R310k5%

C23.3nF100V

R4562k

1%

R59.09k

1%

R610k1%

R10.02Ω

5%

4

OV

GND

3

2

–48V

OV = 71V

UV = 37V UV

5 6

8

7

1

GATE DRAIN

LUCENTJW050A1-E

PWRGD

95V

1640 TA01

8

7

6

5

1

2

4

+

VOUT+

SENSE+

TRIM

SENSE–

VOUT–

VIN+

ON/OFF

VIN–

Figure 1. Typical LT1604 application with a power module

Figure 2. Inrush-current control waveforms

Figure 3. Short-circuit protection waveforms


DESIGN FEATURES

resistor between the VEE and SENSEpins, the circuit breaker will be trippedwhenever the voltage across the senseresistor is greater than 50mV for morethan 3µs, as shown in Figure 3.

When the circuit breaker trips, theGATE pin is immediately pulled toVEE and the external N-channel MOS-FET is turned off. The GATE pin willremain low until the circuit breaker isreset by pulling UV low then high orcycling power to the part. A circuitthat automatically resets the circuitbreaker after a current fault is shownin figure 4.

Transistors Q2 and Q3, along withR7, R8, C4 and D1, form a program-mable one-shot circuit. Before a shortoccurs, the GATE pin is pulled highand Q3 is turned on, pulling node 2 toVEE. Resistor R8 turns off Q2. Whena short occurs, the GATE pin is pulledlow and Q3 turns off. Node 2 starts tocharge C4, and Q2 turns on, pullingthe UV pin low and resetting thecircuit breaker. As soon as C4 is fully

charged, R8 turns off Q2, UV goeshigh and the voltage on the GATEstarts to increase. Q3 turns back onand quickly pulls node 2 back to VEE.Diode D1 clamps node 3 one diodedrop below VEE. The duty cycle is setto 10% to pr event Q1 f romoverheating.

Undervoltage andOvervoltage DetectionThe UV (3) and OV (2) pins can beused to detect undervoltage and over-voltage conditions at the power supplyinput. The UV and OV pins are inter-nally connected to analog comparatorswith 20mV of hysteresis. When theUV pin falls below its threshold or theOV pin rises above its threshold, theGATE pin is immediately pulled low.The GATE pin will be held low untilUV is high and OV is low.

The undervoltage and overvoltagetrip voltages can be programmed usinga 3-resistor-divider, as shown inFigure 5.

VEE

VDD

LT1640L PWRGD

SENSE

C133nF25V

C41µF

100V C3100µF100V

Q1IRF530

R210Ω5%R8

510k5%

R310k5%

C23.3nF100V

R4562k

1%

R71M5%

R59.09k

1%

R610k1%

Q3ZVN3310

Q22N2222

D11N4148

R10.02Ω

5%

4

3

2OV

GND

–48V

UV

5 6

8

7

1

GATE DRAIN

1640 F10a

3

2

+

With R4 = 562k, R5 = 9.09k and R6= 10k, the undervoltage threshold isset to 37V and the overvoltage thresh-old is set to 71V.

PWRGD/PWRGD OutputThe PWRGD/PWRGD output can beused to directly enable a power mod-ule when the input voltage to themodule is within tolerance. TheLT1640H has a PWRGD output formodules with an active-high enableinput, and the LT1640L has a PWRGDoutput for modules with an active-low enable input.

When the DRAIN pin of theLT1640H is more than VPG (1.4V)above VEE (see Figure 6), internaltransistor Q3 is turned off and R7and Q2 clamp the PWRGD pin onediode drop (~0.7V) above the DRAINpin. Transistor Q2 sinks the module’spull-up current and the module turnsoff.

VEE

VDD

LT1640HLT1640L

R4

R5

R64

1640 F11

OV

GND

3

2

–48V

UV

8

VUV = 1.223R4 + R5+ R6

R5 + R6( )VOV = 1.223

R4 + R5+ R6R6( )

+

VEE

VDDLT1640H

SENSE

C1

C3

Q1

R2R3

C2

R4

R5

R6

R1

4

3

2OV

GND

–48V

UV

5 6

8

1

7

GATE

ACTIVE HIGHENABLE MODULE

VOUT+

VOUT–

VIN+

1640 F12

VIN–

PWRGD

DRAIN–

+

VEE

R75k

Q2

Q3

ON/OFF

+

–1.4V

Figure 4a. This circuit resets the circuit breaker after a current fault.Figure 4b. Waveforms of Figure 4a’s circuit

Figure 5. Undervoltage and overvoltagesensing Figure 6. Active-high enable module


DESIGN FEATURES

+

VEE

VDDLT1640L

SENSE

C1

C3

Q1

R2R3

C2

R4

R5

R6

R1

4

3

2OV

GND

–48V

UV

5 6

8

1

7

GATE

ACTIVE LOWENABLE MODULE

VOUT+

VOUT–

VIN+

1640 F13

VIN–

PWRGD

DRAIN–

+ VEE

Q2ON/OFF

+ –1.8V

VEE

VDD

LT1640L PWRGD

SENSE

C133nF25V

C3100µF100V

Q1IRF530

R210Ω5%

R751k5%

R310k5%

C23.3nF100V

R4562k

1%

R59.09k

1%

R610k1%

R10.02Ω

5%

4

3

2OV

GND

–48V

UV

5 6

8

7

14N25

GATE DRAIN

1640 F14

PWRGD

+

When the DRAIN pin drops belowVPG, Q3 will turn on, shorting thebottom of R7 to VEE and turning Q2off. The pull-up current in the modulethen flows through the R7, pulling thePWRGD pin high and enabling themodule.

When the DRAIN pin of the LT1640Lis more than VPG (1.4V) above VEE, theinternal pull-down transistor, Q2, isoff and the PWRGD pin is in highimpedance state (see Figure 7). ThePWRGD pin will be pulled high by themodule’s internal pull-up currentsource, turning the module off. Whenthe DRAIN pin drops below VPG , Q2will turn on, and the PWRGD pin willbe pulled low, enabling the module.

The PWRGD signal can also beused to turn on an LED or optoisola-tor to indicate that the power is good,as shown in Figure 8.

Figure 7. Active-low enable module

Figure 8. Using PWRGD to drive an optoisolator

Gate Pin Voltage RegulationWhen the supply voltage to the chip ismore than 15.5V, the GATE pin volt-age is regulated at 13.5V above VEE. Ifthe supply voltage is less than 15.5V,

the GATE voltage will be about 2Vbelow the supply voltage. At the mini-mum 10V supply voltage, the gatevoltage is guaranteed to be greaterthan 6V and no greater than 18V forsupply voltages up to 80V.

ConclusionLT1640 provides a simple and flexiblesolution for hot swap applications. Itis the first part that allows systemdesigners to connect an 80V supplydirectly to the chip without any volt-age step-down circuitry. It can beprogrammed to control the outputvoltage slew up rate and the inrushcurrent. It has programmable under-voltage and overvoltage protection, andthe PWRGD/PWRGD output can betied directly to a power-module enablepin. The LT1640 simplifies the designof high voltage hot-swap control sys-tems and combines all of these featuresin an 8-pin SO/PDIP package.


forthe latest information

on LTC products, visit

www.linear-tech.com


DESIGN FEATURES

Hot Swapping the PCI BusUsing the LTC1643 by Robert ReayHot Circuit InsertionWhen a circuit board is inserted intoa live PCI slot, the supply bypasscapacitors on the board can drawhuge transient currents from the PCIpower bus as they charge. The tran-sient currents can cause permanentdamage to the connector pins andcreate glitches on the power bus, caus-ing other boards in the system toreset.

The LTC1643 is designed to turn aboard’s supply voltages on and off ina controlled manner, allowing theboard to be safely inserted into orremoved from a live PCI slot withoutcausing glitches on the system powersupplies. The chip also protects thePCI supplies from short circuits andmonitors the supply voltages.

The LTC1643H is designed formotherboard applications, whereasthe LTC1643L is designed for Com-pactPCI applications where the chipresides on the plug-in board. The partis available in the space saving 16-pinSSOP package.

LTC1643 Features❏ Allows safe insertion into and

removal from either amotherboard (LTC1643H) orCompactPCI board (LTC1643L)

❏ Controls all four PCI supplies:–12V, 12V, 3.3V and 5.0V

❏ Programmable foldback currentlimit: a programmable analogcurrent limit with a value thatdepends on the output voltage. Ifthe output is shorted to ground,the current limit drops to keeppower dissipation and supplyglitches to a minimum.

❏ Programmable circuit breaker: ifa supply remains in current limittoo long, the circuit breaker willtrip, the supplies will be turnedoff and the FAULT pin will bepulled low.

❏ Current-limit power up: thesupplies are allowed to power upin current limit. This allows thechip to power-up boards withwidely varying capacitive loadswithout tripping the circuitbreaker. The maximum allowablepower-up time is programmableusing the TIMER pin.

❏ 12V and –12V power switcheson-chip

❏ Power-good output: monitors thevoltage status of the four supplyvoltages.

PCI Power RequirementsAll PCI connectors require four powerrails: 5V, 3.3V, 12V and –12V. Sys-tems implementing the 3.3V signalingenvironment are always required toprovide all four rails in every system.Systems implementing the 5V signal-ing environment may either ship the3.3V supply with the system or pro-vide a means to add it afterward. Thetolerance of the supplies as measuredat the plug-in card is summarized inTable 1.

Power-Up SequenceThe power supplies are controlled byexternal N-channel pass transistorsQ1 and Q2 in the 3.3V and 5.0Vpower paths and internal pass tran-sistors for the 12V and –12V powerpaths (see Figure 1).

Resistors R1 and R2 providecurrent-fault detection and R7 andC1 provide current-control loop com-pensation. Resistors R5 and R6prevent high frequency oscillations inQ1 and Q2.

When the ON pin (5) is pulled high,the pass transistors are allowed toturn on and a 20µA current source isconnected to the TIMER pin (4) (seeFigure 2). The current in each passtransistor increases until it reachesthe current limit for each supply.

R20.005Ω

Q2IRF7413

Q1IRF7413

R10.007Ω

R510Ω

R610Ω

R71k

C10.047µF

5V5A

3.3V7.5A

GND

–12V100mA

12V500mA*

R310k

*R4

10k

11 3 13 12 14

16

15

4

109

812

5

6

7

3VINGND12VINVEEIN

ON

FAULT

PWRGD

3VSENSE 5VSENSE3VOUT

LTC1643H

5VOUT

12VOUT

VEEOUT

TIMER

5VINGATE

1643 TA01

SYSTEMPOWERSUPPLY

PCIPOWERSYSTEM

CONTROLLER C20.1µF

RESETLOGIC

RESET*CONNECT TO SYSTEM CONTROLLER LOGIC-LEVEL SUPPLY

liaRrewoP tnerruClacipyT

ecnaticapaC

%5±V5 A5 Fµ0003<V3.0±V3.3 A5.7 Fµ0003<

%5±V21 Am005 Fµ033<%01±V21– Am001 Fµ021<

Table 1. PCI power supply requirements

Figure 1. Typical LTC1643H application


DESIGN FEATURES

Each supply is then allowed to powerup at a rate controlled either by thecurrent limit into the load capaci-tance or dv/dt = 50µA/C1, whicheveris slower. Current limit faults areignored until the voltage on the TIMERpin (4) reaches 1V below the 12VIN pin(1). Once all four supply voltages arewithin tolerance, the PWRGD pin (7)will be pulled low.

Power-Down SequenceWhen the ON pin (5) is pulled low, apower-down sequence begins (seeFigure 3). Internal switches are con-nected to each of the output supplyvoltage pins to discharge the bypasscapacitors to ground. The TIMER pin(4) is immediately pulled low. Thevoltage on the GATE pin (11) is dis-charged by a 200µA current source toprevent the load currents on the 3.3Vand 5.0V supplies from going to zero

instantaneously and causing glitcheson the power supplies. When any ofthe output voltages dips below itsthreshold, the PWRGD pin (7) is pulledhigh.

TimerDuring a power-up sequence, a 20µAcurrent source is connected to theTIMER pin (4) and current limit faultsare ignored until the voltage on thispin increases to within 1V of the12VIN pin (1). This feature allows thechip to power-up PCI cards with widelyvarying capacitive loads on the sup-plies. The power-up time will be:

tON = 2 × (CSUPPLY × VSUPPLY)/(ILIMIT – ILOAD)For CSUPPLY = 2000µF, VSUPPLY = 5V,

ILIMIT = 7A, ILOAD = 5A the turn on timewill be ~10ms. The timer period shouldbe set longer than the maximum sup-ply turn-on time but short enough tonot exceed the maximum safe operat-ing area of the pass transistor duringa short circuit. The TIMER period willbe:

tTIMER = (CTIMER × 11.1V)/22µAFor CTIMER = 0.1µF, the TIMER

period will be ~50ms. The TIMER pin

ON

TIMER

GATE

SUPPLYOUTPUTS

PWRGD

FAULT

dV/dt = 20µA/C2

dV/dt = 50µA/C1 ORSET BY CURRENT LIMIT

ON

TIMER

dV/dt = 200µA/C1GATE

SUPPLYOUTPUTS

PWRGD

FAULT

ON

TIMER

GATE

SUPPLYOUTPUTS

PWRGD

FAULT

1643 F04

Figure 2. Normal power-up sequence Figure 3. Normal power-down sequence

Figure 4. Power-up into a short circuit on a 3.3V output


DESIGN FEATURES

(4) is immediately pulled low whenthe ON pin (5) goes low.

Thermal ShutdownThe internal switches for the 12V and–12V supplies are protected by aninternal current limit and thermalshutdown circuit. When the tempera-ture of the chip reaches 150°C, allswitches will be latched off and theFAULT pin (6) will be pulled low.

Short-Circuit ProtectionDuring a normal power-up sequence,if the voltage on the TIMER pin (4) hasreached its maximum and a supply isstill in current limit, all of the passtransistors will be immediately turnedoff and the voltage on the FAULT pin(6) will be a logic low, as shown inFigure 4. If a short circuit occurs afterthe supplies are powered-up, theshorted supply’s current will dropimmediately to the limit value (seeFigure 5).

If the supply remains in currentlimit for more than 15µs, all of thesupplies will be latched off. The 15µsdelay prevents quick current spikes,for example from a fan turning on,from causing false trips of the circuitbreaker. The chip will stay in thelatched-off state until the ON pin (5)

ON

TIMER

GATE

SUPPLYOUTPUTS

15µs

PWRGD

FAULT

1643 F05

1

2

R20.005Ω

VID

Q2IRF7413

Q1IRF7413

R10.007Ω

R510Ω

R610Ω R7

1k

C10.047µF

*C40.1µF

*C30.1µF

5V5A5V

3.3V

BD_SEL#

HEALTHY#

12V

–12V

GND

3.3V7.5A

–12V100mA

12V500mA

R310k

R810k

VID

R1110k

11 3 13 12 14

16

15

4

8

109

8

5

6

7

3VIN

12VIN

VEEIN

ON

FAULT

PWR

linear technologlinear technology - analog deviceslinear technology magazine • august 1998 3...

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