an 64-1a -- fundamentals of rf and microwave power measurements

8/13/2019 An 64-1A -- Fundamentals of RF and Microwave Power Measurements

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Fundamentals of RF and MicrowavePower Measurements

Application Note 64-1A

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Application Note 64-1A

Fundamenta ls o f RF a nd

Microwa ve Power Measurements


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Table of Contents

I. Introduction

The Importa nce of P ow er . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

A Br ief History o f P ower Measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

II. Power Measurement

U nits a nd Definit ions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4

Three Methods of Sensing Power . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

Key Power Sensor P ara meters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

The Hierarchy of Power Measurement , Nat ional Standa rds and Traceabil i ty . . . . . . . . . . . . . . . 9

A New S ensor for Power Reference Tra nsfer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11

III. Thermistor Sensors and Instrumentation

Thermis tor S ensors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12

Coaxial Thermistor Sensors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

Waveguide Thermistor Sensors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

B ridges, from Wheat stone to Dua l-Compensa ted DC Types . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13

Thermistors as P ower Tra nsfer Sta nda rds . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15

Other DC -Substit ution Meters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15

IV. Thermocouple Sensors and Instrumentation

P rinciples of Thermocouples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17

The Thermocouple Sensor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18

P ower Meters for Thermocouple Sensors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

Reference Oscillator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

HP EP M Series Power Meters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24

V. Diode Sensors and Instrumentation

Diode Detector P rinciples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26

U sing Diodes for Sensing P ower . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28

New Wide-Dyn am ic-Ra nge CW-Only P ower Sensors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30

A New Versa t i le P ower Meter to Exploit 90 dB Range Sensors . . . . . . . . . . . . . . . . . . . . . . . . . . . 31

Tra ceable P ow er Reference . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33

Signa l Waveform Effects on the Measur ement Uncerta inty of Diode Sensors . . . . . . . . . . . . . . . . 34

VI. Measurement Uncertainty

P ower Tra nsfer, Genera tors an d Loads . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

RF C ircuit Descriptions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37

Reflection C oefficient . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39

Signa l Flowgr aph Visualiza t ion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40Mismat ch Uncertaint y . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44

Mismat ch Loss . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45

Other Sensor Un certa inties . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45

Ca libration Fa ctor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46

P ower Meter Inst rumenta tion Uncertaint ies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47

Ca lculat ing Tota l Uncerta inty . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49

P ower Measurement E quat ion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50

Worst Ca se Uncerta inty . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52


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RSS U ncerta inty . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52

New Method of Combining P ower Meter U ncerta int ies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54

P ower Measurement Model for IS O P rocess . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 55

St an dar d U ncerta inty of Misma tch Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57Exa mple of Ca lculat ion of U ncerta inty U sing ISO Model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 58

VII. Power Measurement Instrumentation Compared

Accura cy vs. Power Level . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62

Frequency Ran ge and S WR (Reflection Coefficient) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 64

Speed of Response . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65

Automat ed Power Measurement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 66

Susceptibility to Overload . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 67

Signa l Waveform E ffects . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68

VIII. Peak Power Measurements

A B rief History of P eak P ower Measur ements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 69P eak P ower Ana lyzers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 69

IEE E Video P ulse Sta ndards Adapted for Microwa ve Pulses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 71

P eak P ower Waveform Definit ions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 72

Measuring Complex Waveforms other tha n P ulsed Power . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 73

Glossary and List of Symbols . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75

P ulse ter ms a nd definit ions, Figu res 8-3 and 8-4, reprint ed from IE EE STD 194-1977 and ANSI /IE EE STD 181-1977, Copyright 1977 by the Ins titut e of Electrical and Electronics Eng ineers, Inc. The IEE E disclaims a ny responsibility or liability resultingfrom the placement a nd use in this publicat ion. Informa tion is reprinted with t he permission of the IEE E.


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I. Introduction

This a pplicat ion not e, AN64-1A, is a ma jor revision of t he 1977 edition

of AN64-1, w hich ha s served for ma ny yea rs a s a key reference for RF

an d microwa ve power measurement. It w as w ritten for tw o purposes:

1) to retain some of the original text of the funda menta ls of RF a ndmicrowa ve power m easurements, w hich tends to be timeless, an d

2) to present more modern power m easurement techniques and test

equipment wh ich represents the current st at e-of-the-ar t.

This note reviews t he popular t echniques a nd instr uments used for meas-

uring power, discusses error mechanisms, and gives principles for cal-

culat ing overa ll measurement un certa inty. It describes metrology-oriented

issues, such as th e basic national sta nda rds, round robin intercompar isons

an d tr aceability processes. These will help users to establish a n unbroken

chain of calibra tion actions from the NIS T (U.S . Nat iona l Inst itute of

Sta nda rds a nd Technology) or other na tional sta nda rd bodies, down t o

the fina l measurement setup on a production test l ine or a communicat ion

tower a t a remote mounta intop. This note also discusses new measure-

ment uncerta inty processes such as t he new I SO Gui de to the Expr essionof Un certai nt i es in M easur ement, a nd t he U SA version, ANSI /NCS L 540Z-

2-1996, U .S. Gui de for Expr ession of U ncert ain ty i n M easur ement, which

defines new a pproaches to ha ndling uncerta inty calculations.

This introductory chapter r eviews t he importa nce of power qua ntities.

Cha pter II discusses units, defines terms such as avera ge power and

pulse power, reviews key sensors a nd t heir para meters, an d overviews

the hierarchy of power standa rds an d the path of tra ceability to the

United Sta tes National Reference Sta ndard. Cha pters III , IV, and V

detail instrument at ion for measuring power w ith th e three most popular

power sensing devices: thermistors, thermocouples, and diode detectors.

Cha pter VI covers power tra nsfer, signal flowgra ph an alysis an d mis-

mat ch uncerta inty, along with t he remaining uncertaint ies of power

instrument at ion an d the calculation of overall uncerta inty. Cha pter VIIcompares t he thr ee popular methods for measuring avera ge power.

P eak an d pulse power measur ement and mea surement of signals

with complex modulations ar e discussed in Cha pter VIII.

The Importance of Power

A systems output power level is frequently the critical factor in the design,

an d ultimat ely the purcha se and performance of almost a ll radio frequency

an d microw a ve equipment . The first key factor is the concept of equity in

tr a de. When a customer purcha ses a product wit h specified power perfor-

man ce for a negotiated price, th e final production-line test results need to

ag ree w ith th e customer s incoming inspection dat a . These receiving,

installation or commissioning phases, often occur at different locations,

an d sometimes across national borders. The various measurements must

be consistent w ithin a cceptable uncertaint ies.

Secondly, measur ement uncerta inties cause a mbiguities in realizable

performa nce of a tr an smitter. For example, a ten-wat t tra nsmitt er costs

more than a five-wa tt tra nsmitter. Twice the power output m eans t wice the

geogra phical a rea is covered or 40 percent more radia l ra nge for a commu-

nicat ion system. Yet, i f the overa ll measur ement uncerta inty of the fina l

product t est is on the order of 0.5 dB, t he unit a ctually shipped could ha ve

output power as much a s 10%lower tha n th e customer expects, w ith result-

ing lower hea droom in its operating profiles.


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Because signa l power level is so importa nt to th e overa ll system perform-

an ce, it is a lso critical wh en specifying the components tha t build up t he

system. Ea ch component of a signa l cha in must receive the proper signal

level from th e previous component a nd pa ss th e proper level to the suc-ceeding component. P ower is so importa nt t ha t it is frequently measured

tw ice at each level, once by the vendor a nd a gain a t t he incoming inspec-

tion sta tions before beginning th e next a ssembly level.

It is at the higher operat ing power levels wh ere each decibel increase in

power level becomes more costly in t erms of complexity of design, expense

of active devices, skill in manufacture, difficulty of testing, and degree of

reliability. The increased cost per dB of power level is especially tr ue at

microwave frequencies, where the high-power solid state devices are inher-

ently more costly a nd t he gua rd-ban ds designed into the circuits t o avoid

maximum device stress are also quite costly.

Man y systems a re continuously monitored for output power dur ing

ordinary operation. This large num ber of power measurements an d th eirimporta nce dicta tes tha t the measurement equipment an d techniques

be accura te, repeata ble, tra ceable, and convenient. The goal of this

HP applicat ion note, and others, is to guide the reader in ma king those

measurement qu alities routine.

Because ma ny of th e examples cited above used the term signal level,

the na tura l tendency might be to suggest measur ing voltage instea d of

power. At low freq uencies, below a bout 100 kHz , power is usua lly ca lculat-

ed from volta ge measurements a cross a known impedance. As the frequen-

cy increases, the impedance has la rge varia tions, so power measur ements

become more popular, a nd voltage or current ar e calculat ed para meters.

At frequencies from about 30 MHz on up through the optical spectrum,

the direct mea surement of power is more accura te an d easier. Anotherexample of decreased usefulness is in w aveguide tra nsmission configura -

tions w here volta ge an d current conditions a re more difficult t o define.

A Brief History of Power Measurement

From the ear liest design an d applicat ion of RF a nd microwa ve systems,

it wa s necessary t o determine the level of power output. Some of the

techniques were quite primitive by todays sta nda rds. For example, when

Sigurd a nd Russell Var ian, t he inventors of the klystron microwa ve power

tube in t he lat e 1930s, were in t he early experimenta l sta ges of their

klystron cavity, the detection diodes of the da y w ere not adequa te for those

microwa ve frequencies. The story is t old tha t Ru ssell cleverly drilled a

small hole at the a ppropriate position in the klystron cavity w all , an d

positioned a fluorescent screen a longside. This technique wa s adequa te

to reveal w hether th e cavit y wa s in oscillat ion a nd to give a gross indica-tion of power level changes a s va rious drive conditions w ere adjusted.

Ea rly measur ements of high power syst em signals w ere accomplished

by ar ra nging to a bsorb the bulk of the syst em power int o some sort of ter-

minat ion an d measuring th e heat buildup versus time. A simple example

used for high power rada r systems wa s the wa ter-flow calorimeter. These

were ma de by fa bricat ing a glass or low-dielectric-loss tube t hrough t he

sidewal l of the waveguide at a shal low a ngle. Since the water wa s an

excellent absorber of the microwave energy, the power measurement

required only a mea surement of the heat rise of the wa ter from input to

output, an d a m easure of the volumetric flow versus time. The useful part


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of that technique was t ha t th e wa ter flow a lso carried off the considera ble

heat from the source under test a t th e same time it wa s measuring the

desired pa ra meter.

G oing into World Wa r II , as det ection cryst a l technology grew from th e

early ga lena cat -wh iskers, detectors became more rugged and performed

to higher RF an d microwa ve frequencies. They were better mat ched to

tra nsmission lines, and by u sing comparison techniques with sensitive

detectors, unknown microwa ve power could be measured a gainst known

values of power generated by calibrat ed signal generators.

P ower substitut ion methods emerged wit h th e advent of sensing elements

wh ich w ere designed to couple tra nsmission line power int o the sensing

element.1 Barretters were positive-temperature-coefficient elements,

typically meta llic fuses, but th ey were frustra tingly fra gile an d easy to

burn out. Thermist or sensors exhibited a nega tive temperat ure coefficient

an d were much more rugged. B y including such sensing elements a s one

arm of a 4-arm balanced bridge, DC or low-frequency AC power could bewith dra wn as RF /MW power w as a pplied, maint aining th e bridge balance

an d yielding a substitut ion va lue of power.2

Commercial calorimeters ha d a place in early measurements. Dry

calorimeters absorbed system power a nd by measur ement of heat rise

versus t ime, were a ble to determine syst em power. The 1960s HP 434A

power meter w as a n oil-flow ca lorimeter, with a 10 wa tt top range, an d

also used a heat comparison betw een th e RF load a nd a nother identical

load driven by DC power.3 Wa ter-flow ca lorimeters w ere offered for

medium to high power levels.

This application note will allot most of its space to the more modern,

convenient an d w ider dynam ic ra nge sensor technologies which have

developed since those ea rly da ys of RF a nd microwa ve. Yet, it is hopedtha t some appreciation w ill be reserved for th ose ear ly developers in t his

field for having endured the inconvenience and primitive equipment of

those times.

_________

1. B .P. Ha nd, Direct Reading UH F Power Measurement,

Hew lett-Pa ckard J ourna l, Vol. 1, No. 59 (Ma y, 1950).

2. E.L. G inzton, Microwa ve Measur ements, McG ra w-Hill, Inc., 1957.

3. B.P. Hand, An Automatic DC to X-Ba nd P ower Meter for the Medium

P ower Ra nge, Hew lett-Pa ckar d J ourna l, Vol. 9, No. 12 (Aug., 1958).


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Units and Definitions

WattThe Interna tional Syst em of Units (SI) ha s established the wa tt (W) as

th e unit of power; one wa tt is one joule per second. Int erestingly, electr icalqua ntities do not even enter into this definition of power. In fa ct, other

electrical units a re derived from the wa tt. A volt is one wa tt per a mpere.

B y the use of appropriate sta nda rd prefixes the wa tt becomes the kilowa tt

(1 kW = 103W), milliwa tt (1 mW = 10-3W), microwa tt (1 W = 10-6W),

na nowa tt (1 nW = 10-9W), etc.

dB

In ma ny cases, such a s when measuring gain or a t tenuation, the rat io of

tw o powers, or relative power, is frequently the desired qua ntity r at her tha n

a bsolute pow er. Relat ive power is the ra tio of one power level, P, to some

other level or r eference level, P re f. The ra tio is dimensionless beca use th e

units of both the numerat or and denomina tor are wa tt s. Relat ive power is

usua lly expressed in decibels (dB )

The dB is defined by

The use of dB h as t wo adva nta ges. First, th e range of numbers commonly

used is more compa ct; for example + 63 dB to -153 dB is more concise tha n

2 x 106 to 0.5 x 10-15. The second adva nta ge is apparent w hen it is necessary

to find the gain of severa l ca scaded devices. Multiplicat ion of numeric gain

is then replaced by the a ddition of the power ga in in dB for each device.

dBmP opular usa ge has added an other convenient unit, dBm . The formula for

dB m is simila r t o (2-1) except th e denomina tor, P re f is alwa ys one mil liwat t :

In t his expression, P is expressed in milliwa tt s and is th e only varia ble, so

dBm is used as a measure of absolute power. An oscillator, for example, may

be sa id to ha ve a power output of 13 dB m. B y solving for P in (2-2), the

power output ca n a lso be expressed a s 20 mW. So dBm means dB above

one milliwa tt, (no sign is assumed + ) but a negat ive dBm is to be interpret-

ed as dB below one milliwa tt. The adva nta ges of the term dBm pa ra llel

those for dB ; it uses compact numbers an d aids t he use of addition instead

of multiplicat ion w hen cascading ga ins or losses in a t ra nsmission system.

PowerThe term a verage power is very popular an d is used in specifying a lmost

all RF a nd microwa ve systems. The terms pulse power an d peak envelope

power are more pertinent to radar and navigation systems.

In elementar y th eory, power is said to be the product of volta ge and current.

B ut for a n AC voltage cycle, this product V x I var ies during the cycle as

shown by curve p in Figure 2-1, according to a 2f relat ionship. From tha texample, a sinusoidal genera tor produces a sinusoidal current a s expected,

but th e product of volta ge and current ha s a D C term a s well as a component

a t tw ice th e genera tor frequen cy. The word power, as most commonly

used, refers t o tha t D C component of the power product.

dB m = 10 log10 ( )P1 m W (2-2)

dB = 10 log10 ( )PP re f (2-1)

II. Power Measurement


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All the m ethods of mea surin g power t o be discussed (except for one cha p-

ter on peak power measur ement) use power sensors w hich, by averaging,

respond to the DC component. P eak power instrum ents an d sensors ha ve

time constants in the sub-microsecond region, allowing measurement of

pulsed power w aveforms.

The funda ment a l definition of power is energy per unit t ime. This corr-

esponds with th e definition of a w at t a s energy tra nsfer at t he rat e of one

joule per second. The importa nt q uestion to resolve is over wh a t tim e is the

energy tra nsfer rat e to be averaged wh en measuring or computing power?

From Figure 2-1 i t is clear tha t if a na rrow time interval is shifted ar oundwith in one cycle, var ying answ ers for energy tra nsfer rat e ar e found. B ut

at radio and microwave frequencies, such microscopic views of the voltage-

current product a re not common. For th is applicat ion note, power is defined

as t he energy tra nsfer per unit time avera ged over man y periods of the low-

est frequ ency (RF or microw a ve) involved.

A more ma thema tical a pproach t o power for a continuous w ave (CW) is to

find the avera ge height under the curve of P in Figure 2-1. Avera ging isdone by finding the area un der the curve, tha t is by integrat ing, and th en

dividing by the length of time over w hich th at ar ea is ta ken. The length of

time should be an exa ct number of AC periods. The power of a CW signa l

a t freq uency (l/T0) is:

wh ere T0 is the AC period, ep and ip represent peak values of e an d i ,

is the phase angle between e an d i , an d n is the number of AC periods.

This y ields (for n = 1, 2, 3 . . .):

If th e integrat ion t ime is many AC periods long, then, wh ether n is a precise

integer or not ma kes a van ishingly small difference. This result for large n

is the basis of power measurement.

For sinusoidal signals, circuit t heory shows t he relat ionship between peak

and rms values as :

U sing t hese in (2-4) yields th e fam iliar expression for power :

DC Component

e

P

Amplitude

R

e

i

t

i

P = 1

nT0(2-3)

nT0

e

0

ep sin ( ) ip sin ( ) dt2T0 t2

T0

P = cos epip

2(2-4)

+

Figure 2-1.The product of volt-age and current, P,varies during thesinusoidal cycle.

ep = 2 E rm s and ip = 2 I rm s (2-5)

P = E rm s I rm s cos (2-6)


12/83

6

Average Power

Avera ge power, like the other pow er term s to be defined, places furth er

restrictions on the a veraging time th an just man y periods of the highest

frequency. Average power mea ns tha t th e energy tra nsfer rat e is to beavera ged over man y periods of the lowest frequency involved. For a CW

signal, the lowest frequency and highest frequency are t he same, so aver-

age power and power are the same. For an amplitude modulated wave,

the power must be avera ged over ma ny periods of th e modulation compo-

nent of the signal a s well .

In a m ore mat hemat ical sense, avera ge power can be writ ten as:

where T,

is the period of the lowest fr equency component of e(t) and i(t).

The a veraging t ime for a verage power sensors an d meters is t ypically from

severa l hundr edths of a second to several seconds a nd t herefore this pro-

cess obtains the a verage of most common forms of am plitude modulat ion.

Pulse PowerFor pulse power, the energy tra nsfer rat e is averaged over th e pulse width,

. Pulse width is considered t o be th e time betw een the 50 percent rise-time/fa lltime a mplitude points .

Mat hemat ically, pulse power is given by

B y its very definition, pulse power avera ges out a ny a berrat ions in the

pulse envelope such as overshoot or ringing. For th is reason it is called

pulse power a nd not peak power or peak pulse power a s is done in ma ny

ra dar references. The terms peak power an d peak pulse power a re not

used here for that r eason. See Chapt er VIII for more explana tion of mea-

surements of pulsed power.

The definition of pulse power h a s been extended since the ear ly da ys of

microwave to be:

where duty cycle is the pulse width times the repetition frequency.

This extended definition, which can be derived from (2-7) and (2-8) for

recta ngular pulses, allows calculat ion of pulse power from an avera ge

power mea surement a nd th e duty cycle.

For microwa ve systems w hich are designed for a fixed dut y cycle, peak

power is often calculat ed by use of the dut y cycle calculat ion a long w ith

an a verage power sensor. One reason is tha t th e instr umenta tion is less

expensive, and in a t echnical sense, the a veraging technique integrat es all

the pulse imperfections int o the a verage.

P p =1

(2-8)

e

0

e(t) i(t)dt

P p =P a vg

Duty Cycle(2-9)

P a vg =1

nT,

(2-7)

nT,

e

0

e(t) i(t)dt


13/83

7

The evolution of highly sophisticated ra dar, electronic wa rfar e an d na viga-

tion systems, often based on complex pulsed and spread spectrum technolo-

gy, ha s led to more sophisticated instr umenta tion for chara cterizing pulsed

RF pow er. To present a more inclusive pictu re of a ll pulsed power mea sure-ments, Chapter VIII P eak P ower Instrumentat ion is presented later in this

note. Theory and pra ctice and deta iled wa veform definitions a re presented

for th ose applicat ions.

.

Peak Envelope PowerFor certain more sophisticat ed microwa ve applicat ions, a nd because of the

need for greater accuracy, the concept of pulse power is not totally satisfac-

tory. Difficulties arise when t he pulse is intentionally non recta ngular or

wh en aberra tions do not allow an a ccura te determination of pulse width .Figure 2-3shows a n example of a G au ssian pulse shape, used in certa inna viga tion sy stems, wh ere pulse power, by eit her (2-8) or (2-9), does not

give a tr ue picture of power in the pulse. P eak envelope power is a term

for describing th e ma ximum power. En velope power will first be discussed.

En velope power is measur ed by making th e avera ging time much less tha n

1/fm wh ere fm is the ma ximum frequency component of th e modulation

w a veform. The avera ging time is therefore limited on both ends: (1) it must

be small compared to t he period of th e highest m odulat ion frequency, a nd

(2) it mus t be lar ge enough to be ma ny RF cycles long.

B y continuously d isplayin g th e envelope pow er on a n oscilloscope, (using a

detector operating in its square-law range), the oscilloscope trace will show

the power profile of the pulse shape. (Squa re law means t he detected out-

put voltage is proportional t o the input RF power, that is the squa re of the

input voltage.) P eak envelope power, then, is the ma ximum value of the

envelope power (see Figure 2-3). For perfectly recta ngula r pulses, peakenvelope power is equa l to pulse power a s defined above. P eak power an a-

lyzers are specifically designed to completely characterize such waveforms.

See Chapter VIII .

t

P

Pp

Pavg

Tr =Duty Cycle

1fr

= fr

Figure 2-2.Pulse power Pp isaveraged overthe pulse width.

Peak EnvelopePower

,

,

,

,

PPp using (2-9)

Pp using (2-8)

InstantaneousPower at t=t1

t1 t

Figure 2-3.A Gaussian pulseand the differentkinds of power.


14/83

8

Avera ge pow er, pulse power, and pea k envelope power a ll yield the sa me

an swer for a CW signal. Of all power measurements, avera ge power is the

most frequently measured becau se of convenient m easurement equipment

with h ighly accura te and t ra ceable specificat ions. P ulse power a nd peakenvelope power can often be calculat ed from an avera ge power measurement

by knowing t he duty cycle. Avera ge power mea suremen ts th erefore occupy

the grea test portion of this a pplicat ion n ote series.

Other Waveforms and Multiple SignalsThe recent explosion of new RF and microwave systems which depend on

signal forma ts oth er tha n simple pulse or AM/FM modulat ion h as ma de power

measuring t echniques more critical. Modern systems use fast digita l phase-

shift-keyed m odulat ions, w ide-channel, multiple carrier signa ls, an d other

complex formats which complicate selection of sensor types. In particular,

the popular diode sensors a re in demand becau se of their w ide dynamic ra nge.

B ut t he sophisticated sha ping circuits need careful ana lysis when used in

non-CW signa l environm ents. More explana tion is deta iled in Cha pter V.

Three Methods of Sensing PowerThere are t hree popular devices for sensing a nd mea suring a verage power at

RF a nd microwa ve frequencies. Ea ch of the methods uses a different kind of

device to convert t he RF power t o a mea sura ble DC or low frequency signal.

The devices a re th e therm istor, the th ermocouple, and t he diode detector.

Ea ch of the next t hree chapters discusses in deta il one of those devices a nd

i ts associated instrumenta t ion. Ea ch method has some advan tages and disad-

van ta ges over the others. After the individual measur ement sensors ar e stud-

ied, the overall measurement errors ar e discussed in Cha pter VI. Then the

results of the th ree methods ar e summarized an d compared in Cha pter VII.

The general measur ement technique for a verage power is to at ta ch a properly

calibrated sensor to the tra nsmission line port at wh ich the unknown power

is to be measured. The output from t he sensor is connected to an appropriatepower meter. The RF power to th e sensor is t urned off an d th e power meter

zeroed. This opera tion is often referred t o a s zero sett ing or zeroing.

P ower is then t urned on. The sensor, reacting to the new input level, sends

a signa l to the power meter a nd th e new meter reading is observed.

Key Power Sensor ParametersIn t he ideal measurement case a bove, the power sensor absorbs all th e power

incident upon the sensor. There ar e tw o cat egories of non-ideal beha vior tha t

ar e discussed in detail in Ch apters VI a nd VII, but will be introduced here.

First, t here is likely an impedan ce mismat ch betw een th e cha ra cteristic

impedan ce of the RF source or tr an smission line and the RF input impedance

of the sensor. Thus, some of the power th at is incident on th e sensor is

reflected back towa rd the generat or rat her tha n dissipat ed in the sensor.

The relationship between incident power P i, reflected power P r , and dissipat-

ed power P d, is:

The relationship between P i and P r for a pa rticular sensor is given by the

sensor reflection coefficient ma gnit ude ,

.

P i = P r + P d (2-10)

P r =,

2 P i(2-11)


15/83

9

Reflection coefficient magnitude is a very important specification for a

power sensor because it contributes to the most prevalent source of error,

mismat ch uncerta inty, wh ich is discussed in Cha pter VI. An ideal power

sensor ha s a r eflection coefficient of zero, no misma tch. While a, of 0.05

or 5 percent (equiva lent t o an SWR of approxima tely 1.11) is preferred for

most sit ua tions, a 50 percent reflection coefficient w ould not be suita ble for

most situa tions due to the lar ge measurement uncertaint y it causes. Some

ear ly w a veguide sensors w ere specified at a reflection coefficient of 0.35.

The second cause of non-ideal behavior is that RF power is dissipated in

places other tha n in the power sensing element. Only the actua l power

dissipat ed in the sensor element gets met ered. This effect is defined as

th e sensor s effective efficiency e. An effect ive effi ciency of 1 (100%)means t ha t a ll the power entering the sensor unit is absorbed by the

sensing element and metered no power is dissipated in conductors,

sidewalls, or other components of the sensor.

The most fr equent ly used specification of a pow er sensor is called thecalibration factor, Kb. Kb is a combina tion of reflection coefficient a nd

effective efficiency a ccording to

If a sensor has a Kb of 0.90 (90%) th e pow er meter w ould norm a lly

indicat e a power level tha t is 10 percent lower tha n th e incident power P i.

Most power meters ha ve the a bility to correct the lower-indicat ed reading

by setting a calibration factor dial (or keyboar d or HP -IB on digital meters)

on the power meter to correspond w ith t he calibration fa ctor of th e sensor

at the frequency of measurement. Ca libration factor correction is not capa -

ble of correcting for th e tota l effect of reflection coefficient. There is st ill a

mismat ch uncerta inty th at is discussed in Chapt er VI.

The Hierarchy of Power Measurements,

National Standards and TraceabilitySince power measurement ha s importa nt commercial r am ificat ions, i t is

importa nt th at power measur ements can be duplicated at different times

a nd at different pla ces. This requires well-beha ved equipment , good mea -

surement t echnique, an d common agreement a s to w hat is the sta ndard

wa tt . The agreement in the Un ited St at es is established by the Nat iona l

Inst itute of S ta nda rds a nd Technology (NIST) at B oulder, Colorado, which

ma inta ins a Nat iona l Reference Sta nda rd in the form of var ious microwa ve

microcalorimeters for different frequency bands . 1, 2 When a power sensor

can be referenced back to tha t Na tional Reference St an dar d, the measure-

ment is sa id to be tr a cea ble to NIST.

The usual pa th of tra ceability for an ordina ry power sensor is shown inFigure 2-4. At each echelon, at least one power sta nda rd is maint ainedfor the frequency band of interest. Tha t power sensor is periodica lly sent

to the n ext higher echelon for recalibrat ion, th en return ed to its original

level. Recalibra tion intervals ar e esta blished by observing the stability

of a device betw een successive reca libra tions. The process might sta rt

with recalibra tion every few months. Then, when t he calibra tion is seen

not to change, the interva l can be extended to a year or so.

Kb = e (1 ,2) (2-12)NIST

NIST

CommercialStandardsLaboratory

ManufacturingFacility

User

Working Standards

MeasurementReferenceStandard

Transfer Standard

MicrocalorimeterNational Reference

Standard

General TestEquipment

Figure 2-4.The traceability pathof power referencesfrom the UnitedStates NationalReference Standard.


16/83

10

Ea ch echelon along the tra ceability pat h a dds some measur ement uncer-

ta inty. Rigorous measurement assura nce procedures ar e used at NIS T

because a ny error at tha t level must be included in the total uncerta inty

at every lower level. As a r esult, the cost of calibrat ion t ends to be great estat NIST and reduces at each lower level. The measurement comparison

technique for calibrat ing a power sensor aga inst one at a higher echelon

is discussed in other documents, especially those dealing with round robin

procedures. 3,4

The Nat iona l Pow er Reference St an dar d for the U .S. is a microcalorimeter

ma inta ined at t he NIST in Boulder, CO, for the va rious coaxial an d wa ve-

guide frequency ba nds offered in t heir measurement services progra m.

These meas urement services are described in NIS T SP -250, ava ilable from

NIST on request.5 They cover coa xial m ounts from 10 MHz t o 26.5 G Hz

an d w aveguide from 8.2 GH z to the high millimeter ra nges of 96 GH z.

A microcalorimeter measures the effective efficiency of a DC substitution

sensor which is then used as the tran sfer standa rd. Microcalorimetersoperat e on the principle tha t a fter a pplying a n equiva lence correction,

both DC a nd absorbed microwa ve power genera te the sam e heat. Comp-

rehensive and exhaust ive ana lysis is required to determine the equivalence

correction a nd a ccount for a ll possible therma l an d RF errors, such a s losses

in the tra nsmission lines and the effect of different th ermal pat hs wit hin

the microcalorimeter and the tra nsfer stan dar d. The DC-substitut ion

technique is used because the funda menta l power mea surement can t hen be

based on DC voltage (or current) and resista nce sta nda rds. The tra ceability

pat h lea ds t hrough th e micro-calorimeter (for effective efficiency, a unit -less

correction factor) an d finally back to the national DC sta nda rds.

In a ddition to na tional measur ement services, other industr ial organiza -

tions often participate in comparison processes known as round robins (RR).

A round robin provides measurement reference dat a to a part icipating la bat very low cost compared to primar y calibrat ion processes. For example,

the Na tional Conference of Sta nda rds La boratories, a non-profit a ssociat ion

of over 1400 world-wide organizations, maintains round robin projects for

ma ny measu rement para meters, from dimensiona l to optical. The NCSL

Measurement C omparison Committee oversees those progra ms.4

For RF power, a calibra ted thermistor mount sta rts out at a pivot lab,

usua lly one with overa ll RR responsibility, then t ra vels to many other

reference labs to be measured, returning to the pivot lab for closure of

measured da ta . Such mobile comparisons ar e also car ried out betw een

Nat iona l Labora tories of various countr ies as a r outine procedure to assure

interna tional measurements a t th e highest level.

Microwa ve power measurement services are ava ilable from many Na tionalLa boratories around the world, such as th e NPL in the Un ited Kingdom

an d PTB in G erman y. Ca libration service orga nizat ions are numerous too,

with na mes like NAMAS in t he Un ited Kingdom.

Figure 2-5.Schematic cross-section of the NISTcoaxial microcalori-meter at Boulder,

CO. The entire sen-sor configuration ismaintained undera water bath with ahighly-stable tem-perature so that RFto DC substitutionsmay be made pre-cisely.


17/83

11

A New Sensor for Power Reference TransferAlthough th ermistor sensors ha ve served for decades a s th e primary

porta ble sensor for tr an sferring RF power, they ha ve several dra wba cks.

Their frequency ran ge is limited, an d th ermistor impedan ce matcheswere never as good as most comparison processes would have preferred.

Except for t he most extreme measur ement cases requiring highest accuracy,

few modern procedures call for coaxia l or wa veguide tuners to ma tch out

reflections from mismatched sensors at each frequency.

A new cooperat ive resear ch effort between HP an d NIS T is aimed a t produc-

ing a novel resistive sensor designed for the express purpose of transferring

microwa ve power r eferences. The element is ca lled a resistive power sensor,

an d its DC to 50 G Hz frequency ran ge provides for DC subst itution tech-

niqu es in a sin gle sensor in 2.4 mm coax. The technology is based on micro-

w a ve microcircuit fa bricat ion of a precise 50 resistive element on G allium-

Arsenide. The resistor presents a positive tempera tur e coefficient w hen

heat ed, and can be operated by a n NI ST Type 4 Power Meter.

Future processes for reference power transfer will likely be based on such

a new technology because of its wide frequency coverage and excellent SWR.

____________

1. M.P. Weidma n a nd P.A. Hu dson, WR-10 Millimeterw a ve

Microcalor imet er, NIS T Technica l Note 1044, J un e, 1981.

2. F.R. Clague, A Calibrat ion Service for Coaxial Reference Stan dar ds for

Microw a ve P ower, NI ST Technica l Note 1374, Ma y, 1995.

3. Nat iona l Conference of St an dar ds Laborat ories, Measurement

Compa rison C ommitt ee, Suite 305B , 1800 30th S t. B oulder, CO 80301.

4. M.P. Weidma n, Direct Compa rison Tra nsfer of Microwa ve Power S ensorCa libra tion, NI ST Technical Note 1379, J a nua ry, 1996.

5. Special P ublication 250; NIST Ca libra tion Services, 1991 Edition.

G eneral References

R.W. B eatt y, Int rinsic Att enuat ion, IE EE Tra ns. on Microwa ve Theory a nd

Techn iq ues , Vol. I I, No. 3 (Ma y, 1963) 179-182.

R.W. B eat ty, Insert ion Loss Concepts, P roc. of th e IEE E. Vol. 52, No. 6

(J un e, 1966) 663-671.

S.F. Adam , Microwa ve Theory & Applica tions, P rent ice-Ha ll, 1969.

C.G . Montgomery, Technique of Microwa ve Measu rement s, Ma ssa chusett s

In stit ute of Technology, Ra dia tion La bora tory S eries, Vol. 11. McG ra w-Hill,

In c., 1948.

Mason an d Zimmerma n. Electronic Circuits, Signals and S ystems, J ohn

Wiley and Sons, Inc., 1960.


18/83

12

B olometer sensors, especially t hermistors, ha ve held an importa nt historical

position in RF/microwa ve power measu rement s. How ever, in recent year s

thermocouple and diode technologies have captured the bulk of those app-

licat ions because of their increased sensitivities, wider dyna mic ranges a ndhigher power capa bilities. Yet, ther mistors a re still the sensor of choice for

certa in applicat ions, such as tra nsfer sta nda rds, because of their power

subst itut ion capa bility. So, alth ough this chapt er is short ened from AN64-1,

the remaining ma terial should be adequat e to understa nd the basic theory

an d opera tion of thermistor sensors a nd t heir associat ed dual-bala nced

bridge power meter instru ments.

B olometers a re power sensors tha t operate by changing resista nce due to a

change in t empera ture. The change in t empera ture results fr om converting

RF or microwa ve energy into heat wit hin the bolometric element. There are

tw o principle types of bolometers, barrett ers an d thermistors. A barr etter

is a t hin w ire tha t ha s a positive temperat ure coefficient of resista nce.

Thermistors a re semiconductors w ith a negat ive temperatur e coefficient.

To have a measura ble cha nge in resistan ce for a sma ll amount of dissipated

RF power, a ba rretter is constructed of a very th in a nd short piece of wire,

or alterna tely, a 10 mA instru ment fuse. The ma ximum power tha t can be

measured is limited by t he burnout level of th e barr etter, typically just over

10 mW, an d th ey ar e seldom used a nym ore.

The thermistor sensor used for RF power measurement is a small bead of

meta llic oxides, typically 0.4 mm dia meter w ith 0.03 mm diameter wire

leads. Thermistor chara cteristics of resistan ce vs. power a re highly non-

linear, an d var y considerably from one thermistor to th e next. Thus th e

bala nced-bridge technique alw ays ma inta ins the thermistor element at a

consta nt r esistance, R, by mean s of DC or low fr equency AC bias. As RF

power is dissipated in t he thermistor, tending to lower R , the bias power is

with dra wn by just th e proper a mount to bala nce the bridge and keep R thesa me value. The decrea se in bias power should be identica l to th e increase

in RF power. Tha t decrease in bias power is then displayed on a meter t o

indicate RF power.

Thermistor SensorsThermistor elements a re mounted in either coaxial or w aveguide structures

so they ar e compatible with common t ra nsmission line systems used at

microwa ve and RF frequencies. The thermistor an d its mounting must be

designed to satisfy severa l importan t requirements so that the thermistor

element w ill absorb as much of the power incident on the mount as possible.

First, t he sensor must present a good impedan ce mat ch to the tra nsmission

line over th e specified frequency ra nge. The sensor must a lso have low r esis-

tive a nd dielectric losses w ithin t he mounting structure because only power

tha t is dissipated in t he thermistor element can be registered on th e meter.In a ddition, mecha nical design must provide isolat ion from therma l and

physical shock and must keep leaka ge small so tha t microwave power does

not escape from the mount in a shunt pat h ar ound the thermistor. Shielding

is also importan t t o prevent extran eous RF power from entering the mount.

Modern t hermistor sensors ha ve a second set of compensating thermistors

to correct for a mbient temperatu re varia tions. These compensa ting th ermis-

tors ar e mat ched in their temperat ure-resistance cha ra cteristics to the

detecting t hermistors. The thermistor mount is designed to ma inta in elec-

trical isolat ion betw een the detecting a nd compensat ing thermistors

yet keeping th e thermistors in very close thermal contact.

III. Thermistor Sensors and Instrumentation


19/83

13

Coaxial Thermistor Sensors

The H P 478A an d 8478A thermistor m ounts (thermistor mount wa s t he

earlier nam e for sensor) conta in four mat ched thermistors, and measu re

power from 10 MHz to 10 and 18 GHz. The two RF-detecting thermistors,bridge-ba lan ced to 100 each, a re connected in s eries (200 ) as far as the

DC bridge circuits a re concerned. For the RF circuit, the tw o thermistors

appear to be connected in parallel, presenting a 50 impedan ce to the test

signal. The principle advan ta ge of this connection scheme is tha t both

RF th ermistor leads t o the bridge are at RF ground. See Figure 3-1 (a).

Compensating th ermistors, wh ich monitor cha nges in am bient temperat ure

but not cha nges in RF power, are also connected in series. These therm-

istors ar e also biased to a t otal of 200 by a second bridge in the power

meter, called th e compensating bridge. The compensat ing t hermistors a re

completely enclosed in a cavity for electrical isolation from the RF signal.

B ut t hey ar e mounted on the sa me therma l conducting block as t he detect-

ing thermistors. The therma l ma ss of the block is large enough to prevent

sudden tempera ture gra dients betw een the thermistors. This isolates thesystem from thermal inputs such as huma n ha nd effects.

There is a part icular error, called dual element error, tha t is l imited to coax-

ial thermistor mounts w here the two thermistors a re in par allel for the RF

energy, but in series for D C. If t he tw o thermistors a re not quite identical in

resistance, then more RF current w ill flow in the one of least resistance, but

more DC power will be dissipated in the one of greater resistance. The lack

of equivalence in t he dissipat ed DC an d RF power is a minor source of error

tha t is proportional to power level. For HP thermistor sensors, this error is

less than 0.1 percent a t t he high power end of their measurement r an ge and

is therefore considered as insignifican t in th e error an alysis of Cha pter VI.

Waveguide Thermistor Sensors

The H P 486A-series of wa veguide ther mistor m ounts covers freq uenciesfrom 8 to 40 G Hz. See Figure 3-1 (b). Wa veguide sensors up to 18 G Hzutilize a post-an d-bar mounting a rra ngement for the detecting thermistor.

The HP 486A-series sensors covering t he K a nd R w a veguide ban d (18 to

26.5 GHz and 26.5 to 40 GHz) utilize smaller thermistor elements which

ar e biased to a n operating resistance of 200 , rat her tha n the 100 used

in lower frequency wa veguide units. P ower meters provide for selecting the

proper 100 or 200 bridge circuitry to ma tch th e thermistor sensor being

used.

Bridges, from Wheatstone to Dual-Compensated DC TypesOver the decades, power br idges for monitoring an d regulat ing power sen-

sing thermistors ha ve gone through a m ajor evolution. Ea rly bridges such

as t he simple Wheat stone type were man ua lly bala nced. Automa tically-

balanced bridges, such as the HP 430C of 1952, provided great improve-ments in convenience but stil l ha d limited dynam ic ra nge due to therma l

drift on their 30 W (full scale) ra nge. In 1966, wit h t he intr oduction of

the first temperat ure-compensated m eter, the H P 431A, drift w as reduced

so much t ha t m eaningful measur ements could be made down to 1 W.1

The HP 432A power meter, uses DC an d not audio frequency power to ma in-

ta in bala nce in both bridges. This elimina tes earlier problems perta ining to

th e 10 kHz bridg e drive signa l applied to the th ermist ors. The HP 432A

ha s th e further convenience of an a utoma tic zero set, eliminat ing the need

for the operator to precisely reset zero for each measurement.

Figure 3-1.(a) HP 478A coaxialsensor simplifieddiagram.(b) HP 486A wave-guide sensorconstruction.

(RC) Compensating

Thermistor

(Underneath)

Thermal

Isolation

Disc

Heat

Conductive

Strap

RF BridgeBias

CompensationBridge Bias

Thermal Conducting Block

RF Power

Rc

Rd

Cb

Rd

Cc

Rc

Cb

(a)

(b)


20/83

14

The HP 432A featur es an instrument at ion a ccura cy of 1 percent. It also

provides the ability to externally measure th e internal bridge voltages w ith

higher accuracy DC voltmeters, th us permitting a higher accuracy level for

power transfer techniques to be used. In earlier bridges, small, thermo-elec-tric volta ges were present w ithin t he bridge circuits w hich ideally should

ha ve cancelled in t he overa ll measurement. In practice, however, cancellation

wa s not complete. In certa in kinds of measurements t his could cause an error

of 0.3 W. In th e HP 432A, th e therm o-electr ic volta ges a re so sma ll, com-

pared t o the metered voltages, a s to be insignifican t.

The principal pa rt s of the H P 432A (Figur e 3-2) a re tw o self-bala ncing

bridges, th e meter-logic section, and t he au to-zero circuit. The RF bridge,

wh ich contains t he detecting t hermistor, is kept in ba lance by aut omatically

varying the DC voltage Vrf , w hich dr ives tha t bridge. The compensating

bridge, which conta ins th e compensating thermistor, is kept in bala nce by

automat ical ly varying the DC volta ge Vc, which drives tha t bridge.

The power met er is initia lly zero-set (by pushing t he zero-set but ton) wit hno applied RF power by making Vc equal to Vr fo (Vrf o means Vrf wit h zero

RF power). After zero-setting, if ambient temperatur e variat ions change

thermistor resista nce, both bridge circuits respond by applying the sa me

new volta ge to maintain balance.

If RF power is applied to the detecting thermistor, Vrf decreases so tha t

wh ere P rf is the RF power a pplied and R is th e value of the th ermistor resis-

ta nce at ba lance. B ut from zero-setting, Vrf o= Vc so tha t

Figure 3-2.Simplified diagramof the HP 432Apower meter.

P rf =Vrfo

2

4R(3-1)

Vrf2

4R

P rf =1

4R (3-2)(Vc

2 Vrf2)


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15

wh ich can be written

The meter logic circuitry is design ed to meter t he volta ge product shown in

equa tion (3-3). Ambient tempera tur e chan ges ca use Vc and Vrf to cha nge

so there is zero chan ge to Vc2 Vr f2 an d therefore no change to the indicat -

ed P r f.

As seen in Figure 3-2, some clever an a log circuitry is used t o accomplishthe multiplication of voltages proportional to (Vc Vrf ) and (Vc + Vrf ) byuse of a voltage-to-time convert er. In t hese days, such simple a rith metic

would be performed by th e ubiquit ous micro-processor, but th e HP 432A

predated th at technology, an d performs w ell without it .

The principal sources of instrumentation uncertainty of the HP 432A lie in

the metering logic circuits. But Vrf and Vc are both avai lable at the rear

pan el of the H P 432A. With pr ecision digita l voltmeters a nd proper proce-dure, those outputs a llow t he instrumenta tion uncerta inty t o be reduced to

0.2 percent for ma ny mea suremen ts. The procedure is described in the

operat ing man ual for the H P 432A.

Thermistors as Power Transfer Standards

For special use as t ransfer sta ndards, the U .S. National Inst i tute for

St a nda rds a nd Technology (NIST), Boulder, CO, accepts t hermist or

mounts, both coaxial and w aveguide, to tra nsfer power pa ra meters such

as calibra tion fa ctor, effective efficiency a nd r eflection coefficient in th eir

mea surement services progra m. To provide th ose services below 100 MHz,

NIST instructions require sensors specially designed for th at performa nce.

One example of a special power calibrat ion t ra nsfer is the one required

to precisely calibrat e the interna l 50 MHz, 1 mW power sta nda rd in t heHP 437B a nd 438A power meters, w hich use a family of t hermocouple

sensors. Tha t int erna l power r eference is needed since th ermocouple sen-

sors do not use t he power substitu tion technique. For th e power reference,

a specially-modified HP 478A thermistor sensor with a larger coupling

capacitor is ava ilable for operation from 1 MHz to 1 GHz. It is designated

the HP H55 478A and feat ures an SWR of 1.35 over its ra nge. For an

even lower tra nsfer uncerta inty a t 50 MHz, the HP H55 478A can be

selected for 1.05 SWR at 50 MHz . This selected model is designa ted t he

HP H75 478A.

HP H76 478A thermistor sensor is the H75 sensor which has been special-

ly cal ibrated in the HP Microwave Sta ndards La b with a 50 MHz power

reference tra cea ble to NIS T.

Other coaxia l and wa veguide thermistor sensors are a vailable for metrolo-

gy use.

Other DC-Substitution MetersOther self-bala ncing power meters can also be used to drive HP thermistor

sensors for measur ement of pow er. In pa rt icular, the NIST Type 4 power

meter, designed by the NIST for high-accuracy measurement of microwave

power is well suited for the purpose. The Type 4 meter uses a ut omat ic ba l-

an cing, a long w ith a four-termina l connection t o the th ermistor sensor

an d external high precision DC volta ge instr umenta tion. This permits

lower uncerta inty th an st an dar d power meters a re designed to accomplish.

P rf =1

4R (3-3)(Vc Vr f) (Vc + Vrf )


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16

Conclusions

There are some adva nta ges to thermistor power measu rements tha t ha ve

not been obvious from the above discussion or from data sheet specifica-

tions. Thermistor mounts a re the only present-day sensors which a llowpower substitut ion measurement techniques, and thus reta in importa nce

for traceability a nd absolute reference to na tional sta nda rds an d DC

voltages.

The funda menta l premise in using a thermistor for power measurements

is that the RF power absorbed by the th ermistor has t he same heat ing

effect on th e thermistor as t he DC power. The measurement is sa id to be

closed loop, because t he feedback loop corrects for m inor device irregula r-

ities.

1. R.F. Pra mann, A Microwave Power Meter wi th a Hundredfold

Reduction in Therma l Drift , Hewlett-Pa ckar d J ournal, Vol. 12,

No. 10 (J un e, 1961).

G eneral References

IE EE S ta nda rd Applicat ion Guide for B olometric P ower Meters, IEE E

Std. 470-1972.

IE EE St an dar d for Electrothermic Power Meters, IEE E S td. 544-1976


23/83

17

Thermocouple sensors ha ve been t he det ection t echnology of choice for

sensing RF a nd microwa ve power since their introduction in 1974. The tw o

ma in reasons for this evolution a re: 1) they exhibit higher sensitivity tha n

previous thermistor t echnology, an d 2) they feat ure a n inherent squa re-law

detection chara cteristic (input RF power is proportional to DC volta ge out).

Since thermocouple are heat -based sensors, they a re true a veraging

detectors. This recommends t hem for all types of signal forma ts from CW

to complex digital pha se modulat ions. In addition, they a re more rugged

tha n t hermistors, make usea ble power measur ements down to 0.3 W

(30 dB m, full scale), an d ha ve lower mea surement uncerta inty becauseof better S WR.

The evolution to t hermocouple technology is the r esult of combining th in-

film and semiconductor technologies to give a thoroughly understood, accu-

ra te, rugged, a nd reproducible power sensor. This chapt er briefly describes

th e principles of thermocouples, the constru ction an d design of modern

thermocouple sensors, and t he instrument at ion used to measur e theirra ther t iny sensor DC-output levels.

Principles of Thermocouples

Thermocouples are based on the fa ct tha t dissimilar meta ls genera te a

volta ge due to tempera ture differences at a hot a nd a cold junction of the

tw o meta ls. As a simple exa mple of th e physics involved, imagin e a long

metal rod that is heat ed at t he left end as in Figure 4-1. Because of theincreased therma l agita tion at th e left end, man y additional electrons

become free from their par ent a toms. The increased densit y of free elec-

trons a t th e left causes diffusion towa rd the right. There is also a force

at tempting t o diffuse the positive ions to the r ight but the ions a re locked

into the meta llic structure an d can not migrate. So far, this explana tion

ha s not depended on Coulomb forces. The migra tion of electr ons towa rd

the right is by diffusion, the sa me physical phenomenon th at tends toequalize the partia l pressure of a ga s throughout a space.

Ea ch electron th at migrat es to the right leaves behind a positive ion.

Tha t ion tends t o at tra ct the electron back to the left wit h a force given

by Coulombs law. The rod rea ches equilibrium wh en the rightw ar d force

of heat-induced diffusion is exactly balanced by the leftward force of

Coulombs law. The leftwa rd force ca n be represented by a n electr ic field

pointing t owar d th e right. The electric field, summed up along th e length

of the rod, gives rise to a volta ge source called the Thomson emf. This

explana tion is gr eatly simplified but indicat es the principle.

Figure 4-1.Heat at one end ofa metal rod givesrise to an electricfield.

IV. Thermocouple Sensors and Instrumentation


24/83

18

The same principles apply at a junction of dissimilar meta ls wh ere differ-

ent free electron densities in the two different metals give rise to diffusion

an d an emf. The nam e of this phenomenon is the P eltier effect.

A thermocouple is usually a loop or circuit of two different materials as

shown in Figure 4-2. One junction of the metals is exposed to heat, th eother is not. If th e loop remain s closed, current w ill flow in the loop as

long as the tw o junctions remain a t different t empera tures. If the loop

is broken t o insert a sensitive voltmeter, it w ill measure the n et emf.

The th ermocouple loop uses both t he Thomson emf a nd t he P eltier emf

to produce th e net therm oelectric volta ge. The tota l effect is a lso know n

as the Seebeck emf.

Sometim es man y pairs of junctions or ther mocouples a re connected in

series and fa bricat ed in such a w ay t ha t th e first junction of each pair is

exposed to heat a nd the second is not. In t his wa y the net emf produced

by one thermocouple adds t o tha t of the next, a nd th e next, etc. , yielding

a la rger th ermoelectr ic output . Su ch a series connection of th ermocouples

is called a thermopile.

Ea rly th ermocouples for sensing RF power were frequently constructed

of the meta ls bismuth a nd a ntimony. To heat one junction in t he presence

of RF energy, the energy w as dissipat ed in a resistor constructed of themeta ls makin g up th e junction. The meta llic resistor needed to be sma ll

in length a nd cross section t o form a resistance high enough to be a suit-

able termina tion for a t ra nsmission line. Yet, the junction needed to pro-

duce a m easura ble change in temperat ure for the minimum power t o

be detected a nd mea sured. Thin-film techniques w ere normally used to

build meta llic th ermocouples. These small meta llic th ermocouples tended

to have para sitic reacta nces an d low burnout levels. Furt her, larger ther-

mopiles, w hich did ha ve better sensitivity, tended t o be plagued by rea ctive

effects at microwave frequencies because the device dimensions became

too large for good impedance mat ch at higher microwa ve frequencies.

The Thermocouple Sensor

The modern t hermocouple sensor w a s int roduced in 19741, an d is exempli-

fied by the HP 8481A power sensor. It w as designed to take adva nta ge of

both semiconductor an d microwa ve thin-film technologies. The device,

shown in Figure 4-3, consists of t wo t hermocouples on a single integrated-circuit chip. The main ma ss of ma teria l is silicon.

The principal structural element is the frame made of p-type silicon, which

supports a th in web of n-ty pe silicon. The smoothly sloped sides of the

fra me result from an an isotropic etch acting on the silicon crystal plan es.

The thin web is produced by epitaxia lly growing it on the p-type substra te

an d th en suita bly controlling the etch, wh ich a lso reveals the surfa ce of

the diffused regions in t he w eb.

Figure 4-2.Total thermo-couple output isthe resultant of

several thermo-electrical voltagesgenerated alongthe two-metalcircuit.

Figure 4-3.Photo-micrographof the structure ofthe HP 8481A ther-mocouple chip ona thin silicon web.

Frame

Web0.005 mm

Tantalum Nitride

Ta2N

FrameWebDiffusedRegion

0.81 mmGold Beam

Lead


25/83

19

Figure 4-4 is a cross section th rough one of th e thermocouples. One goldbeam lead t erminal penetra tes the insula ting silicon oxide surface lay er

to conta ct the web over the edge of the frame. This portion of the web ha s

been more heavily doped by diffusing impurities int o it. The connection

between th e gold lead a nd t he diffused region is the cold junction of th e

th ermocouple, an d th e diffused silicon region is one leg of the t hermocouple.

At t he end of the diffused region nea r th e center of the web, a second metal

penetra tion to the web is made by a ta nta lum nitride film. This contact is

the hot junction of the t hermocouple. The ta nta lum nitride, wh ich is

deposited on the silicon oxide surfa ce, cont inues to th e edge of th e fra me,

wh ere it conta cts the opposite beam lead termina l. This tan ta lum nitride

forms t he other leg of the th ermocouple.

The other edge of the resist or an d th e far edge of th e silicon chip have gold

beam-lead contacts. The beam leads not only make electrical contact to th e

external circuits, but a lso provide mounting surfa ces for at ta ching t he chip

to a substrate , and serve as good thermal pat hs for conducting heat a wa y

from the chip. This ta nta lum-nitride resistor is not a t a ll fragile in contr ast

to similar termina tions constr ucted of highly conductive metals likebismut h/a nt imony.

As th e resistor converts the R F energy int o heat, t he center of t he chip,

wh ich is very thin, gets hotter t ha n th e outside edge for two reasons.

First, th e shape of the resistor causes the current density a nd th e heat

generated t o be lar gest a t t he chip center. Second, th e outside edges of the

chip a re thick and w ell cooled by conduction thr ough the beam lead s. Thus,

there is a t herma l gradient a cross the chip which gives rise to the th ermo-

electr ic emf. The hot junction is th e resistor-silicon connection a t t he center

of the chip. The cold junction is formed by t he outside edges of th e silicon

chip betw een the gold an d diffused silicon region.

The thin web is very important, because the thermocouple output is propor-

tional t o the t empera ture difference between the h ot a nd cold junctions.In t his case the web is fabr icated t o be 0.005 mm th ick. Silicon is quite a

good th ermal conductor, so the w eb must be very t hin if rea sonable temper-

at ure differences a re to be obtained from low power inputs.

| | | |,yz|

, yz z

, yz | |

SiliconOxideSiO2

TantaiumNitrideTa2N

HotJunction

Web

DiffusedRegionCold

Junction

SiO2

Frame

SiO2

GoldAu

GoldAu

Figure 4-4.Cross section ofone thermocou-ple. Power dissi-pated in the tanta-lum-nitride resis-tor heats the hot

junction.


26/83

20

The HP 8481A power sensor conta ins tw o identical th ermocouples on one

chip, electrically connected a s in Figure 4-5. The th ermocouples are con-nected in series as fa r a s the DC voltmeter is concerned. For the RF input

frequencies, the tw o thermocouples a re in pa ra llel, being dr iven t hrough

coupling capa citor Cc. Ha lf the RF current flows through each thermocouple.

Ea ch thin-film resistor an d the silicon in series with it h as a tota l resista nce

of 100 . The tw o th ermocouples in pa ra llel form a 50 terminat ion t o the

RF t ra nsmission line.

The lower n ode of th e left th ermocouple is directly connected t o ground a nd

the lower node of the right thermocouple is a t RF ground thr ough bypass

capacitor Cb. The DC volta ges generat ed by the separa te thermocouples

add in series to form a higher DC output volta ge. The principal adva nta ge,

however, of the two-thermocouple scheme is that both leads to the voltmeter

ar e at R F ground; there is no need for an R F choke in the upper lead.

If a choke were needed it would limit th e frequency ra nge of the sensor.

The thermocouple chip is attached to a transmission line deposited on a

sapphire substrate a s shown in Figure 4-6. A coplana r tra nsmission linestructure a llows the t ypical 50 l ine dimensions to ta per down t o the chip

size, while still ma inta ining the sa me chara cteristic impedance in every

cross-sectional plan e. This str uctur e contr ibutes to the very low reflection

coefficient of t he HP 8480-series sensors, its biggest contribut ion, over the

entire 100 kHz t o 50 G Hz fr equency ra nge.

The principal characteristic of a thermocouple sensor for high frequency

power measurement is its sensitivity in microvolts output per milliwa tt of

RF power input. The sensitivity is equa l to the product of tw o other pa ra -

meters of the t hermocouple, the th ermoelectric power an d t he therma l

resistance.

The th ermoelectric power (not rea lly a power but physics texts use tha t t erm)is the thermocouple output in microvolts per degree Celsius of temperature

difference betw een the hot an d cold junction. In t he HP 8481A th ermocouple

sensor, the th ermoelectr ic power is designed t o be 250V/ C. This is ma n-

a ged by cont rolling the density of n-type impurit ies in th e silicon chip.

The t hickness of th e HP 8481A silicon chip w a s selected s o the t hermocouple

ha s a therma l resistance 0.4 C/mW. Thus, th e overall sensitivity of each

th ermocouple is 100 V/mW. Tw o th erm ocouples in series, how ever, yield a

sensitivit y of only 160 V/mW because of th erma l coupling bet ween th e th er-

mocouples; the cold junction of one th ermocouple is heat ed somewha t by t he

resistor of the other t hermocouple giving a somewh at smaller temperat ure

gradient .

Figure 4-5.Schematicdiagram of theHP 8481A thermo-couple powersensor.

Figure 4-6.Sketch of thethermocoupleassembly for theHP 8481A.

BypassCapacitor

ThermocoupleChip

Input BlockingCapacitor

SapphireSubstrate

Housing


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21

The thermoelectric volta ge is almost constan t w ith externa l temperatu re.

It depends mainly on the temperatur e gradients a nd only slightly on the

am bient temperat ure. Still , ambient temperatu re var iations must be pre-

vent ed from esta blishing gra dients. The chip itself is only 0.8 mm long andis therma lly short-circuited by the relat ively ma ssive sapphire sub-stra te.

The entir e assem bly is enclosed in a copper housing. Figure 4-7depicts thesuperior t herma l behavior of a t hermocouple compared to a thermistor

power sensor.

The thermoelectric output va ries somewha t w ith temperat ure. At high

powers, where the a verage th ermocouple temperatur e is raised, the output

volta ge is lar ger tha n predicted by extrapolating da ta from low power levels.

At a power level of 30 mW the output increases 3 percent, a t 100 mW, it is

about 10 percent higher. The circuitry in t he HP power meters u sed with

th ermocouples compensa tes for th is effect. Circuit ry in the sensor itself

compensates for changes in its am bient t empera ture.

The therma l path r esista nce limits th e maximum power tha t can bedissipat ed. If the hot junction rises to 500 C, differentia l thermal expan-

sion causes t he chip to fractu re. Thus, t he HP 8481A is limited to 300 mW

ma ximum avera ge power.

The therma l resistance combines with the th ermal capa city t o form t he

therma l time constan t of 120 microseconds. This means t ha t t he thermo-

couple volta ge falls t o with in 37 percent of its fina l valu e 120 s a fter t he

RF power is removed. Response time for mea surement s, however, is usually

much longer because it is limited by noise and filtering considerations in the

voltmeter circuitr y.

The only significant a ging mecha nism is thermal a ging of the ta nta lum

nit ride resistors. A group of devices were stress test ed, producing the

results of Figure 4-8. These curves predict t ha t if a device is str esseda t 300 mW for one year, the resist a nce should increase by a bout 3.5 percent.

Nine days a t a ha lf wa tt w ould cause an increase in resista nce of 2 percent.

Aging accumulat es. On the other ha nd, aging effects of the ta nta lum-

nitride termina tion ar e compensa ted by use of the power calibra tion

procedure, w hereby a precision 1 mW, 50 MHz source is used to set a

known level on the meter.

Figure 4-7.Zero drift of ther-mocouple and ther-mistor power sen-

sors due to beinggrasped bya hand.

Time

10

8

4

6

0

2Power(microwatts)

Hand Grasp

Typical Thermocouple Sensor (8481A)

1 Minute

2 Microwatts

Typical Thermistor Mount (8478B)

Figure 4-8.

Results of stepstress aging testshow percentchange in thermo-couple resistancewhen left at vari-ous power levelscontinuously forvarious periods oftime.

1.00

1.50

0.75

0.50

0.25

00.01

1.25

0.1 1 10 100 1000

1 Year

Times (days)

Power

(watts)

% Change

510

21


28/83

22

It is relat ively easy t o ada pt this sensor design for other requirements.

For example, cha nging each t an ta lum-nitride resistor to a value of 150

yields a 75 syst em. To enha nce low frequency RF performa nce, lar ger

blocking a nd bypa ss capa citors extend input frequencies down t o 100 kHz.This usually compromises high frequency performance due to increased

loss a nd par a sitic reacta nce of th e ca pacitors. The HP 8482A power sensor

is designed for 100 kHz t o 4.2 GHz operat ion, w hile the sta nda rd

HP 8481A operat es from 10 MHz t o 18 GH z.

Power Meters for Thermocouple SensorsIntroduction of thermocouple sensor technology required design of a new

power m eter ar chitecture w hich could take a dvan ta ge of increased power

sensitivity, yet be able to deal w ith t he very low output volta ges of the sen-

sors. This led to a family of power meter instrument at ion sta rting w ith

the HP 435A analog power meter, to the HP 436A digital power meter. 2,3,4,5

Some years la ter t he dual-channel HP 438A wa s introduced, wh ich pro-

vided for computa tion of power ra tios of channels A and B as well as power

differences of cha nnels A a nd B . The most recent HP 437B power meteroffered digita l data ma nipulations with t he ability to store an d recall sensor

calibration factor da ta for up to 10 different power sensors.

To understa nd t he principles of t he instrum ent a rchitecture, a very brief

description will be given for the first -intr oduced thermocouple meter, the

HP 435A a na log power meter. This will be followed by an int roduction

of HP s new est power meters, HP E4418A (single cha nnel) a nd HP E4419A

(dua l cha nnel) power meters. They will be completely described in Ch a pter

V, aft er a n ew w ide-dyna mic-ra nge diode sensor is intr oduced.

Thermocouple sensor D C out put is very low -level (a pproximat ely 160 nV

for 1 microwa tt applied power), so it is difficult t o tra nsmit in a n ordinar y

flexible connection ca ble. This problem is mult iplied if the user w a nt s a

long cable (25 feet and more) between the sensor and power meter. For thisrea son it w a s decided to include some low-level AC a mplifica tion circuitry

in the power sensor, so only relatively high-level signals appear on the

cable.

One practical w ay to ha ndle such tiny D C voltages is to chop them to

form a squar e wa ve, then am plify with a n AC-coupled system. After

appropriate amplification (some gain in the sensor, some in the meter),

th e signal is synchronously detected at th e high-level AC. This produces

a high-level DC signal w hich is then fur ther processed to provide the

measurement result. Figure 4-9shows a simplified block diagr am of thesensor/meter a rchitectur e.

Cabling considerations led to the decision to include the chopper and part

of the first AC a mplifier inside the pow er sensor. The chopper itself(Figure 4-10) uses FET switches tha t a re in intima te therma l contact.This is essentia l to keep the tw o FETs a t exactly th e same temperat ure to

minimize drift . To elimina te undesir ed therm ocouples only one meta l, gold,

is used throughout the entire DC pa th. All these contributions were neces-

sar y to help achieve the low drift alrea dy shown in Figure 4-7.


29/83

23

The chopping frequ ency of 220 Hz w a s chosen as a result of severa l factors.Fa ctors tha t dicta te a h igh chopping frequ ency include lower 1/f noise a nd

a la rger possible bandw idth, and t hereby faster step response. Limiting

the chopping to a low frequency is the fa ct tha t sma ll tran sition spikes

from chopping inevitably get included with t he main signa l. These spikes

ar e at just the proper ra te to be integra ted by the synchronous detector and

ma squerade a s valid signals. The fewer spikes per second, the smaller this

ma squerading signa l. However, since the spikes are also present during

the zero-setting opera tion, and rema in the sa me value during the measur e-

ment of a signal, the spikes are essentially removed from the meter indica-

tion by zerosett ing a nd cau se no error. The spikes do, however, use up

dyna mic range of the a mplifiers.

One wa y to minimize noise while amplifying small signals is t o limit t he

channel bandw idth. Since the noise generating mechanisms are broad-

band, limiting t he amplifier ba ndw idth reduces the tota l noise power.

The nar rowest ban dwidt h is chosen for the w eakest signals a nd th e most

sensitive ra nge. As the power meter is switched to higher ra nges, the

bandw idth increases so tha t measu rements can be made more ra pidly.

On th e most sensitive ra nge, the time consta nt is roughly 2 seconds, w hile

on the higher ra nges, the time const a nt is 0.1 seconds. A 2-second tim e

const a nt corresponds t o a 0 to 99 percent rise time of about 10 seconds.

HP 435A Power MeterHP 8481A Power Sensor

One Halfof Input

Amplifier

LowLevelAC

One Halfof Input

Amplifier

ChopperDC

Thermocouple

220 HzMultivibrator

Autozero

Zero

Cal Factor

Amplifiersand

Attenuators

50 MHzReferenceOscillator

Synchronous

Detector

DCAmplifier

Range

RFInput

V

Cable

Meter

Figure 4-10.Simplifiedschematic of

chopper amplifier.

To DC Amplifier

ThermocoupleOutput

DC Path (all gold)

Multivibrator

Input

200

Figure 4-9.HP 435A/8481Aarchitecture blockdiagram.


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24

Reference OscillatorA frequent, s ometimes w ell-directed criticism of thermocouple power mea sure-

ments is tha t such measur ements ar e open-loop, an d thus t hermistor power

measurements ar e inherently more accura te becau se of their DC -substitut ion,closed-loop process. The bridge feedba ck of subst itut ed DC power compen-

sat es for differences between thermistor mounts a nd for drift in the t hermistor

resistan ce-power chara cteristic without recalibra tion.

With thermocouples, where there is no direct power substitution, sensitivity

differences betw een sensors or drift in th e sensitivity d ue to a ging or temp-

erat ure can result in a different D C output voltage for the same RF power.

Because there is no feedback to correct for different sensitivities, measure-

ment s w ith t hermocouple sensors a re sa id to be open-loop.

HP thermocouple power meters solve this limitat ion by incorporating a

50 MHz pow er-reference oscilla tor w hose output pow er is cont rolled with

grea t precision ( 0.7 %). To verify the a ccura cy of th e system, or a djust for

a sensor of different sensitivity, the user connects the thermocouple sensor tothe power reference output a nd, using a calibration adjustment , sets the meter

to rea d 1.00 mW. B y a pplying th e 1 mW reference oscilla tor t o the sensor s

input port just l ike the RF to be measured, th e same capa citors, conductors

an d amplifier chain a re used in the sam e wa y for measur ement as for the ref-

erence calibra tion. This featu re effectively tr a nsforms th e syst em to a closed-

loop, substitution-type system, and provides confidence in a traceability back

to company and NIST standards.

HP EPM Series Power Meters

The tw o-decade indust ry a ccepta nce of HP th ermocouple (a nd diode) sensor

technology for RF power measurements h as resulted in tens of thousands of

units in t he insta lled base a round the w orld. Yet new t echnologies now a llow

for design of diode sensors with far larger dyna mic range an d new power

meters with dr am at ically-expan ded user featu res.

Figure 4-11.HP E4418A featuresmany user-conve-niences and a 90 dBdynamic measure-ment range.


31/83

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The H P E4418A (single cha nnel) an d E 4419A (dua l chan nel) power m eters offer

some significan t user featur es:

Menu-driven user interfa ce, w ith softkeys for flexibility

Large LCD display for ease of reading Sensor E EP ROM which stores sensor cal ibrat ion fa ctors a nd other

correction data (HP E series wide-dynamic-range CW sensors)

Dedicat ed hardkeys for frequently-used functions

Faster measurement speed, faster th roughput

Ba ckwar d compat ibility with all previous HP 8480-series sensors

Form, fit , function replacement with HP 437B a nd 438A power meters

(preserves a u

an 64-1a -- fundamentals of rf and microwave power measurements

Documents