simultaneous wireless information and power transfer ...€¦ · power transfer can incur the...

2996 IEEE INTERNET OF THINGS JOURNAL, VOL. 7, NO. 4, APRIL 2020

Simultaneous Wireless Information and PowerTransfer (SWIPT) for Internet ofThings: Novel Receiver Design

and Experimental ValidationKae Won Choi , Senior Member, IEEE, Sa Il Hwang, Arif Abdul Aziz, Hyeon Ho Jang, Ji Su Kim,

Dong Soo Kang, and Dong In Kim , Fellow, IEEE

Abstract—In this article, we propose a novel simultaneouswireless information and power transfer (SWIPT) scheme for theInternet of Things (IoT). Different from the conventional powersplitting (PS) and time switching (TS) schemes, the proposedscheme sends the wireless power via the unmodulated high-powercontinuous wave (CW) and transmits information by using asmall modulated signal in order to reduce the interference andto enhance the power amplifier efficiency. We design a receivercircuit for processing such SWIPT signals, which is designedwith the aim of minimizing the circuit complexity and powerconsumption for information decoding. This goal is achieved byfirst rectifying the received signal and then splitting the powerand information signals. We analyze the proposed receiver circuitand derive the closed-form expression for the energy harvest-ing efficiency and the frequency response of the communicationsignal. We have implemented the proposed receiver circuit andbuilt the real-time testbed for experimenting with simultaneoustransmission of information and power. By experiments, we haveverified the correctness of the receiver circuit analysis and shownthe validity of the proposed SWIPT scheme.

Index Terms—DC-biased orthogonal frequency-divisionmultiplexing (DC-biased OFDM), energy harvesting efficiency,rectifier, radio-frequency (RF) power transfer, simultaneouswireless information and power transfer (SWIPT), wirelesspower transfer (WPT).

I. INTRODUCTION

RECENTLY, the radio-frequency (RF) wireless powertransfer (WPT) has emerged as a candidate technology

for supplying power to remotely located low-power Internetof Things (IoT) devices [1]. Powering a tremendous numberof IoT devices is expected to be a critical challenge in

Manuscript received July 25, 2019; revised December 10, 2019; acceptedDecember 30, 2019. Date of publication January 7, 2020; date of cur-rent version April 14, 2020. This work was supported in part by theBasic Science Research Program through the NRF funded by the KoreanGovernment (MSIP) under Grant NRF-2017R1A2B4010285, and in part bythe National Research Foundation of Korea (NRF) grant funded by theKorean Government (MSIP) (2014R1A5A1011478). (Corresponding author:Dong In Kim.)

The authors are with the Department of Electrical and ComputerEngineering, Sungkyunkwan University, Suwon 16419, South Korea (e-mail:[email protected]; [email protected]; [email protected];[email protected]; [email protected]; [email protected]; [email protected]).

Digital Object Identifier 10.1109/JIOT.2020.2964302

the forthcoming IoT era. The soaring maintenance cost ofreplacing batteries or connecting wired power cords for IoTdevices will render such traditional solutions not viable any-more. The WPT can solve the power charging problem withits capability of wirelessly transferring power to IoT devices.Among the WPT technologies, the RF WPT is characterized astransferring small power over a long distance, which makes theRF WPT the most suitable solution for powering low-powerIoT devices [2].

The concept of the RF WPT is extended to the simulta-neous wireless information and power transfer (SWIPT) thatenables the transmission of data and power via the same elec-tromagnetic (EM) wave. Two popular SWIPT schemes, powersplitting (PS) and time switching (TS), and their integrationswith various wireless technologies have extensively been stud-ied (e.g., [3]). In the PS scheme, the power and data areconveyed by the same modulated signal. The incoming RF sig-nal at the PS receiver is divided by the RF power divider, andthe divided signals are fed into the information decoder and theenergy harvester. On the other hand, the TS scheme switchesbetween the power and data transmission modes over time. Itis known that the PS scheme has a better tradeoff between thedata rate and harvested power than the TS scheme does.

However, the PS scheme has ignored the fact that thereis a radically wide gap between the required amounts ofthe power for data communication and power transfer [4],leading to inefficiency of the PS scheme in a practicalsystem. While the required power for operating an IoT deviceis around 0 dBm, the data communication is possible aslong as the received power is relatively higher than thenoise floor (i.e., −120 dBm). Therefore, we need around1012 times higher power for powering up the IoT devicethan for sending information in the practical IoT system. Forthe PS scheme, only an infinitesimally small fraction of thereceived power should be directed to the information decoder,which is more of coupling with a very small coupling factorthan PS.

The problem of the PS scheme in this situation is that theEM wave for the power transfer, which takes up the mostof the power in the EM wave, is unnecessarily modulatedeven though a continuous wave (CW) can more efficiently

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CHOI et al.: SWIPT FOR IoT: NOVEL RECEIVER DESIGN AND EXPERIMENTAL VALIDATION 2997

(a) (b)

Fig. 1. (a) Power over frequency and the (b) instantaneous power over timeof the RF signals for the PS and proposed SWIPT schemes. These signals aremeasured by the spectrum analyzer with the RBW of 1 kHz. The communica-tion signal is the DC-biased OFDM with 1024 subcarriers and the subcarrierspacing of 12.21 kHz.

carry the power. Specifically, the modulated EM wave for thepower transfer can incur the following two problems. First,the high-power modulated EM wave for the power transferunnecessarily occupies some bandwidth and can cause severeinterference to other wireless communication systems. Second,the high peak-to-average power ratio (PAPR) of the modu-lated signal greatly degrades the efficiency of the high-poweramplifier (HPA) in the transmitter.

To resolve the above-mentioned problems, we propose totransfer power via a high-power CW signal at the carrierfrequency and to transmit information via a small modu-lated signal occupying some bandwidth around the carrierfrequency. In Fig. 1(a) and (b), we show the frequency-domainand time-domain signals for the proposed SWIPT scheme andthe PS scheme, which are obtained by the spectrum analyzerin the real-life testbed. In Fig. 1(a), the power distribution ofthe proposed SWIPT scheme shows that the power signal atthe 920-MHz carrier frequency has relatively higher power(i.e., 1 mW), whereas the total power for the communicationsignal is much smaller (i.e., 1 μW). Since the high-power sig-nal is concentrated on very narrow bandwidth, the interferenceproblem can be minimized. On the other hand, the PS schemein Fig. 1(a) occupies wide bandwidth with thousand times(i.e., 30 dB) higher power than the proposed SWIPT schemedoes to achieve the same power transfer and communicationperformance, resulting in serious interference to other devices.Fig. 1(b) shows the instantaneous power of the proposedSWIPT and the PS signal over time. In the proposed SWIPT,the dc component of the instantaneous power signal carries thepower and the ac component fluctuating around the dc com-ponent carries the data. We can see that the HPA can be effi-ciently utilized because of the very low PAPR of the proposedSWIPT signal. On the other hand, the high PAPR of the PSscheme lowers the DC-to-RF conversion efficiency of the HPAand prevents the HPA to attain its maximum output power.

Some recent works (e.g., [5] and [6]) have proposed theuse of the high PAPR signal for enhancing the RF-to-DC con-version efficiency of the rectifier. Actually, the high PAPRsignal, such as multisine signals has the higher RF-to-DCconversion efficiency as proved by the experiments in [7]and [8]. This is because the multisine signal can generatea pulse-like form, the peak of which is close to the power

Fig. 2. High-level description of receiver circuit functionality.

level having the maximum RF-to-DC conversion efficiency ofthe rectifier circuit, while keeping the average power to thedesired level [6]. These works and our proposed scheme canbe viewed as the pursuit to the same goal of the optimizedSWIPT but with different performance criteria. We put ouremphasis on the spectrum usage and the HPA efficiency whilethe works [5], [6] focus on enhancing the rectifier efficiency. Inour view, the interference due to the modulated power signalis a serious problem that cannot be solved by other methodsthan concentrating the power signal on to a single frequency,especially when there is a tremendous power gap between thepower and communication signals. On the other hand, the RF-to-DC conversion efficiency problem is caused by the fixedrectifier circuit topology designed for a specific receive powerrange, and can be solved by cutting-edge maximum powerpoint tracking (MPPT) techniques, for example, the adaptiveimpedance matching [9], the rectifier stage adaptation [10],and the load control by a dc–dc converter [11].

The core contribution of this article is to design and ana-lyze the receiver circuit suitable for IoT devices, which isable to simultaneously extract power and data in the proposedSWIPT signal. The receiver circuit for IoT devices is requiredto have low complexity and low-power consumption. To thisend, we use the envelope detection for the information decod-ing. The envelope detection does not need power-consumingactive components, such as the local oscillator (LO), thelow-noise amplifier (LNA), and the mixer. Therefore, smallharvested power can be utilized by other parts of the IoTdevice instead of being consumed for communication. Fig. 2shows the high-level functionality of the proposed receiver cir-cuit. The proposed receiver circuit is different from the PSreceiver in that the proposed receiver first rectifies the RFsignal and then splits the rectified dc power signal and accommunication signal at the baseband into the power and com-munication modules, respectively. We have successfully built alinear communication channel inside the receiver circuit withthe nonlinear rectifier. This linear communication channel isimplemented by exploiting the fact that a small communicationsignal can be linearized around a much larger power signal.

We have analyzed the proposed receiver circuit and obtainedthe closed-form results for the energy harvesting efficiencyand the frequency response of the communication signal. Weconsider the N-stage Dickson charge pump as the rectifyingcircuit and derive the nonlinear closed-form equations that bal-ance the dc and RF power signals. Although there have beensome works analyzing the rectifier in the context of the SWIPT(e.g., [12]), most of these works assume a simple rectifier witha single shunt or series diode. After the dc operating point forthe power signal is determined by the rectifier analysis, we



can linearize the small communication signal around the dcoperating point. By the small-signal analysis, we derive thefrequency response of the communication signal in a closedform.

There are few papers that have analyzed the performanceof the Dickson charge pump (e.g., [13] and [14]). Thesepapers use the time-domain analysis in which each cycleis divided into several time periods, and the charge trans-fer between components is analyzed in each time period.However, this time-domain analysis cannot be used for analyz-ing the proposed SWIPT system since it does not reveal thefrequency-domain characteristics of the communication sig-nals. Our proposed analysis method is the frequency-domainanalysis which is quite different from the above time-domainanalysis. The proposed frequency-domain analysis balancesthe frequency-domain voltage and current signals in the base-band and RF frequencies. We have derived the relationshipbetween these voltages and currents around the diodes basedon the Shockley diode model augmented with the parasiticresistance and capacitance. From this starting point, we haveobtained the equations balancing the frequency-domain volt-ages and currents all over the circuits, which leads to theclosed-form expressions for the power transfer efficiency ofthe power signal and the frequency response of the communi-cation signal.

We have implemented the proposed rectifier circuit and builtthe testbed for verifying the performance of the proposedSWIPT scheme. We show that the analysis results on theenergy harvesting efficiency and the frequency response ofthe communication signals match very well with the exper-imental results as well as the simulation results based onthe commercial circuit design software. Moreover, we haveshown that the simultaneous transmission of power and dataover the air works well in the real-time testbed. The videoclip of experimenting the proposed SWIPT scheme can befound in [15].

Although the SWIPT has been a very active area of researchfor the past few years, there have been only a small numberof works that present the experimental results in a real-world testbed (e.g., [16]–[20]). One line of works [16]–[18]has proposed a dual-purpose hardware that combines theenergy harvesting and information receiving functionalitiesand has experimented various modulation techniques, such asthe magnitude ratio modulation with two-tone signals [16],the amplitude-shift keying (ASK) [17], and the multitonefrequency-shift keying (FSK) [18]. Even if the similar enve-lope detection concept is also considered in these works, theyare basically the PS schemes mixing the power and commu-nication signals in the same frequency band. Another line ofworks [19], [20] has built the prototype software-defined radio(SDR) testbed for verifying the signal adaptation techniquesfor boosting the power transfer efficiency. These works are alsobased on the PS scheme which is different from this article.

The remainder of this article is organized as follows.Sections II and III present the overall SWIPT system modeland the receiver circuit model, respectively. The in-depth anal-ysis on the rectifier is given in Section IV. In Section V,we analyze the performance of the power transfer and

Fig. 3. Transmitter architecture.

communication in the proposed receiver based on the rec-tifier analysis. The experimental results from the SWIPTtestbed and the comparison with the analysis and simulationresults are provided in Section VI. This article is concludedin Section VII.

II. SWIPT SYSTEM MODEL

In this section, we briefly explain the high-level transmitter,wireless channel, and receiver model of the proposed SWIPT.Also, we provide the details of the communication schemefor the proposed SWIPT, that is, the DC-biased orthogonalfrequency-division multiplexing (DC-biased OFDM). In addi-tion, we highlight the advantages of the proposed SWIPT overthe PS SWIPT.

A. Overview of the Proposed SWIPT System

The proposed SWIPT system wirelessly transfers the powerand data at the same time from the transmitter to the receiver.Different from the conventional PS and TS SWIPT schemes,the proposed scheme sends the power via a high-power CWsignal at the carrier frequency while the data are transmittedvia a low-power modulated signal. The signal transmitted fromthe antenna of the transmitter is given in the following form:

UT(t) = Re[(uT + uT(t)) exp(j2π fct)

](1)

where fc is the carrier frequency, uT is the power signal, anduT(t) is the communication signal. The power of the powersignal, denoted by pT = |uT |2/2, is much higher than thatof the communication signal [i.e., |uT(t)|2/2] in average. Weconsider the envelope detection for the communication sig-nal to minimize the power consumption and complexity ofthe receiver. Then, the communication signal uT(t) is not acomplex signal but a real-valued signal. We will discuss thecommunication signal in more detail in Section II-B.

An example hardware architecture of the transmitter for gen-erating the signal in (1) is given in Fig. 3. The source signalat the carrier frequency fc is generated by the LO and is splitand fed into the power and communication signal paths. Inthe communication signal path, the baseband modulated sig-nal from the digital-to-analog converter (DAC) is upconvertedto the carrier frequency by the mixer, and then is filteredand amplified for transmission. An HPA in the power signalpath performs the DC-to-RF power conversion, making the



RF signal carry sufficiently high power. The variable attenua-tor is used to control the amount of the power for the WPT.The signals from the power and communication signal pathsare combined by the RF combiner and are transmitted viathe antenna.

The transmitted signal travels over the air and is captured bythe antenna of the receiver. The received signal at the antennaof the receiver is given by

UA(t) = Re[(uA + uA(t)) exp(j2π fct)

](2)

where uA and uA(t) are the received power and communicationsignals, respectively. In (2), we do not consider the noise orinterference received at the antenna because we focus on thereceiver circuit behavior in response to the signal from thetransmitter. Under the frequency-selective channel model, thereceived signals are given by

uA = uT · h (3)

uA(t) = uT(t) ∗ h(t) (4)

where “∗” is the convolution operator, h(t) is the channelimpulse response, and h = ∫ ∞

−∞ h(t) dt.The receiver performs two functions at the same time, one is

to convert the power signal (i.e., uA) into the dc power, and theother is to retrieve the information from the communicationsignal [i.e., uA(t)]. In this article, we focus on designing andanalyzing the receiver circuit for performing such functions.More details about the receiver will be given in Section III.

B. DC-Biased OFDM for Data Communication

Since the receiver adopts the envelope detection, the com-munication channel is considered as an intensity channel,through which only unipolar signal (i.e., nonnegative real-valued signal) can be transferred. Among the modulationtechniques for the intensity channel, we choose to use theDC-biased OFDM technique. The DC-biased OFDM is robustto the frequency-selective linear distortion in the wirelesschannel and the receiver and transmitter circuits, comparedto other candidate techniques [e.g., pulse position modula-tion (PPM) and on–off keying (OOK)]. It is noted that thePPM or OOK can also be adopted by the proposed systemwithout modification.

The DC-biased OFDM has been popular in the free-spaceoptical communication, such as the visible light communi-cation (VLC) by the name of the DC-biased optical OFDM(DCO-OFDM) [21]. Here, we briefly explain the principle ofthe DC-biased OFDM. Suppose that there are (2K+2) subcar-riers, each of which is called subcarrier k (= 0, 1, . . . , (2K +1)). Let s(k) denote the complex number for the signal ofsubcarrier k. Subcarriers 0 and (K + 1) are not used fordata communication and are set to zero, i.e., s(0) = 0 ands(K + 1) = 0. Subcarriers 1 to K carry the quadrature phase-shift keying (QPSK) or the quadrature amplitude modulation(QAM) symbols. Then, s(k) for k = 1, . . . , K is the con-stellation point of the QAM symbol for subcarrier k. In theDC-biased OFDM, the frequency-domain signal is constrainedto have Hermitian symmetry for making the real-valued time-domain signal. Then, subcarriers (K + 2) to (2K + 1) are the

conjugate of the signal on subcarriers 1 to K, that is, s(2K +2 − k) = s∗(k) for k = 1, . . . , K. Note that half of the degreesof freedom (DOF) is lost because of the envelope detection.

The transmitter applies the inverse fast Fourier transform(IFFT) to the frequency-domain signal and attaches the cyclicprefix (CP) to the time-domain signal from the IFFT. Then, thetime-domain signal is converted to the analog baseband signalby using the DAC. Without loss of generality, the transmittedcommunication signal during one symbol period T is given by

uT(t) = 2√

pT,k

K∑

k=1

|s(k)| cos(2π fkt + ∠s(k)) (5)

where 0 ≤ t ≤ T , fk = fSk/(2K +2) is the baseband frequencyof subcarrier k, fS is the sampling frequency, and pT,k is thetransmit power of subcarrier k.

Since uT(t) is a bipolar signal, a sufficiently large dc signalis added to make the signal unipolar in the DC-biased OFDM.In the proposed SWIPT system, the power signal uT works asthe dc bias, and the summation of the power and communi-cation signals (uT + uT(t)) is a unipolar baseband signal asshown in (1).

C. Comparison Between Proposed and PS SWIPT Schemes

In this section, we discuss the main differences between theproposed and PS SWIPT schemes and highlight the poten-tial advantages of the proposed SWIPT in the followingthree aspects: 1) the interference to other communicationssystems; 2) the receiver circuit power consumption; and 3) thetransmitter HPA efficiency.

For WPT, the PS SWIPT emits much higher power than nor-mal communication systems do, causing severe interference toother communications systems. The proposed SWIPT solvesthis interference issue, which will be shown by simple equa-tions in the following. For both the proposed and PS SWIPTschemes, we denote by PT , fc, W, and h the total transmitpower, carrier frequency, bandwidth, and channel gain betweenthe transmitter and receiver. In addition, we define ρ as thePS ratio, which is the ratio of communication power to totalpower. In the proposed SWIPT, the transmitter assigns ρPT

and (1−ρ)PT for communications and WPT, respectively. Onthe other hand, in the PS SWIPT, the receiver splits the receivepower in the ratio of ρ and (1 − ρ) for communications andWPT, respectively.

In the case of the proposed SWIPT, the transmit power den-sity over the whole bandwidth is ρPT/W while the CW powerof (1 − ρ)PT is concentrated on the carrier frequency fc. Inthe proposed SWIPT, the signal-to-noise ratio (SNR) for datacommunications is |h|2ρPT/(N0W) and the harvested power is|h|2(1−ρ)PT . On the other hand, the transmit power density isPT/W for the PS SWIPT since it uses the whole bandwidth fortransmitting the total power. The receiver of the PS SWIPTsplits the total receive power (i.e., |h|2PT ) to the power fordata communications (i.e., ρ|h|2PT ) and the power for WPT[i.e., (1 − ρ)|h|2PT ]. Then, in the PS SWIPT, the SNR is|h|2ρPT/(N0W) and the harvested power is |h|2(1 − ρ)PT .

In summary, the proposed and PS SWIPT schemes havethe same theoretical SNR and harvested power. However,



(a)

(b)

Fig. 4. Receiver architecture comparison. (a) Proposed SWIPT receiverarchitecture. (b) PS SWIPT receiver architecture.

the transmit power densities of the proposed and PS SWIPTschemes are ρPT/W and PT/W, respectively. Since the totalpower PT should be much larger than that of typical commu-nication systems to supply a usable amount of power to thereceiver, the PS SWIPT can cause severe interference to com-munication systems sharing the same bandwidth. However, theinterference from the proposed SWIPT is not different fromtypical communications systems since ρ is a very small value(e.g., 10−3 in our experiments).

The proposed SWIPT receiver is much simpler than thePS SWIPT receiver as shown in Fig. 4. Compared to the PSSWIPT, the proposed SWIPT receiver does not have someactive components, the LNA, LO, and mixer. Removing theseactive components not only reduces the circuit complexityand implementation cost but also minimizes the circuit powerconsumption of the receiver. The end-to-end power transferefficiency of the RF WPT is quite low even with the cutting-edge beamforming techniques in a moderate distance, and theharvested power is expected to be around 1 mW. Some portionof the harvested power is used up by the active componentsof the PS SWIPT circuits, and only the remaining power canbe utilized by the IoT device itself. However, the active com-ponents, LNA, LO, and mixer, of the PS SWIPT typicallyconsume more than 1 mW. Therefore, it is very difficult todesign the self-sustainable PS SWIPT module, which is ableto supply power to the other parts of the IoT device.

The proposed SWIPT has lower power consumptionbecause it adopts the envelope detection while the PS SWIPTuses the IQ-based coherent detection. In [22], the power con-sumption of the cutting-edge low-power IoT communicationschip design is compared. In [22], it is shown that the enve-lope detection consumes as low as 0.016 mW, whereas theIQ-based detection consumes around 2 mW. The IQ-baseddetection consumes as high as 100 times more power thanthe envelope detection does because of the active components.Therefore, we can conclude that the envelope detection used

by the proposed SWIPT is more suitable technique than theIQ-based detection of the PS SWIPT for the IoT device withthe RF WPT.

The HPA performance is very important in the SWIPT sincethe HPA performs the DC-to-RF power conversion, which isone of the factors determining the end-to-end power trans-fer efficiency. The proposed SWIPT signal has low PAPRdue to the large CW power signal while the PS SWIPT hashigh PAPR due to the modulated power signal. The proposedSWIPT shows a stable power level while that of the PS SWIPTsignal severely fluctuates over time. In the case of the PSSWIPT, the HPA constantly consumes high dc power that isable to support the peak RF power. Even if the RF power islow, the dc power consumption is not reduced and the remain-ing dc power is wasted. Therefore, the high PAPR of the PSSWIPT results in low HPA efficiency. The high PAPR can alsodegrade the communication performance. If the RF power ishigh, the PS SWIPT signal suffers from clipping and nonlineardistortion as the instantaneous power frequently exceeds theHPA capacity. However, the small communication signal of theproposed SWIPT is immune to this nonlinear distortion sinceit is linearized around the large power transfer signal. Thenonlinear distortion incurs severe degradation of the commu-nication performance, and in the case of OFDM, it can causethe interference between subcarriers.

III. RECEIVER MODEL

The main goal of this article is to design and analyze thereceiver circuit in the proposed SWIPT system. In this section,we first explain the mathematical representation of the volt-age, current, and power wave signals in the receiver, and thenintroduce the high-level description of the receiver circuit.

A. Signal Model for Voltage, Current, and Power Wave

Within the proposed receiver circuit, the RF and basebandsignals exist together, and each of these RF and baseband sig-nals again consists the power and communication signals. Dueto this complexity of the signal composition, we start withintroducing the generalized signal model used for the receivercircuit design and analysis.

A voltage, current, or power wave signal in this article hasthe following generalized form:

X(t) = X0(t) +∞∑

n=1

Re[Xn(t) exp(j2πnfct)

](6)

where fc is the carrier frequency, X0(t) is the baseband sig-nal, X1(t) is the RF signal, and Xn(t) for n ≥ 2 is the nthharmonic signal. We assume that the bandwidth of Xn(t) ismuch smaller than the carrier frequency. It is noted that X0(t)is a real signal while Xn(t) for n ≥ 1 is a complex signal.The phase and magnitude of Xn(t) are denoted by ∠Xn(t) and|Xn(t)|, respectively. Then, we have

Xn(t) = ∣∣Xn(t)∣∣ exp

(j∠Xn(t)

). (7)

A signal Xn(t) consists of a dc component xn and an accomponent xn(t), that is

Xn(t) = xn + xn(t). (8)



Fig. 5. Receiver architecture.

The ac component is a small signal that carries information fordata communication, while the dc component is a large signalthat conveys power. Henceforth, we will term the ac compo-nent as the communication signal, and the dc component asthe power signal.

B. Receiver Circuit Model

The whole receiver circuit model is decomposed into sixparts: 1) the antenna; 2) the matching network; 3) the recti-fier; 4) the filter; 5) the communication branch; and 6) thepower branch as shown in Fig. 5. The circuit of the antenna isthe Thévenin equivalent of the receive antenna, which consistsof an antenna voltage source and an antenna impedance. Thevoltage across the antenna voltage source, denoted by VA(t),has only the RF signal, that is, VA(t) = Re [V1

A(t) exp(j2π fct)].The antenna impedance is denoted by ZA at the carrierfrequency fc. The normalized power wave signal from theantenna is defined as UA(t) = Re [U1

A(t) exp(j2π fct)], whereU1

A(t) = V1A(t)/(2

√Re [ZA]). Then, the maximum power that

can be delivered from the antenna to the rectifier is |U1A(t)|2/2

when the impedance is perfectly matched. The matchingnetwork is used to match the impedance between the antennaand the rectifier.

The rectifier is a nonlinear circuit that converts the inputRF signal to the output baseband signal. The rectifier is theN-stage Dickson charge pump. We will present an in-depthanalysis on the rectifier in Section IV. The baseband signalfrom the rectifier is shaped by the LC filter with inductanceLf and capacitance Cf . The shaped signal from the filter isdivided into the power and communication branches.

In the power branch, an inductor with inductance LP blocksthe communication signal and passes the power signal to thepower load with resistance RP. The voltage across the powerload is represented by a baseband voltage signal VP(t) =V0

P(t). Then, the harvested power of the receiver is the powerdelivered to the power load, which is V0

P(t)2/RP.In the communication branch, a capacitor with capacitance

CC is used for delivering the communication signal to thecommunication load with resistance RC, while blocking outthe power signal. The voltage across the communication loadis denoted by VC(t) = V0

C(t). The analog-to-digital converter(ADC) samples the voltage across the communication load.The sampled signal goes through the data receiving process,which is removing the CP, applying the fast Fourier trans-form (FFT), and demodulating the QAM signal to retrieve thedata bits.

Fig. 6. Diode model.

IV. RECTIFIER ANALYSIS

In this section, we analyze the rectifier that is the core com-ponent of the receiver circuit. We consider the Dickson chargepump as the rectifier. Since the rectifier consists of a numberof diodes, we start with presenting an analytic model for anonlinear diode. Then, we analyze the Dickson charge pumpbased on the analytic diode model.

A. Diode Model

As shown in Fig. 6, the diode model consists of theShockley diode model [23] with the parallel capacitance Cd

and the series resistance Rd [24]. Let Id(t) denote the currentgoing through the Shockley diode model, and Vd(t) denotethe voltage across the Shockley diode model. For simplicity,we assume that the harmonic components of the voltage andcurrent signals are negligible. Then, the voltage and currentsignals are given by

Id(t) = I0d(t) + Re

[I1d(t) exp(j2π fct)

](9)

Vd(t) = V0d (t) + Re

[V1

d (t) exp(j2π fct)]. (10)

The Shockley diode model defines the voltage–currentrelationship of

Id(t) = β · [exp(αVd(t)) − 1

](11)

where β is the saturation current of the diode. In (11), α isgiven by α = 1/(ηVT), where η is the diode ideality factorand VT is the thermal voltage. We can calculate the relation-ship between the current signals [i.e., I0

d(t) and I1d(t)] and the

voltage signals [i.e., V0d (t) and V1

d (t)]. To this end, we uti-lize the following equation for expanding the exponential partof (11) as:

exp(Re

[X(t) exp(j2π fct)

])

= B0(|X(t)|)+2

∞∑

k=1

Re[Bk(|X(t)|) exp(jk∠X(t)) exp(j2πkfct)

](12)

where Bk(X) is the modified Bessel function of the first kindsuch that

Bk(X) =∞∑

i=0

1

i!(i + k)!

(X

2

)2i+k

. (13)



Since the modified Bessel function in (13) rapidly decreasesas k increases, we can assume that the harmonic componentsin (12) are negligible. By substituting Vd(t) in (11) with (10),we can expand Id(t) as

Id(t) = β · [exp(αVd(t)) − 1

]

= β ·[exp

(αV0

d (t))

exp(

Re[αV1

d (t) exp(j2π fct)])

− 1]

= β exp(αV0

d (t))

B0

(α|V1

d (t)|)

− β

+ 2 Re[β exp

(αV0

d (t))

B1

(α|V1

d (t)|)

exp(

j∠V1d (t)

)

× exp(j2π fct)]. (14)

Equation (14) leads to the following equations for thebaseband and RF current signals:

I0d(t) = β exp

(αV0

d (t))

B0

(α|V1

d (t)|)

− β (15)

I1d(t) = 2β exp

(αV0

d (t))

B1

(α|V1

d (t)|)

exp(

j∠V1d (t)

). (16)

In Fig. 6, the capacitance Cd models the junction and dif-fusion capacitance of the diode. The capacitance Cd is nota fixed parameter but depends on the voltage across thediode. The RF current signal going through the capacitanceCd is j2π fcCdV1

d (t) while no baseband current signal flows.Then, the baseband and RF current signals going throughthe series resistance Rd are I0

d(t) and (I1d(t)+ j2π fcCdV1

d (t)),respectively, which incur the voltage drop across the seriesresistance.

B. Dickson Charge Pump Analysis

We consider an N-stage Dickson charge pump [23] for therectifier circuit. Although there are other choices for the recti-fier circuit, we have chosen the Dickson charge pump becauseof the following reasons. The active rectification in [25] canimprove the efficiency in the low-power region by usingthe actively controlled switches such as MOSFETs. In theactive rectification technique, the switches are closed bythe cross-coupled structure to reduce the threshold volt-age at the exact moment when the forward current flows.However, the active rectification of the RF signal can beimplemented in a complementary metal–oxide–semiconductor(CMOS) chip, and the board-level implementation is not possi-ble. Therefore, the active rectification is not a low-cost, easilyaccessible technique.

There are other rectifier topologies that can be implementedin a board level, for example, single series diode, single shuntdiode, and bridge rectifiers. These rectifiers are comparedand analyzed by the flow-angle equations in [26]. Accordingto [26], the theoretical maximum efficiency of the single seriesand shunt diode topologies is limited to 81.1%. On the otherhand, the full-wave rectifiers, such as the bridge rectifier andDickson charge pump, have the maximum efficiency of 92.3%.Therefore, we can conclude that the Dickson charge pump isa reasonable choice since its efficiency is higher than the sin-gle series and shunt diode topologies and it allows low-costboard-level implementation.

The rectifier consists of 2N diodes and 2N capacitors asshown in Fig. 7. The ith diode will simply be called diode i. All

Fig. 7. N-stage Dickson charge pump.

Fig. 8. N-stage Dickson charge pump from the point of view of a basebandsignal.

Fig. 9. N-stage Dickson charge pump from the point of view of an RFsignal.

the capacitors have the same capacitance CR. The capacitanceCR is large enough that the capacitors have zero impedancefor the RF signal. Therefore, from the point of view of theRF signal, the capacitors in the rectifier are short circuits. Onthe other hand, the capacitors completely block the basebandcurrent signal. Therefore, the capacitors are open circuits forthe baseband signals. In Figs. 8 and 9, we show the rectifier



circuit from the point of view of the baseband and RF signals,respectively.

The input voltage and current signals of the rectifier are theRF signals such that VRin(t) = Re [V1

Rin(t) exp(j2π fct)] andIRin(t) = Re [I1

Rin(t) exp(j2π fct)], respectively, since the inputport is an open circuit at the baseband frequency as shown inFig. 8. From Fig. 9, we can see that the output RF voltage andcurrent signals of the Dickson charge pump are zero since theoutput port is shorted to the ground at the RF frequency. Then,the output voltage and current signals are given by VRout(t) =V0

Rout(t) and IRout(t) = I0Rout(t).

In the Dickson charge pump, each diode is modeled bythe diode model given in Fig. 6, and the ith diode iscalled diode i as shown in Fig. 7. The voltage and cur-rent of the Shockley diode model of diode i are denoted byVd(i)(t) = V0

d(i)(t) + Re [V1d(i)(t) exp(j2π fct)] and Id(i)(t) =

I0d(i)(t) + Re [I1

d(i)(t) exp(j2π fct)], respectively. Equations (15)and (16) hold for V0

d(i)(t), I0d(i)(t), V1

d(i)(t), and I1d(i)(t) for all

i = 1, . . . , 2N.In Fig. 8, we can see that the baseband current signals of

the Shockley diode models of all diodes are the same. In addi-tion, we can assume that the baseband voltage signals of theShockley diode models of all diodes are the same as well.Then, we can introduce the new baseband voltage and currentsignals V0

Dout(t) and I0Dout(t) such that

V0Dout(t) = −2NV0

d(i)(t) (17)

I0Dout(t) = I0

d(i)(t) (18)

for all i = 1, . . . , 2N. From (17), (18), and Fig. 8, the outputvoltage and current signals of the rectifiers are calculated interms of V0

Dout(t) and I0Dout(t) as

V0Rout(t) = −

2N∑

i=1

(V0

d(i)(t) + RdI0d(i)(t)

)

= V0Dout(t) − 2NRdI0

Dout(t) (19)

I0Rout(t) = I0

d(i)(t) = I0Dout(t). (20)

In Fig. 9, we can see that the RF voltage signals on thediodes have the same magnitude with the phase difference ofπ for even and odd numbered diodes. In addition, the base-band voltage of the Shockley diode models for all diodes is thesame from (17). Then, the RF current signals of even and oddnumbered diodes have the same magnitude and the phase dif-ference of π from (16). We can introduce the new RF voltageand current signals V1

Din(t) and I1Din(t) such that

V1Din(t) = (−1)iV1

d(i)(t) (21)

I1Din(t) = 2N(−1)iI1

d(i)(t) (22)

for all i = 1, . . . , 2N. From (21), (22), and Fig. 9, the inputcurrent and voltage signals of the rectifiers are calculated interms of V1

Din(t) and I1Din(t) as

I1Rin(t) =

2N∑

i=1

((−1)iI1

d(i)(t) + j2π fcCd(−1)iV1d(i)

)

= I1Din(t) + j2π fc · 2NCd · V1

Din(t) (23)

Fig. 10. Equivalent circuit of N-stage Dickson charge pump.

V1Rin(t) = (−1)iV1

d(i)(t)

+((−1)iI1

d(i)(t) + j2π fcCd(−1)iV1d(i)

)Rd

= V1Din(t) + I1

Din(t)Rd/(2N) + j2π fcCdV1DinRd

= V1Din(t) + I1

Rin(t) · Rd/(2N). (24)

Now, we calculate the relationship between V0Dout(t),

I0Dout(t), V1

Din(t), and I1Din(t). By substituting V0

d (t), I0d(t), V1

d (t),and I1

d(t) with V0d(i)(t), I0

d(i)(t), V1d(i)(t), and I1

d(i)(t), respec-tively, in (15) and (16), and from (17), (18), (21), and (22),we have

I0Dout(t) = β exp

(−αV0

Dout(t)/(2N))

B0

(α|V1

Din(t)|)

− β

(25)

I1Din(t) = 4Nβ exp

(−αV0

Dout(t)/(2N))

B1

(α

∣∣∣V1Din(t)

∣∣∣)

× exp(

j∠V1Din(t)

). (26)

Finally, in Fig. 10, we build the equivalent circuit of thewhole rectifier, based on (19), (20), and (23)–(26). In this fig-ure, the inner rectifier is the N-stage Dickson charge pumpconsisting of the diodes without the parallel capacitance andthe series resistance. The inner rectifier is a two-port circuit,and the nonlinear relationship between the voltage and currentof the input and output ports is completely determined by (25)and (26). From (23) and (24), we can see that the series resis-tance Rd/(2N) followed by the shunt capacitance 2NCd relatesthe voltage and current inputs of the whole rectifier to thoseof the inner rectifier, as given in Fig. 10. From (19) and (20),the series resistance 2NRd is placed to connect the output ofthe inner rectifier to that of the whole rectifier, as in Fig. 10.

V. WIRELESS POWER TRANSFER AND DATA

COMMUNICATION ANALYSIS

In this section, we analyze the performance of the receiverin terms of the WPT and the data communication. Theperformance metric for the WPT is the energy harvesting effi-ciency, which is the ratio of the harvested power at the powerload to the antenna power. On the other hand, we analyze thefrequency response for the data communication signal. Finally,the receiver procedure for decoding the DC-biased OFDM sig-nal is explained, and the data communication performance isgiven in terms of the frequency response.

A. Receiver Circuit Analysis

Fig. 11(a) shows the receiver circuit in which the recti-fier in Fig. 5 is replaced by the equivalent rectifier circuitin Fig. 10. The shunt capacitance 2NCd can be moved to the



(a)

(b)

(d)

(c)

Fig. 11. Receiver circuit model for WPT and data communication analy-sis. Receiver circuit with the (a) rectifier equivalent circuit and (b) perfectlymatched source. Receiver circuit for the (c) WPT and (d) data communication.

front of the series resistance Rd/(2N) since the voltage changeacross the series resistance Rd/(2N) due to this modificationis very small. Then, for the brevity of the analysis, we assumethat the matching network perfectly matches the impedance tothe antenna. Under this assumption, the antenna, the match-ing network, and the shunt capacitance 2NCd are replaced bya source as in Fig. 11(b). Since the matching network andthe shunt capacitance 2NCd are lossless, the power from theantenna is delivered to the rest of the circuit without any lossas follows:

∣∣∣U1

A(t)∣∣∣2/2 = Re

[V1

S (t)I1S(t)∗

]/2 (27)

where U1A(t) is the power wave at the antenna, and V1

S (t) andI1S(t) are the voltage and current from the source.

All the signals in the receiver circuit model consist of thedc signal for the WPT and the small ac signal for the datacommunication, that is, U1

A(t) = uA + uA(t), V1S (t) = vS +

vS(t), and I1S(t) = iS + iS(t) for the source, V1

Din(t) = vDin +vDin(t), I1

Din(t) = iDin + iDin(t), V0Dout(t) = vDout + vDout(t),

and I0Dout(t) = iDout + iDout(t) for the rectifier, and V0

P(t) =vP + vP(t), I0

P(t) = iP + iP(t), V0C(t) = vC + vC(t), I0

C(t) =iC + iC(t), and U0

C(t) = uC + uC(t) for the voltage, current,and the normalized voltage at the power and communicationloads. Without loss of generality, we assume that all the signalshave real values except for uA(t).

B. DC Analysis for Wireless Power Transfer

The ac signals for data communications vanish in the dcanalysis for the WPT, and the receiver circuit is reduced toFig. 11(c). The circuit in Fig. 11(c) is completely defined bythe following equations:

pA = (uA)2/2 = vSiS/2 (28)

vS = vDin + Rd/(2N)iDin (29)

iS = iDin (30)

iDin = 4Nβ exp(−αvDout/(2N))B1(αvDin) (31)

iDout = β exp(−αvDout/(2N))B0(αvDin) − β (32)

vDout = vP + 2NRdiP (33)

iDout = iP (34)

vP = RPiP (35)

pP = vPiP (36)

where pA and pP are the antenna and harvested power,respectively.

In this section, we will let vDin = x, and solve the equationsin terms of x. By using (32)–(35), we derive vDout and iDoutas a function of x as follows:

vDout(x) = 2N

α(W0(�B0(αx)) − γ ) (37)

iDout(x) = β

γ(W0(�B0(αx)) − γ ) (38)

where W0 is the principal branch of the Lambert W function,and γ and � are constants such that

γ = αβ(Rd + RP/(2N)) (39)

� = γ exp(γ ). (40)

The harvested power is calculated in terms of x by (33)–(37)in the following equation:

pP(x) = RPβ2

γ 2 (W0(�B0(αx)) − γ )2. (41)

Now, we derive the antenna power in terms of x. The inputcurrent to the inner rectifier (i.e., iDin) is calculated from (31)and (37) in terms of x

iDin(x) = 4Nβ exp(−W0(�B0(αx)) + γ )B1(αx). (42)

From (28)–(30) and (42), we calculate the antenna power interms of x as follows:

pA(x) = (x + 2Rdβ exp(−W0(�B0(αx)) + γ )B1(αx))

× 2Nβ exp(−W0(�B0(αx)) + γ )B1(αx). (43)

Based on (41) and (43), we can calculate the closed-formenergy harvesting efficiency according to x as

η = pP(x)

pA(x). (44)

C. Small Signal Analysis for Data Communication

In this section, we analyze the small signal for data commu-nications at the dc operating point analyzed in Section V-B.When uA, vS, iS, vDin, iDin, vDout, and iDout are given by the dcanalysis, we can linearize the whole receiver circuit under the



assumption that the communication signal is small. Fig. 11(d)shows the receiver circuit for the small-signal analysis for datacommunications. All the voltage and current signals are repre-sented in the frequency domain in this figure, and we performthe frequency-domain analysis after linearizing the circuit.

The source in (27) can be rewritten as follows:

|uA + uA(t)|2 = (vS + vS(t))(iS + iS(t)

). (45)

If (28) holds and the second-order terms of small signals areassumed to be negligible, we can linearize the source (45) asfollows:

Re[uA(t)] = iS2uA

· vS(t) + vS

2uA· iS(t). (46)

Since (46) is linear, (46) can be converted to the frequencydomain as in the following equation:

uA(f ) = iS2uA

· vS(f ) + vS

2uA· iS(f )

= (aS)TsS(f ) (47)

where uA(f ), vS(f ), and iS(f ) are the frequency-domain signalsof Re [uA(t)], vS(t), and iS(t), respectively, and aS and sS(f )are the column vectors such that

aS = (iS

/(2uA), vS

/(2uA)

)T(48)

sS(f ) = (vS(f ), iS(f ))T . (49)

We relate the source signal and the input signal of the innerrectifier as follows:

sS(f ) = GR1sDin(f ) (50)

where GR1 is a 2-by-2 ABCD matrix of the series resistanceRd/(2N), and sDin is a column vector of the voltage and currentat the input to the inner rectifier such that

GR1 =[

1 Rd/(2N)

0 1

](51)

sDin(f ) = (vDin(f ), iDin(f ))T . (52)

We now linearize the inner rectifier. By differentiating (25)and (26), we derive the admittance matrix for the rectifier inthe case of small signals as

(iDin(f )

−iDout(f )

)=

[yii yio

yoi yoo

](vDin(f )vDout(f )

)(53)

where

yii = 4Nαβ exp(−αvDout/(2N)) (54)

× (B0(αvDin) − B1(αvDin)/αvDin) (55)

yio = −2αβ exp(−αvDout/(2N))B1(αvDin) (56)

yoi = −αβ exp(−αvDout/(2N))B1(αvDin) (57)

yoo = (2N)−1αβ exp(−αvDout/(2N))B0(αvDin). (58)

Based on the admittance matrix in (53), we derive the rela-tionship between the input and output signals of the rectifierin the following equation:

sDin(f ) = GRectsDout(f ) (59)

where GRect is the ABCD matrix of the rectifier such that

GRect = − 1

yoi

[yoo 1

yiiyoo − yioyoi yii

](60)

and

sDout(f ) = (vDout(f ), iDout(f ))T . (61)

After the rectifier, the signal goes through the series resis-tance, the LC filter, and the dc block as shown in Fig. 11(d).In Fig. 11(d), we can see that the series capacitance CC andshunt inductance LP work as the dc block for the communi-cation load. Then, the following relationship holds betweenthe signal out of the rectifier and the signal delivered to thecommunication load:

sDout(f ) = GR2GLC(f )GDB(f )sC(f ) (62)

where GR1, GLC, and GDB are the ABCD matrices for theseries resistance of 2NRd, the LC filter, and the dc block suchthat

GR2 = [ 1 2NRd0 1

](63)

GLC(f ) =[

1−(2π f )2LFCF j2π fLFj2π fCF 1

](64)

GDB(f ) =[

1 (j2π fCC)−1

(RP+j2π fLP)−1 1+(j2π fCCRP−(2π f )2LPCC

)−1

]

(65)

and

sC(f ) = (vC(f ), iC(f ))T . (66)

The voltage and current at the communication load arewritten in terms of the normalized voltage vC(f ) as follows:

sC(f ) = aCvC(f ) (67)

where aC is a column vector such that

aC =(

1, (RC)−1)T

. (68)

Finally, from (47), (50), (59), (62), and (67), we can cal-culate the voltage signal across the communication load as afunction of the power wave signal for data communication atthe antenna as

vC(f ) = (f )uA(f ) (69)

where (f ) is a frequency response given by

(f ) = ((aS)

TGR1GRectGR2GLC(f )GDB(f )aC)−1

. (70)

D. Receiver Procedure for DC-Biased OFDM and DataCommunication Performance

The receiver procedure for decoding the DC-biased OFDMsignal is not different from that for the typical OFDM. Thereceiver samples the voltage signal across the communicationload [i.e., vC(t)] with the sampling frequency of fS by using theADC. The receiver finds the position of each OFDM symbolof length (2K + 2) by means of the synchronization method,and the symbol is put into the FFT to obtain the frequency-domain samples. The frequency-domain samples with theFFT indices of 1, . . . , K correspond to subcarriers 1, . . . , K.



(a)

(b)

Fig. 12. Receiver circuit fabrication. (a) Receiver board schematic.(b) Receiver board photograph.

The receiver estimates the channel frequency response byusing the reference signals and corrects the frequency-domainsamples based on the channel estimation results. Then, the cor-rected frequency-domain samples are demodulated to recoverthe bit information.

Now, we analyze the data communication performance. Theend-to-end frequency response is derived as follows. The time-domain fading in (4) can be converted into the frequency-domain one as

uA(f ) = H(f )uT(f ) (71)

where uT(f ) and H(f ) are the Fourier transforms of uT(t) andRe [h(t)], respectively. From (69) and (71), we have

vC(f ) = (f )H(f )uT(f ). (72)

Thanks to the CP, uT(t) and vC(t) can be considered asa periodic signal repeating one OFDM symbol. Then, thereceived frequency-domain sample is derived as

yk = (fk)H(fk)√

pT,ks(k) + χ (73)

where yk is the received frequency-domain sample for subcar-rier k, fk is the baseband frequency of subcarrier k, and χ isthe additive noise with variance N0. The SNR of subcarrier kis given by |(fk)H(fk)|2pT,k/N0.

VI. EXPERIMENTAL RESULTS

A. Receiver Circuit Fabrication and Comparison BetweenExperiment, Simulation, and Analysis Results

In this section, we present the experimental results to ver-ify the proposed receiver circuit design and analysis and toshow the performance of the proposed SWIPT scheme. As afirst step, we have designed and fabricated the receiver cir-cuit with the one-stage Dickson charge pump. This receivercircuit is designed for the carrier frequency fc = 920 MHz.In Fig. 12(a) and (b), we show the schematic and photo-graph of the receiver circuit, respectively. In the schematic in

Fig. 13. V–I graph of the receiver circuit for various antenna power.

Fig. 14. Harvested power according to the power load for various antennapower.

Fig. 12(a), the diodes in the rectifier are Skywork SMS7630.The parameters for SMS7630 are the diode ideality factorη = 1.05, the saturation current β = 5 × 10−6 A, and theseries resistance Rd = 20 �. In the Shockley diode model,the thermal voltage VT is calculated to be 0.0257 V at 25 ◦C.The parameters for the capacitors, inductors, and resistors inFig. 12(a) are CR = 22 pF, LF = 10 μH, CF = 680 pF,CC = 10 μF, RC = 100 �, and LP = 100 μH. The capaci-tance and inductance of the matching network in the receivercircuit are 0.5 pF and 9.1 nH, respectively. We have matchedthe input impedance at the input power of 1 mW and the loadof 100 �. Fig. 12(b) shows the photograph of the fabricatedreceiver board. The substrate material is FR4 with the per-mittivity of 4.8 F/m and the dielectric height of 1 mm. Wehave used a commercial simulation tool, the advanced designsystem (ADS), for deriving the simulation results. The analysisresults are based on the equations in Section V.

We first show the power transfer test results in Figs. 13–15.For this test, we directly feed the RF signal generated froma signal generator (Tektronix TSG4100A) into the receiverboard via the antenna connector. A source meter (KeithleySMU2461) connected to the power connector is used to controlthe load and measure the harvested power.



Fig. 15. Energy harvesting efficiency according to the antenna power forvarious power loads.

Fig. 13 shows the V–I graph of the receiver circuit for var-ious antenna power PA. For obtaining experimental results forthis graph, we have varied the voltage across the power con-nector and measured the current by using the source meter. Wecan see that the analysis results very well match with the exper-imental and simulation results for all ranges of the antennapower. A deviation of the analysis results from the experimen-tal and simulation results is observed when the voltage is highand the antenna power is low. This is mainly because the effectof the breakdown voltage (2 V for SMS7630) is not modeled inthe analysis. If the voltage across the diode exceeds the break-down voltage, the reverse current flows through the diode andthe forward dc current is reduced. Therefore, the dc currentcan be overestimated by the analysis without consideration ofthe breakdown voltage.

In Figs. 14 and 15, we show the harvested power accord-ing to the power load, and the energy harvesting efficiencyaccording to the antenna power, respectively. These graphsare derived from the V–I graph in Fig. 13. Fig. 14 shows thatthe harvested power increases as the power load increases. Inthe rectifier, the harvested power generally increases accord-ing to the load until the voltage across the diode reaches thebreakdown voltage. This is because less power is consumedby the diode with the higher load. In Fig. 14, we have notincreased the power load to the higher level since exceedingthe breakdown voltage potentially incurs permanent damageon the diodes. The energy harvesting efficiency in Fig. 15increases with the antenna power until the antenna power of10 mW for all power loads, reaching up to 0.45 at RP = 180 �.Actually, the energy harvesting efficiency can go higher than0.5 with the higher power load at the risk of exceeding thebreakdown voltage. In Figs. 14 and 15, we can clearly see thatthe analysis proposed in this article is a very accurate tool forpredicting the energy harvesting performance of the Dicksoncharge pump.

In Fig. 16, we show the energy harvesting efficiency of themultistage Dickson charge pump when the number of stagesis 1, 2, 3, and 4. The multistage rectifier is useful for handlingthe high antenna power. In the case that the antenna power ishigh, we can increase the number of rectifier stages to keep the

Fig. 16. Energy harvesting efficiency of multistage Dickson charge pump.

Fig. 17. Time-domain sine wave signals received at the communication loadof the proposed receiver circuit.

voltage and current around each individual diode at the bestoperating points. Since we have not implemented the Dicksoncharge pump with more than one stage, we only compare thesimulation and analysis results to verify the correctness of theanalysis in the multistage case. The power load RP is fixed to600 �, and the perfect input matching is used in the simula-tion. In this figure, the simulation results show that the energyharvesting efficiency drops when the breakdown voltage ofthe diode is exceeded because of too high antenna power. Onthe other hand, the analysis does not describe this efficiencydrop since it does not incorporate the breakdown voltage intothe diode model. However, the simulation and analysis resultsshow good agreement as long as the breakdown voltage isnot exceeded.

In Figs. 17 and 18, we show the communicationperformance of the proposed receiver circuit. For this test, themodulated signal put into the receiver circuit is generated byan SDR platform (NI USRP-2942R). The power of the powersignal is fixed to 1 mW. We apply a sine wave as a basebandcommunication signal for observing the frequency response ofthe proposed receiver circuit.

Fig. 17 shows the time-domain sine wave signal received atthe communication load for various baseband frequencies of



Fig. 18. Frequency-domain graph of the magnitude of the voltage across thecommunication load.

the sine wave. For this test, the power of the communicationsignal is fixed to 0.01 mW, and the power load is set to200 �. In this figure, we can see that the sine waves areclearly received without any nonlinear distortion. Even if therectifier is a nonlinear circuit, the small communication sig-nal is linearized at the dc operating point of the power signal.Therefore, the interference caused by the intermodulation dis-tortion is not a concern as long as the communication signal issmall enough. In Fig. 17, we can also see that the magnitudeof the sine wave varies according to the baseband frequency,while the power of the communication signal at the antennaport is fixed.

The frequency-domain variation of the magnitude of thereceived communication signal is well described by Fig. 18.This graph shows the magnitude of the voltage across thecommunication load according to the baseband frequency ofthe sine wave for various power loads and various power ofcommunication signals. The analysis results in this graph arebased on (69). We can see that the analysis results well matchwith the experimental and simulation results. The main rea-son of the gap between the analysis and other results in thehigher frequency range is that the capacitors in the rectifierare assumed to be open circuits in the small-signal analy-sis. However, the analysis is accurate enough to be used forpredicting the frequency response of the communication sig-nal. The shape of the frequency-domain signal mainly dependson the LC filter. Although we have used a simple LC filter,more sophisticated filters can be adopted for shaping the com-munication signal. From Figs. 17 and 18, we can see that alinear communication channel is successfully implemented bya nonlinear rectifier.

B. SWIPT System Implementation and Test Results

For this article, we have implemented a full-fledged SWIPTsystem testbed that employs the proposed receiver circuit andcommercial off-the-shelf components, as shown in Fig. 19.Fig. 19(a) depicts the overall layout of the SWIPT systemtestbed, and Fig. 19(b) and (c) shows the photographs of theimplemented SWIPT system testbed. The carrier frequency

(a)

(b)

(c)

Fig. 19. SWIPT system implementation. (a) SWIPT system testbed layout.(b) Overall SWIPT system testbed photograph. (c) Receiver side of the SWIPTsystem testbed photograph.

of this testbed is fc = 920 MHz. In the transmitter, the SDRplatform generates the communication and power signals fromthe same LO and transmits them through two different outputports. The power signal is amplified by the HPA and combinedwith the communication signal by using the RF combiner. Weset the transmit power of the power and communication sig-nals at the antenna port of the transmitter to 1 W and 1 mW,respectively. The combined signal is transmitted and receivedby circularly polarized panel antennas. The antennas at thetransmitter and the receiver face each other, and the distancebetween the antennas is adjusted during experiments.

At the receiver, the received signal is decomposed into thepower and communication signals by means of the proposed



(a)

(b)

Fig. 20. (a) Power over frequency and the (b) instantaneous power over timemeasured by the spectrum analyzer.

receiver board. The received power signal is controlled andmeasured by the source meter. The source meter is set tothe constant voltage mode, which keeps the voltage acrossthe power connector of the receiver board to 0.5 V. Thereceived communication signal is amplified by the voltageamplifier, digitized by the ADC, and processed by the FPGA.We use the Labview software for controlling the wholetestbed. The transmit and receive codes for the DC-biasedOFDM are implemented in the SDR platform and FPGA,respectively, by using the Labview FPGA software for fastsignal processing. The overall control and visualization arecoded as the Labview host software in the PC, which isconnected to the SDR platform and FPGA via the PXIchassis.

The sampling rate of the communication signal is 25 MS/s.The FFT size for the DC-biased OFDM is 2048, and thesubcarrier spacing is 12.21 kHz. We use 512 subcarriersamong the 2048 available subcarriers, and 96 subcarriersaround the center and 1440 subcarriers around the edge ofthe bandwidth are unused. Therefore, the actual bandwidth

Fig. 21. Time-domain DC-biased OFDM signal appears at the communica-tion load of the proposed receiver circuit.

of the communication signal is about 6.25 MHz. The actualDOF of one OFDM symbol is 256 due to the Hermitiansymmetry constraint of the DC-biased OFDM. The trans-mit power is equally distributed to all active subcarriers. Wehave implemented the QPSK, 16QAM, 64QAM, and 256QAMmodulation schemes. For the channel estimation, one wholeOFDM symbol is used to send the reference signals over all thesubcarriers. The transmitter periodically sends such an OFDMsymbol with only reference signals, and the receiver estimatesthe frequency response based on such reference signals. Notethat the combined frequency response of the wireless channeland receiver circuit is estimated by this method.

In Fig. 20(a) and (b), we show the power over frequencyand the instantaneous power over time, respectively, mea-sured at the distance of 2 and 4 m away from the transmitter.These results are measured by the spectrum analyzer directlyconnected to the receive antenna. The resolution bandwidth(RBW) of the spectrum analyzer is set to 1 kHz. In Fig. 20(a),we can see that the CW power signal is surrounded by thelow-power communication signal. Since the high power forthe power transfer is concentrated on the carrier frequency,the interference to other devices is minimized. The instanta-neous power over time in Fig. 20(b) shows that the poweris transferred by the constant-envelope power signal whilethe small communication signal fluctuating around the powersignal delivering the data.

At the receiver, the communication signal is extracted bythe receiver board and appears as the voltage across the com-munication load as shown in Fig. 21. In this figure, we canclearly see that the DC-biased OFDM signal is well receivedby the receive board. We can also see that the power of thecommunication signal when the distance between the antennasis 4 m is lower than that of 2-m distance due to the higherattenuation. The communication signal from the receiver boardis digitized by the ADC and is processed by the FPGA forthe FFT and the channel estimation and equalization. Then,the constellation diagram can be constructed as shown inFig. 22. In Fig. 22, we show the constellation diagrams of theQPSK, 16QAM, 64QAM, and 256QAM modulation schemes



(c)

(e)

(h)(g)

(f)

(d)

(b)(a)

Fig. 22. Constellation diagram. (a) QPSK, 2 m. (b) QPSK, 4 m. (c) 16QAM,2 m. (d) 16QAM, 4 m. (e) 64QAM, 2 m. (f) 64QAM, 4 m. (g) 256QAM,2 m. (h) 256QAM, 4 m.

when the distances are 2 and 4 m. We can see that the eachconstellation point is distinguishable even with the 256QAMmodulation.

The bit error rate (BER) of the modulation schemes overthe distance is shown in Fig. 23. We do not show the BER of

Fig. 23. BER of 16QAM, 64QAM, and 256QAM modulation schemesaccording to the distance.

Fig. 24. Antenna and harvested power according to the distance.

the QPSK because it is zero up to 4-m distance. In this figure,we can see that all modulation schemes are usable within thegiven distance range with just 1-mW transmission power ofthe communication signal. The measured SNRs of 16QAM,64QAM, and 256QAM are 16.2, 14.6, and 17 dB at 2 mand 12.3, 13.1, and 14.8 dB at 4 m, respectively. From themeasured SNR, we can calculate the theoretical BER basedon the well-known BER formula. The theoretical BERs cal-culated from the measured SNRs for 16QAM, 64QAM, and256QAM are 3.85 × 10−9, 0.0014, and 0.0075 at 2 m and1.14 × 10−4, 0.0052, and 0.023 at 4 m, respectively. This the-oretical BER generally well agrees with the experimental BERin Fig. 23.

In Fig. 24, we show the antenna and harvested power ofthe power signal over the distance. The antenna power ismeasured by the RF power meter directly connected to thereceive antenna. On the other hand, the harvested power isthe dc power measured by the source meter connected to thepower connector of the receiver board. In Fig. 24, we cansee that there is a power loss due to the rectification. Theenergy harvesting efficiency is relatively low at 2-m distancesince the voltage of the source meter is fixed to 0.5 V. It is



possible to further enhance the energy harvesting efficiencyby adaptively controlling the power load. Fig. 24 shows thatthe harvested power at 4-m distance is higher than 0.5 mW,which is sufficient power for charging an IoT device. FromFigs. 23 and 24, we can see that the proposed SWIPT systemtestbed is capable of simultaneously transmitting informationand power.

We have shot a video clip that shows the experiments inthe proposed SWIPT testbed [15]. This video demonstrates thereal time and simultaneous operation of the WPT and wirelesscommunications.

VII. CONCLUSION

In this article, we have proposed a novel SWIPT schemethat sends power via a large unmodulated CW signal whiletransmitting information via a small modulated signal. Theproposed scheme can greatly reduce the interference to otherdevices compared to the existing PS scheme, since it confinesthe high-power signal for the WPT within a single frequency.We have also proposed an integrated receiver circuit that isable to simultaneously receive the power and communicationsignals without any power-consuming active component. Wehave analyzed the proposed receiver circuit and have derivedclosed-form expressions for the energy harvesting efficiencyof the power signal and the frequency response of the com-munication signal. The analysis results are validated by boththe simulation and experiments.

We have fabricated the proposed receiver circuit and haveimplemented the full-fledged SWIPT testbed. In the exper-iments, a thousand times higher power is used for the WPTthan that for the communication, that is, 1 W for the WPT and1 mW for the communication. By experiments, we have shownthat such a high discrepancy in the transmit power leads to thebalanced amounts of the receive power for the dual purposeof the WPT and communication.

In this article, we have demonstrated that the proposedSWIPT scheme works well in a short range up to 4 m. Thedistance is bounded in our experiment because of the relativelylow transmit power and the space limitation. We believe thatthe range can be further extended to the practically meaningfuldistance by using a higher power or a more sophisticated trans-mitter employing the multiple antenna-based beamforming.

REFERENCES

[1] T. D. P. Perera, D. N. K. Jayakody, S. K. Sharma, S. Chatzinotas, andJ. Li, “Simultaneous wireless information and power transfer (SWIPT):Recent advances and future challenges,” IEEE Commun. Surveys Tuts.,vol. 20, no. 1, pp. 264–302, 1st Quart., 2018.

[2] K. W. Choi et al., “Toward realization of long-range wireless-poweredsensor networks,” IEEE Wireless Commun., vol. 26, no. 4, pp. 184–192,Aug. 2019.

[3] X. Zhou, R. Zhang, and C. K. Ho, “Wireless information and powertransfer: Architecture design and rate-energy tradeoff,” IEEE Trans.Commun., vol. 61, no. 11, pp. 4754–4767, Nov. 2013.

[4] K. Huang and X. Zhou, “Cutting the last wires for mobile communi-cations by microwave power transfer,” IEEE Commun. Mag., vol. 53,no. 6, pp. 86–93, Jun. 2015.

[5] Y. Zeng, B. Clerckx, and R. Zhang, “Communications and signals designfor wireless power transmission,” IEEE Trans. Commun., vol. 65, no. 5,pp. 2264–2290, May 2017.

[6] B. Clerckx, R. Zhang, R. Schober, D. W. K. Ng, D. I. Kim, andH. V. Poor, “Fundamentals of wireless information and power trans-fer: From RF energy harvester models to signal and system designs,”IEEE J. Sel. Areas Commun., vol. 37, no. 1, pp. 4–33, Jan. 2019.

[7] A. Collado and A. Georgiadis, “Optimal waveforms for efficient wirelesspower transmission,” IEEE Microw. Wireless Compon. Lett., vol. 24,no. 5, pp. 354–356, May 2014.

[8] M. H. Ouda, P. Mitcheson, and B. Clerckx, “Optimal operation of mul-titone waveforms in low RF-power receivers,” in Proc. IEEE WirelessPower Transf. Conf. (WPTC), Montreal, QC, Canada, Jun. 2018, pp. 1–4.

[9] Z. Liu, Y.-P. Hsu, B. Fahs, and M. M. Hella, “An RF-DC converterIC with on-chip adaptive impedance matching and 307-μW peak outputpower for health monitoring applications,” IEEE Trans. Very Large ScaleIntegr. (VLSI) Syst., vol. 26, no. 8, pp. 1565–1574, Aug. 2018.

[10] M. A. Abouzied, K. Ravichandran, and E. Sánchez-Sinencio, “Afully integrated reconfigurable self-startup RF energy-harvesting systemwith storage capability,” IEEE J. Solid-State Circuits, vol. 52, no. 3,pp. 704–719, Mar. 2017.

[11] Z. Popovic, E. A. Falkenstein, D. Costinett, and R. Zane, “Low-powerfar-field wireless powering for wireless sensors,” Proc. IEEE, vol. 101,no. 6, pp. 1397–1409, Jun. 2013.

[12] R. Morsi, V. Jamali, A. Hagelauer, D. W. K. Ng, andR. Schober, “Conditional capacity and transmit signal designfor SWIPT systems with multiple nonlinear energy harvestingreceivers,” CoRR, vol. abs/1903.09299, 2019. [Online]. Available:http://arxiv.org/abs/1903.09299

[13] J. Yi, W.-H. Ki, and C.-Y. Tsui, “Analysis and design strategy of UHFmicro-power CMOS rectifiers for micro-sensor and RFID applications,”IEEE Trans. Circuits Syst. I, Reg. Papers, vol. 54, no. 1, pp. 153–166,Jan. 2007.

[14] C. Ma, C. Zhang, and Z. Wang, “Power analysis for the MOS AC/DCrectifier of passive RFID transponders,” in Proc. IEEE Asia Pac. Conf.Circuits Syst. (APCCAS), Singapore, Dec. 2006, pp. 1350–1353.

[15] K. W. Choi, S. I. Hwang, A. A. Aziz, J. S. Kim, D. S. Kang, andD. I. Kim. Simulataneous Wireless Information and Power Tranfer(SWIPT) Experiments. Accessed: Jul. 25, 2019. [Online]. Available:https://youtu.be/kWh8MSAtQxU

[16] M. Rajabi, N. Pan, S. Claessens, S. Pollin, and D. Schreurs, “Modulationtechniques for simultaneous wireless information and power trans-fer with an integrated rectifier–receiver,” IEEE Trans. Microw. TheoryTechn., vol. 66, no. 5, pp. 2373–2385, May 2018.

[17] S. Claessens, N. Pan, M. Rajabi, D. Schreurs, and S. Pollin, “Enhancedbiased ASK modulation performance for SWIPT with AWGN chan-nel and dual-purpose hardware,” IEEE Trans. Microw. Theory Techn.,vol. 66, no. 7, pp. 3478–3486, Jul. 2018.

[18] S. Claessens, N. Pan, D. Schreurs, and S. Pollin, “Multitone FSK mod-ulation for SWIPT,” IEEE Trans. Microw. Theory Techn., vol. 67, no. 5,pp. 1665–1674, May 2019.

[19] J. Kim, B. Clercks, and P. D. Mitcheson, “Prototyping and experi-mentation of a closed-loop wireless power transmission with channelacquisition and waveform optimization,” in Proc. IEEE Wireless PowerTransf. Conf. (WPTC), Taipei, Taiwan, May 2017.

[20] J. Kim, B. Clerckx, and P. D. Mitcheson, “Signal and systemdesign for wireless power transfer: Prototype, experiment and val-idation,” CoRR, vol. abs/1901.01156, 2019. [Online]. Available:http://arxiv.org/abs/1901.01156

[21] J. Armstrong, “OFDM for optical communications,” J. Lightw. Technol.,vol. 27, no. 3, pp. 189–204, Feb. 1, 2009.

[22] T. Schumacher, M. Stadelmayer, T. Faseth, and H. Pretl, “A review ofultra-low-power and low-cost transceiver design,” in Proc. AustrochipWorkshop Microelectron. (Austrochip), Linz, Austria, Oct. 2017,pp. 29–34.

[23] B. R. Marshall, M. M. Morys, and G. D. Durgin, “Parametric analysisand design guidelines of RF-to-DC Dickson charge pumps for RFIDenergy harvesting,” in Proc. IEEE Int. Conf. RFID (RFID), San Diego,CA, USA, Apr. 2015, pp. 32–39.

[24] C. R. Valenta and G. D. Durgin, “Harvesting wireless power: Surveyof energy-harvester conversion efficiency in far-field, wireless powertransfer systems,” IEEE Microw. Mag., vol. 15, no. 4, pp. 108–120,Jun. 2014.

[25] A. S. Almansouri, M. H. Ouda, and K. N. Salama, “A CMOS RF-to-DCpower converter with 86% efficiency and −19.2-dbm sensitivity,” IEEETrans. Microw. Theory Techn., vol. 66, no. 5, pp. 2409–2415, May 2018.

[26] T. Ohira, “Power efficiency and optimum load formulas on RF recti-fiers featuring flow-angle equations,” IEICE Electron. Express, vol. 10,no. 11, 2013.



Kae Won Choi (Senior Member, IEEE) receivedthe B.S. degree in civil, urban, and geosystemengineering and the M.S. and Ph.D. degrees inelectrical engineering and computer science fromSeoul National University, Seoul, South Korea, in2001, 2003, and 2007, respectively.

From 2008 to 2009, he was with theTelecommunication Business, Samsung ElectronicsCompany Ltd., Suwon, South Korea. From 2009 to2010, he was a Postdoctoral Researcher with theDepartment of Electrical and Computer Engineering,

University of Manitoba, Winnipeg, MB, Canada. From 2010 to 2016, hewas an Assistant Professor with the Department of Computer Science andEngineering, Seoul National University of Science and Technology, Seoul. In2016, he joined the Faculty Member with Sungkyunkwan University, Suwon,where he is currently an Associate Professor with the College of Informationand Communication Engineering. His research interests include RF energytransfer, metasurface communication, visible light communication, cellularcommunication, cognitive radio, and radio resource management.

Dr. Choi has served as an Editor for the IEEE COMMUNICATIONS

SURVEYS AND TUTORIALS in 2014, the IEEE WIRELESS

COMMUNICATIONS LETTERS in 2015, the IEEE TRANSACTIONS ON

WIRELESS COMMUNICATIONS in 2017, and the IEEE TRANSACTIONS ON

COGNITIVE COMMUNICATIONS AND NETWORKING in 2019.

Sa Il Hwang received the B.S. degree from theDepartment of Mechatronics Engineering, KoreaPolytechnic University, Siheung, South Korea, in2018. He is currently pursuing the M.S. degreewith the College of Information and CommunicationEngineering, Sungkyunkwan University, Suwon,South Korea.

His research interests include energy harvestingand wireless power transfer.

Arif Abdul Aziz received the B.S. degree in elec-tronics and instrumentation engineering from theUniversity of Gadjah Mada, Yogyakarta, Indonesia,in 2014, and the M.S. degree from the Department ofComputer Science and Engineering, Seoul NationalUniversity of Science and Technology, Seoul,South Korea, in 2017. He is currently pursuingthe Ph.D. degree with the College of Informationand Communication Engineering, SungkyunkwanUniversity, Suwon, South Korea.

His research interests include RF energy transfertechnique and RF circuit design.

Hyeon Ho Jang received the B.S. degree ininformation and communication engineering andmathematics from Sungkyunkwan University,Suwon, South Korea, in 2017, where he is currentlypursuing the Ph.D. degree with the Department ofElectrical and Computer Engineering.

His research interests include convexoptimization, machine learning, and RF energyharvesting for low-power IoT network.

Ji Su Kim received the B.S. degree in computerengineering from Sungkyunkwan University, Suwon,South Korea, in 2019, where he is currently pursu-ing the M.S. degree with the College of Informationand Communication Engineering.

His research interest includes machine learningfor communications.

Dong Soo Kang received the B.S. degree inelectronic engineering from Gachon University,Seongnam, South Korea, in 2018. He is cur-rently pursuing the M.S. degree with the Collegeof Information and Communication Engineering,Sungkyunkwan University, Suwon, South Korea.

His research interests include MU-MIMO and RFsystem.

Dong In Kim (Fellow, IEEE) received the Ph.D.degree in electrical engineering from the Universityof Southern California, Los Angeles, CA, USA, in1990.

He was a Tenured Professor with the Schoolof Engineering Science, Simon Fraser University,Burnaby, BC, Canada. Since 2007, he has beenan SKKU-Fellowship Professor with the Collegeof Information and Communication Engineering,Sungkyunkwan University (SKKU), Suwon,South Korea.

Dr. Kim has been a first recipient of the NRF of Korea EngineeringResearch Center in Wireless Communications for RF Energy Harvestingsince 2014. He has been selected as the 2019 recipient of the IEEECommunications Society Joseph LoCicero Award for Exemplary Service toPublications. He is an Executive Chair for the IEEE ICC 2022 in Seoul. From2001 to 2019, he served as an Editor and an Editor-at-Large of WirelessCommunication I for the IEEE TRANSACTIONS ON COMMUNICATIONS.From 2002 to 2011, he also served as an Editor and a Founding Area Editorof Cross-Layer Design and Optimization for the IEEE TRANSACTIONS

ON WIRELESS COMMUNICATIONS. From 2008 to 2011, he was theCo-Editor-in-Chief of the IEEE/KICS JOURNAL OF COMMUNICATIONS

AND NETWORKS. He served as the Founding Editor-in-Chief for IEEEWIRELESS COMMUNICATIONS LETTERS from 2012 to 2015. He is a Fellowof the Korean Academy of Science and Technology, and a member of theNational Academy of Engineering of Korea.


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