15th March 2005 Hermes-Internal 05-008

Radio Frequency Test for the HERMES Silicon

Recoil Detector

Z. Yea, M. Dohlusa, I. .M. Gregorb, Y. Hollera, I. Hristovab, J. Lund-Nielsena

V. Prahla, M. Reineckea, J. Stewartb, A. Vandenbrouckec, M. Wendta

a DESY, D-22603 Hamburg, Germanyb DESY, D-15738 Zeuthen, Germany

c Department of Subatomic and Radiation Physics, University of Gent, B-9000 Gent, Belgium

Abstract

The target cell and its connection to the HERA electron beam pipe was redesignedto reduce the beam-induced RF field in the scattering chamber which may affect thefunctioning of the new silicon recoil detector. A study was done to investigate theinfluence of the remaining RF field on the silicon recoil detector. With electrical RFsignals going through a specially designed aluminum rod mounted inside the targetcell along the beam axis, the beam-induced RF field in the scattering chamber wassimulated.

The field strength of the magnetic field generated with the cell-rod waveguide inthe scattering chamber was measured using a network analyzer and commercial mag-netic field probes. Measurements with the newly designed connection and the old onebetween the target cell and the C3 collimator were both performed to study the im-provement in the shielding effectiveness provided by the new design. Measurementswere done with a prototype module of the silicon recoil detector in the scatteringchamber, in the frequency domain with sine wave signals, and in the time domain withshort pulses going through the cell-rod waveguide to simulate the beam-induced RFfield. Frequency domain measurements were limited in the frequency range below 500MHz by available devices. The influence on the output of the detector’s frontend elec-tronics was observed only when the frequency of the sine waves was below 50 MHz,which could be explained by the low frequency bandwidth of the electronics. From thetime domain measurements with the new target cell and connection, the influence ofthe beam-induced RF field on the silicon recoil detector was expected to be a constantshift of the output at the level of 0.1 MIP signal at maximum. Here a possible influenceof the beam-induced RF field on the detector beyond the studied frequency range wasassumed to be negligible based on the low frequency bandwidth of the electronics.

Contents

1 Introduction 2

2 Silicon Recoil Detector 6

2.1 Detector Structure . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62.2 Pedestal and Common-Mode Noise . . . . . . . . . . . . . . . . . . . . . . . 10

3 HERA Electron Beam 12

4 RF Shielding against Beam 15

4.1 RF Shielding in General . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 154.2 New Design of the RF shield . . . . . . . . . . . . . . . . . . . . . . . . . . 16

5 Simulation of the HERA RF Field 19

5.1 Design Consideration . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 195.2 Transmission Properties of the Cell-Rod Coax . . . . . . . . . . . . . . . . . 20

6 Magnetic Field in the Scattering Chamber 24

7 RF Measurements 28

7.1 Measurement with a Readout Hybrid . . . . . . . . . . . . . . . . . . . . . . 287.2 Measurements with the Prototype Module . . . . . . . . . . . . . . . . . . . 30

7.2.1 Frequency Domain Measurements . . . . . . . . . . . . . . . . . . . 307.2.2 Time Domain Measurements . . . . . . . . . . . . . . . . . . . . . . 32

8 Conclusions 35

A Transmission Line 36

B Grounding and Shielding 38

1

1 Introduction

The HERMES experiment [1] is a fixed target experiment which studies spin physics withthe HERA electron beam1. The target cell [2], an elliptical tube made of aluminum foil,constitutes a part of the HERA-e beam pipe in the target chamber (scattering chamber).The HERA-e beam collides on the polarized atoms or unpolarized molecules distributed inthe target cell. Scattered electrons and particles produced in the interaction are detectedby the forward spectrometer.

To suppress the non-exclusive background, a recoil detector [3] which consists of asilicon detector, a scintillator fibre detector and a photon detector has been designed. Thesilicon recoil detector (SRD) is located in the scattering chamber around the target cellwhile the others are outside of the scattering chamber, as shown in Fig. 1. A new targetcell (see section 4 for details) and a new scattering chamber have been built for the recoildetector. Additionally a new collimator (C3 collimator) [4] has been built to protect therecoil detector from beam loss and high rate background, together with the existing C2collimator.

The SRD consists of double-sided silicon microstrip detectors. TIGRE sensors fromMicron Semiconductors Ltd. are used to detect particles while HELIX 128-3.0 chips [5] areused for analog readout. An important function of the SRD is to detect the recoil protonsfrom deeply virtual Compton scattering (ep → epγ) or other exclusive processes. To detectthese protons with 9 ÷ 750 MeV kinetic energy, the SRD and its frontend electronics

1HERA can also run with positron beam. In this note, only the electron beam is mentioned for simplicity.

Photon Detector

Water Coolingfor Si Det.

SciFi/LightguideConnector Ring

C3 Collimator(Tungsten)

Connectors

(3 layers tungsten/scintillator)

SciFi 2

Si Det. Hybrid

Thin−walledScatteringChamber

SciFi 1

Si Det.

TIGRE Sensors

Si Det. Frame(Aln)

CellTarget

Figure 1: HERMES recoil detector. It consists of a silicon detector, a scintillator fibre detectorand a photon detector. The silicon recoil detector is installed in the scattering chamber (targetchamber) while the others are outside. On the upstream side of the target cell there are C2 and C3collimators protecting the recoil detector.

2

are placed directly in the HERA-e beam vacuum, namely in the scattering chamber, tominimize the amount of material between interaction points and the SRD. An unusualaspect of this application is the direct exposure of the SRD to the HERA-e beam-inducedRF field in the scattering chamber due to the imperfect shielding against the electron beamprovided by the target cell and its connection to the beam pipe.

The target cell (see Fig. 2 and Fig. 12) is an open-ended elliptical tube made of 50÷125µm thick aluminum foil2. Square pumping holes (10×2 mm2) are cut along the ends of thetarget cell to pump away the target gas in the cell. At the upstream side of the target cell,there is the so-called C2 collimator (and in the future the C3 collimator as well) protectingthe target cell and the HERMES spectrometer. At the downstream side, there is the WakeField Suppressor (WFS) [2] made of perforated titanium foil. The WFS is located in avacuum vessel, i.e., a pump cross just downstream of the scattering chamber. The WFSis used to provide a smooth transition from the elliptical cross section of the downstreamend of the target cell (29.1 × 9.9 mm2) to the round cross section of the HERA-e beampipe (30 × 30 mm2). Since the HERA-e beam has to see along its path a continuous andsmooth conducting beam pipe to reduce the wake field in the beam pipe and energy loss ofthe beam, a good electrical connection between the C2 (C3) collimator and the target celland the WFS is important. Spring fingers (see Fig. 3) are used for this purpose. They arealso used to minimize the heat conductivity between the 100 K target cell and the beamline at room temperature and to provide an easy solution for assembly and disassembly.

There is another double-sided silicon microstrip detector in the HERMES experimentcalled the Lambda Wheel (LW) [6]. The LW is located inside the pump cross mentionedbefore and surrounds the WFS. Similar to the SRD, HELIX chips are used in the LW foranalog readout. The experience [7, 8] of the LW showed that its frontend electronics couldbe strongly affected by the beam-induced RF field. Namely with the currently used spring

2Several cells with different thicknesses and cross sections have been used in the HERMES experiment.

Figure 2: Currently used target cell made of 75 µm thick aluminum foil with 2.1× 0.89 cm2 crosssection. The upper right panel shows the downstream end of the target cell where 10 × 2 mm2pumping holes for removing the target gas are seen.

3

fingers (see Fig. 3) and unshielded pumping holes, the HELIX chips of the LW could beprogrammed without problem only when the electron beam current was smaller than 7 mA,while they could not be initialized to work at all when the beam current was larger than 16mA [9]. This problem was solved by installing a RF screen3 (see Fig. 3) between the pumpcross and the scattering chamber to separate the LW from the scattering chamber, and byimproving the contact of the spring fingers at the downstream side of the target cell4.

From the experience of the LW, the necessity to decrease the possible beam-induced RFinfluence on the SRD was foreseen. Efforts were put on improving the effectiveness of theshielding against the electron beam. The target cell and the spring fingers were redesignedto minimize the amount of RF power penetrating out of the cell (see section 4 for details).

To assure the proper functioning of the SRD, it is necessary to verify that the newdesign provides a sufficient shielding. Testing the SRD directly with the HERA-e beamis prohibitively difficult, therefore only a bench test is possible. The RF influence on theSRD depends on the RF field in the scattering chamber generated by the HERA-e beam,and on the response of the detector itself. Moreover, the scattering chamber may act as aresonating cavity and can increase or decrease the RF field strength at certain frequencies.Therefore it is important to perform the RF test in the scattering chamber using the finaldetector design in its final configuration. After the new target cell and the scatteringchamber for the recoil detector have been built, tests with a functional SRD prototypemodule installed in the scattering chamber can be performed if the beam-induced RF fieldin the scattering chamber can be simulated.

The electromagnetic field generated by an ultra-relativistic beam in the beam pipe is

3The RF screen is a 75 µm copper foil etched with 2 mm diameter holes [7].4When the cell was removed, the bottom half of the spring fingers was observed to be 0.5 cm away from

the cell [9].

Figure 3: Currently used spring fingers. They are 2 mm wide and 35.5 mm long with 5 mm spacing.The spring fingers at the upstream side (refer to the HERA-e beam direction) to connect the C3collimator and the target cell are shown in the left picture, while the downstream side spring fingersto connect the target cell and the wake field suppressor are shown in the right picture. The RFscreen for the LW detector is also shown in the right picture.

4

mainly a transverse electrical and magnetic (TEM) field. This field is similar to the oneproduced by a pulse in a coaxial waveguide in the limit of an infinitely thin inner conductor.This reasoning was confirmed by Gluckstern’s analysis of the effects of a wire on a resonantcavity [10]. The arguments can be summarized into a few sentences: The electromagneticfield is determined uniquely by the source (charge and current) and the boundary condition.Since an infinitely thin wire does not change the boundary condition for the TEM field,a short pulse getting through the wire will generate the same field as a bunch of ultra-relativistic charged particles if the pulse has the same shape and amplitude as the bunch ofcharged particles. This principle is applied in the RF test for the SRD, namely the beam-induced RF field in the scattering chamber is simulated in the frequency domain with sinewave signals, and in the time domain with short pulses going through the inner conductorin the target cell. Moreover, in order to investigate the influence of the beam-induced RFfield on the SRD with a simulated RF field, the strength of the simulated RF field shouldbe as large as that caused by the HERA-e beam. This requires that the current throughthe inner conductor in the cell should be comparable to the current of the HERA-e beam.

The SRD is described in the 2nd section. The relevant features of the HERA-e beamare discussed in the 3rd section. Different RF shielding designs are described in the 4thsection. The simulation of the HERA-e beam-induced RF field in the scattering chamber isdiscussed in the 5th section. The magnetic field measurements with commercial magneticprobes are presented in the 6th section. Measurements with a prototype detector moduleare presented in the 7th section together with the bench measurement with a SRD readouthybrid which is not bounded to a silicon sensor. Conclusions are given in section 8.

5

2 Silicon Recoil Detector

The working principle of a silicon detector relies on the fact that a charged particle produceselectron-hole-pairs along its track in the semi-conductor material. As shown in Fig. 4, n-type silicon with high resistivity is used as the base material on which p+ diode stripswith aluminum contacts are implanted. This side of the sensor is called p-side. On theopposite side, the n-side, the n+ electrode with much higher donor density compared withthe base n-type material is implanted. Then in the presence of an electrical field whichreversely biases the p-n diode of the silicon sensor, the charges produced by the particledrift oppositely towards the electrodes where they are collected. The collected chargesproduce a signal on the electrode which is read out by a charge sensitive preamplifer. Withthe electrodes divided into narrow strips, position information is obtained. An introductionto silicon microstrip detectors can be found in [11].

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p+

n+

-+++

+---

particleIncident

n-Oxide

Al strips

~~300 mµ

0

d

Figure 4: Schematic representation of the ba-sic operation of a silicon microstrip detector. Acharged particle will produce electron-hole-pairsalong its track. The charges created in the de-pleted region is collected and converted to signalsby the charge sensitive preamplifiers.

In this section the silicon recoil detector is described, e.g., the detector structure, thefrontend electronics and especially their bandwidths which give a first impression of thesensitive frequency range of the detector to an external RF field. The basic principle ofanalyzing the readout data are also introduced, e.g., pedestal subtraction and common-mode-noise correction. Grounding and shielding concerns for the detector electronics aredescribed in App. B.

2.1 Detector Structure

The SRD consists of 8 modules mounted in two layers symmetrically around the targetcell (Fig. 5). The two layers are separated by 1.5 cm in which the inner ones are 5.75cm away from the center of the scattering chamber. Not shown in Fig. 5 but shown inFig. 6 are the detector readout hybrids, which are the circuit boards carrying the frontendelectronics. They are glued to a 0.5 mm thick aluminum plate and then to a 5 mm thickaluminum heatsink. Each hybrid is connected to a 25-pole Sub-D feedthrough in theelectrical chamber. The electrical connection between the hybrids and the feedthroughsis done by a two-layer polyimide flexfoil, on which all signal lines are realized with 50 Ωimpedance (microstrip setup) to the common ground layer.

Fig. 6 shows the structure of one detector module (Ref. [13]). The main componentsinclude two TIGRE sensors and two readout hybrids with digital control and analog read-out circuits. The double-sided silicon sensors are from Micron Semiconductors Ltd. with

6

an active area of 9.9 × 9.9 cm2. The active thickness of the sensors is 300 µm. A Mini-mum Ionizing Particle (MIP) perpendicularly going through the sensor will generate 24,000electron-hole pairs or 3.84 fc charge on average. On each side of the sensor there are 128strips with a 758 µm wide pitch. The strip directions of the p-side and the n-side in one sen-sor are arranged perpendicularly to each other so that 2-dimensional position informationis available.

The electrical connection from the detector strips to the hybrids is achieved by 50 µm

2.1x8.9 cm

5.75cm

1.5cm

18.6cm

Scattering Chamber

Cell

Two Layers of Modules

Cross-Section

Cell Support Frame

Figure 5: Silicon Recoil Detector with the Target Cell. The SRD consists of 8 modules mounted intwo layers symmetrically around the target cell. The figure at the right hand side shows the relativeposition of the SRD in the scattering chamber with the target cell.

a)

b)

Helix

Radiation.-Mon.

Temp.-Mon.

Pitch Adapter

Line Driver

Flexleads FrameSilicon Sensors :

1 2

Charge DivisionHeatsink

I/O Flexlead

Set-OffDetail B

Detail A3 mm

~ 30 cm

. .

..

. .

. .

.

. .

.

…

. .

..

. .

. . . . . .

......

. . .

. . .

Strips …

Strips

9.89 cm

Hybrid

7 cm

25 cm

......

. . .

. . .

n-side

p-side

Figure 6: n-side (a) and p-side (b) of a SRD Module. It mainly consists of two double-sided siliconsensors (right) and two hybrids with the readout electronics (left). In (b) the hybrid and detectorpart are shown disconnected. [13]

7

thick polyimide foils (flexleads) [14]. The traces have widths and minimum distances of 70µm. The inner modules have their n-side facing towards the target cell and p-side facingtowards outside, while the outer modules have their n-side facing towards outside and p-side facing towards the target cell. This implies that the long flexleads of the n-side face thetarget cell and the wall of the scattering chamber, while the short flexleads on the p-sideare between the 2 module layers. The readout hybrid has a size of 7 × 10 cm2 without theI/O flexlead.

HELIX 128-3.0 readout chips [5] are used to read out the signals from the silicon sensors.They are manufactured in the 0.8µm-CMOS process of AMS [12]. Corresponding to 128strips in one TIGRE sensor, there are 128 channels in each HELIX chip. As depictedin Fig. 7, each channel features a charge sensitive preamplifier frontend and a CR-RCshaper. The rise time of the preamplifier/shaper is 18 ns [16] with the configuration usedwith the SRD. The output of each frontend channel feeds the signal into a capacitor array(pipeline). A resetable amplifier (pipeamp) reads the triggered event from the pipeline.The 128 channel signals are then serialized together with pipeline information (trailer) andbrought off-chip by means of a two-stage multiplexer and a current driver. The outputstage can operate up to 40 MHz readout clock frequency while it is set to be 10 MHz forthe SRD.

-

+

pipeline current bufferMUX

1 of 128+8+5

-

+

pipeampshaperpreamp

buffer

comparator

from/to neighbour

colu

mn

#

1 of 128 channels

Vd

Vdc

lV

offs

et

Vco

mpR

ef

bias generatortest pulsegenerator

interface circuit

Ipre

32+2

32+2

32+2

32+2

4

Isha

pipelinecontrol

Ibuf

Icom

pIp

ipe

cells

Helix128S-2

Isf

readout control

SerClk

SerData

SerLoad

SyncIn Sclk

Rclk

SufixReset

DataValid

In

Error

SyncOut

AnalogOut

TrigIn

notReset

FcsTp

Tp

Idri

ver

Vfp

Vfs

Vfp

Ipre

Vfs

Isha

IcompVcompRef

Ibuf Vdcl Ipipe

VdIsf

Res

et

Rea

d

Wri

te Voffset Idrv

syncronicity monitor

Figure 7: Schematic of HELIX 128-3.0 chips [5].

The dynamic range of the HELIX chip is ±40 fc. However, the slow protons fromthe exclusive reactions can deposit up to ±210 fc. Hence a charge sharing setup [15] wasdeveloped to increase the dynamic range of the readout electronics. Each strip is connectedto two HELIX channels of which one channel has a additional serial coupling capacitance.

8

Two thin film pitch adaptors connect the charge division network to the HELIX chips. Twolow-gain and two high-gain HELIX chips are located on each hybrid which reads out the256 strips for one side of the module. The channels without the serial coupling capacitor(high-gain channels) give a higher signal compared with the signals from the channels withthe serial capacitor (low-gain channels). The ratio of the signals between the high gainchannel and the low gain channel depends on the input capacitance of the HELIX andthe serial coupling capacitance. A ratio of 3.5:1 to 4:1 is observed with prototype detectormodule [16]. The dynamic range of the readout electronics is increased from ±40 fc to±270 fc by this charge dividing scheme.

An analog line driver (MAX435) on the hybrid with a 275 MHz bandwidth is used tocombine the analogue output signals and the analogue dummy signals from the HELIXchips into pure differential signals. The MAX435 chip also acts as a line driver to transmitthe signals to the external electronics.

LVDS receivers (DS90LV019) [17] are used to transfer differential signals (clock andtrigger) to HELIX chips. As shown in Fig. 8. Differential clock or trigger signals arereceived by the receivers and then sent to the clock or trigger bus which is connected toHELIX. The differential propagation delay from high level to low level or verse vice ofDS90LV019 is at minimum 3 ns which correspond to a maximum bandwidth of 116.7 MHz.

Figure 8: Schematics of a clock (trigger) circuit on the readout hybrid. Differential clock or triggersignals from the ACC board arrive at the left side and then they are output to the HELIX chips fromthe right side (CLK BUS/TRIG BUS)of the circuit. The DS90LV019 is a low voltage differentialsignaling driver/receiver [17].

From the rise time of the front stage of HELIX chips (18 ns), the requirement on theminimum differential propagation delay of the DS90LV019 LVDS receivers (3 ns) and thebandwidth of the analog line driver MAX435 (275 MHz), it is clear that the frontendelectronics is a low frequency system. Influence of external RF to the system will decreasesubstantially at frequencies beyond the bandwidth of the chips except the effect of DC

9

rectification which moves the operating point of amplifiers even by RF with frequenciesbeyond the bandwidth of the amplifier.

2.2 Pedestal and Common-Mode Noise

When the HELIX chip receives a trigger signal, the signals of its 128 channels stored in thepipeline corresponding to that trigger will be read out. The HELIX output of one eventcan be regarded as

PHki = Ski + P

k + Nki + CMNi, (1)

in which i denotes the event number, k denotes the HELIX channel number,

• PHki is the ADC output in counts of the kth HELIX channel from the ith event,• P k is the DC offset (pedestal) of the kth channel,• Ski is the real signal of the kth channel from the ith event,• Nki is the random noise of the kth channel from the ith event,• CMNi is a random voltage offset common for all the channels at the chip, which

may be due to a common change of the working environment, such as temperature,low voltage power supply for the chip, or electrical pickup which affects all channelsequally.

To get the pedestal values ’pedestal runs’ are taken when there are no real signals (forexample, there is no particle crossing the detector). The pedestal for each channel canthen be computed:

P k =1

Nevent

Nevent∑

i=1

(

PHki

)

. (2)

The pedestals are first subtracted for each HELIX channel from the raw data before futureanalysis. The CMN for each event is estimated as:

CMNi =1

N∗channel

N∗channel∑

i=1

(

PHki − P k)

, (3)

where N ∗channel denotes the channels of one HELIX chip being selected to calculate theCMN. These channels are usually chosen to be the first 16 channels not hit by particleswithin the first 32 channels of the chip. The variance (rms) of the CMN from a certainnumber of events is then an estimation of the CMN level, which can be calculated as:

CMNrms =

√

∑Neventi=1

(

CMNi − CMN)2

Nevent − 1. (4)

The variance of the common-mode noise (CMN) is a measure of how large the CMN levelis on an event-by-event basis. Ski +N

ki measured by the detector can then be computed by

10

praw(2,1)

020406080

100

120140160

400 450

Constant 116.9Mean 413.4Sigma 8.305

Raw DATA

pcmn(1)

0

20

40

60

80

100

120

140

160

-50 0

Constant 126.7Mean 0.4570E-01Sigma 7.705

CMN

pnoise(2,1)

0

20

40

60

80

100

120

0 10

Constant 80.86Mean -0.1529E-02Sigma 2.362

Corrected DATA

Figure 9: Raw ADC output (left), common-mode noise (middle) and signals (and noises) aftercommon-mode-noise correction and pedestal substraction (right) were measured or computed for apedestal run when there are no particles going through the detector.

subtracting the pedestal value P k for channel k and the CMN CMNi from the measuredADC output PHki as:

Ski + Nki = PH

ki − P k − CMNi, (5)

which is the best estimate of the signal induced by the charge generated in the detector. Theaverage value of N ki must be zero as the pedestal correction P

k corrects for any constantDC offset. The effect of the noise N ki is then to broaden the measured signal. The varianceof the noise Nkrms is then a measure of the broadening of the measured signal due to noise.It is calculated as:

Nkrms =

√

∑Neventi=1

(

PHki − CMNi − P k − Sk)2

Nevent − 1. (6)

Typical data collected for one channel of a HELIX chip is shown in Fig. 9. The leftpanel is a histogram of the raw ADC output values PHki . The mean of the histogram isthe measure of the pedestal for this channel, which is about 413 ADC counts. The middlepanel is a histogram of the CMN computed for all 2500 events in that run. The widthof the CMN distribution is measure of how large are the influences to the detector whichaffect all channels simultaneously on an event-by-event basis. In this example the CMNdistribution has a width of σ =7.7 ADC counts. In the right panel the response of thedetector is shown after the correction for the pedestal offset and the CMN. The average isnow zero as it should be for data where no charge is injected in the readout electronics andthe width of the distribution has reduced from 8.3 ADC counts to 2.4 ADC counts showingthe importance of the CMN correction and also giving a measure of the residual electricalnoise.

11

3 HERA Electron Beam

To simulate the RF field generated by the HERA-e beam, we have to know the properties ofthe beam. The HERA-e beam is a bunched beam with bunch spacing of 96 ns. A maximumof 220 bunches can be stored in the HERA-e beam. In practice only 189 buckets arefilled with electrons during luminosity running when the experiments are taking data. Theelectrons distribute around the center of the bunch in Gaussian distributions, longitudinallyand transversely. The actual beam parameters and bunch pattern depend on the machineconditions. Design parameters of the HERA-e beam during luminosity running are givenbelow [18]:

Storage Ring Circumference CH = 6335.83 mRevolution Frequency fr = c/CH = 47.317 kHzFull Energy E = 27.5 GeVNumber of Buckets 220Number of Filled Bunches 189Bunch Spacing ∆t = 96 nsBunch Length σz = 10.3 mmBunch Length in Time σt = σz/c 34.3 psAverage Beam Current Ibeam = 40 mAPopulation of Filled Bunch Ne = 2.79 × 1010Beam Peak Current Ipeak = (Neq)/(

√2πσt) ' 52 A

Hence during luminosity running, the length of the bunch (σ of the Gaussian distribution)is 10.3 mm or 34.4 ps in time. With an average beam current of 40 mA, the beam peakcurrent is about 52 A.

As introduced in the 1st section, there are two ways to perform an RF test for the SRD,a test in the time domain or a test in the frequency domain. It is obvious that no benchdevice currently available can provide pulses as strong and also short as the HERA electronbunches for the time domain test. Therefore the time domain measurements have to bedone with much longer pulses or pulses with much less amplitude. Decreasing the amplitudeof the pulses and consequently the amplitude of the RF field may make the influence ofthe generated RF field to the SRD not observable, while increasing the length of thepulses corresponds to a loss of high frequency components. The conseequence will dependon the frequency dependence of the SRD’s response to an external RF field. Thereforeit is important to perform a frequency domain test for the SRD which can tell what isthe sensitive frequency range of the SRD readout electronics, and if the high frequencycomponents of the HERA-e generated RF field are important or not. Below the frequencyspectrum of the HERA-e beam is derived which is needed to perform and interpret thefrequency domain test.

The single bunch current Is.b.(t) can be expressed as:

Is.b.(t) =Neq√2πσt

· exp(

−t2

2σ2t

)

, (7)

12

I(t)

t

σt = 34.4 ps

∆t = 96 nsHERA Electron Beam Current

HERA Electron Beam Spectrum

σf

σf = 4.64 GHz

∆f

∆f = 10.4 MHz

In

f

I0

f0 f1 ... ... ... f525

n fn (MHz) In (mA)0 0 401 10.4 80

. . . . . . . . .445 4635 48.6446 4646 48.5. . . . . . . . .891 9281 10.8892 9292 10.8. . . . . . . . .1336 13920 0.891337 13930 0.88. . . . . . . . .

Figure 10: The HERA electron spectrum. In the left-upper panel, the HERA-e beam is shownwith bunch length of 34.4 ps and bunch spacing of 96 ns. In the left-bottom panel the derivedspectrum of the beam current is shown, which is composed of discrete lines following a Gaussianenvelope.

in which Ne denotes the population of the electrons in the bunch, q is the electron charge,and σt is the bunch length in time. The Fourier transform of the single bunch current tellsthe frequency spectrum of a single bunch:

Ĩs.b.(f) =

∫ +∞

−∞

Is.b.(t) exp (−i2πft) · dt

=

∫ +∞

−∞

Neq√2πσt

· exp(

−t2

2σ2t

)

· exp (−i2πft) · dt

=Neq√2πσt

·∫ +∞

−∞

exp

[

−(

t + i2πfσ2t)2

2σ2t− 2π2f2σ2t

]

· dt

=Neq · exp

(

−2π2f2σ2t)

√2πσt

·∫ +∞

−∞

exp

[

−(

t + i2πfσ2t)2

2σ2t

]

· dt

=Neq · exp

(

−2π2f2σ2t)

√2πσt

·∫ +∞

−∞

exp

(

−t2

2σ2t

)

· dt

= Neq · exp

(

−f2

2σ2f

)

, (8)

here σf = (2πσt)−1 = 4.64 GHz for the σt = 34.3 ps bunches. We can see that the frequency

spectrum of the single bunch current is a Gaussian function with a width of 4.64 GHz.Neglecting the inhomogeneity in the population of individual bunches, the HERA-e

beam can be regarded as an infinite sum of single bunches spaced by ∆t = 96 ns. Thecurrent of this bunch train Ibeam(t) is a periodic function with a period of ∆t and the singlebunch current is an even function of time. A periodic even function with a period of ∆t canbe regarded as a superposition of cosine functions with frequencies of fn = n · ∆f (n=0,

13

±1, ...) where ∆f = ∆t−1. Hence for the HERA-e beam:

Ibeam(t) =

+∞∑

n=−∞

Is.b.(t − n∆t) = I0 ++∞∑

n−∞,n6=0

In · cos (2πfnt) , (9)

where ∆f = 1/∆t ' 10.4 MHz, and the coefficients are:

I0 =1

∆t

∫ +∆t/2

−∆t/2Ibeam(t) · dt

=1

∆t

∫ +∆t/2

−∆t/2

+∞∑

n=−∞

Is.b.(t − n∆t) · dt

=1

∆t·

+∞∑

n=−∞

∫ (n−1/2)∆t

(n+1/2)∆tIs.b.(t) · dt

=1

∆t

∫ +∞

−∞

Is.b.(t) · dt

=Neq

∆t,

In,n6=0 =2

∆t

∫ +∆t/2

−∆t/2Ibeam(t) · cos (2πfnt)

=2

∆t

∫ +∞

−∞

Is.b.(t) · cos (2πfnt) · dt

=2

∆t

∫ +∞

−∞

Is.b.(t) · [cos (2πfnt) + i sin (2πfnt)] · dt

=2

∆t

∫ +∞

−∞

Is.b.(t) · exp (2πfnt) · dt

=2Neq

∆t· exp

(

−f2n2σ2f

)

. (10)

In calculated for Ibeam = 40 mA are listed in Fig. 105. We can see that the HERA-e

beam has a spectrum with discrete lines at multiples of 10.4 MHz, namely 10.4 MHz, 20.8MHz, .... The amplitude of the lines decreases when the frequency increases and followsthe Gaussian envelope of the single bunch current spectrum in Eq. 8. The lowest frequencygenerated by the HERA-e beam is 10.4 MHz which has the largest amplitude of 80 mA(160 mW on 50 Ω), and even at a frequency as large as 4.65 GHz the signal lines will stillhave a relatively large amplitude of 48 mA (57.6 mW on 50 Ω). We can see that for thefrequency domain test, the amplitudes of the sine wave signals to be applied are valid fornormal RF devices at least at low frequency.

5Due to the unfilled bunches, the values listed in Fig. 10 are different from the values calculated viaEq. 10 by a factor of 220/189. The amplitude of the other frequency components due to the inhomogeneityof the population of individual bunches and empty bunches are 1 order of magnitude smaller than thedominant ones at fn = n × 10.4 MHz [20].

14

4 RF Shielding against Beam

As discussed in the first section, good RF shielding to protect the SRD and the LW detectorfrom the HERA-e beam is very important. In principle a beam pipe is desired to be solidwithout any apertures to separate the detectors from the HERA-e beam. However, a solidshield is not applicable, as it is necessary to cut holes into the target cell for removingthe injected gas, and as sliding contacts for the up- and downstream ends of the targetcell are needed for assembly and disassembly reasons. There are two solutions for thisproblem. One is to improve the effectiveness of the current shield so that the RF leakagedoes not significantly affect the detector performance. Another possible solution would beto add a second shield around the detector. This was in principle done for the LW detectorwhere a RF screen was installed between the pump cross, where the detector sits, and thescattering chamber, from where most of the RF leakage comes because of the pumpingholes and the apertures between the spring fingers. However this solution requires that theadditional shield can be put outside of the HERMES acceptance, or the added amount ofmaterial is quite small so that the influence to the detector acceptance can be neglected.As space around the SRD inside the scattering chamber is very small and any materialwould significantly affect the low momentum recoil protons, it was decided to attempt toimprove the effectiveness of the shield and then measure if it is sufficient.

In this section, the new target cell for the recoil detector is described based on animproved RF shield that is designed according to the calculations by Dohlus and Wipf[21]. These calculations, which investigate the effectiveness of different shielding designs,are summerized.

4.1 RF Shielding in General

An introduction to the basic concepts of electromagnetic shielding can be found in Ref. [19].Shielding can be specified in terms of the reduction in magnetic and/or electrical fieldstrength caused by the shield. There are 2 prime considerations for the shield: (1) theshielding effectiveness of the shield material itself, and (2) the shielding effectiveness due todiscontinuities and holes in the shield. One can first determine the shielding effectivenessof a solid shield with no holes, and then the effect of discontinuities and holes.

The shielding effectiveness of a solid shield depends on the frequency of the electro-magnetic wave and on the conductivity of the shield material itself which determine theattenuation of the travelling electromagnetic wave. One usually defines the ”skin depth”, adistance that the travelling electromagnetic field strength decreases by e ' 2.71828. With�, µ and ρ being the permittivity, permeability, resistivity of the shield material, and fbeing the frequency of the electrical and magnetic (EM) wave, the skin depth δ of theshield material can be calculated as δ =

√

2ρ/(2πfµ). The skin depth of aluminum for a10 MHz EM wave is obtained as 26.0 µm. It seems that the target cell made of 50 µm alu-minum is not sufficient, or at least for low frequency components of HERA-e beam-inducedRF field. However, this is not the case since one should also take into account anotherimportant mechanism of shielding, namely reflections of the electromagnetic wave at theboundaries between the inner space and the shield material, and the shield material and

15

the outer space. If multiple reflections between the boundaries are neglected, the loss of the

electromagnetic field strength due to reflection is a factor of√

1f ·2×10

8 [19] for aluminum

where f is in units of Hz. This means that at low frequency the loss due to reflections ismuch larger than the attenuation effect. For example, at f=10 MHz the loss factor due toreflections is 7.3 × 104.

In most cases it is the apertures in the shield, and not the shielding effectiveness of theshield material itself, that determines the overall performance of the shield. The amount ofRF leakage is determined by the maximum dimension of the opening, not by its area. Thiscan be visualized by considering the circuit theory model for shielding [19]. In this approachthe original field induces a current in the conductor, and the induced current generates afield which cancels the original field. If the flow of the flow of the current is disturbedby discontinuities in the shield, then the shielding effectiveness is reduced. The more thecurrent is disturbed, the larger will be the RF leakage. Therefore it is very important tokeep the dimension of the gaps along the beam pipe as small as possible.

4.2 New Design of RF shield

The design of the target cell and of the contacts between the target and the beam line wasredesigned to reduce the RF leakage. As the target cell for the recoil detector needs not tobe cooled down to 100K, the spring finger contacts up- and downstream of the target couldbe redesigned. Different designs with improved shielding effectiveness over the pumpingholes were also studied to optimize the design.

The beam line with a bunch of charge going through was modelled in MAFIA [23] byM. Dohlus and S. Wipf [21] to investigate RF leakage. They first studied the RF leakageout of the spring fingers. Comparing the old design with fingers 2 mm wide and 35.5 mmlong with a spacing of 5 mm to the new design with fingers 2 mm wide but 5 mm long witha spacing of 0.8 mm, the RF power leakage was decreased by 5 orders in magnitude.

Five kinds of shielding for the pumping holes were studied [21]; (1) mesh over theholes, additionally surrounded by a cage; (2) cage surrounding open holes; (3) 4 horizontalwires over a hole; (4) mesh over the holes; (5) open holes (see Fig. 11). According tothe calculation Model 4 reduced RF leakage by at least 2 orders of magnitude in powercompared with open holes. Since the RF leakage from spring fingers was much larger thanthe leakage from the pumping holes, it can be concluded that with the new design of thespring fingers and by covering the pumping holes with a mesh, the RF leakage is reducedby at least 5 orders of magnitude in power compared to the original case.

However, as it is mentioned in Ref. [21], the equipments around the beam pipe arenot modelled in the calculation due to the complicated structure inside the scatteringchamber and multiple reflection in this volume is neglected. Later it will become clearfrom the measurement results presented in section 6, that there are resonance modes inthe scattering chamber, which makes it necessary for such kind of study to model theequipments around the beam pipe. Although the effect of the the apertures in the shield isexpected to dominate the total shielding effectiveness, this expectation should be checked.Since this calculation uses material with infinite conductivity to model the target cell,the mesh/wire covering the pumping holes and the spring fingers, the attenuation and

16

library/homepage.html

Figure 11: Five models investigated in [21].

reflection effects in the shielding material itself are not included. Therefore the results ofthe calculations only give a qualitative prediction.

The 3D model of the final design of the new target cell is shown in Fig. 12. This targetcell [3] is made by 50 µm thick aluminum foil [22] and it has an elliptical cross section. Thecross section of the cell is not constant. At the upstream end of the cell it is 2.1×0.9 cm2.The downstream end of the cell has a tapered shape and the cross section of it ranges from2.1×0.9 cm2 to 2.9×0.99 cm2. Six square pumping holes, which are 10 mm along the beamdirection and 2 mm transversely, are distributed at both ends of the cell. These holes arecovered with a mesh made from etched nickel foil. Compared to the old design, the springfingers are now fixed at the downstream end of the cell and not on the WFS.

Fig. 13 shows photographs of the new cell. The right picture shows the downstreamend, from which the mounted spring fingers and the pumping holes covered by etched nickelfoil are shown. The dimples along the cell which provide the necessary expansion zones arealso visible in the photos. Fig. 14 (left) is the photo of the new spring finger design for theupstream end contact of the cell. Also seen is an additional shielding for the upstream endspring fingers taking advantage of the fact that it can be put outside the acceptance of theHERMES spectrometer.

17

Figure 12: The new HERMES target cell. The 2 mm high 10 mm long pumping holes are seenalong the side of the elliptical target cell at the up- and downstream ends. Also shown is the WFSwhich after installation will be inside the pumping cross downstream of the scattering chamber.

Figure 13: Left: picture of the new target cell for the recoil detector. Right: downstream endof the cell with 2 mm wide, 5 mm long spring fingers with 0.8 mm spacing. The pumping holescovered with nickel mesh are also visible in the pictures.

Figure 14: Pictures of the new spring finger design at the upstream end. Spring fingers are 2 mmwide and 5 mm long with 0.8 mm spacing. Left: spring fingers at the upstream end installed at theC3 collimator. Right: spring fingers with the additional shielding.

18

5 Simulation of the HERA RF Field

As already mentioned in the 1st section, in order to simulate the HERA-e beam, electricalpulses or sinusoidal signals will be propagated along a conductor inside the target cell. Toreduce possible power loss due to impedance matching, the diameter of the conductor ischosen so that the coax formed by the conductor and the target cell is 50 Ω. Calculationswere performed to determine the diameter of the rod, and to study possible effects thataffect the required 50 Ω impedance. According to the results, a 5 mm diameter aluminumrod was purchased and placed in the center of the new target cell. Parts which support thealuminum rod in the center of the target cell and provide convenient connection of the rodto standard commercial coaxial cables outside of the cell were designed and manufactured.In this section, the design and the design considerations are discussed.

5.1 Design Consideration

The inner conductor of the transmission line was designed with the help of Microwave Studio[24]. The diameter of a rod within a 2.1 × 0.9 cm2 elliptical tube to produce effectively a50 Ω transmission line was calculated. The calculations give an impedance value of 50.4 Ωfor a 5 mm diameter rod. To match the much smaller diameter of the inner conductor ofnormal commercial coaxial cables, the shape of the ends of the rod was studied to avoidsharp steps which may affect the impedance. A smooth 45◦ tapering of the 5 mm rod tothe 1 mm diameter inner conductors of a cable or connector was chosen. The calculationspredict good transmission with only 10% reflection up to 6 GHz with the 45◦ taper, whichis sufficient for our application.

Calculations were performed to estimate the deflection of the rod due to its own weight.For a 50 cm long and 5 mm diameter rod, the deflection for aluminum was the smallestamong normal metal materials. The deflection is only 20 µm which gives 4 Ω deviationof impedance. The sensitivity of the impedance on the rod diameter was also studied,changing the rod diameter by 20 µm resulted in a change in impedance of 4 Ω. Thereforea tolerance of 20 µm was chosen for the rod diameter yielding an expected impedance of50.4±4 Ω, which is acceptable. So a 50 cm long 5±0.02 mm diameter Al rod was purchased.

The calculations indicate that the main source of the impedance deviation from 50 Ωcomes from the positioning of the rod inside the cell. Shifting the rod vertically from thecenter by 20 µm will approximately have the same effect as a 0.02 mm change in diameter.Therefore, the precision of mounting the rod in the center is a critical issue for the 50 Ωimpedance matching. In the later measurements, it has been found that the transmissionproperties of the cell-rod coax were quite stable after disassembly and reassembly.

Designing a means of holding the rod in place and connecting it to standard coaxialcables at the up- and downstream ends was difficult. As it is shown in Fig. 15, the crosssections of the C2/C3 collimators and of the target cell are not constant. The varyingcross sections of the C2/C3 collimator will make the 50Ω impedance matching much moredifficult. However, as the cross section from the last part of the C3 collimator to the targetcell is approximately constant, the problem can be avoided by connecting the cell/rodwaveguide to a 50 Ω semirigid cable inside the C3 collimator. An elliptic copper block

19

was produced with spring fingers soldered around, as shown in Fig. 15. The block can beinserted into the end of the C3 collimator with the spring fingers providing good electricalcontact to the tungsten C3 collimator. An EZ141-SP semi-rigid cable was used to connectthe aluminum rod to outside RF devices. The outer metal shielding of the semi-rigid cablewas soldered to the copper block and the inner conductor was inserted into the copper endof the rod and soldered. At the downstream side a standard SMA connector with a longpin provides the electrical contact with the other end of the rod, as shown in Fig. 15. Thisconnector is screwed to a copper cap. The cap fits over a machined piece with the innerdimensions being the same as that of the inside of the target cell.

To keep the RF environment in the scattering chamber as close as possible to that ofthe HERMES experiment, a RF screen similar to the one for the LW was built. To findan acceptable perforated foil, Goodfellow [25] and Metaq [26] were contacted. One kind ofMetaq etched foil was chosen as it met our requirements of conductivity. The perforated foilis made of copper, with holes of 1×1 mm2 and 0.05 mm distance between holes (Fig. 16).The foil is put between a ring and a holding wheel with an indium wire inbetween whichprovides good electrical contact.

5.2 Transmission Properties of the Cell-Rod Coax

The Al rod and the surrounding cell form a transmission line, in which the cell acts asthe outer conductor and the rod acts as the inner conductor. The basic principle of atransmission line is introduced in App. A. After assembly the local impedance distributionalong the cell-rod coax was measured. For these measurements a step voltage is appliedto the downstream end of the cell-rod coax. The transmitted and reflected voltages arethen measured as a function of time. From this information the impedance is computedas a function of position along the cell-rod coax. The results of a typical measurementare shown in Fig. 17. The impedance along the cell varied between 38 Ω and 56 Ω. Thedimples along the cell (see section 4) are visible in the plot, they appear as small peaks onthe smoothly changing impedance. The connections at the two ends of the rod appear asa sharp peak or a deep valley, respectively. These sharp impedance changes will contributeto the reflection of high-frequency components. The big and slow valley in the middle isintroduced possibly by the misalignment of the rod from the center of the cell. This smoothimpedance change will contributes to the reflection of low-frequency components. As it isdifficult to tune the position of the rod inside the cell once it is installed, and as it turns outthe overall transmission efficiency is quite good (Fig. 18), no attempts have been made tofurther improve the impedance matching. The measurements are made with both the newspring fingers (shown in Fig. 17) and the old spring fingers (not shown) at the upstreamside. Changes are found observed only near the location of the changed spring fingers,namely the sharp peak and the valley in the ends. They do not affect the low frequencyfeatures of the transmission line.

More detailed information concerning the transmitted and reflected power as a functionof frequency can be obtained using a network analyzer to measure the scattering param-eters, i.e., a set of parameters to describe the transmission and reflection properties of anetwork. For example in a two-ported network, S11 is the ratio of the reflected signal at

20

port 1 over the input signal to port 1, while S21 is the ratio of the transmitted signal atport 2 over the input signal to port 1. The exact definitions can be found in App. A. Theworking principle of the network analyzer is as follows: There are 3 ports at the networkanalyzer and the ports of the analyzer can be connected to the ports of the network undermeasurement. After selecting the output port of the analyzer, the analyzer then can gen-erate calibrated sinusoidal signals at that port. In these measurements an Agilent networkanalyzer 8753ES is used. The frequency of the signals is automatically swept from 30 kHzto 6 GHz, or any frequency below 6 GHz. The reflected signals from the output port andthe transmitted signals coming out of the network under measurement are measured by thenetwork analyzer. Therefore, the scattering parameters of the network under measurementare determined at every frequency. The measured scattering parameters of the cell-rodcoax are shown in Fig. 18. The measurements performed with the cell-rod coax indicatethat S21 (the fraction of voltage transmitted through the cell-rod coax) is more than 90%for frequencies below 500 MHz, which is sufficient for our RF test.

21

target cellC2/C3 collimatorspring fingers

RF screen

Figure 15: Cell-Rod transmission line. See text for description.

Figure 16: Left: RF screen for RF test looking from the downstream side of the HERMES system.Right: RF screen for the LW detector looking from the upstream side.

22

Figure 17: Local Impedance distribution of the Cell-Rod transmission line (with new spring fingers)measured with the HP four-channel test set 54124A. Below the correspondent position dependenceof the cross section of the cell-rod transmission line is shown.

New design Spring fingers with an additional shielding

00.10.20.30.40.50.60.70.80.9

1

0 1 2 3 4 5 6

S11S21

Old design Spring fingers

00.10.20.30.40.50.60.70.80.9

1

0 1 2 3 4 5 6

S11S21

Figure 18: Scattering Parameters of the Cell-Rod transmission line measured with the Agilentnetwork analyzer 8753ES. The top plot was taken with new spring fingers while the bottom onewas taken with old spring fingers.

23

6 Magnetic Field in the Scattering Chamber

The magnetic field in the target chamber generated with the cell-rod coaxial waveguide wasmeasured by using a network analyzer and commercial magnetic field probes. The inputport 1 and the output port 2 of the network analyzer were connected to the cell-rod coaxialwaveguide, while the third port was connected to one of the probes. By measuring S13 (seeApp. A), the ratio of the voltage induced in the probe to the input voltage in the cell-rodwaveguide, it was possible to observe the magnetic field strength at the position of theprobe. To investigate the improvement in the shielding effectiveness provided by the newlydesigned spring fingers, measurements of S13 with both the newly designed spring fingersand the old ones at the upstream side of the target cell were performed and compared.During these measurements the RF screen between the scattering chamber and the pumpcross was removed so that moving and rotating one of the probes inside the scatteringchamber was possible.

Two magnetic field probes from Electro-Magnetics Inc. , namely EHFP-30/1, a 6 cmdiameter loop antenna, and EHFP-30/3, a 1 cm diameter loop antenna, were used in themeasurements. Since the characteristic response curves for these antennas are only availablebetween 0.1 MHz to 100 MHz, no attempt was made to calculate the absolute magneticfield strength. As the input voltage was kept constant, the ratio of the measured S13 withold and new spring fingers equals the ratio of the magnetic field at the position of theprobe. Only the measured S13 with the newly designed spring fingers and the old ones ata given frequency were compared as a measure of the shielding improvement.

The more sensitive probe, EHFP-30/1, was mounted at the center of the detectorsupporting frame (see Fig. 19) and it was sensitive to the magnetic field perpendicular tothe detector surface. To see if there is a possible cross talk between the analyzer and theprobe, S13 was measured when the input port and the output port of the network analyzerwere connected not through the cell-rod coaxial waveguide but directly to each other. Itwas found that in this case the measured S13 was close to zero. Therefore it was concludedthat the cross talk is not an issue in these measurements.

The S13 values measured with EHFP-30/1 are shown in Fig. 20 as a function of fre-quency, from 30 kHz to 1 GHz. Since the peaks of S13 shifted slightly between the twomeasurements, the reduction of the maximum value of S13 is taken as a measure of theimprovement in shielding effectiveness. The maximum S13 measured with the old springfingers is 1.3 × 10−6 and the one measured with the newly designed spring fingers is0.03 × 10−6. Therefore the RF magnetic field in the scattering chamber is reduced byabout a factor of 40 at its dominant frequency6.

The smaller probe, EHFP-30/3, was mounted on a handmade linear translator belowthe cell which could be moved along the z-direction7. The distance between the cell axis

6The output voltage of the magnetic probe can be obtained from the measured S13 values, multipliedby the input voltage on the cell-rod coax, which for example at 550 MHz is 80 mA×50 Ω=4 V. Thereforeat 550 MHz the voltage measured with this 6 cm diameter probe is 0.4 × 10−6 × 4 V=1.6 µV with the oldspring fingers, and 0.015 × 10−6 × 4 V=0.06 µV with the old spring fingers.

7The co-ordinate system used here is the right-handed HERMES co-ordinate system. Namely, the z-axispoints downstream along the electron beam direction, the x-axis is horizontal and the y-axis is vertical.

24

and the antenna was 2.5 cm. The upstream-side spring fingers are located at z=–20 cm.By changing the orientation of the probe, both the horizontal x and vertical y componentsof the magnetic field were measured. The field was measured from z=–25 cm to z=20 cm(the position of the cell downstream end).

S13 values measured with EHFP-30/3 are shown in Fig. 22 as a function of frequencyand z-position. S13x (S13y) is measured with the probe being oriented such that it wassensitive to the x (y) component of the magnetic field. It can be seen that the RF leakage isdecreased significantly with the newly designed spring fingers near the position of the springfingers. S13 measured with EHFP-30/3 at z=–20 cm, the position of the upstream-sidespring fingers, is shown in Fig. 21. It can be seen that the improvement in the shieldingeffectiveness is by about 2-3 orders in magnitude of the field strength, i.e., about 4-6 ordersin magnitude of the power. This result agrees approximately with the calculations byDohlus and Wipf [21], which states that the RF power leakage integrated over the spaceand frequency is decreased by 5 orders of magnitude.

Scattering Chamber

~2.5cm

2.1x8.9 cm

Scattering Chamber

Cell

o45 horizental angle,

EHFP-30/3

EHFP-30/1

EHFP-30/3

EHFP-30/1

~6cm

Cell

in the module center

at the positon of the silicon recoil detector

EndView Side View

Figure 19: Magnetic probes in the target chamber. A 1 cm diameter loop is fixed at the end of analuminum stick and can be moved along the target chamber, 2.5 cm below the center of the cell. A6 cm diameter loop is fixed at the center of the left-bottom (seen from the upstream side) siliconmodule position.

25

-0.2

0

0.2

0.4

0.6

0.8

1

1.2

1.4

0 0.2 0.4 0.6 0.8

Old spring fingers

-0.005

0

0.005

0.01

0.015

0.02

0.025

0.03

0 0.2 0.4 0.6 0.8

New fingers with an additional shielding

Figure 20: S13 as a function of frequency measured by EHFP-30/1 with old spring fingers (left)and with new spring fingers (right). Note the different y-scale.

-0.5

0

0.5

1

1.5

2

2.5

3

3.5

0 0.2 0.4 0.6 0.8

Old spring fingers

0

0.0025

0.005

0.0075

0.01

0.0125

0.015

0.0175

0.02

0 0.2 0.4 0.6 0.8

New fingers with an additional shielding

Figure 21: S13x measured when the probe was placed just below the upstream side spring fingersand oriented such that it was sensitive to the x component of the magnetic field, which correspondsto the left plots at z=–20 in Fig. 22. Note the different y-scale.

26

S31 measured with the old spring fingers

00.2

0.40.6

0.81

-25-20-15-10 -5 0 5 10 15 20 25

-0.50

0.51

1.52

2.53

3.54

z(cm)

S13x(10-6)

f(GHz)

00.2

0.40.6

0.81

-25-20-15-10 -5 0 5 10 15 20 25

-0.50

0.51

1.52

2.53

3.54

z(cm)

S13y(10-6)

f(GHz)

S13 measured with new spring fingers and an additional shielding

00.2

0.40.6

0.81

-25-20-15-10 -5 0 5 10 15 20 25

00.10.20.30.40.50.6

z(cm)

S13x(10-6)

f(GHz)

00.2

0.40.6

0.81

-25-20-15-10 -5 0 5 10 15 20 25

00.10.20.30.40.50.6

z(cm)

S13y(10-6)

f(GHz)

Figure 22: S13 as a function of frequency and z-position measured by EHFP-30/3 with old springfingers (top) and with new spring fingers (bottom). z is the position of the probe. z = 0 correspondsto the center of the cell, the same as in the standard HERMES coordinate system. z = −22corresponds to the position of the upstream side spring fingers. S13x (S13y) was measured whenthe probe was oriented such that it was sensitive to the x (y) component of the magnetic field.

27

7 RF Measurements

The RF influence to the SRD was studied by measuring the influence of the simulated RFfield to a SRD prototype module installed in the scattering chamber. Studies were alsodone on a readout hybrid alone which was not connected to silicon sensors. The latter aredescribed in subsection 7.1, and the former in subsection 7.2.

7.1 Measurement with a Readout Hybrid

The hybrid used in the measurement was a charge-injection hybrid [15], in which the HELIXchips were not connected to silicon sensors. A few channels of these chips were connected toa charge injection system. A picture of the charge injection hybrid is shown in Fig. 23. TheDAQ system in the measurement was the same as in the charge injection studies, where thesoftware triggered the charge injection system and the output digital signals of the HADCwere recorded for later data analysis. 2,500 events were taken at each frequency.

A three-turn 14 cm diameter loop antenna, acting as a transmitter, was used to coupleRF power directly to the readout hybrid. A sketch of the setup is shown in Fig. 24. Thetransmitting antenna was placed 9 cm above the hybrid. A three-turn 1 cm diameter loopantenna was placed 2 cm above the center of the hybrid to monitor the strength of the fieldpicked up by the hybrid. The hybrid and the antennas were kept in a 20 × 16 × 25 cm3box covered with aluminum foil which acts as a Faraday cage to prevent direct couplingbetween the RF generator and the readout electronics.

The transmitting antenna was connected outside of the box to a HP8656B sine-wavegenerator to an amplifier. Sine-wave signals going through the transmitting antenna gen-erated the RF field (mainly magnetic field) within the box. The frequency was variablebetween 0.1 MHz and 990 MHz. The 1 cm loop antenna (receiving antenna) was connectedoutside to an oscilloscope to monitor the voltage induced in the antenna. The voltage inthe receiving antenna was regarded as a measure of the RF field picked up by the hybrid.

The HP8656B generator used in this measurement provided the sine-wave signals witha frequency range of 0.1÷990 MHz. However, the amplifier only works up to 525 MHzwith maximum output amplitude of 15÷10 V. In the frequency range of 525÷990 MHz,the amplitude of the transmitter was limited by the generator to be below 1 V.

An influence of the RF field generated by the transmitting antenna to the hybrid read-out was observed mainly at low frequencies (

receiver were identical, the transmitter’s ability to transmit RF power had the same fre-quency dependence as the receiver’s sensitivity to pick up RF field. Therefore the inducedvoltage in the receiver should be proportional to the root of the receiver’s sensitivity. Thevoltage measured in the receiver in this calibration measurement was then a measure ofits sensitivity to the RF field as a function of frequency. In conjunction with the receiver’svoltages monitored during the measurements with the hybrid, the possibility of the largeRF influence at low frequencies due to large RF field generated by the transmitting antennawas excluded.

The observed frequency dependence of the hybrid’s sensitivity to external RF fieldagrees with the low-frequency bandwidths of the frontend electronics which are describedin subsection 2.1. This measurement also agrees with the measurements of the prototypemodule which will be described in the next subsection.

C inVin

HELIXInput

Charge Injection System

50Ω

Figure 23: Left: schematic of the charge-injection system. Right: charge-injection hybrid used inthe RF measurement (right). Vin is a signal from the pulse generator, attenuated to get the requiredinput charge. Cin is the charge-injection capacitance. It defines the amount of charge dq = Cin ·Vincollected by the charge-injection capacitance.

Figure 24: Schematic diagram of the setup to determine the sensitive frequency range of thefrontend electronics.

29

7.2 Measurements with Prototype Module

To investigate the influence of the beam-induced RF field to the SRD, measurements weredone with a SRD prototype module, Hamlet, in the scattering chamber. The beam-inducedRF field was simulated in the frequency domain by sine-wave signals, and in the timedomain by short pulses going through the cell-rod coaxial waveguide. The copper meshwas installed between the scattering chamber and the pump cross. With its n-side facingthe cell, Hamlet was installed in the left-top side (looking from the the upstream end) ofthe inner layer of the detector frame. Measurements were done with the newly designedspring fingers and also with the old ones between the target cell and the C3 collimator.

7.2.1 Frequency Domain Measurements

The purpose of the RF tests with Hamlet in the frequency domain was to study the sensitivefrequency range of the SRD to external RF field. This is also important for the time domaintest because, as discussed in section 3, longer pulses have to be used to simulate the HERA-ebeam in the time domain which do not include the high frequency components.

The test setup is shown in Fig. 25. Sine-wave signals generated by the HP 8656B,which are not synchronized to the clock signals of Hamlet, are amplified by an EIN 525LARF amplifier (25 W, 1÷500 MHz). The amplified signals then pass through the rod. Thefrequency spectrum of the transmitted signals is measured by an Agilent 8563EC spectrumanalyzer with a 40 dB attenuator in front. The frequency of the sine-wave signals is sweptfrom 2 MHz to 500 MHz in 2 or 5 MHz steps. The current along the cell-rod coax as afunction of frequency is plotted in Fig. 26. With the amplifier we used, the achieved currentsare 9÷15 times the Fourier coefficients of the HERA-e beam spectrum (In in Eqn. 10 andFig. 10).

At each frequency, 2500 events were taken with Hamlet when RF signals were goingthrough the rod, and 2500 events without RF. During the measurements, it was observedthat every time when the RF was turned off, Hamlet returned back to initial performancewithout the need of reprogramming the HELIX chips, which means that the RF did notpermanently change the settings of the electronics. Therefore the measurement resultswithout RF will not be discussed.

Plots demonstrating the influence of a 10 MHz unsynchronized sine wave signal toHamlet are shown in Fig. 27 (p-side). In the top-left panel is the histogram of the commonmode noise of the 1st chip (low gain chip) without RF; in the top-right panel is the histogramof the noise of the 2nd channel of that chip without RF. The CMN value for a given eventrepresents a shift in the output signal common to all channels while the noise after CMNcorrection represents fluctuations in a given channel. From Fig. 27 it is seen that without RFthe common shifts in the output were on the order of 8 ADC counts, while the fluctuationsof a single channel about this common shift were on the level of 2 ADC counts. In thebottom panels of Fig. 27 are the corresponding histograms with RF. It can be seen fromthe plots that, due to the external RF influence, the initial gaussian distributed CMN andnoise were changed to the following functional form:

f(x,A) =1

π·

1√A2 − x2

. (11)

30

R&S Attenuator

40dB, 0-1GHz

25W , 1-500MHz

525LA RF Amp.EIN

0.1-990MHzgenerator 8656BHP synthesized signal

Agilent 8563EC spectrum analyzer

RF off -> 5MHz RF -> RF off -> 10MHz ->RF off

-> ... ... ...

-> 495MHz -> RF off -> 500MHz RF

Hamlet

cell-rod transmission line

GPIB control

GPIB control

VME,HADC,HLCU,Helix ...

Figure 25: Schematic diagram of the RF test setup in the frequency domain.

600

800

1000

1200

1400

0 50 100 150 200 250 300 350 400 450 500

I(mA

)

f(MHz)

Spectrum in FDM

Figure 26: The current along the cell-rod coax as a function of frequency achieved in the frequencydomain measurement. Each point corresponds to one measurement.

This shape can be understood if the sinusoidal RF field is producing a sinusoidal shift inthe output values y = A sin 2πft where A denotes the maximum influence. As the detectortrigger was not synchronized to the RF generator, the sine-wave function will be sampledrandomly yielding an offset with the distribution of Eq. 11. The CMN collection based onthe first 16 channels of a chip does not completely correct for the RF disturbance so thata remaining influence is seen in the noise of the single channels with the same functional

31

form. The parameter A of the blue curves in Fig. 27 were fitted manually. In the bottom-left panel of Fig. 27, the blue curve corresponds to A'85. In the bottom-right panel theblue curve corresponds to A'15.

The results of the frequency domain test are plotted in Fig. 28. The plots show:

The RMS of the CMN of the high gain chip for N=2500 events with RF,

CMNrms =√

1N−1

∑Ni=1 CMNi,

and the RMS of the noise of all the 128 channels of that chip for 2500 events with RF,

Nrms =√

1M∗N−1

∑Mk=1

∑Ni=1 N

ki ,

where i denotes the event number, k denotes the HELIX channel number (k=1,M; M=128).

As it can be seen from the left panel (old design), the RF influence to Hamlet is confinedto frequencies below 50 MHz. The frequency dependence of RF influence can be explainedby the low frequency bandwidth of the electronics as discussed in section 2.1. From theright panel it can be seen that the improvement of the RF shielding at low frequencies ofthe newly designed spring fingers compared with the old ones is about a factor of 10, andthere is no special frequency range anymore where RF constitutes a real problem.

7.2.2 Time Domain Measurements

RF measurements were also done in the time domain with Hamlet. The RF field in thescattering chamber generated by the HERA-e beam was simulated in the time domain byshort pulses going through the cell-rod coax. The setup was similar to the one shown inFig. 25, but a HP 8082A pulse generator was used to generate short pulses instead of HP8656B. The short pulses were phase-locked to the 10 MHz detector clock. The relative timedelay between the clock and the pulses was measured with an oscilloscope. Because of thedifficulty to determine the absolute timing of the detector, the relative delay between theclock and the pulse was scanned from 0 to 95 ns in 5 ns steps. It is assumed that the largestmeasured effect of the RF field occurred when the timing was adjusted so that the pulsearrived at the electronics sufficiently in advance of the trigger signal to produce the largestsignal at the shape output. This timing should be the same as in the final experiment whenparticles are produced during a electron bunch crossing and the trigger timing is adjustedto produce the largest signal. It was learned from the frequency domain measurements thatthe dominant RF influence to Hamlet is below 50 MHz, mainly at 10, 20, 30, and 40 MHz.The pulse shape was optimized to achieve the maximum amplitude at 10 MHz, which wasabout a factor of 15 times larger than the first harmonics of the HERA-e beam spectrum.The measured spectrum is plotted in Fig. 29.

The synchronized short pulses to Hamlet mainly influenced the pedestals. In Fig. 30 thepedestal shift averaged over the 128 channels of the HELIX chip are shown in dependenceof the delay between the clock and the pulse. The improvement of the spring finger designis again about a factor of 10, which agrees with the frequency domain measurements.Moreover, the maximum RF influence (pedestal shift) also agrees with that found fromthe frequency domain measurements. For example, for the 1st HELIX chip of n-side, the

32

pcmn(1)

020406080

100120140160

-40 -20 0 20 40

MeanRMS

0.1120E-01 8.059

1st Chip CMN without RF

pnoise(2,1)

0

20

40

60

80

100

120

-5 0 5 10

MeanRMS

-0.4640E-02 2.431

2nd Channel Noise without RF

pcmn(1)

0

20

40

60

80

100

-100 0 100

MeanRMS

-0.4600E-01 67.82

1st Chip CMN with RF

pnoise(2,1)

0

20

40

60

80

100

120

-40 -20 0 20 40

MeanRMS

-0.2720E-01 11.90

2nd Channel Noise with RF

Figure 27: RF influence of 10 MHz unsynchronized sine-wave signals to the p-side of Hamlet. Thedata was analyzed with CMN substraction. Top-left: histogram of the common mode noise of the1st chip (low gain chip) without RF; top-right: histogram of the noise of the 2nd channel of thatchip without RF. In the bottom panels the corresponding histograms with RF are shown.

0

20

40

60

80

100

120

140

160

180

200

0 20 40 60 80 100 120

Old DesignNew Design

HG Chip Common Mode Noise

AD

C C

ount

s

f(MHz)

0

20

40

60

80

100

0 20 40 60 80 100 120 140

Old DesignNew Design

HG Chip Noise

AD

C C

ount

s

f(MHz)

Figure 28: CMNrms (left) and Nrms (right) as a function of frequency measured with the highgain chip.

maximum influence is approximately 100 ADC channels which agrees with Fig. 27 whereA=85 was obtained from the fit. From the measurements with the newly designed spring

33

fingers, the RF influence to the readout of Hamlet, due to synchronized short pulses withthe same current as the HERA Fourier coefficient, was expected to be 0.1 MIP.

0

200

400

600

800

1000

1200

1400

0 20 40 60 80 100

I(mA

)

f(MHz)

Spectrum in TDM

Figure 29: The current along the cell-rod coax as a function of frequency achieved in the timedomain measurement.

-25

-20

-15

-10

-5

0

5

10

15

20

25

0 20 40 60 80 100

HG Chip Common Mode Noise

New Design

AD

C C

ount

s

-250

-200

-150

-100

-50

0

50

100

150

200

250

0 20 40 60 80 100

Old Design

delay(ns)

Figure 30: Results of time domain measurement with newly designed spring fingers (top) and theold ones (bottom). The pedestal shifts averaged over the chips were plotted via the delay betweenthe detector clock and the pulses through the Al rod.

34

8 Conclusions

Measurements were done to investigate the influence of the RF field of the HERA-e beamon the silicon recoil detector. In a specially designed test set-up the electrical RF signalswere generated and sent through an aluminum rod mounted inside the target cell alongthe beam axis. In this way, the HERA-e RF field in the target chamber was simulatedin the frequency domain by sine-wave signals, and in the time domain by short pulsessynchronized to an external 10 MHz clock.

Measurements with the newly designed spring fingers and the old ones, between thetarget cell and the C3 collimator, were both done to study the improvement in the shieldingeffectiveness provided by the new design. The field strength of the magnetic field generatedwith the cell-rod waveguide in the scattering chamber was studied with a network analyzerand commercial magnetic field probes. Measurements were done at the position of thesupport frame of the silicon recoil detector and along the target cell as well. The reductionof the RF induced voltages in the magnetic probes with the newly designed spring fingerscompared with the old ones, depends on the position of the probes and on the frequency.It was found that the reduction is 2-3 (4-6) orders of magnitude in field strength (power)at the position near the spring fingers. This is compatible with calculations of Dohlus’ andWipf’s for the RF leakage outside the target cell, integrated over space and frequency.

Tests were also done with a prototype module of the SRD in the scattering chamber.Due to the limited time scale of the project, only a small fraction of the HERA-e spectrumhas been investigated, i.e., components with frequency below 500 MHz. Hence no definiteguarantee for the SRD can be drawn from this study. However, it was found that RFinfluence to the prototype detector module is mainly from RF field below 50 MHz, whichcan be explained by the low frequency bandwidth of the electronics. In this frequencyrange, the observed effect of RF influence was decreased by about a factor of 10 with thenew design spring fingers compared to the old ones. Therefore, if the possible influenceof the beam-induced RF field beyond the investigated frequency range, based on the lowbandwidth frequency of the SRD electronics, can be assumed to be negligible, the beam-induced RF influence on the SRD can be expected to constitute a constant shift of theoutput at the level of a 0.1 MIP signal at maximum.

35

A Transmission Line

Here the basic concepts of transmission line theory are introduced, i.e., impedance andscattering parameters of a transmission line, to be used in section 5.2.

The key difference between circuit theory and transmission line theory is the electricalsize of the system compared to the wavelength of the RF signals. Circuit analysis assumeslumped elements since the physical dimensions of a network are much smaller than theelectrical wavelength, while a transmission line may have a considerable size compared tothe wavelength. Thus a transmission line is a distributed lumped-element network wherevoltage and current can vary in magnitude and phase over its length. As shown in Fig. 18,a transmission is schematically represented by two wire line (for transverse electrical andmagnetic (TEM) wave).

v(z,t)

∆z

-

(a)

z

i(z,t)

-

++

-v(z,t)

(b)

∆

∆

zz

R∆∆

∆

i(z+ z,t)

v(z+ z,t)∆zG CL∆z

z

+

i(z,t)

Figure 31: Voltage and current definitions and equiv-alent circuit for an incremental length of transmissionline. (a) Voltage and current definitions. (b) Lumped-element equivalent circuit.

The short piece of the transmission line can be modelled by four lumped elements:

R= series resistance per unit length, for both conductors, in Ω/m.

L= series inductance per unit length, for both conductors, in H/m.

G= shunt conductance per unit length, in S/m.

C= shunt capacitance per unit length, in F/m.

The series inductance L represents the total self-inductance of the two conductors, andthe shunt capacitance C is due to the close proximity of the two conductances. The seriesresistance R represents the resistance due to the finite conductivity of the conductors, andthe shunt conductance G is due to the dielectric loss in the material between the conductors.A finite length of transmission line can be viewed as a cascade of sections of the form ofFig. 31b. From Kirchhoff’s voltage law and Kirchhoff’s current law follows:

v(z + ∆z, t) − v(z, t) = −R∆zi(z, t) − L∆zi(z, t)

t

i(z + ∆z, t) − i(z, t) = −G∆zv(z, t) − C∆zv(z, t)

t(12)

36

For the sinusoidal steady-state condition with cosine-based phasers, travelling wave solu-tions to Eq. 12 can be found as:

V (z) = V +o e−γz + V −o e

γz

I(z) = I+o e−γz + I−o e

γz =γ

R + jωL

[

V +o e−γz + V −o e

γz]

(13)

where γ = α + jβ =√

(R + jωL)(G + jωC) and the e−γz (e+γz) term represents wavepropagation in the +z (−z) direction. The characteristic impedance of the transmissionline, Z0, which relate the voltage and current one the line can be written as:

Z0 =V +oI+o

=−V −oI−o

=

√

R + jωL

G + jωC(14)

It is assumed that an incident wave of the form V +o e−jβz is generated from a source

(z < 0) and travelling along a transmission line with characteristic impedance Z0. Whenthere is a place z = 0 with impedance of ZL differing from Z0, a reflected wave will beexcited for z < 0:

V (z) = V +o e−γz + V −o e

γz ,

I(z) =V +oZ0

e−γz −V −oZ0

eγz , (15)

and at z = 0:

ZL =V (0)

I(0)=

V +o + V−o

V +o − V −oZ0. (16)

The reflected voltage normalized to the incident voltage is known as the voltage reflectioncoefficient Γ:

Γ =V −oV +o

=ZL − Z0ZL + Z0

(17)

The scattering parameter concept for an N -port network, the scattering matrix, pro-vides a complete description of the network at its N ports. The scattering matrix relatesthe voltage waves incident on the ports to those reflected from the ports. The scatteringmatrix is defined in relation to these incident and reflected voltage waves as:

V −1V −2. . .V −N

=

S11 S12 . . . S1N

S21...

...SN1 . . . SNN

V +1V +2. . .V +N

(18)

where V +n is the amplitude of the voltage wave incident on port n and V−n is the amplitude

of the voltage wave reflected from port n.

37

B Grounding and Shielding

The electrical setup of the silicon recoil detector concerning shielding and grounding isshown in Fig. 32. On the left-hand side (green) the double-sided silicon sensor (TIGRE) isshown. The sensor is biased via 6 MOhm polysilicon resistors, the analog output signals arecoupled via 1000 pF capacitors to the readout hybrids (n-Hybrid and p-Hybrid, black). Onthe hybrids, the analog signals are amplified, filtered and stored in an analog pipeline insidethe HELIX 3.0 IC. The HELIX uses +2 V (VDD), -2 V (VSS) and a GND potential inbetween VDD and VSS as supply voltages. In order to protect the integrated capacitors onthe sensors, the hybrid’s GND potentials are lifted to one half of the necessary sensor biasvoltage. Therefore, for a bias voltage of 60 V, the n-Hybrid is operated at a ground potentialof +30 V (= GND N), the p-Hybrid is operated at -30 V (= GND P). Between hybrids andthe beampipe (brown), the electrical connection is done by a two-layer polyimide flexfoil,on which all signal lines are realised with 50 Ohm impedance (microstrip setup). Thesupply voltage traces have a large width (e.g. 2 cm for VSS, 4.2 mm for VDD). For eachhybrid, there is one 25-pole Sub-D feedthrough in the beampipe (not shown). All signalsto and from one detector module are interfaced by one ACC module (brown). BetweenACC and beampipe, all signal lines are realised as individually-shielded, twisted-pair lines(CAT 7 Ethernet cables), while the power supplies are transferred through cables with alarge diameter (0.7 mm) and an overall shield. On the ACC, all necessary control signalsare lifted by transformes (TTL) or capacitors (LVDS) to the respective bias voltage. Theconnection to the VME Crate (blue) with the HLCU and HADC modules is again realisedwith shielded, twisted-pair lines (CAT 7 Ethernet, 30 m long). The analog output signalsare converted from bias level to ground level by an ac-coupling at the HADC input. Thehybrid’s supply voltages are generated by the NIKHEF LV Power Supplies (brown) closeto the ACC. The sensor’s bias voltages (+30 V and -30 V) are produced by high-voltagemodules (not shown) that are connected via a filter box to the bias lines on the ACC.

Since the GND lines on both hybrids are lifted to one half of the bias voltage, ananalog ground reference (0 V A, red line on hybrids and ACC) has been introduced. Allanalog voltages, especially BIAS and GND, are connected via a block capacitor to 0 V A.Measurements have shown, that the noise can be significantly reduced by connecting the 0V A between n-Hybrid and p-Hybrid by a short cable and also on the ACC. 0 V A shouldbe completely dc-current free. In order to achieve a high common-mode-noise rejection,all sensitive signals (LVDS and analog signals) are fully differential on the complete signalpath.

The silicon detector is fully floating referred to the beampipe ground. Even the safetyground that is connected to the ACC rack is provided from the Electronic Trailer 30 metresaway. The mechanical setup is isolated by ceramic distance pieces. In order not to havethe mechanical support structure of the silicon modules fully floating, they are connectedvia a large resistor to 0 V A on the hybrids.

The shields of the cables between ACC and beampipe are connected on the ACC sideto ground (SCR on ACC, blue) and not to the beampipe ground. On the hybrid’s side, theshields can be connected via 10 kOhm to 0 V A. The shields of the cables are not connectedto the shell of the respective connectors, but to signal pins which are then connected to

38

signal ground (SCR). SCR is also connected to the VME ground via the shields of the cablesbetween ACC and VME Crate. Different concepts concerning shielding and grounding canbe tested in the final environment at HERMES in a simple way by jumpers on the ACC.

References

[1] K. Rith, Prog. Part. Nucl. Physics 49 (2002) 245-324

[2] C. Baumgarten et al., Nucl. Instrum. and Meth. A, 496 (2003), 277

[3] The HERMES collaboration, The HERMES Recoil Detector Technical Design Report,DESY PRC 02-01 and HERMES 02-003

[4] T. Chen, et al., Report on Monte Carlo Studies for a HERMES C3 Collimator, HER-MES Internal 03-035

[5] W. Fallot-Burghardt, et al., Helix128-x User Manual V2.1 (1999), availableon http://wwwasic.kip.uni-heidelberg.de/ feuersta/projects/Helix/helix/helix.html;W. Fallot-Burghardt, Ph. D Thesis, Ruprecht-Karls-University Heidelberg, (1998);U. Trunk, Ph. D Thesis, Rupertus Carola University of Heidelberg, (2000);J. J. Velthuis, Ph. D Thesis

[6] A Large Acceptance Recoil Detector for HERMES, DESY PRC 01-01 and HERMES01-017; J. J. M. Steijger, Nucl. Instr. Methods A 447 (2000) 55-60; J. J. M. Steijger,Nucl. Instr. Methods A 453 (2000) 98-102; J. J. M. Steijger, Nucl. Instr. Methods A453 (2000) 98-102; J. Visser, et al., submitted to Nucl. Instr. Methods A

[7] J. J. M. Steijger, private communication

[8] M. van Beuzekom, private communication

[9] Lambda Wheel detector logbook

[10] R. L. Gluckstern and R. Li, Part. Accel. 29 (1990) 159

[11] A. Peisert, Silicon Microstrip Detectors, edited by F. Sauli, Instrumentation in HighEnergy Physics, World Scientific (1992), ISBN 981-02-0597-X

[12] AMS, 0.8µm CMOS Design Rules and Process Parameters, AMS (1995)

[13] M. Reinecke et al. , A silicon strip recoil detector for momentum measurement andtracking at HERMES, IEEE Trans. Nucl. Sci. , 51 (2004) 1111-1116

[14] B. Krauss, Optimisation of the Flex-Foil Layering for the HERMES Recoil Detector

[15] M. Kopytin et al, Decision on the readout chip for the new HER