long-range ultra-wideband radar sensor for industrial ... · ahmed abbas hussein ameri kassel,...

Ahmed Abbas Hussein Ameri

Long-Range Ultra-Wideband Radar Sensor for Industrial Applications

Ahm

ed A

bbas

Hus

sein

Am

eri

Long

-Ran

ge U

ltra-

Wid

eban

d Ra

dar

Sens

or fo

r Ind

ustr

ial A

pplic

atio

ns

ISBN 978-3-86219-442-1



kasseluniversity

press

This work has been accepted by the faculty of electrical engineering / computer sciences of the University of Kassel as a thesis for acquiring the academic degree of Doktor der Ingenieurwissenschaften (Dr.-Ing.). Supervisor: Prof. Dr.-Ing. G. Kompa Co-Supervisor: Prof. Dr.-Ing. A. Bangert Defense day: 28th November 2012 Bibliographic information published by Deutsche Nationalbibliothek The Deutsche Nationalbibliothek lists this publication in the Deutsche Nationalbibliografie; detailed bibliographic data is available in the Internet at http://dnb.dnb.de. Zugl.: Kassel, Univ., Diss. 2012 ISBN print: 978-3-36219-442-1 ISBN online: 978-3-8621944-3-8 URN: http://nbn-resolving.de/urn:nbn:de:00023443 © 2013, kassel university press GmbH, Kassel www.uni-kassel.de/upress Printed in Germany

http://dnb.dnb.de/

III

Acknowledgments

All praise to God who has been bestowing me the ability to complete

this thesis. I would like to express my deepest gratitude and appreciation to

my supervisor Prof. Dr.-Ing. G. Kompa. I am very much grateful for his

sustained guidance, strong motivation and remarkable inspiration in the

course of this work and for giving me the opportunities to work in different

research projects over the past several years.

I wish to express my special gratitude to the Prof. Dr.-Ing. A. Bangert, head

of the Department of Microwave Electronics Lab (MiCEL), University of

Kassel, for giving me the intellectual stimulation and academic support

through the continuation of this research work.

I am very much grateful to Prof. Dr. sc. techn. B. Witzigmann and Prof.

Dr.-Ing. habil. P. Lehmann, for their accepting to be members of the

disputation committee.

I am most thankful to the Otto-Braun-Fonds, who supported me financially

throughout the last 2 years.

My sincere thanks go to all of the colleagues Mr. R. Hadi, Mr. R. Chatim,

Mrs. R. Ghahremani, Dr. A. Zamudio, Dr. S. Dahmani, Dr. M. Monsi, Dr.

M. Rui, Mr. J. Weide, and Mr. C. Sandhagen, Mrs. H. Nauditt and all other

members of the Department of Microwave Electronics Lab, University of

Kassel, for their friendly cooperation and outstanding spirit teamwork.

My greatest acknowledgments go certainly to Erfurth Family for their

supports during my doctoral work in the University of Kassel.

I would like to offer my deep gratitude to my mother and my father for

their continuous guidance, encouragement, support, and prayers during my

life. I would also like to offer my gratitude to my brothers and my sisters

for their love and encouragement.


Kassel, November 2012

IV

Contents

1 Introduction 1

References 7

2 Overview of Non-Contact Sensors 9

References 17

3 Radar Equation for Level Sensing 19

References 29

4 UWB Antenna 31

4.1 Definition of UWB Antenna 31

4.2 Determining UWB Antenna Bandwidth 32

4.3 Design Challenging of UWB Antenna 33

References 35

5 Si-Based Avalanche Pulse Generator 36

5.1 Description of Ultra-Short Pulse Generator 37

5.1.1 Avalanche Transistor Circuit 39

5.1.2 Attenuator Circuit 48

5.1.3 Signal Divider 49

5.1.4 SRD Pulse Sharpener Circuit 51

5.1.5 Signal Combiner 58

5.2 Fabrication and Measurement Results 60

References 65

6 GaAs-Based Avalanche Pulse Generator 68

6.1 Breakdown Measurement 68

6.1.1 Two-Terminal Measurement 69

6.1.2 Three-Terminal Measurement 78

6.1.3 Experimental Avalanche Breakdown Analysis 79

V

6.1.4 Ultra-Short High-Power Pulse Generation 87

References 99

7 UWB Antenna Design 101

7.1 Design of TEM Horn Antenna 102

7.1.1 Design Procedure 102

7.1.2 Simulation and Measurement Results 104

7.2 Design of Double-Ridge Horn Antenna 109

7.2.1 Simulation Results 117

7.2.2 Size Reduction 122

7.2.3 Fabrication and Measurements 125

References 139

8 Ranging Measurements 142

8.1 Measurement Setup 142

8.2 Minimum Detectable Signal 144

8.3 Distance Measurement to Metal Plate 146

8.4 Distance Measurement to Bricks Wall 152

8.5 Water Level Control Measurement 156

8.6 Range Uncertainty 161

References 163

9 Conclusion and Future Work 164

Appendix 168

Appendix A 168

Appendix B 171

References 173

Own Publications 174

VI

List of Figures

Figure Entitled Page

Figure 1.1 FMCW radar transmitted signal. 3

Figure 1.2 Transmitted and received signal of FMCW radar. 4

Figure 2.1 Measurement signals of (a) pulse radar and (b) CW radar. 12

Figure 2.2 Peak and average power of CW and pulse signals. 13

Figure 2.3 Federal Communication Commission (FCC) masks for UWB

transmission systems. (a) power spectral density, (b) average

power. 16

Figure 3.1 Schematic of extended target detection. 19

Figure 3.2 Level control measurement with (a) wide beam width, (b) narrow

beam width. 20

Figure 3.3 Gaussian picosecond pulse. (a) time domain, (b) power spectrum. 24

Figure 3.4 Simulated received power as a function of range at frequency

bandwidth of 3.4 GHz. (a) for different values of transmitted power

(G = 7 dB), (b) for different values of antenna gain (PT = 20W). 26

Figure 3.5 Time-domain response of pulsed radar. 28

Figure 4.1 Determination of UWB antenna bandwidth. 33

Figure 5.1 Block diagram of proposed pulse generator. 38

Figure 5.2 Two terminal p-i-n junction. 38

Figure 5.3 Circuit diagram of avalanche transistor generator. 39

Figure 5.4 Photograph of used Si bipolar power transistor (chip). 40

Figure 5.5 IC-VCE avalanche breakdown characteristic of a bipolar transistor. 41

Figure 5.6 IC-VCE characteristic of a bipolar transistor for avalanche mode of

operation. 42

Figure 5.7 Amplitude of output pulse of avalanche transistor circuit versus

supply voltage VCC with CCC = 110 pF. 43

VII

Figure 5.8 FWHM of output pulse of avalanche transistor circuit as

a function of supply voltage VCC with CCC = 110 pF. 43

Figure 5.9 Rise-time of output pulse of avalanche transistor circuit as

a function of supply voltage VCC with CCC = 110 pF. 44


CCC. 44

Figure 5.11 Rise-time and FWHM of output pulse of avalanche transistor

circuit versus CCC. 45

Figure 5.12 Amplitude of output pulse of avalanche transistor circuit as a

function of RL. 46

Figure 5.13 Peak power of output pulse of avalanche transistor circuit

versus RL. 46


PRF. 47

Figure 5.15 Output waveform of avalanche transistor circuit. 48

Figure 5.16 Circuit diagram of attenuator circuit. 48

Figure 5.17 Output waveform of the attenuator circuit. 49

Figure 5.18 Circuit diagram of balun transformer (signal divider). 49

Figure 5.19 Output waveforms of the balun transformer (signal divider). 51

Figure 5.20 Circuit diagram of SRD sharpener circuit. 52

Figure 5.21 SRD in SOT-23 package (a), Schottky diode in SOD-323

package (b). 53

Figure 5.22 Signal flow diagram of SRD sharpener pulser. 54

Figure 5.23 Circuit diagram of the SRD pulser at circuit points V1, V2, V3

and V4. 54

Figure 5.24 Amplitude of output pulse of SRD sharpener circuit as a function

of input pulse amplitude. 56

Figure 5.25 FWHM and tr of output pulse of SRD sharpener circuit as

a function of input pulse amplitude. 56

VIII

Figure 5.26 Balanced output signals of the SRD sharpener circuits. 58

Figure 5.27 Circuit diagram of the transmission line transformer

(signal combiner). 59

Figure 5.28 Output waveforms of the signal combiner. 59

Figure 5.29 Complete ultra-short pulse generator. (a) avalanche transistor

circuit and attenuator circuit, (b) signal divider, SRD sharpener

circuit and signal combiner, and (c) photograph 61

Figure 5.30 Short discharge path of the avalanche transistor circuit. 62

Figure 5.31 Output waveform of the realized ultra-short pulse generator. 62

Figure 5.32 Normalized power spectrum of the pulse in Figure 3.30. 63

Figure 6.1 Photograph of the 0.8-m GaAs-MESFET device under test

manufactured by Mitsubishi (chip inside the package). 69

Figure 6.2 Schematic circuit diagram of the two-terminal breakdown

measurement.(a) gate-drain breakdown, (b) gate-source

breakdown, and (c) gate-drain and gate source breakdown. 70

Figure 6.3 Two-terminal pulsed breakdown measurement (drain-gate) at

different temperatures. 72

Figure 6.4 Two-terminal DC breakdown measurement (drain-gate) at


Figure 6.5 Two-terminal pulsed breakdown (source-gate) measurement at


Figure 6.6 Two-terminal DC breakdown measurement (source-gate) at


Figure 6.7 Breakdown voltage as a function of temperature obtained from

two-terminal pulsed I-V measurements. 74

Figure 6.8 Comparison of DC and pulsed breakdown voltages as a function

of temperature. 75

Figure 6.9 Two-terminal breakdown measurement (drain-gate and source-gate)

at T = 300K. 76

IX

Figure 6.10 Pulsed I-V measurement of gate-source. (a) forward and reverse

regions, (b) forward region. 77

Figure 6.11 Drain-current injection technique DC measurement of GaAs-

MESFET (MGF-1601B) device with injected drain-current ID = 1

mA/mm. 78

Figure 6.12 Schematic circuit diagram for pulsed breakdown measurement

(VGG = -4V, RDD = 525). 80

Figure 6.13 Pulsed breakdown measurement at VGG = -4V for different

temperatures (RDD = 525,VDD ≤ 40V). 80

Figure 6.14 Drain-source breakdown voltage as function of temperature. 81

Figure 6.15 Three-terminal pulsed I-V measurement for T = 300K

(VGG = -4V, RDD = 525). 82

Figure 6.16 Schematic circuit diagram for pulsed breakdown measurement

with gate current control (RGG = 1 k, RDD = 525, VGG = -4,

VDD ≤ 60V). 83

Figure 6.17 Measured pulsed breakdown characterization with snap-back

effect for T = 300K (VGG = -4V, RGG = 1 k, RDD = 525). 84

Figure 6.18 Pulsed breakdown measurement at T = 300K for different values

of RGG. 84

Figure 6.19 Schematic circuit diagram for pulsed breakdown measurement,

showing gate-drain and gate-source diodes in anti-series

(RGG = 1 k, RDD = 525 and VGG = -4V). 85

Figure 6.20 I-V characteristic gate-source diode of D1. 86

Figure 6.21 Pulsed breakdown measurement for different temperatures

(VGG = -4V, RGG = 1 k, RDD = 525). 87

Figure 6.22 Pulse generator circuit schematics based on GaAs-MESFET. 87

Figure 6.23 Photograph of pulse generator shown in Figure 6.23. 88

Figure 6.24 DC I-V characteristice of transistor in breakdown region. 89

X

Figure 6.25 Waveforms of output pulse of the circuit in Figure 6.23

(VDD = 12.4V). 89

Figure 6.26 Waveforms of output pulse of the circuit in Figure 6.23 for

different values of VDD (Vsupply). 90

Figure 6.27 (a) Charging, and (b) discharging circuit schematics of ultra-

short high amplitude pulse generator. 91

Figure 6.28 Complete circuit schematic of ultra-short high amplitude pulse

generator. 92

Figure 6.29 Waveforms of output pulse for different values of DC supply

voltage (VDD), (a) up to VDD = 55V and (b) up to VDD = 330V. 93

Figure 6.30 Amplitude of output pulse as a function of DC supply voltage

for T = 300K. 94

Figure 6.31 Normalized output pulses for different values of DC supply

voltage. 94

Figure 6.32 Rise time (tr) of output pulse as a function of DC supply voltage. 95

Figure 6.33 Pulse width (FWHM) of output pulse as a function of DC supply

voltage. 95

Figure 6.34 Amplitude of output pulse as a function of temperature. 96

Figure 6.35 Rise-time of output pulse as a function of temperature. 97

Figure 6.36 Pulse width of output pulse (FWHM) as a function of temperature. 97

Figure 6.37 Off-state and on-state breakdown I-V characteristic of transistor. 98

Figure 6.38 Waveforms of output pulse with floating gate and feeding gate

(VGG = -4, RGG = 1k) at VDD = 200V. 98

Figure 7.1 TEM horn antenna with (a) linearly tapered structure, and

(b) exponentially tapered structure. 103

Figure 7.2 Structure of TEM horn antenna with exponentially tapered plate. 103

Figure 7.3 Modified TEM horn antenna with cylindrically shaped aperture. 104

Figure 7.4 Photograph of the realized TEM horn antenna. 105

XI

Figure 7.5 Simulation and measurement result of input reflection coefficient

of TEM horn antennas. 106

Figure 7.6 Simulated and measured gain radiation pattern in E-plane of the

antenna at 1 GHz. 106

Figure 7.7 Simulated and measured gain radiation pattern in H-plane of the

antenna at 1 GHz. 107

Figure 7.8 Time-domain response of a first radar system with the brick wall

located at 16m from the radar sensor. (a) reference pulse and

reflected pulse, (b) reflected pulse. 108

Figure 7.9 Double-ridge horn antenna structure. (a) perspective view,

(b) side view. 109

Figure 7.10 Double-ridge waveguide structure. 110

Figure 7.11 TEM horn antenna with linearly tapered structure. 112

Figure 7.12 Horn section of the double-ridge horn antenna. (a) side view,

(b) top view. 113

Figure 7.13 Waveguide and horn sections structure (side view). 115

Figure 7.14 Configuration of the coaxial line to waveguide transition.

(a) top view, (b) side view. 117

Figure 7.15 The geometry of the antenna structure for HFSS simulation.

(a) 3D, (b) side view, (c) top view. 118

Figure 7.16 Simulated magnitude of S11 of designed antenna versus frequency. 119

Figure 7.17 3D view of radiation pattern of designed antenna at (a) 1.5 GHz,

(b) 3.2 GHz, (c) 6 GHz. 120

Figure 7.18 Current distribution on the surface of the designed antenna at

(a) 1.5 GHz, (b) 3.2 GHz, (c) 6 GHz. 121

Figure 7.19 Shorted double-ridge horn antenna. 122

Figure 7.20 Simulated magnitude of S11 of shorter antenna versus frequency. 123

Figure 7.21 Shorted antenna with circular ridge section. 123

Figure 7.22 Simulated magnitude of S11 of modified antenna versus frequency. 124

XII

Figure 7.23 Simulated magnitude of S11 of conventional and modified

double-ridge horn antennas versus frequency. 125

Figure 7.24 Realization of modified double-ridge horn antenna. (a) antenna

parts, (b) photograph. 126

Figure 7.25 VNA instrument setup for characterization of the antenna. 127

Figure 7.26 Simulation and measurement of the magnitude of input reflection

coefficient S11 of the modified double-ridge horn antenna versus

frequency. 127

Figure 7.27 Simulation and measurement of VSWR of the modified double-

ridge horn antenna versus frequency. 128

Figure 7.28 Simulation and measurement of the input resistance of the

modified double-ridge horn antenna versus frequency. 129

Figure 7.29 Simulation and measurement of the input reactance of the modified

double-ridge horn antenna versus frequency. 129

Figure 7.30 Transient radiation pattern measurement setup: (a) broadside

radiation, (b) edge-on radiation. 131

Figure 7.31 Measurement results of the far-field radiation of the antenna:

(a) broadside, (b) edge-on. 132

Figure 7.32 Gain radiation pattern measurement setup. 133

Figure 7.33 Simulation and measurement results of gain radiation pattern of

antenna at 1.8 GHz, (a) H-plane, (b) E-plane. 135


antenna at 4 GHz, (a) H-plane, (b) E-plane. 136


antenna at 6 GHz, (a) H-plane, (b) E-plane. 137


antenna at 7.5 GHz, (a) H-plane, (b) E-plane. 138

Figure 8.1 Measurement setup of the bi-static radar sensor. 142

Figure 8.2 Reference measurement point between two antennas of the

bi-static radar sensor. 143

XIII

Figure 8.3 Time-domain response of the radar sensor without any target. 144

Figure 8.4 Measured time significant points in pulsed laser radar. 146

Figure 8.5 Distance measurement setup for metal plate. (a) side-view,

(b) top-view. 147

Figure 8.6 Photograph of measurement setup. 148

Figure 8.7 Time-domain response of the radar sensor measurement towards

an aluminum plate (70 cm x 70 cm x 0.2 cm) located at

5.6m. (a) reference pulse and reflected pulse, (b) reflected

pulse and target ringing. 149


an aluminum plate (70 cm x 70 cm x 0.2 cm) located at 11.75m.

(a) reference pulse and reflected pulse, (b) reflected pulse and

(b) target ringing. 150


an aluminum plate (70 cm x 70 cm x 0.2 cm) located at 20m from

the radar sensor. (a) reference pulse and reflected pulse,

(b) reflected pulse and target ringing. 151


a brick wall located at 10.3m from the radar sensor. (a) reference

pulse and reflected pulse, (b) reflected pulse and target ringing. 153







Figure 8.13 Normalized received signal as function of distance. 156

Figure 8.14 Radar sensor measurement setup for water level control.

(a) side view, (b) front view. 157

Figure 8.15 Photograph of measurement setup for water level control. 158

XIV


water surface level of (a) 0 cm (empty tank), (b) 45 cm, and

(c) 75 cm. 159-160

Figure 8.17 Measured water levels as a function of the actual water levels. 161

Figure 8.18 Scattering of measured range data. 162

Figure 8.19 Distance error probability of radar range measurements. 162

Figure A.1 Types of attenuator configuration: (a) T-attenuator,

(b) Pi-attenuator. 168

Figure A.2 Bridge-t attenuator circuit. 169

Figure B.1 Measurement results of transformer ADT1-1WT. (a) insertion

loss, (b) input reflection coefficient. 171

Figure B.2 Measurement results of transformer TC1-1-43A+. (a) insertion

loss, (b) input reflection coefficient. 172

XV

List of Tables

Table Entitled Page

Table 2.1 Overview of Non-Contact Sensor Principles. 10

Table 2.2 Radar Systems in Comparison. 11

Table 3.1 Dielectric Constant and Reflection Coefficient of selected

Materials at 5.6 GHz. 22

Table 5.1 Data Sheet of the SRDs Manufactured by M-Pulse Microwave. 39

Table 5.2 Output Pulse Parameters of the Balun Transformer. 50

Table 5.3 Pulse Parameter for Different Lengths of Delay Line. 57

Table 5.4 Output Pulse Parameters of the SRDs Sharpener Circuits. 58

Table 5.5 Pulse Generator Parameters. 64

Table 7.1 Comparison of Antenna Types. 101

Table 7.2 Initial Dimensions of Waveguide Section Parameters. 111

Table 7.3 Initial Dimensions of Horn Section. 114

Table 7.4 Optimized Dimensions of Waveguide Section Parameters

(Figure 7.10). 115

Table 7.5 Optimized Dimensions of the Horn Section. 116

Table 7.6 Directional Radiation Beam of Antenna. 134

Table 8.1 Amplitude of Received Pulses. 145

Table 9.1 Summary of Generation Approaches of Ultra-Short High-Power

Pulses. 166

Table A.1 Resistive Element Equations of Attenuator Types. 169

Table A.2 Resistive Element Values of Attenuator Types for Different Loss

Values. 170

XVI

List of Symbols

BVDG Drain-gate breakdown voltage

BVDS Drain-source breakdown voltage

BVSG Source-gate breakdown voltage

BVCBO Collector-base breakdown voltage with open emitter

BVCEO Collector-emitter breakdown voltage with open base

AR Effective aperture of the receiving antenna

bw Fractional bandwidth

c Velocity of light

CB Series capacitor (DC block)

CCC Charging capacitor

fC Center frequency

fH High frequency limit

fL Low frequency limit

GT Gain of the transmitting antenna

GR Gain of the receiving antenna

ID Drain current

IG Gate current

Pav Average power

Ppeak Peak power

PT Power of transmitted signal

PR Power of received signal

RBE Base-emitter resistor

RDD Drain resistor

RL Load resistor

S11 Scattering coefficient at port 1

tf Fall time

tr Rise time

VCC Collector voltage

VCE Collector-emitter voltage

VDS Drain-source voltage

VGG Gate voltage

XVII

Vout Output voltage

f Difference frequency

Input reflection coefficient

Permittivity

r Relative permittivity

Free space permittivity

Free space wave impedance

Wave impedance )/( ro

Wavelength

Free space permeability

r Relative permeability

Permeability )( or

Conductivity

p Pulse width

Angular frequency

XVIII

List of Abbreviations and Acronyms

3D Three dimensions

BJT Bipolar junction transistor

BW Frequency bandwidth

CW Continuous wave

DC Direct current

FCC Federal Communication Commission

FMCW Frequency modulated continuous wave

FWHM Full width half maximum

GaAs Gallium arsenide

HFSS High-frequency structure simulator

HPBW Half-power beam width

IEEE Institute of Electronic and Electrical Engineers

LADAR Laser detection and ranging

MESFET Metal-semiconductor field effect transistor

PML Perfectly matched layer

PRF Pulse repetition frequency

SD Schottky diode

Si Silicon

SMA Sub-miniature version A

SNR Signal-to-noise ratio

SODAR Sound detection and ranging

SRD Step recovery diode

TEM Transverse electromagnetic

UWB Ultra-wideband

VCO Voltage-controlled oscillator

VNA Vector network analyzer

VSWR Voltage standing wave ratio

WRC World Radiocommunication Conference

TFE Thermionic field emission

XIX

Zusammenfassung

Die vorliegende Forschungsarbeit befasst sich mit dem Entwurf eines

weitreichenden Ultrabreitband (UWB)-Radarsensors für industrielle

Anwendungen. Er beinhaltet den Aufbau eines Impulsgenerators zur

Erzeugung extrem kurzer Impulse hoher Leistung sowie die Bereitstellung

einer Ultrabreitband-Antenne. Der entwickelte Radarsensor basiert auf dem

bi-statischen Systemkonzept.

Das Ziel dieser Arbeit ist es, Objekte mit einem niedrigeren

Reflexionskoeffizienten wie Backsteine mit einer relativen

Dielektrizitätszahl von etwa 4,5 bis auf eine maximale Entfernung von 20

m meßtechnisch zu erfassen. Um dieses Ziel zu erreichen, wurde ein

neuartiger (Radar-) Pulsgenerator entwickelt, der ultrakurze

Hochleistungsimpulse erzeugt. Der Pulsgenerator besteht aus einer

Transistorentladeschaltung mit einem Silizium-Transistor im Avalanche-

Betrieb und einer Impulsversteilerungsstufe unter Verwendung von Step-

Recovery-Dioden (SRD). Der entwickelte Pulsgenerator erzeugt elektrische

Impulse mit einer Anstiegszeit von 112 ps, einer Impulsdauer (FWHM)

von 155 ps und einer Amplitude von 34,5 V.

Über die Erzeugung von ultrakurzen Impulsen mit Hilfe von

Avalanche-Transistoren auf Silizium-Materialbasis und anschließender

SRD- Impulsversteilerung hinaus, wurde in dieser Arbeit auch die

Erzeugung von ultra-kurzen Hochleistungsimpulsen mit Hilfe von

modernen GaAs-MESFETs untersucht. Dazu wurden Strom-

/Spannungsmessungen (Zwei-Elektroden- und Drei-Elektroden-

Messungen) durchgeführt, um Durchbrucheffekte des gewählten

Transistortyps zu studieren. Mit der Aufspaltung eines konventionellen

Avalanche-Impulsschaltkreises in eine separate Ladeschaltung und einen

separaten Entladekreis wurde eine neue Technik gefunden, ultrakurze

Hochleistungsimpulse zu erzeugen. Es konnten Impulse mit einer

Anstiegszeit von 136 ps, einer Impulsbreite (FWHM) von 420 ps und einer

Amplitude von 169 V erzielt werden.

XX

Um die erzeugten ultrakurzen Impulse zu übertragen und zu

empfangen, wurden zwei verschiedene TEM-Hornantennentypen

entwickelt. Die erste Antenne ist eine Hornantenne, welche eine

zylinderförmige Anpassungsapertur besitzt. Diese Antenne weist eine

Bandbreite von etwa 1,45 GHz auf, welche sich von 250 MHz bis zu 1,7

GHz erstreckt. Um die Größe der TEM- Hornantenne zu verringern, wurde

eine weitere neuartige Antenne „Double-Ridge Horn Antenna“ entworfen.

Die Frequenzbandbreite der neuen Antenne erstreckt sich von 1,4 GHz bis

7 GHz. In Bezug auf die TEM-Hornantenne konnte die Länge dieser

Antenne und ihre Apertur um etwa 50 bis 70 Prozent reduziert werden.

Mit dem entwickelten Ultrabreitband-Radarsensor wurden

Testmessungen durchgeführt. Diese betrafen Entfernungsmessungen gegen

Metallplatten und Backsteinwände sowie Füllstandsmessungen in Wasser-

Behältern. Die Meßunsicherheit des Radarsensors ergab sich zu 14 mm

(Meßdatenstreuung bei festem Ziel: ± 14 mm).

XXI

Abstract

This research work deals with the design of long-range ultra-wideband

(UWB) radar sensor for industrial application. The design details include

the design of ultra-short, high amplitude pulse generator and ultra-wideband

(UWB) antenna. The developed radar sensor was built in a bi-static

configuration.

The goal of this work is to cover a maximum detection range of 20m

towards targets with lower reflection coefficients such as bricks with a

dielectric constant r of about 4.5. To achieve this goal, a novel ultra-short,

high power pulse generator (radar pulser) has been developed. The new

pulser consists of a transistor discharge circuit with silicon transistor

operating in the avalanche mode and a new step recovery diode (SRD) pulse

sharpening circuit. The developed pulse generator produces electrical pulses

with an amplitude of 34.5V, a rise-time of 112 ps and a pulse width

(FWHM) of 155 ps.

In addition to the generation of ultra-short pulses based on silicon

avalanche transistor and SRD pulse shaping circuit, the generation of ultra-

short high power pulse based on modern GaAs MESFET devices was also

investigated in this work. Two-terminal and three-terminal I-V

measurements were carried out to study the breakdown phenomenon of this

transistor type. By splitting the conventional avalanche pulse generator

circuit into separated charging circuit and separated discharging circuit, a

new technique to generate ultra-short high-power pulses was found.

Through this approach, very fast high amplitude pulses with rise-time of

136 ps, pulse width (FWHM) of 420 ps and amplitude of 169V were

obtained.

In order to transmit and receive the generated ultra-short pulses, two

different types of TEM horn antenna have been analyzed. First, a horn

antenna with cylindrical matching aperture was developed. This antenna

exhibits a bandwidth of about 1.45 GHz extending from 250 MHz up to 1.7

GHz. Then, to reduce the size of the TEM horn antenna a novel antenna

‘‘double-ridge horn antenna’’ has been designed. The frequency bandwidth

XXII

of the new antenna extends from 1.4 GHz up to 7 GHz. With respect to the

TEM horn antenna, the length and aperture size of the antenna could be

reduced by approximately 50 and 70 percent, respectively.

Using the developed ultra-wideband radar sensor, test measurements were

performed. These included distance measurements towards metal plates

and brick walls as well as water level control in tanks. The uncertainty of

the radar sensor has been found to be 14 mm (measured data scattered

within ±14 mm for a fixed target).

Chapter 1 Introduction

1

Chapter 1

Introduction

Non-contact radar-based level control sensors are widely used in industrial

processes such as liquid and solid level control measurement. Microwave,

optical (laser) and ultrasonic radar systems compete with each other. Any

measurement principle has its strength and application limitation [1].

Laser radar also termed LADAR (laser detection and ranging) has become

matured in recent years. It provides a narrow beam being able to measure

through small holes and openings [2]. The high beam spatial resolution can

also be used to image the 3D surface of a subject [3]. However, special

protection against dust and vapor must be taken into account. As shown in

[2], beam intensity attenuation by turbulences in the laser path can be

prevented by appropriate gas washing.

Ultrasonic radar also termed SODAR (sound detection and ranging) is

widely accepted as a low-cost technology. As the ultrasonic wave is a

material wave, the attenuation increases strongly with the operation

frequency [4]. Therefore, ultrasonic sensors for larger ranges will operate at

low frequencies.

Due to high air turbulence, there is a risk that the ultrasonic signal is blown

away so that receiving signal detection fails. This was already observed

with floodgate control [5]. Furthermore, because of high attenuation, the

transmitter of the ultrasonic radar must provide sufficient power to handle

large ranges, for example 20 meter as discussed in this thesis. This is

commonly accomplished by the combination of several ultrasonic

transducers, which make the sensor head more bulky [4]. As already


2

mentioned, propagation of ultrasonic wave is connected with elasticity of

gas molecules. Because the intensity of vibrations of the gas molecules

(mostly air) is temperature dependent, the velocity of ultrasonic waves

depends on temperature. And this can vary along the measured path.

Therefore, to calibrate the measured results, a temperature sensor is

frequently installed in front of the ultrasonic radar [6].

Microwave radar level sensors are widely accepted to be less prone to

industrial environment influencing parameters, such as dust, heat, vapor,

and deposition of mud. The antenna can be positioned in some distance

from the transmit/receive unit by waveguide interconnection.

However, beam focusing is a challenge at low frequencies of the ISM

(industrial, scientific and medical) bands [7], where the system price is still

affordable. Thus, with respect to system compactness, we have similar

restriction as was discussed with ultrasonic radar systems.

Regarding microwave systems in use (e.g. [8]-[9]), it is seen that FMCW

(frequency modulated continuous wave) radars are frequently offered by

the suppliers. They operate, for example, in X-band (from 8.5 to 9.9 GHz

and from 9.7 to 10.3 GHz [10]) and in K-band (from 24 to 26 GHz [6]).

They operate continuously with microwave power (in the range of 100 mW

[11]) and provide a focused beam of 40 degree and 18 degree in X-band

and K-band, respectively [12].

The FMCW radar transmits a continuous signal. The frequency of this

signal changes linearly with time during a time interval T, which

corresponds to a frequency difference f as shown in Figure 1.1. A voltage

controlled oscillator (VCO) is needed to ramp the signal between lower

frequency f1 and higher frequency f2. If the reflected signal is related to a

single target the distance between radar and target (R) is simply calculated

from the frequency difference (f) between the transmitted and received

(reflected) signal as follows.

From Figure 1.2, we find the relationship


3

T

f

t

f B

where

12 fffB

Solved for t, we get

Tf

ft

B

The distance R is calculated as

tc

R 2

By substituting (1.3) in (1.4), R is written as

Bf

fcTR

2

T is the time period and c is velocity of light. Range detection in industrial

tanks or containers is frequently hindered by obstructions inside the tank

like agitator blades, inlet valves and ladders. These obstructions can

provide additional reflections (false echoes). In this case, due to multiple

target detection, the evaluation of FMCW-radar-based receiving signals

becomes more complex.

Figure 1.1 FMCW radar transmitted signal.

(1.5)

(1.1)

(1.3)

(1.4)

(1.2)


4

Figure 1.2 Transmitted and received signal of FMCW radar.

It is noteworthy that the ultra-wideband (UWB) radar level sensor is less

discussed in the literature and far away from practical use [13]. The

principle of such a system is very simple. A switch provides a short

electrical pulse, which is directly radiated against the subject (solid or

liquid surface). The reflected pulse is recorded by an envelope (threshold)

detector. Thus, the operation principle needs no continuously operating

microwave source as a signal carrier. An oscillator is not needed, which

had to be stabilized with respect to temperature variation, and which would

operate within the allowed specified frequency bands [WRC: World

Radiocommunication Conference (formally WARC [14])].

Nevertheless, there are many problems to be solved yet to establish an

UWB system for practical use. Main issues are system compactness

(reducing antenna size and simultaneously maintaining narrow beam

width), high peak transmitted power, sharp pulse transients for high

measurement accuracy, and ultra-short pulses for low average radiated

power.

This thesis is aimed to analyze, design and fabricate novel components for

a pulsed radar systems and to combine them for an advanced radar system.

The completed system has been tested, and it has been proven that the

characteristic data are very promising for further industrial implementation.

The thesis is organized as follows:


5

In Chapter 2, an overview of non-contact ranging sensors is presented. The

objective and the goal of the current work are described.

In Chapter 3, the radar equation for level sensing is derived, and the effect

of radar equation parameters on the radar detection range is discussed.

In Chapter 4, the definition of UWB antenna and design challenges of the

antenna are presented.

In Chapter 5, a Si-based avalanche pulse generator is presented. First, a

description of the pulse generator is presented. It consists basically of an

avalanche transistor circuit and a pulse sharpener circuit. Then, effects of

the external parameter variation of the avalanche transistor circuit and

sharpener circuit on the generated pulse are discussed. Finally, the

fabrication of the proposed ultra-short pulse generator is presented.

Chapter 6 describes a new approach for the generation of ultra-short high

power pulses using GaAs-based transistors. First, conventional two-

terminal and three-terminal pulsed I-V measurements of GaAs-MESFET

device are performed to recognize the breakdown phenomena of this

transistor type. Then, a new device modulation technique is described,

which provides powerful spike-like pulses. Finally, the measurement of

generated pulse parameters as a function of supply voltage is discussed.

Chapter 7 deals with the design of UWB antennas. The design of different

types of TEM horn antenna is presented. First, the design procedure of

antenna structure is described. The antennas are optimized for reducing size

and maintaining sufficient bandwidth. Finally, the realization of the

antennas is discussed; measurement and simulation results are compared.

In Chapter 8, the measurement setup and measured results for the

developed UWB radar sensor are presented. First, choosing of the detection

threshold (minimum detectable signal) is described. Then, experiments

regarding distance and water level control measurements are discussed.


6

Distance measurement is performed with different targets such as metal

plates and brick walls. In addition, time-dependent measurement accuracy

of the radar sensor is presented.

Finally, conclusions are given in Chapter 9 with additional

recommendations for future work.


7

References

[1] D. Patrick, Industrial Process Control Systems, 2nd

Edition, Fairmont Press, Inc.,

Indian, 2009.

[2] G. Kompa, ‘‘Optical Short-Range Radar for Level Control Measurement,’’ IEE

Proceeding, Vol. 131, June 1984, pp. 159-164.

[3] A. Biernat, Erzeugung und Anwendung von ultrakurzen Laserradarimpulsen mit

hoher Leistung, Doctoral Thesis, University of Kassel, 1998.

[4] V. Magori, ‘‘Ultrasonic Sensor in Air,’’ IEEE Ultrasonic Symposium, October

1994, pp. 471-481.

[5] G. Kompa, Private Communications.

[6] Various Technics of Liquids and Solids Level Measurements, Indumart Inc.,

http://www.indumart.com.

[7] G. Kompa, High Frequency Sensors, Lecture Notes, University of Kassel, April

2002.

[8] Level Measurement with Radar, VEGA Grieshaber KG. www.vega.com.

[9] FMCW Radar Level Transmitter, PSM Instrumentation, www.psmmarine.com.

[10] P. Devine, Radar Level Measurement, VEGA Controls, England, 2000.

[11] D. Daniels, Ground Penetrating Radar, 2nd

Edition, Institution of Engineering

and Technology, UK, 2004.

[12] P. Heide, ‘‘24 GHz Short-Range Microwave Sensors for Industrial and

Vehicular Applications,’’ Workshop, University of Ilmenau, July 1999.

[13] C. Paulson, J. Chang, C. Romero, J. Watson, F. Pearce, N. Levin, ‘‘Ultra-

Wideband Radar Methods and Techniques of Medical Sensing and Imaging,’’

International Symposium on Optics, October 2005, pp. 1-12.

http://www.indumart.com/

http://www.psmmarine.com/


8

[14] F. Lyall, International Communications: The International Telecommunication

Union and the Universal Postal Union, Ashgate Publishing Limited, England,

2011.

Chapter 2 Overview of Non-Contact Sensors

9

Chapter 2

Overview of Non-Contact Sensors

A sensor is a transducer that converts energy from one form into another

[1]-[2], e.g. thermal energy is converted into electrical or chemical energy

into optical. More precisely, sensors are translators of a generally non-

electrical quantity into an electrical one [3]. They represent the connecting

link between environment and electronics, where they collect the

information about variables in environment, and provide the results as

electrical signals [4]. The application fields of sensors are very broad.

Regarding industrial use, the sensors play very important role in the

process technology. They are used to monitor the process sequences, or to

collect information about the current process state in order to guarantee

production quality.

In the industrial measurement technique, the sensors can be classified into

two kinds: Contact and non-contact sensors [5]. A contact sensor is a

sensor which physically touches the subject to be inspected. The direct

contact between the sensor probe and measured subject increases the risk of

mechanical abrasion, and thus provides the following disadvantages [2]:

Material wasting by abrasion

Material damaging by collision and vibration

Sensor damaging (deformation, surface defect)

Injury of functioning by mud or corrosion

Conversely, non-contact sensor is a sensor that does not touch the subject

to be inspected. Regarding industrial measurement approaches, non-contact


10

measuring principles are of increasing importance for surveillance and

control of automated processes [6]. An overview of non-contact sensors

with regard to the interaction field is presented in Table 2.1.

Table 2.1 Overview of Non-Contact Sensor Principles [2].

Time

variation of

interaction

field

Constant or slowly varying Fast varying

Energy

carrier

Electric field Magnetic

field

Electromagnetic wave

Acoustic

wave/Ultra-

sonic

Ultrasonic

method

Microwave Optical wave

Method Capacitive

method

Inductive

method

Microwave

method

Optical method

Sensor

examples

Capacitive

proximity

sensor

Inductive

proximity

sensor

Pulse radar

CW radar

Pulse radar

Camera systems

Pulse radar

Interaction

range

a few mm a few mm a few 100 m a few 100 m a few 10 m

Distance

resolution

mm mm mm ... dm mm ... dm mm

Cost Low Low Low ... high Low ... high Low ... mid

The capacitive and inductive sensors are classical near-field sensors [3].

They are robust, matured and low cost systems [2]. But the measurement

range of these sensors is restricted to the mm-range [7].

Regarding radar sensors, it generally represents a remote sensing approach,

which provides information by sending electromagnetic or sound waves

and receiving their reflection from objects [8]-[9]. Radar systems can be

distinguished according to the used kind of wave as follows:

Ultrasonic radar

Microwave radar


11

Laser radar

Comparison between radar system types is presented in Table 2.2.

Table 2.2 Radar Systems in Comparison [10].

Property Ultrasonic Microwave Laser

Dust influence + ++ -

Vapor influence + ++ -

Temperature influence - + +

Range attenuation - + +

Compactness - - +

Ultrasonic radar systems are known to be relatively cheap. But the high

value of ultrasonic attenuation limits the measurement range compared

with laser and microwave radars [4]. Practically, the measurement range of

ultrasonic radar systems is limited to about 20m [2]. Also, the

environmental noise, vibration, temperature and air motion can influence

the performance of ultrasonic systems.

Regarding laser radar systems, these systems provide a high focused beam

for measurement. But, they are more sensitive with respect to dust and

vapor [11].

On the other hand, the microwave radar systems involve some relevant

properties, which are very advantageous for industrial use. Dust, fog,

temperature gradient, vapor and mud may not significantly influence the

reliability of the system [10]. Microwave systems are therefore superior to

aforementioned approaches.

The microwave radar systems can be distinguished due to the signal

waveform used [2], [12]. The pulse radar emits and receives short pulses

(typically in millisecond or nanosecond region) with a low pulse repetition

frequency (PRF, typically in kHz region), while the CW radar transmits

and receives a CW (continuous wave) signal [2] as shown in Figure 2.1.


12

Figure 2.1 Measurement signals of (a) pulse radar and (b) CW radar.

By transmission of regularly spaced short pulses (instead of continuous

waves), the following benefits occur [12]:

The transmitted power (peak power, Ppeak) could be increased, while

keeping the average power (Pav) rather low. The average power of a

pulse train and CW signal are given by

TPP

p

peakav

2

peak

av

PP

where T is the pulse repetition time and p is the pulse width. From

(2.1), it can be seen that Pav of pulse signal depends on Ppeak, T, and

p. If the Ppeak is increased, the Pav can be kept low by increasing T or

(2.1) (Pulse train)

(2.2) (CW)


13

decreasing p as shown in Figure 2.2. The Pav of CW depends only on

Ppeak as presented in (2.2).

The distances of the number of targets can be easily distinguished by

measuring the time difference (d1, d2) between transmission and

reception pulses as shown in Figure 2.1.

In case of presence of several targets, the evaluation of the receiving

signals of CW radar systems becomes more complex. In the industrial

process, disturbances of any kind must be taken into account. Therefore,

the pulse microwave radar should be preferred for multiple target detection

in industrial applications [2].

Figure 2.2 Peak and average power of CW and pulse signals.

The Department of Microwave Electronics (MICEL), University of Kassel,

formerly High Frequency Engineering (HFT), has long experience in near-

field laser and microwave radar technology. With respect to microwave

radar, a first pulsed radar sensor for near-range detection and ranging was

built in 2001 [13]. Due to the excellent antenna performance, the antenna

was rebuilt by a research group of University of Helsinki for mobile

communications [14]. The microwave sensor was developed in bi-static

configuration based on the developed pulsed laser radar sensor described in

[15]. The key point of the development was to replace the measuring

optical head of the laser radar by an ultra-wideband (UWB) antenna to


14

make ultra-short pulse (in picosecond region) operation possible. The

UWB antenna in [13] covers a frequency range of 1 GHz to 5 GHz. Ultra-

fast electrical pulses with pulse width of 150 ps, rise-time of 130 ps, and

peak power of 600 mW had been used as transmitted signal. In 2007, an

UWB radar sensor for near-field detection was developed [16] with the key

point to reduce the size of the antenna and to search for a mono-static radar

configuration solution. The designed antenna operated in a frequency range

of 0.65 GHz to 20 GHz [16]. The antenna had a size of about 50% of

antenna size in [13]. Further goal was to establish extremely short

transmitted pulses. Short pulse with pulse width of 75 ps and rise-time of

50 ps was generated in [16]. However, the peak power of the transmitted

pulse was only 50 mW.

The radar sensors designed in [13] and [16] exhibited high ranging

accuracy of about 6 mm, but the maximum available detection range was

only about 1m using objects with high reflection coefficient, such as metal

plate and water.

Near-field radars with measurement range in the meter region and

measurement accuracy in the millimeter range are very attractive for

applications in the production area. Such systems can be used, among

others, for level control measurement of solids and liquids.

The goal of this thesis is to develop an UWB radar sensor which covers an

extended measuring range of about 20m with an accuracy in the mm range

towards targets with lower reflection coefficients such as bricks with a

dielectric constant r of about 4.5 [17].

As discussed in [2], the detection range of a radar depends on a number of

factors, i.e. power of the transmitting pulse, gain of the transmitting and

receiving antennas, reflection coefficient of the target, and power of the

reflected pulse. One possibility to increase the detection range is to increase

the power of the transmitting pulse.

As reported in [16], the measurement accuracy (range error) depends on a

number of factors, i.e. rise time of the transmitted pulse, signal to noise


15

ratio (SNR), and number of measurement events. One possibility to

increase the range accuracy is to reduce the rise time of the transmitted

pulse [18].

The emission of an UWB radar must comply with the power spectral

density regulation [19] as shown in Figure 2.3(a). It can be seen that the

radiation mask for both indoor and outdoor systems should not exceed the

-75 dBm/MHz and -41 dBm/MHz limit for the frequency range from 0.96

GHz to 1.61 GHz and from 3.1 GHz to 10.6 GHz, respectively. For the

frequency from 1.99 GHz to 3.1 GHz, the radiation mask for indoor and

outdoor system are -51 dBm/MHz and -61 dBm/MHz, respectively. The

maximum allowed emission power as a function of frequency is shown in

Figure 2.3(b) [20]. It can be seen that maximum average power, which is

allowed to emit is about 0.8 mW at 10.6 GHz.

For transmitting and receiving ultra-short pulses, UWB antennas must be

available. The size minimization of the antenna is one of the key challenges

in UWB radar sensor design because the antenna is frequently the largest

component in the system [21].


16

(a)

(b)

Figure 2.3 Federal Communication Commission (FCC) masks for UWB transmission

systems [19]. (a) power spectral density, (b) average power [20].


17

References

[1] I. Sinclair, Sensor and Transducers, Newnes, Elsevier Inc., Linacre House,

Jordan Hill, Oxford, UK, 2001.


2002.

[3] J. Fraden, Handbook of Modern Sensors, Physics, Designs, and Applications,

Springer-Verlag New York, Inc., 2004.

[4] V. Magori, ‘‘Ultrasonic Sensor in Air,’’ IEEE Ultrasonic Symposium, October

1994, pp. 471-481.

[5] B. Colosimo, and N. Senin, Geometric Tolerances, Springer-Verlag London,

2011.

[6] J. Wilson, Sensor Technology Handbook, Newnes, Elsevier Inc., Linacre House,

Jordan Hill, Oxford, UK, 2005.

[7] S. Saha, Introduction to Robotics, Tata McGraw-Hill Publishing Company

Limited, New Delhi, 2008.

[8] P. Zahng, Advanced Industrial Control Technology, Elsevier Inc., Oxford, UK,

2010.

[9] J. Webster, The Measurement Instrumentation and Sensor Handbook, CRC

Press LLC, USA, 1999.

[10] Pulse Radar Type, Matsushima Machinery Laboratory Co., LTD., 2007.

[11] M. Monsi, Laser Radar for Precise Vehicle Velocity Measurement, Doctoral

Thesis, University of Kassel, April 2009.

[12] R. Locher, and A. Pathak, ‘‘Use of BiMOSFETs in Modern Radar

Transmitters,’’ IEEE International Conference on Power Electronics and Drive

Systems, 2001, pp. 711-717.

[13] A. Duzdar, Design and Modeling of an UWB Antenna for a Pulsed Microwave

Radar Sensor, Doctoral Thesis, University of Kassel, July 2001.


18

[14] P. Eskelinen, and T. Tarvinen, ‘‘Improving An Inverted Trapezoidal Antenna for

Mobile Communication,’’ 13th

IEEE International Symposium on Personal,

Indoor and Mobile Communications, September 2002, pp. 1266-1269.

[15] G. Kompa, “Extended Time Sampling for Accurate Optical Pulse Reflection

Measurement in Level Control,’’ IEEE Transactions on Instrumentation and

Measurement, vol. IM-33, 1984, pp. 97-100.

[16] A. Ruengwaree, Design of UWB Radar Sensor, Doctoral Thesis, University of

Kassel, November 2007.

[17] A. Molisch, Wireless Communications, 2nd

, Wiley-IEEE Press, UK, 2011.

[18] A. Wehr, and U. Lohr, ‘‘Airborne Laser Scanning,’’ Journal of Photogrammetry

and Remote Sensing, 1999, pp. 68-82.

[19] Federal Communications Commission, Notice of Inquiry in the Matter of:

Revision of Part 15 of the Commission´s Rules Regarding Ultra-Wideband

Transmission Systems, Document # 02-48, April 2002.

[20] C. Corral, S. Emami, and G. Rasor, ‘‘Ultra-Wideband Peak and Average Power

Limits,’’ Consumer Communications and Networking Conference, January

2006, pp. 478-481.

[21] G. Cheng, T. Ho, W. Wang, C. Chang, and S. Chung, ‘‘Highly Integrated

Automotive Radar Sensor,’’ Electronics Letters, vol. 43, August 2007, pp. 993-

994.

Chapter 3 Radar Equation for Level Sensing

19

Chapter 3

Radar Equation for Level Sensing

Radar technology had been basically developed for military purposes.

Search radar systems provide long range air traffic control and surveillance.

In this classical radar application, the target (aircraft) has small dimension

compared to the radar beam dimension, i.e., angle and range resolution of

the radar. In this case the target is denoted as ‘‘single-point’’ target. The

received power PR of a point target decreases with distance according to PR

~ 1/R4 (point target radar equation) [1]. In contrast, regarding level

measurement of solids and liquids, the material measuring plane totally

covers the radar beam. Such radar targets are denoted as distributed,

extended or area target [2], as shown in Figure 3.1.

Figure 3.1 Schematic of extended target detection.


20

Level control measurement in tanks and silos can suffer from pulse

reflection of radar signals at container walls or other obstructions inside the

tank [3], as illustrated in Figure 3.2(a). Therefore, it is important that the

radar measuring beam is sufficiently focused so that the incident radar

signals impinge completely onto the material surface [Figure 3.2(b)].

(a) (b)

Figure 3.2 Level control measurement with (a) wide beam width, (b) narrow beam

width.

The radar range equation of an extended target can be derived as follows

[4].

Under assumption that specular reflection occurs at the target surface, the

power density S(R) at the receiving antenna (at a distance 2R from the

transmitting antenna) can be calculated as [5]

224 R

PRS T

where PT is the power of transmitted signal and R is the range. Taking into

account the gain of the transmitting antenna (GT) and the reflection

(3.1)


21

coefficient () at the object surface, the power density at the receiving

antenna can be written as

2

224

TT G

R

PRS

The reflection coefficient of the target surface can be expressed by [6]

0

0

with

j

and

0

00

where and are the wave impedance of the target (solid, liquid) and free

space, respectively. , , and are the conductivity, permeability, and

permittivity of the target material. and are the permeability and

permittivity of free space, respectively. denotes the angular frequency.

For conductive surfaces, such as metal plate and highly conductive liquids,

the conductivity goes to infinity and with (3.4) becomes 0. In this case

the reflection coefficient of the target surface () is equal to -1.

For non-conductive surfaces, is equal to 0. Then in (3.4) becomes

For non-magnetic materials with and = r 0, can be written

as

(3.2)

(3.3)

(3.4)

(3.5)

(3.6)


22

0

0

r

where r is relative permittivity. By substituting (3.7) and (3.5) in (3.3),

becomes

r

r

1

1

Table 3.1 shows the dielectric constant (r) and reflection coefficient () of

selected materials, which frequently occur in industrial processes.

Table 3.1 Dielectric Constant and Reflection Coefficient of Selected Materials at 5.6

GHz [7].

Material r

Water 73 (200 C) -0.79

Bricks 4.5 -0.33

Cement 2.6 -0.23

Coal 2.5 -0.22

Oil (petroleum) 2.2 (200 C) -0.2

The received power PR at the radar receiver is dependent on the power

density S at the receiving antenna and on the effective aperture of the

receiving antenna AR [8]. PR is calculated as [9]

SAP RR

The effective aperture of antenna is a measure of how much power the

antenna captures from the power density of a plane wave incident upon the

antenna [8], and it depends on the direction of the incident wave [9]. In

case that the incoming wave is in the direction of antenna directivity, AR is

given by [10]

(3.8)

(3.7)

(3.9)


23

4

2R

R

GA

where GR is the gain of the receiving antenna and is the wavelength. By

substituting (3.2) and (3.10) in (3.9), PR can be written as

2

22

2

24

R

GGPP RTT

R

With GT = GR = G, (3.11) becomes

2

22

22

44

R

GPP T

R

It can be seen that the received power is, in contrast to the point target, only

inversely proportional with the square of R (PR 1/R2). From (3.12), the

radar range equation of the extended target can be written as

8G

P

PR

R

S

To estimate the needed values of radar range equation parameters, required

to achieve the intended minimum detection range of 20m, (3.13) has been

analyzed. Before starting with simulation of radar range equation, the range

of parameters values of (3.13) is defined according to available radar

source and antenna.

For UWB microwave radar, the rise-time of the transmitted pulse typically

ranges from tens of picosecond to a hundreds of picosecond [11], [12]. A

measurement-based ultra-short Gaussian pulse with rise-time of 90 ps

shown in Figure 3.3(a) has been used as transmitted signal in the simulation

of radar range equation. Such pulse can be generated using a SRD (Step

Recovery Diode) sharpener circuit as discussed in [13]. The power of the

pulses based on SRD sharpener circuits ranges from tens of mW to tens of

watts as discussed in [11], [12]. Therefore, the amplitude and the peak

(3.13)

(3.11)

(3.10)

(3.12)


24

power of the pulse shown in Figure 3.3(a) are chosen to be 36V and 26W

(under 50 matching conditions), respectively. The power spectrum of this

pulse is shown in Figure 3.3(b). From this figure and based on the

bandwidth defined by 20 dB power drop of the transmitted pulse, which

includes 90% of the total pulse energy [12], it can be seen that the pulse

exhibits a bandwidth of around 4 GHz.

(a)

(b)

Figure 3.3 Simulated Gaussian picosecond transmitted pulse. (a) time domain, (b)

power spectrum.


25

To transmit the energy of the transmitted pulse shown in Figure 3.3, a TEM

antenna is needed. The TEM horn antenna represents a very attractive

option for radar application because this type of antenna has some special

features such as high gain, low VSWR, and relatively simple construction.

Therefore, in the simulation of radar range equation, horn antenna has been

used. In general, the bandwidth of the antenna is determined using

impedance bandwidth approach as will be discussed in the next Chapter. In

this approach, the bandwidth of the antenna is defined for a frequency

range where the reflection coefficient () at the antenna feed is equal or

less than -10 dB (see Figure 4.1).

To cover mostly the bandwidth of the transmitted pulse shown in Figure

3.3(a), a horn antenna with a low-limit frequency of about 600 MHz and a

bandwidth of about 4 GHz is suggested.

The TEM horn antenna with moderate aperture size (in region of hundreds

of cm2) and bandwidth of about 4 GHz (0.6 – 4.6 GHz) exhibits gain values

varying between 3-12 dB over a frequency range of 1 to 4 GHz as shown in

Figure 3.4(b) [14].

Two different targets have been used in this simulation. First target is a

metal plate ( = -1). Second target is a brick material with a dielectric

constant of r = 4.5.

Regarding minimum detectable signal (MDS), it depends on the noise level

(N) of receiver and signal-to-noise ratio (SNR). It can be calculated as [15]

NSNRMDS )(

For radar sensor application, the SNR value is suggested to be about 8 dB

[4]. For the receiver (sampling oscilloscope DSO81204B type) with the

noise level of -33 dBm, the MDS can be calculated to be -55 dBm.

After estimation of system parameters, simulation of radar range equation

is performed. In Figure 3.4, the received powers as a function of range for a

frequency bandwidth of 3.4 GHz (0.6 GHz - 4 GHz) for different targets,

(3.14)


26

different values of transmitted power, and different values of antenna gain

are presented.

(a)

(b)

Figure 3.4 Simulated received power as a function of range for a frequency bandwidth

of 3.4 GHz. (a) for different values of transmitted power (G = 7 dB), (b) for different

values of antenna gain (PT = 20W).


27

Figure 3.4(a) shows the received power for an antenna gain of 7 dB and

different values of transmitted peak power. It can be seen that the

maximum detection range of radar towards bricks target (r = 4.5) increases

from 18m to more than 20m with increase of transmitted peak power of

pulse from 10W to 30W, respectively.

The influence of antenna gain on the radar range is shown in Figure 3.4(b).

In this figure, the received powers are calculated with 20W peak

transmitted power and different values of antenna gain. It can be seen that

maximum detection range of radar towards bricks target (r = 4.5) increases

from 13m to more than 20m with increase of antenna gain from 4 dB to 10

dB, respectively.

Regarding pulse radar, the transmitted pulses radiated by the transmitting

antenna, are reflected by the target and return in the direction of the radar.

The reflected pulses are collected by the receiving antenna and detected by

the receiver. The time interval between the transmitted and received pulses

is a measure for the distance of the target [4].

The elapsed time for a microwave pulse transmitted from the transmitter

antenna, reflected by the target and received by the receiver antenna is used

to determine the range of the target (R), which reads

2

tcR

where t is the elapsed time between transmitted and target return pulses and

c is the velocity of light in free space. The maximum distance of a target

from the pulsed radar at which the detection is possible refers to the

maximum range of radar. As shown in Figure 3.5, the maximum elapsed

time between the transmitted and reflected pulse must be less than or equal

to T (pulse repetition time) to avoid multiple transmitted pulses and

ambiguity in range measurement. The maximum measureable range (Rmax)

can be written as

(3.15)


28

2max

TcR

Figure 3.5 Time-domain response of pulsed radar.

(3.16)

Time

Time

Am

plit

ud

e


29

References

[1] M. Skolnik, Introduction to Radar Systems, New York: McGraw-Hill, 1962.

[2] W. Wiesbeck, Radar System Engineering, Lecture Notes, IHE, University of

Karlsruhe, October 2007.

[3] P. Devine, Radar Level Measurement, VEGA Controls, England, 2000.


2002.

[5] H. Schantz, The Art and Science of Ultra-Wideband Antennas, Norwood: Artech

House Inc., 2005.

[6] Y. Ju, K. Inoue, and M. Saka, ‘‘Contactless Measurement of Electrical

Conductivity of Semiconductor Wafers Using the Reflection of Millimeter

Waves,’’ Applied Physics Letters, vol. 81, November 2002, pp. 3585-3587.

[7] W. Telford, Applied Geophysics, 2nd

Edition, Cambridge University Press, US,

1990.

[8] R. Yadava, Antenna and Wave Propagation, PHI Learning Private Limited,

2011.

[9] R. Chatterjee, Antenna Theory and Practice, New Age International Limited,

New Delhi, 1998.

[10] L. Barclay, Propagation of Radiowaves, 2nd

Edition Institution of Engineering

and Technology, UK, 2003.





[13] P. Protiva, J. Mrkvica, and J. Machac, “Universal Generator of Ultra-Wideband

Pulses,” Radioengineering 17, December 2008, pp. 74-78.


30

[14] Horn Antennas, A. H. Systems Inc., www.ahsystems.com/catalog/horns.php.

[15] A. William, Space Antenna Handbook, John Wiley, UK, 2012.

Chapter 4 UWB Antenna

31

Chapter 4

UWB Antenna

The antenna acts as a transducer which converts the signal on a

transmission line into electromagnetic waves in free space in the

transmitting phase and vice versa in receiving phase [1].

In the UWB radar systems, the antennas play a crucial role. It is used to

transmit and receive very short-time duration pulses (in picosecond range)

[2], [3]. In these systems, the function of the antenna during transmission

phase is to concentrate the radiated energy into a shaped beam that points

in the desired direction and illuminates only the selected target and during

reception phase, it collects the energy reflected by the target and delivers it

to the receiver.

In the following sections the definition, bandwidth calculation, and design

challenges of an UWB antenna are presented.

4.1 Definition of UWB Antenna

The major difference that distinguishes the UWB antenna from a

narrowband antenna is the large operation bandwidth. The bandwidth can

be described in different ways. In simple way, the frequency bandwidth

(BW) is defined as the difference between the upper (fH) and lower (fL)

operation frequencies [4],

LH ffBW (4.1)


32

In addition, the frequency bandwidth of the system is often described based

on the center frequency (fC), which is defined as the arithmetic average of

the upper and lower frequencies [4],

LHC fff 2

1

Another method for describing the frequency bandwidth of the antenna

defines the bandwidth as a ratio (Br) of the upper frequency to the lower

frequency [2],

L

Hr

f

fB

According to DARPA [5] and FCC [6], an UWB antenna is defined with

fractional bandwidth greater than 0.25 and 0.2, respectively. The fractional

bandwidth (bw) is defined as the ratio of bandwidth (BW) to the center

frequency (fC) [2],

Cf

BWbw

An alternative definition has been provided by FCC. It considers any

antenna with a bandwidth greater than 500 MHz as an UWB antenna [6].

4.2 Determination of UWB Antenna Bandwidth

In order to determine the bandwidth of the antenna, the values of upper and

lower operation frequencies should be known as shown in the equations in

section 4.1. There are many approaches to determine the bandwidth of the

antenna. This can be based on impedance or gain [4]. However, the

approach that takes into account all antenna properties which are important

to a particular application is a preferred approach to define the bandwidth

of the antenna. Thus, the Institute of Electrical and Electronics Engineers

(IEEE) standard says: “The bandwidth of an antenna is defined as: the

range of frequencies within which the performance of the antenna, with

(4.2)

(4.3)

(4.4)


33

respect to some characteristics, conforms to a specified standard”. In this

work, the impedance bandwidth approach was used and defined with the

goal that the reflection coefficient () at the antenna feed is equal or less

than -10 dB, within the defined bandwidth. Using this approach, the upper

and lower operating frequencies are defined as the endpoints of frequency

range across which the antenna meets the impedance goal as shown in

Figure 4.1.

Figure 4.1 Determination of UWB antenna bandwidth.

4.3 Design Challenges of UWB Antenna

One of the key problems to design UWB radar sensor is the availability of

an antenna, which transmits and receives very short-time duration pulses.

The following requirements should be taken into account designing an

antenna for UWB radar sensors:

In order to cover the bandwidth of the ultra-short pulses, the UWB

antenna should be able to yield a large bandwidth. According to the

FCC's definition, the frequency bandwidth of UWB antenna should

be greater than 500 MHz.

To minimize the pulse reflection at the input of the antenna, the

impedance of the antenna should be matched to the impedance of

fL fH


34

transmission line over the entire operational band. For good

impedance match, the reflection coefficient () at the antenna feed

should be equal or less than -10 dB.

In the application of pulsed radar sensor, the amplitude of the

radiated pulses in a desired radiation direction should be maximized

and in undesired radiation directions should be minimized.

Therefore, a directional high gain antenna is preferred for such

application.

The size of the antenna is a great challenge in the UWB antenna

design. A suitable antenna needs to be small enough to be

compatible with a practical UWB radar sensor.


35

References

[1] R. Chatterjee, Antenna Theory and Practice, 2nd

Edition, New Age International

(P) Ltd., 2004.






House Inc., 2005.

[5] C. Foster, “Assessment of Ultra-Wideband (UWB) Technology,” IEEE

Aerospace and Electronic System Magazine, November 1990, pp. 45-49.

[6] Federal Communication Commission, Notice of Inquiry in the Matter of:

Revision of Part 15 of the Commission´s Rules Regarding Ultra-Wideband

Transmission Systems, Document # 02-48, April 2002.

Chapter 5 Si-Based Avalanche Pulse Generator

36

Chapter 5

Si-Based Avalanche Pulse Generator

A key problem in the design of UWB radar is the availability of a pulse

source with sufficient peak power and rise-time as short as possible. The

characteristic of the radar source determines the overall performance of

radar system.

Many approaches have been published for the design of a pulse

generator for radar sensor application. In addition to the classical GUNN

element [1]-[2], a transistor operating in the avalanche mode (in the

following simply referred to as avalanche transistor) has been proven as a

powerful pulse source. Pulse generation with an avalanche transistor is

based on the well-known avalanche phenomenon [3]-[4] occurring in a

bipolar junction transistor (BJT) when the transistor experiences

breakdown after application of a very high bias voltage. In order to achieve

ultra-short (sub-nanosecond) rise time of the transmitted pulse, special

semiconductor devices, for example tunnel diode [5] and step recovery

diode (SRD) [6] are used as pulse sharpener. In case of tunnel diodes, these

diodes offer the fastest transition time (in range of sub-picosecond) at very

low power levels (in range of mW) [7]. The SRDs are a compromise

alternative for these devices and offer ultra-short transition time (typically

in the range from tens of picosecond to a hundreds of picosecond) at

moderate power levels (ranging from hundreds of mW to tens of watt) [8].

Therefore, these diodes are very appropriate to be used in sharpener

network [7]. The SRD works as a charge controlled switch, which can

change from a low impedance to a high impedance state very rapidly (in


37

the order of picoseconds [9]). This ability of the SRD is used to sharpen the

slow waveform edges (in nanosecond range). The rise-time and maximum

reverse voltage of the SRD type (MA44769-287T) used in this work is

specified as 90 ps and 30V, respectively [10].

In this work, a low-cost high voltage picosecond radar pulser using silicon

(Si) BJT circuit with SRD based pulse sharpening circuit is designed and

fabricated.

In the following sections, the design details of the proposed pulse

generator are presented.

5.1 Description of Ultra-Short Pulse Generator

A block diagram of the proposed ultra-short pulse generator using Si-BJT

device with SRD is shown in Figure 5.1. It consists of the following

circuits:

- Avalanche transistor circuit

- Attenuator circuit

- Signal divider (balun transformer)

- Two equal SRD pulse shaping circuits

- Signal combiner (transmission line transformer)

The SRD is a two terminal p-i-n junction [9] as shown in Figure 5.2. The

breakdown voltage of SRD depends on the electrical field in the i-region

and the width of the i-region. As discussed in [11], the breakdown voltage is

directly proportional to the width of the i-region. Also, the transition time of

SRD depends on the width of the i-region and is directly proportional to the

i-region width as shown in [12]. Therefore, the SRDs with ultra-fast

transition time (tens of picosecond) have commonly a low breakdown

voltage (tens of volts) as shown in Table 5.1.


38

Figure 5.1 Block diagram of proposed pulse generator.

Figure 5.2 Two terminal p-i-n junction.

The output pulses of the avalanche transistor circuit drive the SRD

waveform edge sharpener. To match the amplitude of the avalanche pulse

to the power limitation of the connected SRD sharpener circuits, the

powerful avalanche pulse with a high voltage of about 180V is first

reduced and split into two pulses using attenuator circuit and signal divider

(balun transformer), respectively. The output pulses of the balun

transformer are two pulses with opposite polarities (balanced pulses).

These pulses are fed into equal SRD pulse shaping circuits. The purpose of

the SRDs are to sharpen the leading edge of the balun transformer output

pulses. The sharpened pulses are then processed in a pulse-forming circuit

to produce Gaussian-like pulses. Finally, the output pulses of the SRDs

sharpener circuits are combined using a transmission line transformer


39

(signal combiner). The output signal of the transmission line transformer

will stimulate the UWB transmitter antenna.

Table 5.1 Data Sheet of the SRDs Manufactured by M-Pulse Microwave [13].

SRD Breakdown voltage (Vr) Transition time (tr)

MP402 20 V 50 ps

MP403 30 V 70 ps

MP404 40 V 120 ps

MP406 60 V 240 ps

5.1.1 Avalanche Transistor Circuit

A simplified circuit diagram of the avalanche transistor circuit is shown

in Figure 5.3. A Si-BJT (see Figure 5.4), which is an equivalent type of that

reported in [14], is used as an ultrafast switch in the circuit. Fabrication of

the pulser is described in section 5.2.

Figure 5.3 Circuit diagram of avalanche transistor generator.


40

Figure 5.4 Photograph of used Si bipolar power transistor (chip).

Figure 5.5 shows schematically the output current-voltage characteristics

(IC–VCE) of a bipolar transistor for avalanche mode of operation. Without

trigger at the transistor base, the transistor is in off-state, i.e. non-

conducting. At low voltage value VCC, the collector current IC, as shown in

Figure 5.5, is very low and increases only slightly with increasing supply

voltage. When the voltage approaches BVCER (collector-emitter breakdown

voltage with given base-emitter resistor value RBE), then the electrical field

in the collector-emitter path becomes very high so that carrier

multiplication effect occurs; the current increases disproportionately.

Exceeding BVCER, the transistor breaks down and may reach point B

(dependent on the dynamic load line), which may not be a stable operation

point, thus, reaching point B`.

When the high current in B` exceeds the rated dissipation power of the

transistor (mostly temperature effect), second breakdown will happen,

which means that the transistor is destroyed [15]. As can be seen in Figure

5.5 the IC-VCE characteristic depends on the value of base resistor. The

lower the value of RBE, the higher the breakdown voltage BVCER.

C

E

C

C: Collector

E: Emitter

B: Base B


41

Figure 5.5 IC-VCE avalanche breakdown characteristic of a bipolar transistor [15]-[16].

Figure 5.6 illustrates the avalanche mode of operation. After triggering and

after the capacitor has been discharged, the transistor is switched into its

off-state. The capacitor is recharged again by the supply voltage VCC

through RCC and RL with a time constant

)()( CCLCCCCLCCCC RRRCRRC

until the quiescent point A is reached. In this point, the IC current is very

low, typically in the A-region. The electrical field in the collector-base

path is extremely high. When the base is triggered, current flow starts and

electrons are accelerated and gain high kinetic energy to generate multiple

electron-hole pairs, which indicates the avalanche breakdown.

The current increases rapidly. However, the capacitor discharges, the

voltage VC decreases, and thus, the electric field in the collector-base path;

the breakdown can no longer be maintained. Therefore, after avalanche

collapse, the current switches to a low off-state value. Then the charging of

the capacitor begins again, and having reached the quiescent point A, the

circuit is waiting for the next trigger signal.

(5.1)


42

Figure 5.6 IC-VCE characteristic of a bipolar transistor for avalanche mode of operation.

Because of the very fast effect of avalanche multiplication, the switching

time of the avalanche transistor lies on the order of 1 ns or less. The shape

and the amplitude of the output pulse Vout depend on the values of VCC, CCC

and RL.

The values of CCC,VCC and RL of the avalanche transistor circuit shown in

Figure 5.3 were varied to investigate the effect on the output pulse

parameters such as rise-time (tr), pulse width (FWHM) and amplitude.

Each measurement has been repeated 15 times. The average value of these

measurements has been recorded.

To evaluate the variation of the pulse characteristics due to the change of

the supply voltage (VCC), the circuit parameters shown in Figure 5.3 have

been chosen as CCC = 110 pF, RCC = 50 k, RB = 50, RBE = 50, RL =

50and CB = 2 nF. Based on the measurement, it is evaluated that the

avalanche breakdown action begins at VCC = 233V. Therefore, VCC was

changed between 233V and 330V.

The amplitude of the output pulse is shown in Figure 5.7 as a function of

VCC. It is evident from Figure 5.7 that the amplitude of the pulse increases

with increase of VCC.


43

The variation of pulse width (FWHM) and rise-time (tr) with respect to VCC

are presented in Figure 5.8 and 5.9, respectively. It can be seen that the

pulse width and rise-time decrease with the increase of VCC.

Figure 5.7 Amplitude of output pulse of avalanche transistor circuit versus supply

voltage VCC with CCC = 110 pF.

Figure 5.8 FWHM of output pulse of avalanche transistor circuit as a function of supply

voltage VCC with CCC = 110 pF.


44

Figure 5.9 Rise-time of output pulse of avalanche transistor circuit as a function of

supply voltage VCC with CCC = 110 pF.

Considering the pulse amplitude, pulse width (FWHM) and rise-time, it can

be summarized that the best selection of VCC, which provides fast, high

amplitude pulse would be 330V.

To evaluate the variation of the pulse characteristics due to the change of

the CCC, the circuit parameters have been chosen as VCC = 330V and RL =

20. The value of CCC has been changed in 50 pF step. The effect of

changing CCC value on the pulse parameters rise-time, FWHM and

amplitude of the output pulse are presented in Figure 5.10 and 5.11.

Figure 5.10 Amplitude of output pulse of avalanche transistor circuit versus CCC.


45

It is evident from these figures that the rise-time, pulse width and amplitude

of the pulse increase with the increase of CCC.

Figure 5.11 Rise-time and FWHM of output pulse of avalanche transistor circuit versus

CCC.

The evaluation of pulse amplitude variation due to the change of RL has

been performed by choosing the following element values: VCC = 330V and

CCC = 85 pF. RL was changed between 1 and 50. The amplitude of the

output pulse is presented in Figure 5.12 as function of load resistance RL. It

can be seen that the amplitude of the output pulse increases with the

increase of RL.

The peak power of the output pulse is presented in Figure 5.13 as function

of load resistance RL. It can be seen that the power of the output pulse

increases with the increase of RL up to RL = 5 and decreases with

increase of RL between 5 and 50. From Figure 5.13, it is illustrated that a

maximum pulse power of 650W can be obtained with an optimum load of

5. In this work, RL has been chosen as 50. The reason is that the

proposed pulse generator has been implemented in 50 environment to

match the input impedance of 50 of the connected UWB antenna. By

terminating the avalanche transistor circuit with 50, 302W peak power

can be obtained.


46

Figure 5.12 Amplitude of output pulse of avalanche transistor circuit as a function of

RL.

Figure 5.13 Peak power of output pulse of avalanche transistor circuit versus RL.

In order to complete the pulse generation analysis, it is important to

measure the output pulse parameters e.g. its amplitude, as a function of

pulse repetition frequency (PRF). The PRF of the avalanche transistor

circuit is limited by two factors [17]. First factor is the charging time of

CCC. It is very important to ensure that CCC is fully recharged before the

next trigger pulse comes. The second factor is the transistor temperature.

With increasing PRF, the average dissipation power increases


47

proportionally. This is also seen in Figure 5.14 for lower value of PRF.

However, above PRF ≈ 50 kHz, the influence of temperature seems to

decrease. This can be understood as follows.

The charge time constant is calculated as = CCC ∙ RCC = 110 pF x 50 k

= 5.5 s. Then the time to completely charge the capacitors is

approximately 5 = 27.5 s, corresponding to a frequency of 36.4 kHz

(shadowed region in Figure 5.14). Above PRF ≈ 36.4 kHz, the charge time

period is too short to completely charge the capacitor. This means that the

average dissipation power decreases, which is reflected by the smoother

decrease of Vout with PRF in Figure 5.14.

Figure 5.14 Amplitude of output pulse of avalanche transistor circuit as a function of

PRF.

The waveform of the output pulse of the avalanche transistor circuit

(Figure 5.3) is presented in Figure 5.15. This pulse has been measured by

choosing the following parameter values: VCC = 330 V, CCC = 100 pF, RCC

= 50 k, RBE = 50CB2 nF and RL = 50. The rise-time (tr), fall-time

(tf), pulse width (FWHM) and amplitude of the plotted pulse are 1.15 ns,

2.2 ns, 2.05 ns and -183V, respectively.


48

Figure 5.15 Output waveform of avalanche transistor circuit.

5.1.2 Attenuator Circuit

The amplitude matching of the avalanche pulse to the power limitation of

the connected SRD sharpener circuits has been performed in two stages.

The first stage is done using a matched attenuator circuit. A resistive

attenuator circuit with 9 dB attenuation and 50 characteristic impedance,

constructed in ‘‘pi’’ configuration as shown in Figure 5.16, has been

designed.

Figure 5.16 Circuit diagram of attenuator circuit.

The attenuator circuit parameters (R1, R2 and R3) which provide 9 dB

attenuation are calculated and resulted in R1 = 105, R2 = 61 and R3 =

105 (see Appendix A). The output pulse of the attenuator circuit applying

avalanche transistor pulse in Figure 5.15 is shown in Figure 5.17. From


49

Figure 5.15 and 5.17, it is realized that the amplitude of avalanche

transistor pulse was reduced from 183V to 65V using attenuator circuit.

Figure 5.17 Output waveform of the attenuator circuit.

5.1.3 Signal Divider

The second stage of amplitude matching of the avalanche pulse to the

power limitation of the connected SRD sharpener circuits has been done

using signal divider. In the simplest case, at sufficient low frequencies, a

balun transformer can be used as signal divider. Figure 5.18 shows the

circuit diagram of balun transformer.

Figure 5.18 Circuit diagram of balun transformer (signal divider).


50

The implemented transformer of type ADT1-1WT+ manufactured by Mini-

Circuits with a turn’s ratio 1:1 is specified for a frequency range from 0.4

MHz up to 800 MHz, defined for an input reflection coefficient S11 < -10

dB [18] (see S-parameter measurement of the transformer in Appendix B).

Coupling factor of the transformer is about 94%. The transformer

comprises a small ferrite core with a primary and secondary winding. By

applying the output pulse of the attenuator circuit (Figure 5.17) to the input

of balun transformer, two nearly identical pulses with opposite polarities

are obtained at the output of the transformer as shown in Figure 5.19. The

rise-time (tr), fall-time (tf), pulse width (FWHM) and amplitude of the

output pulses of the balun transformer are summarized in Table 5.2.

Table 5.2 Output Pulse Parameters of the Balun Transformer.

Positive pulse Negative pulse

Amplitude (V) 30 -31

FWHM (ns) 2.25 2.23

tr (ns) 1.25 1.28

tf (ns) 2.3 2.3


51

Figure 5.19 Output waveforms of the balun transformer (signal divider).

The outputs of the transformer are terminated by 50 microstrip

transmission lines, which are connected to the input of the SRD pulse

shaping circuits.

5.1.4 SRD Pulse Sharpener Circuit

The next stage of the pulse generator is the pulse shaping stage using step

recovery diodes (SRD). SRDs are used primarily to generate very fast rise-

time (in the picosecond region) pulses in frequency comb generators,

harmonic frequency multipliers and samplers [9]. In most applications, the

SRD works as a charge controlled switch. During the forward bias

condition, a large amount of charge is injected into the diode making the

impedance low. In case of reverse biasing, the device continuous as low

impedance until all the charge is totally removed, at the point where the

diode rapidly switches from the low to a high impedance [19]. The ability

of the SRD to store charge and change its impedance level rapidly is used

to sharpen the slow waveform edges (in the nanosecond region). The


52

transition time (tr) of SRD depends on the device structure and the external

circuit that is connected to the diode [20].

To sharpen the slow edges of the balun transformer (signal divider) output

pulses, SRD sharpener circuit consisting of two equal SRD pulse shaping

circuits, has been used. As shown in Figure 5.20, each circuit comprises

SRD, Schottky diode (SD), delay line, LC bias network and a coupling

capacitor C [8].

Figure 5.20 Circuit diagram of SRD sharpener circuit.

The SRDs are used to sharpen the fall time of the negative pulse of the

balun output and the rise time of the positive pulse of the balun output. The

sharpened pulses are then processed in pulse-forming circuits, which

consist of Schottky diodes and delay lines, to produce Gaussian-like pulses.

The MA44769-287T SRD in the SOT-23 package [10], shown in Figure

5.21(a), and BAT 62-03W Schottky diode in the SOD-323 package [21],

shown in Figure 5.21(b), are used in the SRD pulse shaping circuit.


53

(a) (b)

Figure 5.21 SRD in SOT-23 package (a), Schottky diode in SOD-323 package (b).

The working principle of the SRD sharpener circuit is considered in Figure

5.22. When no driving pulse from the avalanche transistor is present

(steady-state), the SRD is forward biased by a constant bias current (90

mA). The Schottky diode (SD) is reverse biased and does not influence the

circuit. After triggering the avalanche circuit, a negative voltage pulse is

generated. Then, the balun transformer provides two pulses of (ideally)

equal amplitudes with opposite polarities. The positive pulse (Figure 5.22)

will pass through the coupling capacitor and the delay line to the SRD.

Once the SRD is turned off, fast rising edge voltage steps propagate in both

directions away from the SRD. The first step [V2 in Figure 5.22 (blue line)]

propagates unchanged through the coupling capacitor to the output which is

terminated by a 50 microstrip transmission line, whereas the second one

propagates along the delay line back to the input [V3 in Figure 5.22 (red

line)]. However, the shunt Schottky diode is now opened by the positive

driving pulse (balun transformer) sufficiently to short-circuit the

transmission line. Therefore, the incoming step waveform is reflected back

with inverted polarity. It propagates to the output again, where it

contributes to the output waveform as shown in Figure 5.23. By

superposition of the delayed inverted step with the unchanged forward

waveform, a Gaussian pulse is generated at the output (V4 in Figure 5.22).


54

Figure 5.22 Signal flow diagram of SRD pulse sharpener.

Figure 5.23 Circuit diagram of the SRD pulser at circuit points V1, V2, V3 and V4.


55

The output pulse of the SRD sharpener circuit shown in Figure 5.20

depends only on the input pulses and delay lines between the step recovery

diodes and Schottky diodes. The SRD pulse sharpener circuit shown in

Figure 5.23 has been verified by varying the values of input pulse

amplitude and delay line length to investigate the effect on the output pulse

characteristic such as rise-time (tr), pulse width (FWHM) and amplitude.

Each measurement has been repeated 15 times and the average of these

measurements has been recorded. For all measurements, the rise-time (tr) of

the input pulse (avalanche pulse) is 1.15 ns. Its pulse repetition frequency

(PRF) was chosen as 1 kHz. The change in tr of the input pulse and its PRF

has no significant effect on the output pulse characteristic.

The length of the delay line is firstly set to 7 mm and the input pulse

amplitude is varied between 10 to 30V. Figure 5.24 shows the output pulse

amplitude as a function of input pulse amplitude for the SRD pulse

sharpener circuit. As can be observed, the amplitude of the output pulse

increases with the increase of input pulse amplitude. It can be seen that for

high input pulse of about 30V, output pulse with an amplitude of 24V is

obtained.

The rise-time (tr) and pulse width (FWHM) of the output pulse are

presented in Figure 5.25 as a function of the input pulse amplitude. It is

realized that the rise-time (tr) increases from 98 ps to 108 ps while the pulse

width (FWHM) increases from 128 ps to 155 ps as the amplitude of the

input pulse increases from 10 to 30V.

The increase of the output pulse rise-time and pulse width with the

increase of input pulse amplitude is due to the increase in the amount of

injected charges in SRD under forward bias conditions as the amplitude of

the input pulse is increased [22]. This leads to the fact that the SRD needs

longer time to release all stored charges under reverse bias conditions.


56

Figure 5.24 Amplitude of output pulse of SRD sharpener circuit as a function of input

pulse amplitude.

Figure 5.25 FWHM and tr of output pulse of SRD sharpener circuit as a function of

input pulse amplitude.

The delay line between step recovery diode and Schottky diode shown in

Figure 5.23 was implemented as a section of microstrip line. To evaluate

the variation of the output pulse characteristics due to the change of the

delay line length, the amplitude of the input pulse is set to 30V. Table 5.3

shows the amplitude and pulse width (FWHM) of the measured output

pulses of the SRD sharpener circuit for different lengths (l) of the delay


57

line. It can be seen that the maximum observed amplitude of the generated

pulses is 28V with FWHM of about 185 ps for l = 10 mm. Minimum

amplitude of the generated pulse is 13V with FWHM of 130 ps for l = 5

mm.

Table 5.3 Pulse Parameter for Different Lengths of Delay Line.

Line length l=10 mm l=7 mm l=5 mm

Amplitude (V) 28 24 13

FWHM (ps) 185 155 130

The line length (l) was chosen as 7 mm, which is a good compromise

between high amplitude and narrow pulse width.

By applying the output pulses of balun transformer (Figure 5.19) to the

SRD sharpener circuits (Figure 5.20), two Gaussian-shaped pulses are

obtained at the output of the SRD sharpener circuits as shown in Figure

5.26. The rise-time (tr), fall-time (tf), pulse width (FWHM) and amplitude

of the two pulses are tabulated in Table 5.4.


58

Figure 5.26 Balanced output signals of the SRD sharpener circuits.

Table 5.4 Output Pulse Parameters of the SRDs Sharpener Circuits.

Positive pulse Negative pulse

Amplitude (V) 25 -26

FWHM (ns) 0.135 0.14

tr (ns) 0.096 0.102

tf (ns) 0.1 0.109

5.1.5 Signal Combiner

Figure 5.27 shows the circuit diagram of the transmission line transformer

[18]. This type of the transformer exhibits large frequency bandwidth up to

4 GHz [18]. In this work, this type of the transformer has been used as

signal combiner, which combines the output pulses of SRDs sharpener

circuits. The transmission line transformer of type TC1-1-43A+


59

manufactured by Mini-Circuits with a turn’s ratio 1:1 is specified for a

frequency range from 0.65 GHz up to 3.7 GHz, which is defined for an

input reflection coefficient S11 < -10 dB (see S-parameter measurement of

the transformer in Appendix B). The coupling factor of the transformer is

about 75%. The transformer comprises a small ferrite core with a primary

and secondary winding. By applying both output pulses of the SRDs

sharpener circuits (Figure 5.26), the transformer provides one pulse with an

amplitude of 38V, a rise-time of 110 ps and a pulse width (FWHM) of 145

ps as shown in Figure 5.28.

Figure 5.27 Circuit diagram of the transmission line transformer (signal combiner).

Figure 5.28 Output waveforms of the signal combiner.


60

5.2 Fabrication and Measurement Results

The complete pulse generator is shown in Figure 5.29(a) and (b). The pulse

generator circuit was realized in microstrip technique using RO4003C

substrate with dielectric permittivity (εr) of 3.38, dielectric substrate

thickness (h) of 0.81 mm, and conductivity thickness (t) of 18 m. Figure

5.29(c) shows the fabricated ultra-short pulse generator circuit with a

physical size of 86 mm x 37 mm.

A 50 microstrip transmission line was used to realize the circuit. To avoid

any parasitic effect, the avalanche transistor was implemented as a chip as

shown in Figure 5.4. A Hameg 8035 pulse generator delivered the trigger

pulses with amplitude of 5V, pulse width of 50 ns, and repetition frequency

of 1 kHz to trigger the avalanche transistor. The waveforms of the output

pulses have been measured with a 12 GHz sampling oscilloscope

DSO81204B type with 50 input impedance manufactured by Agilent.

The discharge current of the avalanche transistor circuit flows from the

capacitor CCC through the avalanche transistor and the resistor RL as shown

in Figure 5.30. It is very important to keep this discharge path as short as

possible to reduce potential parasitic inductive effects in the discharge

circuit; otherwise these may have strong effect on the shape, width, rise-

time and fall-time of the output pulse.


61

(c)

Figure 5.29 Complete ultra-short pulse generator. (a) avalanche transistor circuit and

attenuator circuit, (b) signal divider, SRD sharpener circuit and signal combiner, and (c)

photograph of fabricated circuit.

1

2

3

5


62

Figure 5.30 Short discharge path of the avalanche transistor circuit.

Two DC power supply sources were used to bias the step recovery

diodes. The bias current of each source was set to 90 mA during the

measurement.

The waveform of the output pulse of the fabricated pulse generator is

shown in Figure 5.31. This pulse has been measured by choosing the

following parameter values: VCC = 330 V, CCC = 100 pF, RCC = 50 k, RBE

= 50CB2 nF, R1 = 101R2 = 61R2 = 101 l (length of delay

line) = 7 mm and RL = 50. The rise-time (tr), fall-time (tf), pulse width

(FWHM) and amplitude of this pulse are 112 ps, 150 ps, 155 ps and 34.5V,

respectively.

Figure 5.31 Output waveform of the realized ultra-short pulse generator.

C: Collector

E: Emitter

B: Base

B

E

C C

RL


63

The normalized power spectrum of the pulse is measured using a sampling

oscilloscope of type DSO81204B and is presented in Figure 5.32. It is seen

that regarding a signal limit of -20 dB, which includes 90% of the total

pulse energy [23], the pulse exhibits a bandwidth of about 3.5 GHz.

Figure 5.32 Normalized power spectrum of the pulse in Figure 5.31.

A summary of the specifications of the fabricated ultra-short pulse

generator is shown in Table 5.5.


64

Table 5.5 Pulse Generator Parameters.

Parameter Value

Trigger pulse

PRF

Pulse width

Pulse amplitude

1 kHz

50 ns

5 V

Avalanche transistor circuit

Supply voltage Vcc

Output pulse tr

tf

FWHM

Pulse amplitude

330 V

1.15 ns

2.20 ns

2.05 ns

-183 V

Attenuator

Output pulse tr

tf

FWHM

Pulse amplitude

1.20 ns

2.20 ns

2.08 ns

-65 V

Signal divider (transformer)

Output pulse tr (+)

tr (-)

tf (+)

tr (-)

FWHM (+)

FWHM (-)

Pulse amplitude (+)

Pulse amplitude (-)

1.25 ns

1.28 ns

2.30 ns

2.30 ns

2.25 ns

2.23 ns

30 V

-31 V

SRD sharpener circuit

Output pulse tr (+)

tr (-)

tf (+)

tr (-)

FWHM (+)

FWHM (-)

Pulse amplitude (+)

Pulse amplitude (-)

96 ps

102 ps

100 ps

109 ps

135 ps

140 ps

23 V

-24 V

Signal combiner (transformer)

Output pulse tr

tf

FWHM

Pulse peak

112 ps

150 ps

155 ps

34.5 V


65

References

[1] G. Kompa, “Sensoren im MHI-Bereich—Entwicklungsstand und Trends,” VDI-

Z, vol. 130, 1988, pp. 42–54.

[2] Y. Tao, J. Nin, and G. Deliste, “Ka-Band Solid-State Pulsed Gunn Oscillator and

Power Combiner,” International Journal of Infrared and Millimeter Waves, vol.

16, 1995, pp. 1769-1772.

[3] R. J. Baker, “High Voltage Pulse Generation Using Current Mode Second

Breakdown in a Bipolar Junction Transistor,” Review of Scientific Instruments,

vol. 62, April 1991, pp. 1031–1036.

[4] A. Kilpelä, Pulsed Time-of-Flight Laser Range Finder Techniques for Fast,

High Precision Measurement Applications, Doctoral Thesis, University of Oulu,

Finland, January 2004.

[5] E. Miller, Time-Domain Measurements in Electromagnetics, Springer, New

York, 1986.

[6] A. Ruengwaree, R. Yowuno, and G. Kompa, “Ultra-Fast Pulse Transmitter for

UWB Microwave Radar, ” European Microwave Conference Proceedings,

September 2006, pp. 1833-1836.

[7] P. Protiva, J. Mrkvica, and J. Machac, “A Compact Step Recovery Diode

Subnanosecond Pulse Generator,” Microwave and Optical Technology Letters,

February 2010, pp. 438-440.

[8] A. Ameri, G. Kompa, and A. Bangert, “Balanced Pulse Generator for UWB

Radar Sensor,” European Microwave Conference Proceedings, October 2011,

pp. 198-201.

[9] Hewlett Packard, Pulse and Waveform Generation with Step Recovery Diodes,

Application Note 918, October 1986.

[10] M/A-COM Technology Solutions http://www.macomtech.com/.

http://ieeexplore.ieee.org/xpl/RecentIssue.jsp?punumber=4915264

http://ieeexplore.ieee.org/xpl/mostRecentIssue.jsp?punumber=4638692

http://www.macomtech.com/


66

[11] J. Agrawal, Power Electronic System: Theory and Design, Prentice Hall, US

2001.

[12] J. Liu, Photonic Devices, Cambridge, 2005.

[13] MP40 Step Recovery Diodes, M-Pulse Microwave, http://www. mpulsemw.com/

SRD_Diode.htm.

[14] G. Kompa, “Recent Advances in Laser Radar Technology with New Facilities for

Quality Control,” Proc. of Conference on Modern Design, Manufacturing and

Measurement (MODMM), May 6 - 8, 1993, Tsinghua University (Beijing, China),

Paper D-3, pp. 242 - 247.

[15] A. Kilpelä, and J. Kostamovaara, “Laser Pulser for a Time-of-Flight Laser

Radar,” Review of Scientific Instruments, June 1997, pp. 2253-2258.

[16] T. Buchegger, G. Ossberger, A. Reisenzahn, A. Stelzer, and A. Springer, “Pulse

Delay Techniques for PPM Impulse Radio Transmitters,” IEEE Conference on

Ultra Wideband Systems and Technologies, November 2003, pp. 37–41.

[17] E. Fulkerson, D. Norman, and R. Booth, ‘‘Driving Pockels Cells Using

Avalanche Transistor Pulsers,” IEEE International Pulsed Power Conference,

July 1997, 1341-1346.

[18] RF Transformers, Mini-Circuits, http://www.minicircuits.com/products/ transfor

mers_sm_a.shtml.

[19] Z. Jianming, G. Xiaowei, and F. Yuanchun, “A New CAD Model of Step

Recovery Diode and Generation of UWB Signals,” IEICE Electronics Express,

vol. 3, December 2006, pp. 534-539.

[20] X. Xu, Characterization and Modeling of SRD Diodes for the Computer Aided

Design of a Generator of Ultrashort Pulses, Master Thesis, University of Kassel,

November 1999.

[21] BAT 62-03W Schottky Diode, Infineon, http://www.infineon.com/cms/

en/product/findProductTypeByName.html?q=BAT62.







http://www.jstage.jst.go.jp/search/?typej=on&typep=on&typer=on&d3=au&dp3=Zhou+Jianming&ca=999999&alang=all&rev=all&pl=20&search=%8C%9F%8D%F5%8E%C0%8Ds

http://www.jstage.jst.go.jp/search/?typej=on&typep=on&typer=on&d3=au&dp3=Gao+Xiaowei&ca=999999&alang=all&rev=all&pl=20&search=%8C%9F%8D%F5%8E%C0%8Ds

http://www.jstage.jst.go.jp/search/?typej=on&typep=on&typer=on&d3=au&dp3=Fei+Yuanchun&ca=999999&alang=all&rev=all&pl=20&search=%8C%9F%8D%F5%8E%C0%8Ds

http://www.infineon.com/cms/%20en/product/findProductTypeByName.html?q=BAT62

http://www.infineon.com/cms/%20en/product/findProductTypeByName.html?q=BAT62


67



Chapter 6 GaAs-Based Avalanche Pulse Generator

68

Chapter 6

GaAs-Based Avalanche Pulse Generator

In addition to the pulse generation based on bipolar junction transistor

(BJT), a study of generation of ultra-short high power pulse based on the

avalanche breakdown voltage of modern GaAs-MESFET is presented in

this chapter. In this study, GaAs-MESFET device specified with low

breakdown voltage (a few volts) has been used to generate pulses with high

voltage amplitude (tens of volts).

In the following section, the investigation details of the new study are

presented.

6.1 Breakdown Measurement

Packaged GaAs-MESFET (MGF-1601B) device with gate-width wG = 740

m (4x185 m), gate-length Lg = 0.8 m and gate-drain spacing Lgd = 2.2

m has been used. The device is manufactured by Mitsubishi and shown in

Figure 6.1. The study is performed in two parts. In the first part, two-

terminal and three-terminal I-V measurements are carried out to measure

the gate-drain breakdown voltage and drain-source breakdown voltage

using DC and pulsed I-V measurements. In the second part, the procedure

of generation of ultra-short high amplitude pulses based on the avalanche

breakdown is described.


69

Figure 6.1 Photograph of the 0.8-m GaAs-MESFET device under test manufactured

by Mitsubishi (chip inside the package).

6.1.1 Two-Terminal Measurements

In order to characterize the breakdown behavior of the GaAs-MESFET

device, a two-terminal reverse diode technique [1] is carried out to measure

gate-to-drain and gate-to-source breakdown voltages. The two-terminal DC

and pulsed I-V measurements are performed using IVCAD-3 Pulsed

system [2]. The I-V measurement of gate-drain breakdown voltage is

carried out by biasing the gate, grounding the drain and keeping the source

floating as shown in Figure 6.2(a). The measurement of gate-source

breakdown voltage is performed by biasing the gate, grounding the source

and keeping the drain floating as shown in Figure 6.2(b). Finally, the

measurement of breakdown voltage can be performed by biasing the gate

and grounding both drain and source as shown in Figure 6.2(c). Regarding

pulsed I-V measurement, the width of the pulse and duty cycle are set to

200 ns and 0.02%, respectively. The reason of choosing short pulse width

with low duty cycle is to minimize the self-heating effect [3]. The

amplitude of the pulse is varied from 11 to 16V at 0.1V step intervals.

A discussed in [4]-[5], the impact ionization (avalanche breakdown),

tunneling and thermionic field emission (TFE) or some combination

thereof are responsible for breakdown. In the most devices, one can easily

determine breakdown mechanism through temperature dependent

measurements [4].


70

(a)

(b)

(c)

Figure 6.2 Schematic circuit diagram of the two-terminal breakdown measurement. (a)

gate-drain breakdown, (b) gate-source breakdown, and (c) combined gate-drain and gate

source breakdown.

As reported in the literature [1], [4], [5], the breakdown voltage due to

impact ionization has a positive temperature coefficient. On the contrary,

the breakdown voltage due to tunneling and TFE mechanism has a zero and

negative temperature coefficient, respectively [5]-[6]. In this work, the two-


71

terminal breakdown voltage measurements have been carried out at

different temperatures to analyze the breakdown mechanism.

The two-terminal DC and pulsed I-V measurements are shown in Figure

6.3 to Figure 6.6. From these figures, it can be seen that the drain-gate and

source-gate breakdown voltages (BVDG, BVSG) determined at drain-gate

current (IDG = 1 mA/mm) and source-gate current (ISG = 1 mA/mm),

respectively, increase with increase of temperature. Therefore, it is

supposed that impact ionization is the dominant mechanism for breakdown.

It can also be seen that for low VDG and VSG, IDG and ISG increase with

increase of temperature. This indicates that tunneling and TFE should be

the origin of IDG and ISG increase in the lower bias range (below the

crossing point of the curves) [7]. On the other hand, for high VDG and VSG,

IDG and ISG decrease with increase of temperature, which is believed that

impact ionization is the dominant IDG and ISG source [7].

Figure 6.7 shows the breakdown voltage BVDG and BVSG determined from

pulsed I-V measurement at different defined values of IDG and ISG as a

function of temperature. It can be seen that for high current (above 0.6

mA/mm for IDG and above 0.77 mA/mm for ISG), the breakdown voltage

has a positive temperature coefficient. This means that impact ionization is

responsible for breakdown. For low current value (below 0.6 mA/mm for

IDG and below 0.77 mA/mm for ISG), the breakdown voltage has a negative

temperature coefficient. This indicates that the tunneling and TFE is the

dominant mechanism for breakdown. For IDG = 0.6 mA/mm and ISG = 0.77

mA/mm, the temperature dependence of the breakdown voltage is

negligible. This indicates that only pure tunneling is the dominant

breakdown mechanism [1].


72

Figure 6.3 Two-terminal pulsed breakdown measurement (drain-gate) at different

temperatures.

Figure 6.4 Two-terminal DC breakdown measurement (drain-gate) at different

temperatures.


73

Figure 6.5 Two-terminal pulsed breakdown (source-gate) measurement at different

temperatures.

Figure 6.6 Two-terminal DC breakdown measurement (source-gate) at different

temperatures.


74

Figure 6.7 Breakdown voltage as a function of temperature obtained from two-terminal

pulsed I-V measurements.

In the two-terminal breakdown measurement, a reverse bias is applied to

the Schottky gate. Electrons are injected from the gate through the potential

barrier via TFE into the high-field drain–gate region (in case of two-

terminal gate-drain) or into high-field source-gate region (in case of two-

terminal gate-source) [7]. The high-electric field in this region (gate-drain

region or gate-source region) heats up the injected electrons to energies

where they start impact ionizing. The generated electron-hole pairs are

seperated by the electric field [8]. The electrons flow to the drain (in case

of two-terminal gate-drain) or to the source (in case of two-terminal gate-

source) [9], while the holes flow to the gate. Both electron and hole

currents constitute the negative gate current seen in the measured data.

In Figure 6.8, drain-gate breakdown voltages based on DC and pulsed

measurements are compared. Evaluation has been done with drain-gate

current value IDG = 1mA/mm. From this figure, it is realized that the

breakdown voltage obtained from DC I-V measurement is higher than the

breakdown voltage obtained from pulsed I-V measurement. It is observed

that with increase of measured current, the drop between pulsed and DC I-


75

V characteristic becomes lower as shown in Figure 6.9. It is supposed that

the increasing voltage drop is due to the increasing resistance value of the

series resistor of gate-drain and gate-source diodes with increasing

temperature.

Figure 6.8 Comparison of DC and pulsed breakdown voltages as a function of

temperature.

From Figure 6.9, it can be seen that BVDG (determined at IDG = 1 mA/mm)

is higher than BVSG (determined at ISG = 1 mA/mm). This may be due to

length difference between gate-drain space and gate-source space (gate-

source space of the device was unknown) as reported in [10], the

breakdown voltages (BVDG and BVSG) are directly proportional to the space

length Ldg and Lsg, respectively.


76

Figure 6.9 Two-terminal breakdown measurement (drain-gate and source-gate) at T =

300K.

For completeness, also the forward characteristic of the transistor has been

measured. As will be shown, this will help to describe the new method of

pulse generation in section 6.1.4. Figure 6.10 shows the measured pulsed I-

V characteristic of the gate-source diode. The threshold voltage Vth is

found by the crossing of the straight line approximating the gate-forward

conduction branch with the VGS-axis. It is found to be 0.67V.


77

(a)

(b)

Figure 6.10 Pulsed I-V measurement of gate-source. (a) forward and reverse regions,

(b) forward region.


78

6.1.2 Three-Terminal Measurements

Three-terminal (source grounded) measurement technique is used to obtain

drain-source breakdown voltage (BVDS) and BVDG [11].

A classical approach for three-terminal breakdown voltage was performed

by Bahl et al. [11], which was followed in this thesis to get an overview of

the breakdown properties of the GaAs-MESFET (MGF-1601B) device.

Following this technique a fixed current is injected into the drain of the

device, while sweeping gate-source voltage (VGS) from above threshold

voltage to below it. The drain-gate voltage VDG when ID = - IG (i.e., source-

current Is = 0) is termed the BVDG. The drain-source breakdown voltage

BVDS is defined as the maximum VDS attained. The drain-current injection

technique has been performed by DC measurements. The current injection

measurements are carried out using IVCAD-3 system [2]. The injected

drain-current is set to 1 mA/mm. Figure 6.11 shows the measurement

results of drain-current injection technique, which are very comparable

with the published results in [11].

Figure 6.11 Drain-current injection technique DC measurement of GaAs-MESFET

(MGF-1601B) device with injected drain-current ID = 1 mA/mm.


79

From Figure 6.11, it is realized that the breakdown voltage BVDG and BVDS

occur at VDG = 15.67V and VDS = 11.9V, respectively. By comparison of

the BVDG values obtained from two-terminal DC measurement (see Figure

6.9) and three-terminal measurement technique (Figure 6.11), both values

are approximately same.

To determine which mechanism dominates three-terminal off-state

breakdown, the three-terminal pulsed I-V measurements are carried out at

different temperatures as discussed in the next section.

6.1.3 Experimental Avalanche Breakdown Analysis

In the literature, the measurement techniques have been proposed to

characterize the breakdown behavior of transistor, without the risk of

damage [10]-[11].

However, as will be discussed in section 6.14, the impact ionization with

avalanche multiplication can be used to generate very short pulses with

high peak power. Such ultra-short powerful pulses are very useful for near-

field UWB radar systems. Therefore, in this section, the avalanche effect in

the GaAs-MESFET (MGF-1601B) device is studied in more details. I.e. the

breakdown measurements have to be performed very carefully to avoid any

device failure. This, in particular can be taken into account by the testing

instrument, which can control the permissible line current and which is

deduced from the device rating value. In the data sheet of the MGF-1601B

device, the rating drain current is specified as 250 mA. Therefore, in the

following experiments, the line value of the maximum drain current has

been fixed to 80 mA.

The (three-terminal) breakdown measurements have been performed by

biasing the gate at a fixed voltage and sweeping the drain-source voltage

VDS. The measurement has been done with the channel-off (off-state

breakdown). Therefore, the value of gate-source voltage VGS has been

chosen to be equal to -4V (deep pinch-off), whereas, the pinch-off voltage

(VP) of the transistor is about -2.4V. The three-terminal pulsed I-V

measurements are carried out using IVCAD-3 system. The width of the


80

pulse and duty cycle are set to 200 ns and 0.02%, respectively Figure 6.12

shows the schematic circuit diagram for the measurement.

Figure 6.12 Schematic circuit diagram for pulsed breakdown measurement (VGG = -4V,

RDD = 525).

The pulsed breakdown measurements are shown in Figure 6.13.

Figure 6.13 Pulsed breakdown measurement at VGG = -4V for different temperatures

(RDD = 525,VDD ≤ 40V).

From Figure 6.13, it can be observed that drain-source breakdown voltage

BVDS defined at ID = 1 mA/mm increases with increase of temperature.

This means that the main breakdown mechanism in this device is impact


81

ionization as was already found with two-terminal measurements. It can

also be seen that for low VDS (up to 9.7V), the drain current ID increases

with increase of temperature. This indicates that TFE is the origin of ID in

this bias range [7]. On the other hand, for high VDS, ID decreases with

increase of temperature, consistently with the hypothesis that impact

ionization is the dominant ID source [7].

Figure 6.14 shows the drain-source breakdown voltage BVDS determined

from pulsed I-V measurement at different fixed values of ID as a function

of temperature. It can be seen that for high current (above 0.7 mA/mm), the

breakdown voltage has a positive temperature coefficient. This means that

impact ionization is responsible for breakdown. For low current (below 0.7

mA/mm), the breakdown voltage has a negative temperature coefficient.

This indicates that pure tunneling and TFE is the dominant mechanism for

breakdown. For ID = 0.7 mA/mm, the breakdown voltage has negligible

temperature dependence. This indicates that pure tunneling is the dominant

breakdown mechanism [7].

Figure 6.14 Drain-source breakdown voltage as function of temperature.

To analyze the breakdown mechanism, the currents through the source,

drain and gate terminals are simultaneously monitored during the off-state

pulsed I–V measurement. The result is shown in Figure 6.15.


82

Figure 6.15 Three-terminal pulsed I-V measurement for T = 300K (VGG = -4V, RDD =

525).

According to temperature-dependence measurement discussed in Figure

6.13, the increase of IG and ID shown in Figure 6.15 for VDS < 9.7V is due

to the TFE mechanism.

For VDS > 9.7V, both ID and IG increase with increase of VDS. As discussed

with temperature-dependence measurement shown in Figure 6.13, the

impact ionization is the origin of ID and IG increase in the higher bias range

[7].

From Figure 6.15, it is realized that ID ≈ -IG. This leads to the conclusion

that breakdown occurs between drain and gate.

Electrons are injected from the gate through the potential barrier via field

emission induced tunneling (TFE) into the high-field drain–gate region [7].

The high-electric field in this region heats up the injected electrons to

energies where they start impact ionizing. The generated electron-hole

pairs are seperated by the electric field [8]. The electrons flow to the drain

[9], so that they contribute to an increase in the drain current [12]. The

holes can follow a number of paths [13]: They can flow to the gate terminal

and add to the gate current; they can escape into the substrate, or they can


83

flow to the source. Due to the proximity of the gate to the region of impact

ionization most of the holes are swept towards the gate [14]. So they

contribute to an increase in the gate current

From Figure 6.15, it can also be seen that there is a small current flow to

the source. This is due to some of the holes generated by impact ionization

and entering the source.

In a further experiment, it was tried to suppress the gate breakdown current.

This was established by introducing a high-value resistor (RGG) in the gate-

source bias circuit (gate current control). RGG was chosen as 1k. The

modified circuit for breakdown measurement is shown in Figure 6.16.

Figure 6.16 Schematic circuit diagram for pulsed breakdown measurement with gate

current control (RGG = 1 k, RDD = 525, VGG = -4, VDD ≤ 60V).

The results of the pulsed I-V measurement of the new circuit is shown in

Figure 6.17. It can be seen that now ID ≈ -IS, i.e. that due to the high

impedance of the gate bias resistor RGG, the gate current could efficiently

be stabilized at a low level. The gate-drain breakdown mechanism has

change to drain-source breakdown mechanism.

From Figure 6.17, it can be seen that a snap-back effect [13] occurs at VDS

= 12.7V.


84

Figure 6.17 Measured pulsed breakdown characterization with snap-back effect for T =

300K (VGG = -4V, RGG = 1 k, RDD = 525).

The effect of changing RGG on the pulsed breakdown measurement is

shown in Figure 6.18.

Figure 6.18 Pulsed breakdown measurement at T = 300K for different values of RGG.


85

From Figure 6.18, it can be seen that the drain-source breakdown voltage

(BVDS) decreases with increase of RGG. This can be explained based on the

Figures 6.19 and 6.20.

We consider just the point that diode D2 breaks down and assume that the

diode current through D2 is still zero. At this moment the pulse voltage

VDD drops across diode D1. As is indicated in Figure 6.20, diode D1

switches from the quiescent voltage VQ to the voltage point VD2, th (VD2, th is

defined as the voltage VD1, when D2 breaks through). Since we have

assumed VD2 = 0 and ID = 0 at breakdown of D2, VDS, th ≈ VDD, th ≈ VD2, th –

VQ, whereby VQ is dependent on RGG. For the bias points Q and Q` we can

write

GGGGGQ RIVV

)̀( GGGGGGGQ IIRIVV

Thus, IVQ`I < IVQI for RGG` > RGG. Therefore, an increase of RGG leads to a

reduced VDD, th ≈ VDS, th as shown in Figure 6.20.

When the applied VDD is higher than VDD, th, the drain current pulse (ID) will

flow to the source. This leads to an increase of ID and reduction of VDS and,

therefore, to a snap-back effect as shown in Figure 6.18.

Figure 6.19 Schematic circuit diagram for pulsed breakdown measurement, showing

gate-drain and gate-source diodes in anti-series (RGG = 1 k, RDD = 525 and VGG = -

4V).

(6.2)

(6.1)


86

Figure 6.20 I-V characteristic of gate-source diode D1.

Figure 6.21 shows the temperature-dependent I-V measurement of the

circuit shown in Figure 6.16. It can be seen that for high current (above 0.8

mA/mm), the drain-source breakdown voltage BVDS increases with

increase of temperature. Therefore, it is supposed that impact ionization is

the dominant mechanism for breakdown as discussed with two-terminal

measurements (section 6.1.1). For low current (below 0.8 mA/mm), the

BVDS decreases with increase of temperature. This indicates that tunneling

and TFE should be the responsible mechanism of breakdown voltage.


87

Figure 6.21 Pulsed breakdown measurement for different temperatures (VGG = -4V,

RGG = 1 k, RDD = 525).

6.1.4 Ultra-Short High-Power Pulse Generation

The procedure of pulse generation based on the avalanche breakdown of a

GaAs-MESFET device is presented in this section. Based on pulse

generation, which is performed using a bipolar junction transistor discussed

in Chapter 5, a new circuit with a GaAs-MESFET device (see Figure 6.22)

has been tested to generate ultra-short pulses.

Figure 6.22 Pulse generator circuit schematic based on GaAs-MESFET.

A photograph of the pulse generator shown in Figure 6.22 is presented in

Figure 6.23.


88

Figure 6.23 Photograph of pulse generator shown in Figure 6.22.

Regarding the circuit in Figure 6.22, the transistor is initially in pinch-off

state (because of high negative voltage applied on the gate-terminal VGG = -

4V). Figure 6.24 shows the measured DC I-V breakdown characteristics

and load line (RDD = 525). As long as the bias point (intersection point of

load line and DC I-V characteristics) is located below the snap-back point,

the circuit will remain in a stable condition. With increasing bias voltage

VDD, the load line is shifted upwards, drain-source voltage and drain

current increases, and the voltage across the discharging capacitor CL

follows VDD. With a positive trigger pulse at the gate terminal causes the

transistor to switch on and the CL capacitor discharges with a current pulse

through the load resistor RL. This leads to a negative voltage pulse across

RL. The waveform of the output pulse of the new circuit (Figure 6.22) is

presented in Figure 6.25. This waveform has been measured using 350

MHz sampling oscilloscope 54641A type manufactured by Agilent with

1M input impedance.

1

4

2

3


89

Figure 6.24 DC I-V characteristic of transistor in breakdown region.

The rise-time (tr), fall-time (tf), pulse width (FWHM) and amplitude

peak of the plotted pulse are 0.9 ns, 2.11 ns, 1.6 ns and -0.52V,

respectively.

Figure 6.25 Waveforms of output pulse of the circuit in Figure 6.22 (VDD = 12.4V).


90

As discussed in the three-terminal measurement (section 6.1.3), for high

VDS > 12.7V, the transistor will trigger by itself. This leads to a limit of the

output pulse amplitude as shown in Figure 6.26.

Figure 6.26 Waveform of output pulses of the circuit in Figure 6.22 for different values

of VDD (Vsupply).

To increase the amplitude of the output pulse, separate charging of CL is

needed. Therefore, we proposed a new circuit as shown in Figure 6.27.

From Figure 6.27, it can be seen that the new circuit consists of two parts.

The first part is a charging circuit which consists of charge capacitor CL =

5.7 pF and charging resistor of RC = 30 k. The second part is a discharge

circuit which consists of GaAs-MESFET device biased at high negative

gate voltage of VGG = -4V with high value resistor RGG = 1 kconnected

in series to the gate terminal. The output pulse is measured across the 50

resistor (input impedance of oscilloscope) connected to source terminal. In

the first circuit, the load resistance is connected in series with CL to drain

terminal as shown in Figure 6.22. To protect the output pulse measurement

equipment (oscilloscope) from high DC voltage during charging period, the

load resistance is connected to source terminal in the new circuit [Figure


91

6.27(b)]. In the new circuit, the trigger pulse is not needed because the

transistor will trigger by itself at high supply voltage (VDS > 12.7V).

(a)

(b)

Figure 6.27 (a) Charging and (b) discharging circuit schematics of ultra-short high

amplitude pulse generator.

The working principle of the new circuit can be explained as follows: First,

the transistor is initially in pinch-off state (because of high negative voltage

applied at the gate-terminal). After CL is charged up to VDD value, both

circuits (charging circuit and discharging circuit) are manually connected

as shown in Figure 6.28. If the capacitor voltage (VC) is high enough, the

gate-drain avalanche breakdown will occur and transistor will trigger itself.

Therefore, the capacitor will discharge with a pulse current through the

load resistor RL. This leads to a positive voltage pulse across RL. The


92

output pulse waveform is measured using 12 GHz sampling oscilloscope

DSO81204B type with 50 input impedance manufactured by Agilent.

Figure 6.28 Complete circuit schematic of ultra-short high amplitude pulse generator.

The measurement of the output pulse has been repeated 15 times for each

VDD value and the average value of these measurements is recorded. The

waveforms of the output pulses for different values of VDD are shown in

Figure 6.29. An output pulse with maximum amplitude of 169V, rise-time

of 136 ps and pulse width (FWHM) of 420 ps is obtained at a supply

voltage of VDD = 330V (330V is the highest available DC voltage). From

Figure 6.29, it is realized that the amplitude of the output pulse increases

with the increase of supply voltage (VDD). The amplitude of the output

pulse as a function of VDD is shown in Figure 6.30.


93

(a)

(b)

Figure 6.29 Waveforms of output pulse for different values of DC supply voltage

(VDD), (a) up to VDD = 55V and (b) up to VDD = 330V.

169V


94

Figure 6.30 Amplitude of output pulse as a function of DC supply voltage for T =

300K.

Figure 6.31 shows the waveform (normalized value) of the output pulse for

different values of VDD. It can be seen that the rise-time of the pulse

becomes shorter with the increase of the supply voltage (VDD). This is due

to charge carrier velocities which are high enough to produce the complete

avalanche multiplication action at high supply voltage [15].

Figure 6.31 Normalized output pulse for different values of DC supply voltage.

169V

330V


95

The dependence of rise-time (tr) and pulse width (FWHM) of the output

pulse as a function of supply voltage (VDD) are presented in Figure 6.32 and

Figure 6.33, respectively.

Figure 6.32 Rise time (tr) of output pulse as a function of DC supply voltage.

Figure 6.33 Pulse width (FWHM) of output pulse as a function of DC supply voltage.

tr min = 122 ps

p min = 270 ps


96

It can be seen that the pulse width and rise-time exhibit nonlinear

dependence with supply voltage. The highest nonlinear variation in the

pulse width and rise-time are observed at supply voltage of 17V. It is

because of the beginning stage of avalanche breakdown [15]. It can also be

seen that a minimum rise-time of 122 ps has been obtained at a supply

voltage of 200V. For pulse width, a minimum value of 270 ps has been

measured at VDD = 70V

The results of temperature dependent measurement of the output pulse

characteristic are presented in Figure 6.34 to Figure 6.36. Amplitude of the

output pulse as a function of the temperature is shown in Figure 6.34. It can

be seen that the amplitude decreases with increase of temperature. The

temperature coefficient is calculated

(168.3V(T2) – 169.2V(T1))/ 40K (T2 – T1) = 0.0225 V/K (6.3)

This is due to drain current (output pulse current), which decreases with

increase of temperature as discussed in three-terminal measurement (Figure

6.21).

Figure 6.34 Amplitude of output pulse as a function of temperature.

Temperature (K)


97

Rise-time and pulse width of output pulse as a function of temperature are

shown in Figure 6.35 and Figure 6.36, respectively. It can be seen that both

rise time and pulse width increase with increase of temperature.

Figure 6.35 Rise-time of output pulse as a function of temperature.

Figure 6.36 Pulse width of output pulse (FWHM) as a function of temperature.

All the previous measurements have been performed with gate biased to

high negative voltage below the threshold (off-state breakdown). In a

further experiment, it was tried to measure the output pulse of the circuit

shown in Figure 6.28 with floating gate (on-state breakdown) [16]. Figure

6.37 shows the off-state and on-state I-V characteristics of the transistor. It


98

can be seen that at high supply voltage (VDD), the I-V characteristics of off-

state and on-state are identical. Figure 6.38 shows the waveforms of the

output pulse both with floating gate (on-state breakdown) and feeding gate

(off-state breakdown). From Figure 6.38, it is realized that the amplitudes

of the output pulse with floating gate and feeding gate are similar.

Figure 6.37 Schematic breakdown I-V characteristic of transistor under on-state (solid

line) and off-state (dashed line) condition.

Figure 6.38 Waveforms of output pulse with floating gate and feeding gate (VGG = -4,

RGG = 1k) at VDD = 200V.


99

References

[1] G. Arumilli, RF Breakdown Effects in Microwave Power Amplifiers, Master

Thesis, Massachusetts Institute of Technology, June 2007.

[2] http://www.amcad-engineering.fr/-Pulsed-IV-RF-system-.html.

[3] E. Zanoni, G. Meneghesso, D. Buttari, M. Maretto, and G. Massari, “Pulsed

Measurement and Circuit Modeling of a New Breakdown Mechanism of

MESFETs and HEMTs,” IEEE International Reliability Physics Symposium,

2000, pp. 243-249.

[4] M. Somerville, C. Putnam, and J. Alamo, “Determining Dominant Breakdown

Mechanism in InP HEMTs,” IEEE Electron Device Letters, vol. 22, December

2001, pp. 565-567.

[5] C. Putnam, Power Limiting Mechanism in InP HEMTs, Master Thesis,

Massachusetts Institute of Technology, June 1997.

[6] H. Czichos, T. Saito, and L. Smith, Metrology and Testing, Springer-Verlag,

New York, USA, 2011.

[7] H. Li, O. Hartin, and M. Ray, “An Updated Temperature-Dependent Breakdown

Coupling Model Including Both Impact Ionization and Tunneling Mechanisms

for AlGaAs/InGaAs HEMTs,” IEEE Transactions on Electron Devices, vol. 49,

September 2002, pp. 1675-1678.

[8] A. Sleiman, A. Carlo, P. Lugli, G. Meneghesso, E. Zanoni, and J. Thobel,

“Channel Thickness Dependence of Breakdown Dynamic in InP-Based Lattice-

Matched HEMTs,” IEEE Transaction on Electron Devices, vol. 50, October

2003, pp. 2009-2014.

[9] C. Tsironis, “Prebreakdown Phenomena in GaAs Epitaxial Layers and FET´s,”

IEEE Transaction on Electron Devices, vol. 27, January 1980, pp. 277-282.

[10] M. Somerville, J. Alamo, and P. Saunier, “Off-State Breakdown in Power

pHEMT’s: The Impact of the Source,” IEEE Transaction on Electron Devices,

vol. 45, September 1998, pp. 1883-1889.


100

[11] S. Bahl, J. Alamo, J. Dickmann, and S. Schildberg, “Off-State Breakdown in

InAlAs/InGaAs MODFETs,” IEEE Transaction on Electron Devices, vol. 42,

January 1995, pp. 15-22.

[12] E. Zanoni, M. Manfredi, S. Bigliardi, A. Paccagnella, P. Pisoni, C. Tedesco, and

C. Canali, “Impact Ionization and Light Emission in AlGaAs/GaAs HEMT´s,”

IEEE Transaction on Electron Devices, vol. 39, August 1992, pp. 1849-1857.

[13] J. Walker, Handbook of RF and Microwave Power Amplifiers, Cambridge

University Press, UK, 2012.

[14] K. Hui, C. Hu, P. George, and P. K. Ko, “Impact Ionization in GaAs

MESFET´s,” IEEE Transaction on Electron Devices Letter, vol. 11, March

1990, pp. 113-115.



[16] J. Kuzmik, D. Pogany, E. Gornik, P. Javorka, and P. Kordo, “Electrical

Overstress in AlGaN/GaN HEMTs: Study of Degradation Processes,” Solid-

State Electronics, vol. 48, 2004, pp. 271-276.

Chapter 7 UWB Antenna Design

101

Chapter 7

UWB Antenna Design

Antennas with wideband impedance matching, high gain and high

directivity are one of the most important devices for UWB radar

applications [1]. Such antennas are required to transmit and receive ultra-

short pulses [2]. Different configurations of antennas are being used for

UWB radar applications such as Vivaldi antenna [3], bow-tie antenna [4],

trapezoidal antenna [5] and TEM horn antenna [6]. The comparison of

some antenna types is shown in Table 7.1. It can be seen that the horn

antenna exhibits high gain and high directivity performance, large

bandwidth and acceptable half-power beam width. These features make

TEM horn antenna a very attractive option for radar application.

Table 7.1 Comparison of Antenna Types [7].

Antenna type Bandwidth

ratio

Typical gain

(dB)

Radiation

pattern

Dipole 10:1 2 Omni-direction

Bow-tie 5:1 0 - 4 Omni-direction

TEM horn 18:1 4 - 20 Beam

Vivaldi 10:1 3-10 Beam

In this work, two different types of horn antenna, which are transverse

electromagnetic (TEM) horn antenna and double-ridge horn antenna have

been designed to meet the design criteria of the pulsed radar sensor, as


102

discussed in Chapter 4. The design criteria included the ability to yield a

large bandwidth (greater than 500 MHz according to FCC definition) to

cover the bandwidth of radar pulser, and the capability to radiate and

receive picosecond electrical pulses with minimum distortion and high

gain. These conditions were successfully met in the design of these

antennas.

In the following sections, the design details of TEM horn antenna types are

described.

7.1 Design of TEM Horn Antenna

In general, the TEM horn antenna has a linearly tapered structure or an

exponentially tapered structure. A linearly tapered structure is used more

frequently because it is easy to construct [Figure 7.1(a)]. On the other hand,

an exponentially tapered structure [Figure 7.1(b)] delivers a smaller input

reflection coefficient over a narrow frequency bandwidth [8]. In this work,

an exponentially tapered structure has been chosen to design a TEM horn

antenna.

7.1.1 Design Procedure

First, a basic TEM horn antenna consisting of two exponentially tapered

plates [9]-[10] (Figure 7.2) has been designed as shown in [8]. Commonly,

the input impedance of the TEM horn antenna at the feeding point is 50Ω

[9]-[10]. Regarding the horn antenna as a TEM waveguide, the

characteristic impedance at the aperture should be matched to 377Ω.

Therefore, the exponentially tapered structure is designed to match the

input impedance of the antenna at the feed point (50 Ω) to 377Ω at the

aperture.


103

Figure 7.1 TEM horn antenna with (a) linearly tapered structure, and (b) exponentially

tapered structure.

Figure 7.2 Structure of TEM horn antenna with exponentially tapered plate.

The axis length of the horn antenna is generally selected as half of

wavelength at lowest frequency [9]. Regarding a frequency range from 270

MHz to 1.7 GHz, which was adopted to cover the bandwidth of radar

source discussed in [8] (which is similar to the radar source discussed in


104

Chapter 5, with rise-time of 400 ps and amplitude of 48V) the axis length

of the antenna is calculated as 56 cm. The aperture size (W x H) of the

antenna is determined to be (50 cm x 50 cm) as described in [8].

To reduce the size of antenna, the axial length has simply been shortened to

a length L = 42 cm (L/L = 0.375, where /L = 112 cm). The new aperture

size is w = 24 cm and h = 12 cm. In this case, the characteristic impedance

of the antenna at the feed point cannot be matched to the free space

impedance at the aperture. Therefore, high reflection would be expected at

the antenna aperture.

To minimize the reflection at the aperture, additional means for matching

has been considered connecting a cylindrical section as shown in Figure

7.3. As discussed in [8], the optimum radius of the cylindrical section is

found to be 6 cm (D = 12 cm) as shown in Figure 7.3. The new dimensions

of the antenna with cylindrically shaped aperture are L = 46 cm (L/L =

0.41), h = 36 cm and w = 24 cm. With respect to basic TEM horn antenna,

it is shown that the modified antenna can be reduced in length and aperture

size by approximately 18 and 35 percent, respectively.

Figure 7.3 Modified TEM horn antenna with cylindrically shaped aperture.

7.1.2 Simulation and Measurement Results

The reduced TEM horn antenna with cylindrically matching section was

investigated using both simulation and measurement. The antenna has been


105

simulated using Ansoft High-Frequency Structure Simulator (HFSS)

version 12.1 [11]. The manufactured antenna is shown in Figure 7.4.

Aluminum plates with a thickness of 1.5 mm have been used for the

antenna conducting plates. For supporting and fixing the plates,

polyethylene has been used. The antenna is fed through a coaxial cable

with a SMA connector.

Figure 7.4 Photograph of the realized TEM horn antenna.

In Figure 7.5, the simulated and measured input reflection coefficients for

the fabricated TEM horn antenna are presented. It can be seen that there is

very good agreement between simulation and measurement. For a

reflection coefficient S11 less than -10 dB, the antenna bandwidth exhibits a

frequency range from 0.25 GHz up to 1.7 GHz.

The simulated and measured gain radiation pattern in E-plane and H-

plane of the antenna at 1 GHz are shown in Figure 7.6 and 7.7,

respectively. It can be seen that the peak gain of about 9 dBi is obtained

at broadside direction. It can also be seen that the HPBW (half power

beam width) of the antenna in E-plane and H-plane is about 76o and 47

o,

respectively.


106

Figure 7.5 Simulation and measurement result of input reflection coefficient of TEM

horn antenna.

Figure 7.6 Simulated and measured gain radiation pattern in E-plane of TEM horn

antenna at 1 GHz.


107

Figure 7.7 Simulated and measured gain radiation pattern in H-plane of TEM horn

antenna at 1 GHz.

The fabricated TEM horn has been used to realize a first bi-static UWB

radar system in this work. Using Figure 8.1, the system comprises a

picosecond pulse generator, the fabricated TEM horn antennas and a

sampling scope. A Gaussian-like pulse with a rise-time of 112 ps, a pulse

width (FWHM) of 155 ps, and peak power of 24.5W (see Figures 5.30 in

Chapter 5) was obtained at the output of pulse generator and fed directly

into the transmitting antenna (TEM horn antenna).

Distance measurement was performed towards a brick wall. Figure 7.8

shows the time-domain response of the realized radar to the brick wall in a

distance of 16m. It can be seen that pulse reflection from the wall could be

detected. The distance between radar sensor and wall has been calculated

from elapsed time between the reference pulse and reflected pulse

according to (3.15) to be 16m.

As explained above, high detection range of about 16m towards brick wall

which has low reflection coefficient (r = 4.5) has been achieved using the

fabricated TEM horn antenna. But the size of the antenna is too large for a

compact radar sensor. Therefore, a new double ridge horn antenna has been

designed with high potential of dimension reduction.


108

(a)

(b)

Figure 7.8 Time-domain response of a first radar system with the brick wall located

at 16m from the radar sensor. (a) reference pulse and reflected pulse, (b) reflected

pulse.


109

As will be shown the length and aperture size of the new antenna could be

reduced about 55% and 70%, respectively, with respect to the fabricated

TEM horn antenna.

In the following sections, the design details of a new double-ridge TEM

horn antenna are presented.

7.2 Design of Double-Ridge Horn Antenna

The conventional double-ridge horn antenna [12] can be decomposed into

following parts: The feed section, the waveguide section and the horn

section as shown in Figure 7.9.

(a)

(b)

Figure 7.9 Double-ridge horn antenna structure. (a) perspective view, (b) side view.


110

The feed section consists of a coaxial line and a cavity. The waveguide

section consists of a rectangular waveguide with two ridges (double-ridge

rectangular waveguide). The two tapered ridges with two flares (upper and

lower) represent the horn section. The design of antenna is divided into

three parts. The first part is the design of the double-ridge waveguide

section. The second part is the design of the horn section. The last part is

the design of the feed section. In the following, design details for each part

will be described.

A. Design of Double-Ridge Waveguide Section

The double-ridge waveguide consists of a pair of ridges symmetrically

placed in the center of the rectangular waveguide, parallel to the side wall.

In the double-ridge horn antenna, the waveguide section interconnects the

horn section with the feed section. Figure 7.10 shows the general structure

of the waveguide section. The parameters a, b and l signify the width,

height and length of waveguide section, respectively. The width of ridges

and the distance between them are expressed by d and s, respectively.

Figure 7.10 Double-ridge waveguide structure.

Because of coaxial feeding line match, the characteristic impedance of

the waveguide is chosen as 50. The corresponding ratios d/a, s/b and b/a

have been taken from published results in [13] to be equal to d/a = 0.25, s/b

= 0.1 and b/a = 0.5. The normalized cutoff wavelengths c10/a and c30/a,

which are related to the fundamental and second higher-order mode, TE10


111

and TE30 respectively, have been taken from published results in [14] to be

c10/a = 5 and c30/a = 0.7.

A frequency range from 0.65 GHz to 5.3 GHz was adopted for the design

to cover the bandwidth of 3.5 GHz of the radar emitter discussed in

Chapter 5. Based on the bandwidth of the ridge waveguide, determined by

the cut-off wavelengths of the fundamental mode TE10 and second higher-

order mode TE30 [15], the cutoff wavelengths c10 and c30 have been

calculated as 0.46m and 0.057m, respectively. With c10 and c30 known

and based on the ratios (d/a, s/b, b/a, c10/a and c30/a) in [13] and [14], the

initial values of the waveguide section parameters (a, b, s, d) can be

obtained as shown in Table 7.2. The length (l) has been chosen to be 20

mm.

Table 7.2 Initial Dimensions of Waveguide Section Parameters.

Parameters Dimension (mm)

a 86

b 43

d 30

s 4.3

B. Design of Horn Section

The design of horn section consists of three parts: Determination of axial

length, calculation of aperture size and design of shape ridges. The axis

length of the horn section is generally given by [8]

2

LL

where L is the axial length of horn section and L is the wavelength at the

lowest operating frequency of antenna [8]. For a frequency range of 0.65

GHz to 5.3 GHz the length (L) is calculated to be 230 mm. Regarding an

(7.1)


112

output power of 24.5W of the radar emitter discussed in Chapter 5, and

taking into account the demanded maximum range of 20m, calculations

discussed in Chapter 3 delivered a minimum antenna gain of 7.7 dBi. The

antenna gain determines the aperture size, which is found by simulation of

the TEM horn section without ridges using Ansoft HFSS. TEM horn

section with linear tapered and axial length of 230 mm has been structured

as shown in Figure 7.11. The dimensions of TEM horn section at feed side

(a and b) are defined as shown in Table 7.2. After several simulations and

trials, the optimal width of the upper and lower flares (W) and the distance

between them (H) at the aperture side are found to be 210 mm and 206

mm, respectively, which deliver a peak gain of 7.7 dB at broadside, needed

to cover the specified maximum range of 20m.

Side view Top view

Figure 7.11 TEM horn antenna with linearly tapered structure.

The last part in the design of the horn section is the tapering of the two

equal ridges. Regarding the ridge design, different types of profiles such as

exponential, sinusoidal, binomial can be used [16]. The exponential profile

offers better match between impedance of the waveguide section and the

free-space [17]. The exponentially tapered ridges act as a wideband

impedance transformer, the impedance varying from Zo at the feed point of

the horn section (double-ridge waveguide) to ZL at the aperture of the


113

antenna. The characteristic impedance at any point (y) along the

exponential taper is written as [18]

00 ln

1,0,

Z

Z

LLyeZyZ Ly

The separation between two ridges k(yi) is determined by an exponential

function [17]; it is given by

imy

i enyk

where n and m are constants to be determined using the separation between

the ridges at input k0 [separation between ridges of waveguide section (s

from Table 7.2)] and output aperture kL (the height of the aperture H). In

order to synthesize the exponentially tapered ridges, the axial length of the

horn section is divided into 24 sections as shown in Figure 7.12.

(a) (b)

Figure 7.12 Horn section of the double-ridge horn antenna. (a) side view, (b) top view.

The initial values of the height of the exponentially tapered ridges

[s(yi)] and spacing between them [k(yi), i=1, 2, …,24] at each section has

been obtained using (7.3) as shown in Table 7.3.

(7.2)

(7.3)

y


114

Table 7.3 The Initial Dimensions of Horn Section.

Section y (mm) s(y) (mm) k(y) (mm)

1 0 12.85 4.30

2 0 16.28 5.08

3 20 19.64 6.01

4 30 22.91 7.12

5 40 26.09 8.42

6 50 29.14 9.97

7 60 32.05 11.79

8 70 34.80 13.96

9 80 37.34 16.51

10 90 39.66 19.54

11 100 41.69 23.12

12 110 43.40 27.36

13 120 44.72 32.37

14 130 45.58 38.30

15 140 45.90 45.32

16 150 45.57 53.62

17 160 44.49 63.45

18 170 42.50 75.07

19 180 39.45 88.83

20 190 35.14 105.10

21 200 29.34 124.36

22 210 21.77 147.14

23 220 12.12 174.10

24 230 0 205.99

After the initial dimensions of waveguide and horn sections have

been obtained (Table 7.2 and 7.3), both sections are connected together as

shown in Figure 7.13. The new structure has been simulated using Ansoft


115

HFSS (one-port S-parameter simulation). After several simulations and

trials, the optimized dimensions of waveguide and horn sections have been

obtained as shown in Table 7.4 and 7.5, respectively.

Figure 7.13 Waveguide and horn sections structure (side view).

Table 7.4 Optimized Dimensions of Waveguide Section Parameters (Figure 7.10).

Parameters Dimension (mm)

a 73

b 30

d 12

s 2.6


116

Table 7.5 Optimized Dimensions of the Horn Section.

Section y (mm) s(y) (mm) k(y) (mm)

1 0 13.70 2.6

2 10 16.84 4.96

3 20 19.88 5.53

4 30 22.81 7.33

5 40 25.60 9.39

6 50 28.25 11.75

7 60 30.72 14.45

8 70 33.00 17.55

9 80 35.05 21.10

10 90 36.85 25.16

11 100 38.34 29.82

12 110 39.50 35.16

13 120 40.27 41.27

14 130 40.60 48.27

15 140 40.41 56.29

16 150 39.64 65.50

17 160 38.20 76.02

18 170 36.00 88.09

19 180 32.91 101.91

20 190 28.82 117.75

21 200 23.57 135.84

22 210 17.00 156.68

23 220 8.92 180.50

24 230 0 206

C. Design of Feed Section

In order to excite the antenna, a coaxial line with SMA adapter is

mounted to the waveguide as shown in Figure 7.14. The inner connector of


117

the coaxial line (electric field probe) is led through a hole in the lower ridge

and is connected to the upper ridge. The shield of the coaxial line is

connected to the lower ridge. The transition between the coaxial line and

waveguide is important with respect to the return loss of the antenna [19].

It is very common to use a cavity for matching to obtain low return loss for

the coaxial-to-ridge waveguide transition [19]. The transition provides

mode conversion of the TEM-mode in the coaxial line to the TE-mode in

the waveguide. The length of the cavity (t) and distance between coaxial

line and ridged edge (r) are obtained through the simulation of the antenna

as shown in the next section.

(a) (b)

Figure 7.14 Configuration of the coaxial line to waveguide transition. (a) top view, (b)

side view.

7.2.1 Simulation Results

Ansoft HFSS is used to analyze the prototype of the antenna. The

geometry of the antenna structure in 3D form is shown in Figure 7.15. The

dimensions of the antenna are defined as shown in Table 7.4 and Table 7.5.

Waveguide port with characteristic impedance of 50 is used to model the

coaxial line.


118

(a)

Figure 7.15 The geometry of the antenna structure for HFSS simulation. (a) 3D, (b)

side view, (c) top view.


119

To prevent any distortion in the radiation and impedance characteristics

of the antenna, an air box which represents the radiation boundary was

drawn around the structure of the antenna [20]. The air box was terminated

by using a perfectly matched layer (PML) type of absorbing boundary. The

distance from the radiating source to the radiation boundary is set to L/10

[21], where L is the wavelength at the lowest operating frequency of

antenna. For lower frequency of 1.2 GHz, the distance from the antenna

structure to the radiation boundary is determined to be 25 mm (L/10). To

calculate the return loss at the feed point of the antenna, one-port S-

parameter simulation is applied. After several HFSS simulations and trials

of the proposed antenna, the optimized length of the cavity (t) and the

coaxial line spacing from ridged edge (r) shown in Figure 7.14 have been

obtained to be t = 7.5 mm and r = 5 mm.

These values deliver a minimum return loss at the antenna feed point.

Figure 7.16 shows the simulation of reflection coefficient (with HFSS) at

the feed point (S11). For the reflection coefficient S11 less than -10 dB, the

antenna bandwidth [22] covers a frequency range from 1.2 GHz up to 6.5

GHz.

Figure 7.16 Simulated magnitude of S11 of designed antenna versus frequency.

The 3D far-field radiation pattern at the frequencies 1.5, 3.2 and 6 GHz,

which is obtained from HFSS simulation, are shown in Figure 7.17. It can


120

be concluded that the radiation patterns have one major lobe in the y-axis.

With increasing frequency, the width of main lobe (beam width) decreases

and the amplitude of the gain increases.

(a)

(b)

(c)

Figure 7.17 3D view of radiation pattern of designed antenna at (a) 1.5 GHz, (b) 3.2

GHz, (c) 6 GHz.

Theta

Phi

Theta

Phi

Phi

Theta


121

The current distribution on the antenna surface, which is obtained from

HFSS simulation, is shown in Figure 7.18.

(a)

(b)

(c)

Figure 7.18 Current distribution on the surface of the designed antenna at (a) 1.5 GHz,

(b) 3.2 GHz, (c) 6 GHz.


122

It is observed that the maximum current is concentrated on the ridge

surfaces.

7.2.2 Size Reduction

To reduce the size of radar sensor system, compact antenna is needed

because the antenna is frequently the largest component in the radar

systems [23]. As discussed in the design of TEM horn antenna, the size

reduction of new double-ridge horn antenna is accomplished by reducing

the axial length and aperture size. First, the length of the horn section has

simply been shortened to a length L = 120 mm (L/λ = 0.26, where λL = 460

mm) as shown in Figure 7.19. No further optimization has been done. The

new aperture has a size of width (w) = 140 mm and height (h) = 120 mm.

Figure 7.19 Shorted double-ridge horn antenna.

The simulation result from HFSS of the input reflection coefficient of the

shorter antenna is shown in Figure 7.20. It can be seen that the bandwidth

with S11 ≤ -10 dB is approximately 3.3 GHz (from 3 to 6.6 GHz).

Comparing with the results in Figure 7.16, it can be noticed that the

bandwidth of the antenna is reduced by 20%.


123

Figure 7.20 Simulated magnitude of S11 of shorter antenna versus frequency.

To increase the bandwidth of the shorted antenna, the shape of the ridges

has been modified by adding circular section at the end of the ridges, as

shown in Figure 7.21.

Figure 7.21 Shorted antenna with circular ridge section.

The circular section controls the opening of the ridges, which improves the

matching between feeding line impedance and free space impedance.

Through several HFSS simulation, the optimum radius of the circular


124

section is obtained as r = 65 mm (r/λL = 0.141). The new dimensions of the

modified antenna with circular ridge profile are: L = 174 mm (L/λL =

0.378), h = 160 mm and w = 160 mm. The input frequency bandwidth of

the modified antenna, using the condition of S11 < -10 dB, is approximately

5.42 GHz (from 1.38 to 6.8 GHz) as shown in Figure 7.22. The input

reflection coefficients of conventional and modified antenna are compared

in Figure 7.23. From this figure, it can be noticed that the bandwidth (for

S11 < -10 dB) of the modified type is approximately equal to the

conventional type. It can also be seen that there is good agreement between

simulation results of conventional and modified types in the frequency

range between 2.5 GHz and 6 GHz. However, for lower frequencies up to

2.5 GHz, disagreement between simulation results of both types occurs.

This might be attributed to the length different of both antenna types.

With respect to the conventional antenna, the axial length and aperture size

of the modified antenna could be reduced by approximately 21 and 40

precent, respectively.

Figure 7.22 Simulated magnitude of S11 of modified antenna versus frequency.


125

Figure 7.23 Simulated magnitude of S11 of conventional and modified double-ridge

horn antennas versus frequency.

7.2.3 Fabrication and Measurements

The procedure to build up the modified antenna and the measurement

setups for antenna characterization is presented in this section.

7.2.3.1 Manufacturing of the Antenna

In order to construct the modified antenna in a simple way, the antenna has

been divided into several parts as illustrated in Figure 7.24(a). The parts of

the fabricated antenna are upper cover (1), lower cover (2), upper ridge (3),

lower ridge (4), right side (5) and left side (6) edges of the waveguide

section, and back edge of the cavity (7). Aluminum plates with the

thickness of 2 mm, 12 mm and 6 mm are used to fabricate upper and lower

covers, upper and lower ridges, sides and back edges, respectively. The

antenna is constructed by combining antenna parts using screws as shown

in Figure 7.24(b). To feed the antenna, SMA connector is mounted to the

waveguide section. The inner conductor penetrates the lower ridge and is

connected to the upper ridge. The outer conductor is connected with the

lower ridge.


126

(a) (b)

Figure 7.24 Realization of modified double-ridge horn antenna. (a) antenna parts, (b)

photograph.

7.2.3.1 Measurement Setups and Measurement Results

A. Return Loss (S-Parameter)

The S-parameter measurement setup of the fabricated antenna is shown in

Figure 7.25. The PNA-X Vector Network Analyzer (VNA) from Agilent is

used to measure the input reflection coefficient under matched load

conditions. Before the measurement is started, a full one-port VNA

calibration is applied. The calibration method used in this work is called

SOL (short, open, load), which requires a short, open and matched load

standard. The calibration procedure has been automatically accomplished

using N4960 electronic calibration (Ecal) module. The reference plane of

the measurement is set at the SMA connector under the waveguide section.

The calibration procedure is performed over frequency bandwidth of 7.035

GHz (from 1 to 8.035 GHz) with frequency step size of 35 MHz. After the

calibration of VNA is carried out, the manufactured antenna is connected to

the VNA port as shown in Figure 7.25. The input reflection coefficient

(S11) data is measured over the specified frequency bandwidth.


127

Figure 7.25 VNA instrument setup for characterization of the antenna.

The magnitude of S11 of the modified antenna obtained from both VNA

measurement and Ansoft HFSS simulation is shown in Figure 7.26. It can

be seen that there is good agreement between simulation and measurement.

The usable bandwidth of the fabricated antenna which is defined from the

S11 data, namely |S11| < -10 dB, exhibits a frequency range from 1.4 GHz

up to 7.0 GHz. The ratio bandwidth (Br) and fractional input bandwidth

(bw) of the antenna, defined by (4.3) and (4.4) in Chapter 4 are calculated

to be 1.33 and 5.0:1, respectively.

Figure 7.26 Simulation and measurement of the magnitude of input reflection

coefficient S11 of the modified double-ridge horn antenna versus frequency.

In addition, the voltage standing wave ratio (VSWR) at the input of the

antenna is calculated from the |S11| data as [24]


128

20

20

11

11

101

101dBS

dBS

VSWR

For the modified antenna, the simulation results of VSWR obtained from

HFSS is compared with the measurement result obtained with VNA as

shown in Figure 7.27. It can be seen that there is very good agreement

between the simulation and measurement. In case of VSWR results, the

bandwidth of the antenna is defined as the frequency range for which the

VSWR is less than 2. From Figure 7.27, it can be noticed that the antenna

bandwidth covers a frequency range from 1.4 GHz to 7.0 GHz.

Figure 7.27 Simulation and measurement of VSWR of the modified double-ridge horn

antenna versus frequency.

Both measured and simulated input resistance and reactance of the antenna,

which are calculated from S-parameter results are shown in Figures 7.28

and 7.29, respectively.

(7.4)


129

Figure 7.28 Simulation and measurement of the input resistance of the modified

double-ridge horn antenna versus frequency.

Figure 7.29 Simulation and measurement of the input reactance of the modified double-

ridge horn antenna versus frequency.

The measurement results of input resistance of the antenna are oscillating

around 50 for frequency range between 1.5 GHz to 7.3 GHz. In this

frequency range, the input resistance varies between a maximum of 90

and a minimum of 32 value. The measurement results of input reactance

of the antenna are also oscillating around 0 for the same frequency range.


130

This oscillatory shape of measurement results of both, input resistance and

input reactance are due to multiple reflection of the antenna.

B. Radiation Measurements

The setups for measurement of the transient radiation pattern and gain

radiation pattern of the antenna are presented in this section. The radiation

characteristics of the antenna are calculated for far-field (Fraunhofer)

region. Therefore, the receiving test antenna is located in far-field region of

the emitting antenna. To fulfill the far-field condition, the two antennas

have to be separated by a distance (R) [25]

22DR

where D is the largest dimension of antenna aperture and is the

wavelength at operational frequency of antenna. For D = 160 mm (largest

dimension of antenna aperture) and for a frequency range extending from

1.4 GHz to 7 GHz, the minimum distance between two antennas (Rmin)

should be set to 23 cm. These measurements have been performed inside

the lab of Microwave Electronics Department, University of Kassel.

B.1 Transient Radiation Pattern

The measurement setups for the transient radiation pattern are shown in

Figure 7.30. A pulse with rise time of 112 ps, pulse width (FWHM) of 155

ps and amplitude of 35V, which is provided by the pulse generator

discussed in Chapter 5, is fed directly to antenna under test (AUT). At the

receiving side, the receiver antenna (test antenna) is connected to the

sampling oscilloscope (Agilent DSO81204B) with 50 input impedance to

measure the waveforms of the receiving pulses. The measurements of the

transient radiation pattern are performed with the distance of 2m between

transmitter and receiver antennas. The transient radiation pattern of the

antenna is investigated in two different radiation planes. The first plane is

(7.5)


131

at an azimuth angle = 90o (broadside) and the second plane is at an

azimuth angle = 0o (edge-on) as shown in Figure 7.30(a) and 7.30(b),

respectively.

(a)

(b)

Figure 7.30 Transient radiation pattern measurement setup: (a) broadside radiation, (b)

edge-on radiation.

The measurements of the transient radiation pattern are carried out by

fixing the receiving antenna in broadside direction (azimuth angle of 90o)

with elevation angle of 90o and rotating the radiating antenna at different

elevation angles (from 0o to 90

o) for both, broadside (azimuth angle of

90o) and edge-on (azimuth angle of 0

o) orientation.

The measurement results of the broadside and edge-on far-field radiation of

the antenna are shown in Figure 7.31. The far-field radiation from the


132

broadside of the antenna (azimuth angle of 90o) is presented in Figure

7.31(a). It can be seen that the maximum pulse amplitude occurs at an

elevation angle equal to 90o. For the edge-on far-field radiation of the

antenna, the maximum pulse amplitude is also attained at an elevation

angle equal to 90o as shown in Figure 7.31(b).

(a)

(b)

Figure 7.31 Measurement results of the far-field radiation of the antenna: (a) broadside,

(b) edge-on.


133

The investigated radiation planes (broadside and edge-on) are sufficient to

characterize the far-field time-domain radiation of the antenna.

Furthermore regarding a bistatic radar concept as discussed in this thesis,

the edge-radiated field is used to provide a reference pulse between

transmitting and receiving antennas, while broadside radiation is used to

detect targets and receive their returns.

B.2 Gain Radiation Pattern

The measurement setup of the gain radiation pattern is shown in Figure

7.32. A signal generator (HP83650B), which provides RF signals with an

output power of 5 dBm is connected directly to antenna under test (AUT).

At the receiving side, the receiver antenna (test antenna) is connected to the

spectrum analyzer (FSV-Rohde & Schwarz) to measure the output power

of the receiver antenna.

The measurements of the gain radiation pattern are performed in the E-

plane (elevation angle of 90o and azimuth angle of 0

o to 360

o) and H-

plane (elevation angle of 0o to 360

o and azimuth angle of 90

o).

Figure 7.32 Gain radiation pattern measurement setup.

The measurement and simulation (HFSS) results of gain radiation pattern

in E-plane and H-plane of the antenna are presented in Figure 7.33 to

Figure 7.36 at 1.8, 4, 6 and 7 GHz, respectively. The half-power beam

width (HPBW) in the E-plane and H-plane is shown in Table 7.6.


134

Table 7.6 Directional Radiation Beam of Antenna.

It can be seen that the directional radiation beam of the antenna (HPBW) in

both planes (E-plane and H-plane) becomes narrow with increase of

operation frequency. It can also be seen that by increasing the operation

frequency from 1.8 GHz to 6 GHz, the beam width of the antenna reduces

of about 42% and 12% in H-plane and E-plane, respectively. A minimum

beam width of about 42 degree and 32 degree has been measured at 7 GHz

in E-plane and H-plane, respectively. The small variation of beam width in

E-plane with increase of frequency is due to the circular matching section.

From Figures 7.33 to 7.36, it becomes evident that the radiation with

maximum amplitude occurs at broadside direction (elevation angle of 90o

and azimuth angle of 90o).

It can also be seen that a maximum gain of 12.66 dBi has been obtained at

frequency of 7 GHz.

Frequency

(GHz)

Directional radiation beam (degree)

E-plane

[HPBW (3 dB)]

H-plane

[HPBW (3 dB)]

Simulation Measurement Simulation Measurement

1.8 46 50 78 80

4 46 48 56 62

6 42 44 40 46

7 36 42 28 32


135

(a)

(b)

Figure 7.33 Simulation and measurement results of gain radiation pattern of antenna at

1.8 GHz, (a) H-plane, (b) E-plane.

Ga

in (

dB

i)

Ga

in (

dB

i)


136

(a)

(b)


4 GHz, (a) H-plane, (b) E-plane.

Ga

in (

dB

i)

Ga

in (

dB

i)


137

(a)

(b)



Ga

in (

dB

i)

Ga

in (

dB

i)


138

(a)

(b)



Ga

in (

dB

i)

Ga

in (

dB

i)


139

References

[1] A. Andriianov, “Generators, Antennas and Registrator for UWB Radar

Application,” International Workshop on Ultrawideband Systems and

Technologies, May 2004, pp. 135-139.

[2] R. Jongh, A. Yarovoy, L. Lightart, I. Kaploun, and A. Schukin, “Design and

Analysis of New GPR Antenna Concepts,” Conference on Ground Penetrating

Radar, May 1998, pp. 1-6.

[3] C. Rusch, J. Schäfer, T. Klein, S. Beer, and T. Zwick, “W-Band Vivaldi Antenna

in LTCC for CW-Radar Nearfield Distance Measurements,” Proceedings of the

5th European Conference on Antennas and Propagation, April 2011, pp. 2124-

2128.

[4] F. Congedo, and L. Tarricone, “Modified Bowtie Antenna for GPR

Applications,” International Conference on Ground Penetrating Radar, June

2010, pp. 1-5.

[5] P. Eskelinen, “Improvements of an Inverted Trapezoidal Pulse Antenna,” IEEE

Antennas and Propagation Magazine, vol. 43, June 2001, pp. 82-86.

[6] A. Jamali, and R. Marklein, “Design and Optimization of Ultra-Wideband TEM

Horn Antennas for GPR Applications,” General Assembly and Scientific

Symposium, August 2011, pp. 1-4.

[7] P. Foster, “Performance of Ultra-Wideband Antennas”, Proceedings SPIE

Ultrawideband Radar, vol. 1631, pp. 134-145, 1992.

[8] A. Ameri, G. Kompa, and A. Bangert, “Study About TEM Horn Size Reduction

for Ultra-Wideband Radar Application,” German Microwave Conference

(GeMiC), 2011, pp. 1-4.

[9] H. Choi and S. Lee, “Design of Exponentially Tapered TEM Horn Antenna for

the Wide Broadband Communication,” Microwave and Optical Technology

Letters, vol. 40, 2004, pp. 531-534.





http://ieeexplore.ieee.org/xpl/RecentIssue.jsp?punumber=74




140

[10] K. Chung, S. Pyun, and J. Choi, “Design of an Ultrawide-Band TEM Horn

Antenna with a Microstrip-Type Balun,” IEEE Trans. Antenna and Propagation,

vol. 53, 2005, pp. 3410-3413.

[11] High Frequency Structure Simulator (HFSS), Ansoft Corporation, Version 12.1,

2009.

[12] A. Mallahzadeh, and A. Imani, “Modified Double-Ridged Antenna for 2-18

GHz,” Journal of Applied Computational Electromagnetics Society, vol. 25,

2010, pp. 1-7.

[13] J. Helszajn and M. McKay, “Voltage-Current Definition of Impedance of

Double Ridge Waveguide Using the Finite Element Method,” IEE Proceedings

Microwaves, Antennas and Propagation, vol. 145, 1998, pp. 39-44.

[14] S. Hopfer, “The Design of Ridged Waveguides,” IRE Transaction-Microwave

Theory and Techniques, October 1955, pp. 20-29.

[15] J. Qiu, Y. Suo, and W. Li, “Design and Simulation of Ultra-Wideband Quad-

Ridged Horn Antenna,” International Conference on Microwave and Millimeter

Wave Technology, April 2007, pp. 1-3.

[16] M. Ghorbani, and A. Khaleghi, “Wideband Double Ridged Horn Antenna:

Pattern Analysis and Improvement,” European Conference on Antennas and

Propagation, April 2011, pp. 865-868.

[17] F. Karshenas, A. Mallahzadeh, and A. Imani, “Modified TEM Horn Antenna for

Wideband Applications,” International Symposium on Antenna Technology and

Applied Electromagnetics and the Canadian Radio Sciences Meeting, February

2009, pp. 1-5.

[18] D. M. Pozar, Microwave Engineering, 2nd

Edition, New York: Wiley, 1998.

[19] A. R. Mallahzadeh and A. A. Dastranj, “Modified Double-Ridged Antenna for

2-18 GHz,” Applied Computational Electromagnetics Society Journal, vol. 25,

2010, pp. 137-143.








141

[21] I. Bardi and Z. J. Cendes, “New Directions in HFSS for Designing Microwave

Devices,” Microwave Journal, Horizon House Publications Inc, August 1998.


House Inc., 2005.

[23] G. Cheng, T. Ho, W. Wang, C. Chang, and S. Chung, “Highly Integrated

Automotive Radar Sensor,” Electronics Letters, vol. 43, August 2007, pp. 993-

994.

[24] R. Haupt, Antenna Arrays: A Computational Approach, John Wiley and Sons,

Inc., 2010.

[25] R. Yadava, Antenna and Wave Propagation, PHI Learning Private Limited,

2011.

Chapter 8 Ranging Measurements

142

Chapter 8

Ranging Measurements

8.1 Measurement Setup

The radar components, which are discussed in Chapters 5 and 7 have been

used to build a complete radar sensor. The designed system has been tested

for distance measurement (towards metal plate and brick wall) and for

water level control measurements. The developed radar sensor was

constructed in bi-static configuration. The measurement setup of the sensor

is shown in Figure 8.1.

Figure 8.1 Measurement setup of the bi-static radar sensor.

In all measurements, high voltage pulses with an amplitude of 183V (at

50 load) are generated using avalanche transistor circuit (see Figure

5.15). This pulse was attenuated and split into two pulses using attenuator


143

circuit and signal divider, respectively. Two balanced pulses with

amplitudes of 30V and -31V (see Figures 5.17 and 5.19) were obtained at

the output of the signal divider. These pulses were used to feed two equal

SRD pulse shaping circuits. The output of the SRD sharpener circuits were

two Gaussian-shaped pulses with rise-time of 102 ps and 96 ps, pulse width

(FWHM) of 140 ps and 135 ps, and amplitude of -24V and 23V (see

Figures 5.26). A Gaussian-shaped pulse with a rise-time of 112 ps, a pulse

width (FWHM) of 155 ps, and an amplitude of 35V was obtained at the

output of the signal combiner (see Figure 5.31).

The output of the signal combiner was fed directly to the transmitting

antenna [double-ridge horn antenna (TX)]. The receiving antenna (RX)

receives first the reference pulse directly from TX (using edge radiation

characteristic of the antenna flares) and secondly the target return pulse

(using the broadside radiation characteristic of antenna ridge). RX was

connected to the sampling oscilloscope (Agilent DSO81204B) with 50

input impedance to downconvert the picosecond received pulses. For data

acquisition during measurements, general purpose interface bus (GPIB)

was used to connect PC and oscilloscope.

In all measurements, the distance between the two antennas of radar

sensor (TX and RX) was chosen to be 40 cm as shown in Figure 8.2.

Figure 8.2 Reference measurement point between two antennas of the bi-static radar

sensor.


144

The reference point of the radar sensor is taken as virtual center point

between the two apertures of the antennas as shown in Figure 8.2. This was

done to obtain the shortest range measurement possible.

8.2 Minimum Detectable Signal

At the beginning of the radar signal processing, the thresholds of decisions

(minimum detectable signal) were chosen before measuring the radar

received signals.

To determine the thresholds of decisions, the peaks of received pulses are

measured in absence and presence of different targets. Figure 8.3 shows the

time-domain response of radar sensor in absence of any target. These

measurements have been done in the corridor inside the building of

Electrical Engineering Department, University of Kassel. From Figure 8.3,

it can be seen that the measured data involved noise and unwanted peaks

which occur due to antenna ringing [1]. The ringing of the antenna is

caused due to energy storage or multiple reflections in the antenna [2].

Figure 8.3 Time-domain response of the radar sensor without any target.


145

The amplitudes of received signals are presented in Table 8.1 which

includes the amplitude of target return pulses for different targets, noise

level and unwanted peaks.

Table 8.1 Amplitude of Received Pulses.

It is evident from Table 8.1 that the unwanted pulses (due to the antenna

ringing) are detected after the reference pulse is received. These pulses

exist in time domain after reference pulse as shown in Figure 8.3.

The amplitudes of target return pulses [for both targets (metal plate and

brick wall)] at distance of 20m are lower than the unwanted peaks.

Therefore, the detection range of radar sensor was divided into two regions.

First region is up to 3m and second region is from 3m up to 20m. The

thresholds of decisions are taken for each region. In this case, if the

received pulse peaks are higher than the decision threshold of the region,

then the pulses are considered for further signal processing. On the

contrary, if the peaks of received pulses are less than the decision threshold

of the region, then it is supposed that no target is detected. To study the

level of decision threshold, several tests have been conducted. It has been

found that 60 mV and 10 mV are the optimum thresholds of decisions for

first region (up to 3m) and second region (from 3m up to 20m),

respectively.

Target Amplitude (mV)

Brick wall [300 cm x 310 cm x 27 cm (at 20m distance)] 21

Metal plate [70 cm x 70 cm x 2 mm (at 20m distance)] 14.3

Unwanted peaks [antenna ringing (at 0.6m)] 43

Noise level (at 20m) 6


146

After definition of the threshold for each region, the measurement of

detected pulses could be started. The target range is measured using the

standard time of flight (TOF) method [3]-[5]. The time difference between

the detected pulse and reference pulse is used in conjunction with (3.15) in

Chapter 3 to calculate the distance.

To evaluate the time difference (td) between the reference pulse and

detected pulse, the time significant points at 50% of pulse amplitudes are

evaluated [5] as shown in Figure 8.4.

Figure 8.4 Measured time significant points in pulsed laser radar [5].

8.3 Distance Measurement to Metal Plate

The first experimental test for the radar sensor was to measure the

distance of metal plate with a size of (70 cm x 70 cm x 2 mm). The

measurement setup of pulsed radar sensor for the measurement distance of

the metallic target is shown in Figure 8.5. The two antennas were mounted

on two single-legged stands with height of 90 cm above the ground. The

target (metal plate) was fixed on a wooden box which has a height of 35

cm. The antennas were oriented such that they were facing the target (the

target is parallel to the antenna broadside). The target was placed in the

vertical position at distances of 5.6m, 11.75m, and 20m from the antenna.

These measurements have been performed in the corridor (height = 3.1m


147

and width = 2.7m) inside the building of Electrical Engineering

Department, University of Kassel. In this experiment, radar pulse with a

rise-time of 112 ps, pulse width of 155 ps and amplitude of 35V was fed

into the transmitting antenna (with 50 input impedance).

(a)

(b)

Figure 8.5 Distance measurement setup for metal plate. (a) side-view, (b) top-view.


148

Figure 8.6 shows a photograph of the measurement setup with radar

sensor. Figure 8.7 to Figure 8.9 show the time-domain responses of radar

sensor with aluminum plate.

Figure 8.6 Photograph of measurement setup.

In addition to reference pulse, reflected pulse and unwanted peaks

which are labeled in Figure 8.7 to Figure 8.9, some ringing in the time-

domain response is noticed. This ringing occurred after the target return

pulse is received.


149

(a)

(b)

Figure 8.7 Time-domain response of the radar sensor measurement towards an

aluminum plate (70 cm x 70 cm x 0.2 cm) located at 5.6m. (a) reference pulse and

reflected pulse, (b) reflected pulse and target ringing.

80 mV


150

(a)

(b)


aluminum plate (70 cm x 70 cm x 0.2 cm) located at 11.75m. (a) reference pulse and

reflected pulse, (b) reflected pulse and target ringing.


151

(a)

(b)


aluminum plate (70 cm x 70 cm x 0.2 cm) located at 20m from the radar sensor. (a)

reference pulse and reflected pulse, (b) reflected pulse and target ringing.

14.3 mV


152

Regarding the radar equation discussed in Chapter 3, the received power of

a point target and an extended target decreases with the distance R

according to PR ~ 1/R4 and PR ~ 1/R

2, respectively. As the radar measuring

beam is only moderately focused, the question arises whether the used

metal plate with an areal dimension of 70 cm × 70 cm represents a point or

an extended target. From Figures 8.7 and 8.9 we obtain a voltage signal

amplitude of 80 mV for a distance of 5.6m; at a distance of 20m we get

14.3 mV. Thus, the voltage and power ratios turn out to be 5.59 and 31.25,

respectively. The distance ratios (R2/R1)2 and (R2/R1)

4 are calculated as

12.75 and 162.7, respectively. Thus, as a good approximation, the metal

plate can be considered as an extended target within the measured range up

to 20m.

8.4 Distance Measurement towards Brick Wall

The next experimental test was performed by using brick wall as target. In

this experiment, the used wall was a 27 cm thick brick wall inside the

building of Electrical Engineering Department, University of Kassel. The

antenna setup and transmitted pulse were the same as used for distance

measurement to metal plate. The antennas were positioned in front of the

wall at distances of 8.8m, 11.7m and 19.9m. Figure 8.10 to Figure 8.12

show the time-domain response of radar sensor to the brick wall.


153

(a)

(b)

Figure 8.10 Time-domain response of the radar sensor measurement towards a brick

wall located at 8.8m from the radar sensor. (a) reference pulse and reflected pulse, (b)

reflected pulse and target ringing.


154

(a)

(b)


wall located at 11.7m from the radar sensor. (a) reference pulse and reflected pulse,

(b) reflected pulse and target ringing.


155

(a)

(b)


wall located at 19.9m from the radar sensor. (a) reference pulse and reflected pulse,

(b) reflected pulse and target ringing.


156

Figure 8.13 shows the measured normalized received power of target return

pulse (for both targets, metal plate and brick wall) as a function of distance.

It can be seen that at target distances of 0 to 30 cm, the received power of

reflected signals increases with increase of distance. After 30 cm distance,

the received power of target return pulse is inversely proportional to the

square of the distance. The reason is that at near distance, the transmitting

and receiving radiation pattern overlap only partly in case of given bi-static

radar configuration. 1/R2 ratio is valid at larger range when both radiation

pattern overlap completely over the given target area.

Figure 8.13 Normalized received signal as function of distance.

8.5 Water Level Control Measurement

In addition to measurement towards metal plates and brick walls, water

level control measurement was also performed. The measurement setup of

the bi-static radar sensor for water level control measurement is shown in

Figure 8.14.

A 200 liters plastic rainwater tank was used in this measurement. The water

level inside the tank was changed in steps of 5 cm till a level of 75 cm was

reached. The antennas were mounted on a single-legged stand (T-shape,


157

see Figure 8.14(b). With this stand, it is possible to change the elevation of

the antennas from the ground (from 1m up to 2.3m).

(a)

(b)

Figure 8.14 Radar sensor measurement setup for water level control. (a) side view, (b)

front view.


158

Figure 8.15 shows a photograph of the water level control measurement

setup of the radar sensor.

Figure 8.15 Photograph of measurement setup for water level control.

Figure 8.16 shows the time-domain response signal of the bi-static radar

sensor for water levels of 0 cm (tank is empty), 45 cm and 75 cm,

respectively. The distances between the antennas and water surfaces are

calculated as 210 cm, 165 cm and 135 cm, respectively. From the results in

Figure 8.16, it can be realized that the amplitude of return pulse from

surface of water is relatively large. This is due to the high value of

reflection coefficient of water [6] as discussed in Chapter 3.


159

(a)

(b)

Figure 8.16 Time-domain response of the radar sensor measurement towards water

surface level of (a) 0 cm (empty tank) and (b) 45 cm.


160

(c)

Figure 8.16 Time-domain response of the radar sensor measurement towards water

surface level of (c) 75 cm.

Figure 8.17 shows the measured water level as function of the actual water

level. Measurements have been repeated 15 times for each level step (10

cm). The average value of these measurements is recorded. The

measurement error is defined as the difference between actual and

measured distances. A maximum error of 1.5 cm occurred at a water level

of 50 cm (R = 160 cm).


161

Figure 8.17 Measured water levels as a function of the actual water levels.

8.6 Ranging Uncertainty

The time-dependent measurement accuracy of the radar sensor was also

tested. The measurements were accomplished by keeping a target [metal

plate (70 cm x 70 cm x 0.2 cm] at a fixed position from the antennas and

record the measured data several hundred times. The ranging deviations are

given as the difference of actual and measured distances. Figure 8.18 shows

the measured range deviations for 250 distance measurements for a target

at a fixed position of 5m from the antennas. The values scatter within ±14

mm, corresponding to a measurement uncertainty of 14 mm. The statistical

occurrence of distance error was investigated by plotting the distance error

as a function of its probability as shown in Figure 8.19.


162

Figure 8.18 Scattering of measured range data.

Figure 8.19 Distance error probability of radar range measurements.


163

References

[1] A. Ruengwaree, A. Ghose, and G. Kompa, ‘‘A Novel Rugby-Ball UWB

Antenna for Near-Range Microwave Radar System,’’ IEEE Transaction on

Microwave Theory and Techniques, vol. 54, June 2006, pp. 2774-2779.

[2] W. Wiesbeck, G. Adamiuk, and C. Sturm, ‘‘Basic Properties and Design

Principles of UWB Antennas,’’ IEEE Proceedings, vol. 97, February 2009, pp.

372-385.





[5] A. Ghose, Pulsed Measurement Based Nonlinear Characterization of Avalanche

Photodiode for the Time Error of 3D Pulsed Laser Radar, Doctoral Thesis,

University of Kassel, July 2005.



Chapter 9 Conclusion and Future Work

164

Chapter 9

Conclusion and Future Work

In this work, a UWB radar sensor for long ranging has been presented

and discussed. The proposed radar sensor was built in a bi-static

configuration.

Main focus of this work was to cover a measuring range of about 20m

with a measurement accuracy in the mm range towards targets with lower

reflection coefficients such as bricks with a dielectric constant r of about

4.5.

A pulse generator with ultra-short high amplitude electrical pulses

including a silicon avalanche transistor circuit with a new SRD pulse

sharpening circuit was developed. Because of limited rated voltage of high-

voltage ultra-fast SRD, the output pulse of an avalanche pulse generator

with an amplitude of -183V was first reduced and split into two pulses using

an attenuator circuit and a signal divider (balun transformer), respectively.

The output pulses of the signal divider drive two separate SRD pulse

shaping circuits comprising a SRD sharpener and a pulse-forming network

(Schottky diodes and delay lines) to sharpen and form the balun transformer

output pulses. A transmission line transformer (as signal combiner) was

used to combine the output pulses of the SRD sharpener circuits. Ultra-

short, high power pulses with a rise-time of 112 ps, a fall-time of 150 ps, a

pulse width (FWHM) of 155 ps and an amplitude of 34.5V were obtained.

In addition, the effects of the external parameters variation of the avalanche

transistor and SRD sharpener circuits on the pulse characteristics were

discussed.


165

In addition to the generation of ultra-short pulses using a silicon avalanche

circuit with SRD pulse shaping circuit, the generation of ultra-short high

power pulses based on the avalanche breakdown phenomenon of modern

GaAs MESFET was also studied. This study was performed in two parts.

In the first part, conventional two-terminal and three-terminal I-V

measurements were carried out for different temperature values to

recognize the breakdown phenomena of the transistor. In the second part, a

new device modulation technique is used to generate ultra-short high-

power pulses. In this technique, two separate circuits (charging circuit and

discharging circuit) were used to generate ultra-short pulses. Through this

study, very fast high amplitude pulses with rise-time of 136 ps, FWHM of

420 ps and amplitude of 169V were obtained.

In order to transmit and receive ultra-short pulses, two different types of

TEM horn antenna were developed. The first antenna consists of two

exponential tapered plates with an axial length of 42 cm. The measured

bandwidth of this antenna exhibits a frequency range from 0.25 GHz up to

1.7 GHz. This antenna was used to realize a first complete bi-static UWB

radar system in this work. With this antenna, measurement distance of

about 16m towards brick wall was achieved.

To reduce the size of TEM horn antenna, a new compact double-ridge horn

antenna was designed. Different kinds of double-ridge horn antennas were

studied. In the first type, a conventional double-ridge horn antenna was

designed and optimized. Then, the length of the conventional double-ridge

horn antenna was simply shortened (second type). Subsequently, the shape

of the shorted antenna ridges was optimized by adding circular section at

the end of the ridges to improve the bandwidth of the antenna. The

modified antenna was constructed using aluminum plate. The bandwidth of

the antenna was defined by the input reflection coefficient S11 having a

value less than -10 dB. Based on this condition, the antenna bandwidth

covered a frequency range from 1.4 GHz up to 7 GHz. The radiation of the

antenna with maximum amplitude occurs at broadside direction (elevation


166

angle of 90o and azimuth angle of 90

o). With respect to the TEM horn

antenna, the modified double-ridge horn antenna could be reduced in length

and aperture size by approximately 50 and 70 percent, respectively.

The discussed radar components have been used to build a complete

radar sensor. The designed system has been tested for distance measurement

(towards metal plate and brick wall) and for water level control

measurements. Before measuring the radar received signals, the thresholds

of decisions (minimum detectable signal) were chosen. The target range was

performed using the standard time of flight concept. In this method, the

target distance was computed from the time difference between the detected

pulse and reference pulse. The time-dependent measurement accuracy of the

radar sensor was investigated. A measurement uncertainty of 14 mm was

obtained.

Further research may be started to enhance radar sensor performance. It has

been demonstrated that a modern GaAs-MESFET device can provide ultra-

short high power pulses based on the avalanche breakdown phenomenon.

In comparison with the avalanche pulse generation using silicon

technology, the future employments of III-V devices will simplify the radar

transmitter concept as illustrated in Table 9.1.

Table 9.1 Summary of Generation Approaches of Ultra-Short High-Power Pulses.

As discussed in this thesis, conventional Si-BJT devices need SRD-circuits

for pulse sharpening. High impulse amplitudes require additional balancing


167

of the avalanche pulse prior to pulse sharpening. In case of the GaAs-

MESFET, pulse sharpening is no longer necessary. The avalanche pulse is

extremely high (169V at 50) and ultra-short (rise-time of 136 ps).

The following subjects could be treated in future:

Searching for a suitable switch in the GaAs-MESFET-avalanche

pulser.

Investigating the breakdown phenomenon of further MESFET and

HEMT types on GaAs material basis as well as GaN material

basis.

Investigating of the reliability of the III-V avalanche pulse

(influence of PRF, stress test).

Regarding compactness of radar sensor, a monostatic radar concept

would be desirable.

Appendix

168

Appendix A

Configurations and Parameter Calculation of

Attenuator Circuit

A.1 Attenuator Circuit Configurations [1]

Attenuators are passive resistive circuits which are used to reduce the

power of the measured signal to match the power limitation of the

measurement equipments without introducing distortion. There are three

common types of attenuator used in microwave circuits. These types are:

T-attenuator (T), pi-attenuator (Pi) and bridged-t attenuator. Figure A.1 and

A.2 show the circuit schematics of the attenuators types.

(a)

(b)

Figure A.1 Three common types of attenuator: (a) T-attenuator, and (b) Pi-attenuator.

Appendix

169

Figure A.2 Bridge-t attenuator circuit.

A.2 Calculation of the Attenuator Resistive Elements

The required equations for calculating the resistive elements of attenuators

types are presented in Table I [2].

Table A.1 Resistive Element Equations of Attenuator Types.

Attenuator

type R1 R2 R3

T 2

10

10

110

110RZ inL

L

110

10**2

10

10

L

L

outin ZZ 2

10

10

110

110RZoutL

L

Pi

210

10 1

110

110

1

RZ

L

in

L

10

10

10

*110

2

1L

outin

LZZ

210

10 1

110

110

1

RZ

L

out

L

Bridged t

110 20

0

L

Z 110 20

0

L

Z

0Z

Zin is the input impedance, Zout is the output impedance and Z0 is the

characteristic impedance. The values of the resistive elements of

attenuators types for different values of loss (L) (where Zin = Zout = 50)

are shown in Table II.

Appendix

170

Table A.2 Resistive Element Values of Attenuator Types for Different Loss Values.

Loss (dB)

Attenuator type

T Pi Bridged t

R1 R2 R1 R2 R1 R2

0 0 open open 0 0 open

1 2.9 433.3 869.5 5.8 6.1 409.8

2 5.7 215.2 436.2 11.6 12.9 193.1

3 8.5 141.9 292.4 17.6 20.6 121.2

4 11.3 104.8 221 23.8 29.2 85.5

5 14 82.2 178.5 30.4 38.9 64.2

6 16.6 66.9 150.5 37.4 49.8 50.2

7 19.1 55.8 130.7 44.8 61.9 40.4

8 21.5 47.3 116.1 52.8 75.6 33.1

9 23.8 40.6 105 61.6 90.9 27.5

10 26 35.1 96.2 71.2 108.1 23.1

11 28 30.6 89.2 81.7 127.4 19.6

12 29.9 26.8 83.5 93.2 149.1 16.8

13 31.7 23.6 78.8 106.1 173.3 14.4

14 33.4 20.8 74.9 120.3 200.6 12.5

15 34.9 18.4 71.6 136.1 231.2 10.8

16 36.3 16.3 68.8 153.8 265.5 9.4

17 37.6 14.4 66.4 173.5 304 8.2

18 38.8 12.8 64.4 195.4 347.2 7.2

19 39.9 11.4 62.6 220 395.6 6.3

20 40.9 10.1 61.1 247.5 450 5.6

30 46.9 3.2 53.3 789.8 1531.1 1.6

40 49 1 51 2499.8 4950 0.5

50 49.7 0.3 50.3 7905.6 15761.4 0.2

100 50 0 50 open open 0

Appendix

171

Appendix B

Measurement Results of Transformers

B.1 Balun Transformer (ADT1-1WT+)

(a)

(b)

Figure B.1 Measurement results of transformer ADT1-1WT+. (a) insertion loss, (b)

input reflection coefficient.

Appendix

172

B.2 Transmission Line Transformer TC1-1-43A+

(a)

(b)

Figure B.2 Measurement results of transformer TC1-1-43A+. (a) insertion loss, (b)

input reflection coefficient.

Appendix

173

References

[1] J. Carr, Microwave and Wireless Communications Technology, Butterworth

Heinemann, US, 1996.

[2] G. Ballou, Handbook for Sound Engineers, 3rd

Edition, Elsevier Inc., Oxford,

UK, 2002.

Publication

174

Own Publications

[1] A. Ameri, G. Kompa, and A. Bangert, ‘‘Study about TEM Horn Size Reduction for

Ultra-wideband Radar Application’’, German Microwave Conference (GeMiC), pp. 1-4,

Darmstadt, Germany, March 2011.

[2] A. Ameri, G. Kompa, and A. Bangert, ‘‘650W Pulse Generator for Ultra-Wideband

(UWB) Radar Application’’, German Microwave Conference (GeMiC), pp. 1-4,

Darmstadt, Germany, March 2011.

[3] A. Ameri, G. Kompa, and A. Bangert, ‘‘Balanced Pulse Generator for UWB Radar

Application’’, 8th European Radar Conference (EuRAD), pp. 198-201, Manchester,

United Kingdom, October 2011.



Ahm

ed A

bbas

Hus

sein

Am

eri

Long

-Ran

ge U

ltra-

Wid

eban

d Ra

dar

Sens

or fo

r Ind

ustr

ial A

pplic

atio

ns

ISBN 978-3-86219-442-1

long-range ultra-wideband radar sensor for industrial ... · ahmed abbas hussein ameri kassel,...

Documents