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ECSE 536, RF MICROELECTRONICS, MCGILL, APRIL 2016 1

Analysis of an UWB CMOS Transceiver forMBOA Mode-1 OFDM

Mido Assran, 260505216, and Michael Kleinman, 260529700

Abstract—This paper describes the LNA and mixer in theRX path of a direct-conversion (homodyne) ultra-wideband(UWB) transceiver for Mode 1 OFDM designed in 0.13-µmCMOS technology. The LNA and mixer under study incorporatethree resonant networks and three phase-locked-loops used forfast band hopping in compliance with the MBOA’s Mode 1UWB OFDM standard. The LNA uses a common gate cascodeconfiguration, with band-select inputs, and a 16-dB gain switch.Overall the LNA achieves a gain of 16-18 dB over the threeMode 1 bands; a noise figure of 3.15 dB and consumes 4 mWof power from a 1.5-V supply. The mixer uses a tracking circuitto halve the biasing current flowing through the LO MOSFETS.Simulations predict a voltage conversion gain of 14.4dB and aNoise Figure of 17.6dB with 3.825mW of power consumed.

I. INTRODUCTION

THE origins of Ultra-Wideband (UWB) systems are rootedin their utility as an analytical tool in the characterization

of microwave networks by means of short carrier-free pulsesoperating based on the principles of time-domain electro-magnetics [6]. UWB carrier-free pulses were used early on(∼1960s) for the characterization of the intrinsic properties ofmaterials; their primordial utility transitioned to the analysis ofradiating and receiving elements as experimental technologyadvanced, with recent time-domain applications concentratingon free-space time-domain reflectrometry for BAseband Radar(BAR). The spectral content of UWB carrier-free systems isconcentrated from zero frequency to the microwave region,and provides an apparent insensitivity to incidental jamming -ideal for military applications [6].

Potential for high data rates has sparked recent commercialinterest in UWB systems - their future applications have beenwidely debated; the most probable potential employments forUWB in communication systems is likely to be personalhome-networking; however there has also been some abitiousresearch in utilizing UWB in 5G systems which incorporatevery dense networks of small cells working at non-traditionalcellular spectrum in the mmWave region [3][4][5]. Nonethe-less, with the most likely use of UWB systems being personal-home networking, multi-band OFDM is the likely choice forfuture UWB systems - a spin-off of tradional OFDM orMIMO-OFDM used in the common 802.11 WLANs.

This paper describes the design of an LNA and a mixerused in the RX path of an UWB CMOS transceiver forMBOA Mode-1 OFDM presented by Razavi et. al shown inFigure 1 [1]. Section II presents the constraints set forth bythe aforementioned standard, and discusses briefly how theproposed architecture deals with said constraints. Section IIIand Section IV describe the LNA design and simulations, andthe mixer design and simmulations respectively. Section V

Fig. 1. Transceiver Architecture [1]

concludes with a critical analysis of the designed system, andilluminates areas of future work for potential alternatives tothe paper’s design.

II. MBOA MODE-1 OFDM

The Multi-Band Orthogonal frequency division multiplex-ing Alliance (MBOA) has created a standard utilizing vanillaorthogonal frequency division multiplexing (OFDM) over mul-tiple bands simultaeneously.

A. OFDM for UWB

OFDM is a well-established technique for digital multicar-rier modulation using many closely spaced subcarriers. OFDMdivides a band into many smaller channels, and transmitsdata in parallel along each channel using a low-spectral ratemodulation scheme, namely quadrature phase shift keying(QPSK) in the case of the MBOA’s standard. OFDM lever-ages frequency diversity (one of the many kinds of diversityachievable in communication systems), to provide robustnessto low channel condition numbers (sever channel conditions),hence its widespread popularity in preference to high-spectralrate modulations.

B. MBOA Bands & Channelization

The MBOA standard partitions the spectrum between 3-10GHz into 528 MHz bands, with each band employing regular


Fig. 2. MBOA band structure and channelization [1]

Fig. 3. Local oscillator frequency hopping [2]

OFDM to achieve an aggregate transmission data rate of 480Mbs.

More precisely, the spectrum is divided into 14 bands from3168 MHz to 10560 MHz, with each band further subdividedinto 128 OFDM channels each employing QPSK (permittingthe use of an ADC with low resolution). The band structureand channelization outlined by the MBOA standard are shownin Figure 2.

C. MBOA Standard Mode-1

Mode-1 of the MBOA standard specifies the interleavingof symbols amongst the first three bands shown in Figure2, thereby further exploiting frequency diversity beyond thatprovided by vanilla OFDM [1]. The band hopping improvesthe system’s capabilities of dealing with multipath effects andinterference. In reception, the MBOA standard requires listen-ing to a symbol on one band for ∼312 ns, and then switching tolisten on the subsequent band in less than 9.47ns. Switchingbands in less-than 9.47ns places very critical constraints onthe PLLs used for down conversion of the signal in the RXpath; concequently the principal challenge in designing such asystem is the rapid LO-hopping. Figure 3 illustrates the band-hopping/bit-interleaving procdure required.

D. Transceiver Architecture Band Hopping

The transceiver architecture (Figure 1) addresses the 9.47nsLO-hopping constraint by simply adding three PLL frequency-synthesizes each driving the LO port of one of the three mix-ers; each one of the three mixers performing downconversion

Fig. 4. LNA architecture [1]

in one (static) band. The RF port of each of the three mixers isdriven by a corresponding resonant tank of the LNA (whichhas three resonant tanks in total - one for each band). Thespecific resonant network corresponding to the desired bandis made "selectable" by the addition of a band-select input tothe LNA.

III. LNA

A. Architecture

The LNA architecture to be analyzed, taken from Razaviet. al, is provided in Figure 4. As mentioned in Part D ofSection II, the LNA has three resonant structures, one for eachband. It uses a common gate configuration to provide a 50Ω input impedance for matching with the antenna, and usesthe inductive degeneration to improve linearity, and providea lossless input-matching, but critically, the inductor (mustbe large) provides a high Q, and keeps the input return loss(S11) below -10 dBm. An input return loss below -10 dBmis relatively standard in most designs. The gain switch, whenenabled, turns off transistor M1, decreasing the current sunkin the lower plane of the cascode, and turns on transistor M6to compensate for the decrease in input impedance.

B. Fixed Parameters

Since the executive purpose of this paper is to perform acomparison to the results obtained by Razavi et. al, the sameparameters held fixed in Razavi et. al[1], are also held fixedin our LNA design (shown in Table I).

C. Bias

With 1.5V at the rails, there was not much choice for Vb andthe band select voltage, and so they were chosen reasonablyat 0.7V, and 1.25V respectively (to keep the transistors insaturation). Then to achieve the bias current of 2.5 mA, thewidth of transistor M1 (W1) was swept over a range of values,and plotted versus the DC drain current. Subsequently thetransistor width on the curve was chosen to achieve the desired


TABLE IFIXED LNA DESIGN PARAMETERS

Parameter Value

L (transistor length) 0.13µm

iD (drain current) 2.5mA

W2 W1/8

VDD 1.5V

L1 20nH

Fig. 5. Width of transistor M1 swept and plotted vs drain current (DC)

bias current, in this case, the width was taken to be 27µmas shown in the marker in Figure 5. The widths of M3,M4,and M5 were all chosen to be 1.6 times the value of (W1)in order to ensure sufficient capability to source the currentrequirements of the lower plane.

D. Simulation

The remaining part of the design involved carefully calli-brating the inductive-loads to resonate at the center frequencyof their respective band; to have a large enough gain at thecenter frequency, and to have a low-enough Q to minimizeattenuation at the edges of the band and keep a near-flat gainover the band.

The noise figure, input return loss, gain, and input-impedance of the LNA are shown in Figures 15, 16, 17, and18 respectively - here shown for the first band in the Mode-1MBOA standard. Table IV summarizes our simulation resultsfor the Razavi et. al LNA and our proposed LNA. *All thecalculations (i.e gain, input impedance, ...) are conducted witha 1150 Ω port 2 - output - termination.

E. LNA Discussion

The LNA described in Razavi et. al is recreated and simu-lated to provide a baseline for comparison to our proposed

Fig. 6. Noise figure of the LNA over Band-1 for a 1150Ω termination

Fig. 7. Input return loss of the LNA

Fig. 8. Gain of the LNA over Band-1 for a 1150Ω termination

LNA (shown in Appendix A). Our proposed LNA (samearchitecture), provides less noise figure, an improved inputreturn loss, and a flatter in-band gain at the expense of a gainreduction of 5 dB on average. As will be shown, this typeof LNA is proposed to complement the mixer designed inthe Section IV which will be shown to have a much higher


Fig. 9. Variation of input impedance of the LNA with input frequency

TABLE IILNA SIMULATION RESULTS

Parameter Based on Razavi Proposed

Noise Figure 3.15 dB 2.83dB

S(1,1) -10 dB -18.5 dB

Gain 18 dB 13 dB

Bias 2.5 mA 2.5 mA

Input Impedance 41+j2 : 50-j15 Ω 50-j7 Ω

conversion gain, but also a larger noise figure. The principaladvantage of this design is to flatten the LNA gain over theband, achieving only 0.19 dB of gain variation over a 528MHz band in contrast to the 6 dB gain variation of the baselinemethod shown in Appendix A. The proposed design is alsomuch more optimal for the sharing of the antenna with the TXpath since the return loss is improved by nearly 10 dB, andthe input impedance (much more easily matched) is constantover the band. In contrast the baseline design presented byRazavi et. al requires an input matching network to performoptimally over a 528 MHz band with a frequency variant inputimpedance, a non-trivial task.

Fig. 10. Architecture of a single arm of one mixer [1]

IV. MIXER

A. Architecture

The mixer described in Razavi et. al is a standard single-ended mixer with two unique modifications. First, there is atracking circuit which is used to halve the biasing currentflowing through the LO MOSFETS. This has the advantageof reducing noise figure. In addition, the load consists ofa variable resistance in order to provide variable gain. Thearchitecture of a single arm (there are I and Q arms) of oneof the mixers (there are three in total) is shown in Figure 10.

B. Tracking Circuit Operation

The operating principle of the tracking circuit is as follows:KCL at Node X revealts that 0.25IB flows through 2RH . Thisfixes the voltage at Node X at VDD −0.25IB · 2RH . AssumingM5 and M3 have an equivalent Vgs , the voltage at Node Pshould equal the voltage at Node X. Therefore, the currentflowing from VDD to Node P is equal to VDD− (VDD−0.25IB ·2RH ) = 0.5IB , thereby halving the current in the LO plane.

C. Simulation

Initial simulation attempts using the Razavi et. al mixerarchitecture resulted in inconsistent circuit behaviour whenthe RH was set to 50Ω. The current flowing flow VDD toNode P was not equal to half of the biasing current IB . Uponfurther inspection, a current probe was placed between theLO MOSFETS and Node P in order to measure the biasingcurrent through these MOSFETS, and indirectly measure thecurrent flowing from VDD to Node P. This current was sweptagainst RH , as shown in Figure 11. Obsrvations showed thatthe afformentioned Tracking Circuit Operation only appliedwhen RH was greater than 1.5kΩ, that is, the LO plane currentwas only half of the total sunk bias current when RH wasabove 1.5kΩ.

Additionaly, the design process depicted gross dependenceof the gain and noise figure of the mixer on the width of M1and the value of RH . Concequently, these two aforementionedparameters were the main tool used to achieve our proposedmixer alterations to be discussed in Subsection Comparison ofMixers.


Fig. 11. Net LO plane current plotted versus RH

TABLE IIIMIXER SIMULATION RESULTS

Parameter Based on Razavi Proposed

Noise Figure 16 dB 17.6 dB

Voltage Conversion Gain 10 dB 14.4 dB

Power Consumption 3.75 mW 3.825 mW

Fig. 12. Input voltage at RF for the first band

D. Comparison of Mixers

The critical figures to measure the performance of a mixerare the gain, noise figure, and power consumption. The com-parison of these simulation values from Razavi et Al and oursimulation are shown in Table III.

The power consumption of the mixer (3.825 mW) corre-sponds to a VDD of 1.5 V. Plots of the fundamental inputvoltage (at RF) and the corresponding fundamental outputvoltage (at IF) are shown in Figures 12 and 13(similar analysisis performed in the other two bands). Lastly the noise figureof the mixer is shown in Figure 14.

Fig. 13. Output voltage at IF for the second band

Fig. 14. Noise figure of the mixer

E. Discussion

Since the LNA sacrificed gain in order to improve noisefigure, input matching, and gain consistency over the band, itwas desired that the mixer have improved gain to compensate.Our simulations show that gain was able to be significantlyimproved by sacrificing 1.5dB of noise figure.

V. CONCLUSION

The LNA and mixer presented by Razavi et Al. in [1] wereanalyzed and simulated in concert (as is typically the casewith modern mixer design). Subsequent modifications weremade, and new designs were proposed while maintaining thecurrent architecture of both building blocks. The new designof the LNA still maintained three resonant structures, butincorporated an improved input return loss, an improved noisefigure, improved gain concsistency over each band, and a moreeasily matched input-impedance, at the expense of a lower


gain. The lower gain was compensated for by the alteration ofthe single-balanced to achieve higher gain at the expense of ahigher noise figure. The mixer design still maintained all thesame architectural components as that presented in Razavi etAl. (most noteably the tracking circuit). ’

For future work, it is suggested to explore the architecturaldesigns of UWB CMOS transceivers using only one resonant-network in the LNA and only one mixer. This would improvethe noise figure of the LNA by reducing the capacitance atthe intersection of the upper and lower-planes in a cascodeconfiguration (thereby decreasing the effect of the input-reffered noise at the band-select transistors). Such a designwould not only improve performance, but also reduce chiparea. Such a design would also require new and improvedfrequency synthesizers capable of fast switching.

APPENDIXLNA REFERENCE: SIMULATION OF THE LNA DESIGN

PRESENTED IN Razavi et. al [1]

Fig. 15. Noise figure of the LNA over Band-1 for a 100Ω termination

Fig. 16. Input return loss of the LNA

Fig. 17. Gain of the LNA over Band-1 for a 100Ω termination

Fig. 18. Variation of input impedance of the LNA with input frequency

TABLE IVLNA SIMULATION RESULTS

Parameter Value

Noise Figure 3.15 dB

S(1,1) -10.5 dB

Gain 18 dB

Bias 2.5 mA

Input Impedance 45 + j0 Ω

REFERENCES

[1] B. Razavi et al., "A UWB CMOS transceiver", Journal of Solid-StateCircuits, vol. 40, pp.2555-2562, Dec. 2005

[2] R. Van De Beek, D. Leenaerts and G. Van Der Weide, "A Fast-HoppingSingle-PLL 3-Band MB-OFDM UWB Synthesizer", IEEE J. Solid-StateCircuits, vol. 41, no. 7, pp. 1522-1529, 2006.

[3] "5G ultra-wideband enhanced local area systems at millimeter wave


| Nokia", Nokia Solutions and Networks, 2016. [Online]. Available:http://networks.nokia.com/news-events/insight-newsletter/articles/5g-ultra-wideband-enhanced-local-area-systems-at-millimeter-wave.[Accessed: 01- Apr- 2016].

[4] "Bridging the spectrum gap with 5G | Nokia", Nokia Solutions andNetworks, 2016. [Online]. Available: http://networks.nokia.com/news-events/insight-newsletter/articles/bridging-the-spectrum-gap-with-5g.[Accessed: 04- Apr- 2016].

[5] M. Ghavami, L. Michael and R. Kohno, Ultra wideband signals andsystems in communication engineering. Chichester: Wiley, 2007.

[6] C. L. Bennet and G. F. Ross, "Time-domain electromagnetics and itsapplications," Proc. IEEE, vol. 66, no. 3, pp. 299-318, Mar. 1978.

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