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FACULTY OF ENGINEERING AND SUSTAINABLE DEVELOPMENT .
Design and Development of Gigahertz Range VCO Based
on Intrinsically Tunable Film Bulk Acoustic Resonator
Danial Tayari
June 2012
Master’s Thesis in Electronics
Master’s Program in Electronics/Telecommunications
Examiner: Prof. Daniel Rönnow
Supervisor: Prof. Spartak Gevorgian
i
PREFACE
The Master‟s thesis is the outcome of my research at Terahertz and millimeter wave laboratory
of Chalmers University of Technology, Sweden, for the Master‟s program in Electronics/
Telecommunication Engineering at University of Gävle, Sweden.
Professor Spartak Gevorgian at Chalmers University of Technology supervised me during the thesis.
The thesis is examined by Professor Daniel Rönnow at University of Gävle.
The main focus of the thesis is on design and fabrication of voltage controlled oscillators by the
use of tunable film bulk acoustic resonator. The design was done in Advanced design system
(ADS) and fabrication was performed at Chalmers clean room at the department of Micro
technology and Nano Science, MC2.
ii
Abstract
The purpose of this thesis is to des ign and fabricate Gigahertz range voltage controlled oscillator
based on intrinsically tunable film bulk acoustic resonator.
Modified Butterworth Van Dyke (MBVD) model was studied and implemented to simulate
FBAR behavior. Advanced designed system (ADS) was used as the simulation tool.
Oscillator theory is studied and an oscillator based on non-tunable FBAR at 2GHz is simulated
which shows -132 dBc/Hz phase noise @ 100 kHz offset frequency.
A 5.5 GHz Voltage controlled oscillator based on intrinsically tunable FBAR is designed.
Frequency tuning of 129 MHz with phase noise of -106 dBc/Hz @ 100 kHz is achieved. The
circuit is designed on a novel carrier substrate which includes integrated resonators and passive
components. Bipolar junction transistors are mounted on the carrier substrate by silver epoxy.
The thesis describes the design, development and processing of the carrier substrate, BSTO
based resonators, and the oscillator circuit.
iii
Acknowledgments
I would like to thank Professor Spartak Gevorgian for considering me as a member of his
research group.His constant supervision during the thesis period guided me through the right
pass to achieve my goal.
Dr.John Berge who helped me in the fabrication process of the VCO,without his help the
fabrication wouldn‟t have been done in the proposed time.
My special thanks to Professor Daniel Rönnow for accepting to be the examiner of my work.
To my friends at Gävle and Göteborg who always helped and supported me during my stay in
Sweden.
Finally I am thankful to my family for encouraging and supporting me in life.
iv
Notations and Abbreviations
Notations
C Capacitance
Cm Motional capacitance
foff , ωoff Offset (angular) frequency
g Coplanar wave guide gap width
kt2 (effective) piezoelectric coupling coefficient
Phase-noise relative to carrier at offset frequency
Lm Motional inductance
PDC DC power
Qp Parallel resonance quality factor
Qs Series resonance quality factor
Rm Motional resistance
s Coplanar wave guide strip width
Zo Characteristics impedance
𝜀eff Effective permittivity
α Attenuation constant
φ Acoustic phase
v
Abbreviations
AC Alternating Current
AIN Aluminum Nitride
BAW Bulk Acoustic Resonator
BJT Bipolar Junction Transistor
BSTO Barium Strontium Titanate
CPW Coplanar Waveguide
DC Direct Current
FBAR Film Bulk Acoustic Resonator
FOM Figure of Merit
GSG Ground Signal Ground
HFO Hafnium Oxide
IF Intermediate Frequency
LO Local Oscillator
MBVD Modified Butterworth Van-Dyke
PLD Pulsed Laser Deposition
PN Phase Noise
RF Radio Frequency
SAW Surface Acoustic Wave
VCO Voltage Controlled Oscillator
ZnO Zinc Oxide
vi
Contents 1 Introduction ..................................................................................................................... 1
1.1 Introduction and Motivation .......................................................................................... 1
2 Film bulk acoustic resonators............................................................................................. 3
2.1 Characteristics ............................................................................................................. 6
2.1.1 Quality factor ....................................................................................................... 6
2.1.2 Effective Coupling coefficient................................................................................. 6
2.2 Modeling .................................................................................................................... 7
2.3 Tunable FBAR ............................................................................................................ 8
3 Oscillator theory ..............................................................................................................11
3.1 Types of oscillators .....................................................................................................11
3.1.1 Relaxation oscillators ...........................................................................................11
3.1.2 Harmonic oscillators.............................................................................................12
3.2 Oscillation criteria.......................................................................................................12
3.2.1 Reflection oscillator .............................................................................................14
3.2.2 Transistor oscillator ..............................................................................................15
3.3 Oscillator Phase noise ..................................................................................................16
3.4 Oscillator figure of merit ..............................................................................................17
4 FBAR Oscillators .............................................................................................................18
4.1 Fixed frequency FBAR Oscillator ..................................................................................19
4.2 Voltage controlled oscillator (VCO) Based on Non-tunable FBARs. ...................................22
5 Tunable FBAR VCO ........................................................................................................25
5.1 Design.......................................................................................................................25
5.1.1 ADS momentum design ........................................................................................26
5.1.2 Substrate definition ..............................................................................................26
5.1.3 Coplanar waveguide .............................................................................................27
5.1.4 Grounding capacitors............................................................................................29
5.1.5 Tunable FBAR resonator.......................................................................................31
5.1.6 Decoupling capacitor ............................................................................................32
5.1.7 Meandered circuit ................................................................................................33
5.1.8 Co-Simulation .....................................................................................................34
6 Device Fabrication and Measurement ................................................................................37
6.1 BSTO film growth by the PLD......................................................................................38
vii
6.1.1 PLD overview .....................................................................................................38
6.1.2 Laser-target interaction. ........................................................................................39
6.1.3 The plume...........................................................................................................39
6.1.4 Pulsed laser .........................................................................................................39
6.2 Measurements ...........................................................................................................42
6.2.1 Test resonator measurement...................................................................................42
6.2.2 Oscillator measurements .......................................................................................44
7 Conclusion and future work ..............................................................................................45
8 References .......................................................................................................................46
Appendix1: Tunable FBAR Resonator MBVD model Extracted Parameters ...............................49
Appendix2: Transmission line parameter extraction ..................................................................50
Appendix3: Fabrication Steps and recipe ..................................................................................53
1
1 Introduction
The motivation for the work presented in the thesis and thesis organization are provided in this
chapter.
1.1 Introduction and Motivation
Oscillators are devices which create a periodic AC signal at a defined frequency. All RF and
microwave devices containing transmitters or receivers use oscillators. Usually oscillators are
deployed in receivers and transmitters with mixers in which the signal can be up or down
converted in frequency. In contrast to fixed frequency oscillators, - the voltage control oscillators
(VCOs) can produce a range of frequencies enabling the radio device using them to operate on
many different frequencies.
Communication systems frequency range can be determined by the frequency range of the
oscillator. So this frequency range is desirable to be large enough for covering the operating
communication band.
In today‟s communication systems, staying within the allocated frequency band and not
disturbing other users of adjacent frequencies is very important. Therefore the frequency stability
of the devices should be very high. In case of oscillators the frequency stability is defined by its
phase noise and it is a critical issue since it determines the frequency stability of the complete
radio transceiver.
There are number of parameters affecting the oscillator phase noise. One of the most important
one is the reactive components quality factors. In LC oscillators reactive components such as
inductors and capacitors are employed in the resonator part .The LC resonator defines the
oscillation frequency and the quality factor of the resonator has significant impact on oscillator
phase noise. So the concern is to have resonators with higher quality factor. LC resonator quality
factor is generally very low (around 20) making it a challenge for designers to design low phase
noise oscillators based on LC resonators. However designing broad frequency range oscillators is
easier when using these low quality factor components.
To have high quality resonators, surface acoustic wave (SAW) and bulk acoustic wave (BAW)
devices can be used. The film bulk acoustic resonator (FBAR) is a type of recently developed
BAW devices with very high quality factor.
The main focus of this thesis work is to present an oscillator based on quite recently developed
tunable FBAR, based on BSTO material in which presents tunable characteristics under dc bias.
2
These types of FBARs make it possible for the oscillator to have relatively high tuning range
compared to traditional AlN FBAR oscillators and are suitable in wireless communication
systems and sensor applications, e.g. bio sensing.
The contents of this thesis are organized as follows.
Chapter 2 discusses the film bulk acoustic resonator, its characteristics and modeling. Tunable
FBARs are also briefly introduced in this chapter. Chapter 3 gives some background on
oscillation theory and important criteria for oscillators. Design of fixed frequency FBAR
oscillators is presented in chapter 4 while chapter 5 gives detailed explanation of a VCO based
on tunable FBAR. Chapter 6 argues about the fabrication process and presents the measurement
results. The conclusions and future work are given in the final chapter.
3
2 Film bulk acoustic resonators
Film bulk acoustic resonator (FBAR) consists of a piezoelectric thin film which is sandwiched
by a pair of electrodes. Depending on the applied frequency of the electrical signal to the FBAR;
its piezoelectric material may expand or contract resulting in generation of the acoustic waves.
When the thickness of the piezoelectric layer is the same as an integer of half acoustic
wavelength the resonance occurs which means the resonance frequencies are determined by
thickness and are independent from lateral dimensions.
At some frequency the impedance of FBAR reaches its minimum magnitude, which means the
generated acoustic wave travels in the most efficient way through the physical material. This
frequency is referred to the series resonant frequency fs . On the other hand when FBAR
impedance magnitude reaches its maximum there is no response from the piezoelectric meaning
that no acoustic wave transfers energy through the FBAR. This happens at parallel resonance
frequency or anti resonance frequency fp [1].
Fig. 2.1 FBAR structure under bias voltage
Piezo-electric material
4
Fig. 2.2 illustrates the relation between the FBAR Impedance and its resonance frequencies
In comparison to LC resonators FBAR offers very high quality factor. In recent years FBARs
using different material with very high quality factors of more than 2000 at a few gigahertzes
have been introduced and FBAR duplexer filters for mobile phones now are commercially in
use. Integrated LC resonators with varactor typically have Q factor of around 20.
In order to have high quality factor of the FBAR, the resonator must be acoustically isolated
from the substrate. Depending on the Type of isolation FBARs are categorized as solidly
mounted or membrane mounted, Fig. 2.3.
The first type uses an acoustic reflector, a Bragg reflector, consisting of λ/4 layers with high and
low acoustic impedance alternatively. The second type based on an air cavity formed below the
bottom electrode.
Series resonance
Parallel resonance
Fig. 2.2 A typical FBAR frequency response. Parallel and series resonances are shown
5
(a) (b)
Fig 2.3(a) solidly mounted and (b)membrane mounted FBAR
Dimensions of FBAR are relative to acoustic wavelength. Acoustic propagation speed in solid
materials is about 103_10
4 m/s. For e lectromagnetic waves however it is of order 10
7_10
8 m/s. So
acoustic wavelength is four to five order of the magnitude lower than electrical wavelength,
consequently FBARs are much smaller than electromagnetic resonators based on transmission
line segments, for example. In addition acoustic loss is fairly low for piezoelectric materials at
gigahertz range making them useful for high quality factor resonators at those frequencies. For
example AlN_FBARs with quality factor of 280 demonstrated at 20 GHz at [2]. Some reported
FBARs are given in Table 2.1.
Table 2.1: some recently reported FBARS
Reference fs[GHz] Qs Size[mm2]
[3] 1.1 386 0.058
[4] 1.9 832 -
[5] 1.9 1200 0.01
[6] 5 290 -
[7] 1.9 1025 -
[8] 4.9 300 -
[9] 1.8 8000 -
Air
Top-electrode
Piezo-electric material
Silicon
bottom-electrode
Top-electrode
Piezo-electric material
Silicon
bottom-electrode
Z1
Z2
Z2
Z1
6
2.1 Characteristics
The important characteristics of FBAR relevant for Oscillators are the Q-factor (quality factor)
and the resonance frequencies. Regarding these, the coupling coefficient is important in that the
quality factors and impedance response depend on it.
2.1.1 Quality factor
For a resonator the quality factor is defined as
(2.1)
Where the maximum Energy stored in the resonator and is the energy dissipated in
the lossy sections in one resonance period.
In simple resonators Q is commonly obtained from 3_dB bandwidth of the impedance. In
FBARs however, the Q can be determined by the equation given in [10].
|
| (2.2)
2.1.2 Effective Coupling coefficient
The coupling coefficient shows the percentage of the energy converted from mechanical to
electrical and vice versa. [11]
(2.3)
7
2.2 Modeling
One of the most commonly used models for FBAR is the modified Butterworth-Van Dyke
(MBVD) model. [12] Which is shown in Fig. 2.4
Rm Cm Lm
Rs
R0 C0
Fig. 2.4 MBVD model
In this model C0 represents the parallel plate capacitance, Cm, Lm, and Rm show the acoustic
resonance, Rs represents the ohmic loss of the electrodes while R0 defines the dielectric loss.
Typically the MBVD model is extracted from the measurements and used as a model in circuit
simulation.
The measured reflection coefficient of an FBAR resonator and its equivalent MBVD model is
plotted in Fig. 2.5. The parameters of MBVD model should be tuned in a way that simulated and
measured plots fit each other excellently.
Fig. 2.5 measured and equivalent MBVD model reflection coefficient
8
2.3 Tunable FBAR
Traditional FBARs are not tunable, however in case of intrinsically tunable FBAR a DC voltage
is used to tune the resonance frequency of FBAR [13]. A tunable FBAR can be obtained from
the DC field dependency of dielectric constant, acoustic velocity and electromechanical coupling
coefficient
As an example of tunable FBAR, presented in [14] is shown in Fig. 2.6(a). For this FBAR,
BSTO as the ferroelectric material and HfO2 and SiO2 as the layers of the Bragg reflector are
used.
As the bias voltage increases the resonance loop grows. For comparison Fig. 2.6(b) represent a
plot of traditional FBAR. Although the resonance frequencies are not the same, we can see the
difference between these two. For a given resonance frequency and capacitance, the resonance
loop size is governed by the effective coupling coefficient and acoustic quality factor.
This shows that FBAR Based on AlN gives much higher quality factor than the BSTO tunable
one.
Table 2.2 gives the resonance frequencies of the resonator for different bias voltages. The
resonator is then connected to the rest of the circuit.
(a) (b) Fig. 2.6 (a) Tunable FBAR resonating from 5.5GHz to 5.7 GHz (b)ALN non tunable FBAR resonating at 2 GHz
9
„Table 2.2: Extracted resonance frequencies from resonator MBVD model
Bias voltage(V) fs(GHz) fp(GHz)
5 5.729 5.752
10 5.658 5.743
15 5.612 5.722
20 5.58 5.733
25 5.553 5.712
Extracted parameters of the MBVD model are given in Figure 2.7.
(a)
Motional Inductance vs. DC Bias
0
2
4
6
8
10
12
0 5 10 15 20 25 30
Voltage(V)
Mo
tio
nal
Ind
ucta
nce(n
H)
10
(b)
(c)
Fig. 2.7 Extracted MBVD model parameters vs. bias voltage (a) motional inductance L m (b) motional resistance Rm (c) motional
and static capacitance Cm and C0
Motional Resistance vs. DC Bias
1.5
1.6
1.7
1.8
1.9
2
2.1
2.2
2.3
2.4
2.5
0 5 10 15 20 25 30
Voltage(V)
Mo
tio
nal
Resi
stan
ce (
Oh
ms)
Motional & Static capacitance vs. DC Bias
0
1
2
3
4
5
6
0 5 10 15 20 25 30
Voltage(V)
C0
(pF
)
0
20
40
60
80
100
120
140
160
Cm
(fF
)
C0(pF)
Cm(fF)
11
3 Oscillator theory
An oscillator is a circuit which uses DC power to generate periodic AC signal at its output. In an
oscillator there is no need for any input signal except for the DC power supply. This chapter
discusses briefly the two main types of oscillators, criteria to have the oscillation and the
important merits to evaluate the oscillator performance. More explanations about working
principle of the oscillators can be found in [15].
3.1 Types of oscillators
Oscillators are divided into two main groups, relaxation oscillators and harmonic Oscillators.
3.1.1 Relaxation oscillators
These oscillators switch repetitively between two states e.g. charging or discharging a capacitor
or inductor. The amplitude of the charging current and the time constant determines the
frequency of oscillation.
A simple relaxation oscillator is shown in Fig. 3.1
Fig. 3.1 Schematic of a relaxation oscillator [15]
12
3.1.2 Harmonic oscillators
A harmonic oscillator consists of a resonator and an active device. The active device cancels the
losses in the resonator, resulting constant oscillation amplitude at the frequency defined by the
resonator.
In the oscillators the resonator is made of an inductor and a capacitor. The resonance
frequency of a circuit is dependent on the values of the inductor and capacitor and defined as
√
Where is the value of the inductor and is the value of the capacitor.
For microwave frequencies, harmonic oscillators are preferred due to better phase noise
performance. The oscillators in this work are of harmonic type.
3.2 Oscillation criteria
Harmonic oscillators use positive feedback. If a transistor is sufficiently biased it can provide
enough feedback from the output for oscillation. This type of design is known as reflection
oscillation which is explained later in this chapter. A schematic of a feedback oscillator is shown
in Fig. 3.2
An amplifier is shown by block α while β represents feedback block connecting output to the
input. A small input signal is used to understand the oscillation.
α
β
𝑉𝑜𝑢𝑡 𝛿𝑣
Fig. 3.2 feedback oscillator schematic
13
The output signal is having a transfer function given in (3.2)
(3.2)
Where α is the forward gain and a function of the amplitude of the signal, while dependence of
the feedback β, is typically on the signal frequency .The open-loop gain is identified as
(3.3)
Analyzing small-signal closed loop transfer function can be done to see if the circuit has
necessary condition for oscillation. It is important that the function has a pair of poles in the
right–hand plane (RHP), or
Number of poles in the RHP + Number of poles in LHP > 0
Identifying the closed-loop transfer function and its poles is usually a difficult task. Instead small
signal open-loop gain can be analyzed. To do this in simulation one can break open the
circuit appropriately for open loop analyses.
The Nyquist criterion represents an oscillation criterion which is based on the open-loop gain
analyses. According to this criterion when the small signal loop gain encircles the point 1+j 0 in
the clock wise direction with increasing frequency, the closed-loop system is unstable.
An example of the Nyquist plot is shown in Fig. 3.3
Fig. 3.3 (a)Nyquist plot for a circuit showing instability (b) Magnitude and phase of the open loop gain
14
The oscillation is guaranteed by Nyquist criterion, but since this criterion is based on the small-
signal loop gains, no information regarding steady state frequency and amplitude
oscillation can be obtained from it.
To have stable oscillation, loop gain must decrease and eventually be equal to 1+j 0.This is
known as Barkhusen criterion [16]
In practice after the oscillation has been started the magnitude of the loop gain, due to the device
non-linearity will be reduced to 1 for stable amplitude.
Having a zero phase in the loop gain means all the signals are summed together, producing a sum
that is greater than any of the single signals. If they were in opposite phase for example, they
would have cancelled out, resulting in no oscillation.
3.2.1 Reflection oscillator
Reflection oscillators are common topology for microwave oscillators since the necessary
feedback to make the circuit unstable can be provided by parasitic elements of the amplifier.
Fig. 3.4 shows RF-circuit for one port negative-resistance oscillator.
𝛤𝑖𝑛 𝑅𝑖𝑛− 𝑍0 + 𝑗𝑋𝑖𝑛 𝑅𝑖𝑛+ 𝑍0 + 𝑗𝑋𝑖𝑛
𝛤𝐿 𝑅𝐿− 𝑍0 + 𝑗𝑋𝐿 𝑅𝐿+ 𝑍0 + 𝑗𝑋𝐿
𝑋𝑖𝑛
𝑅𝑖𝑛
𝑋𝐿
𝑅𝐿
Fig.3.4 one port negative-resistance oscillator
15
Where Zin = Rin+ jXin is the input impedance of the active device and ZL = RL+ jXL is the
impedance of the passive load.
For the oscillation to occur the following conditions must be satisfied:
+
+
For a passive load , indicating . So the negative resistance refers to an energy
source. When the magnitude of the signal which causes the negative resistance and the loss of
the passive component (resonator) are balanced, steady state oscillation happens.
3.2.2 Transistor oscillator
In this type of oscillator the negative resistance is provided by terminating a potentially unstable
transistor. The circuit model is shown in Fig. 3.5.
Negative resistance
Transistor
Terminating
Network
Load
Network
Fig. 3.5 Oscillator circuit model of negative resistance topology
16
3.3 Oscillator Phase noise
Phase noise is an important parameter for performance of an oscillator and is a way to qualify
frequency stability. Short term random fluctuations in the output signal are referred as phase
noise. For a given signal as
0 +
both the amplitude and phase are constant but in realistic oscillators either external or
internal noise-sources to the oscillator cause both quantities to have fluctuations.
Output spectrum of a realistic oscillator is given in Fig. 3.6
Amplitude variation is limited and due to circuit non-linearities for steady state oscillation has
less impact on the oscillator performance. Phase variation on the other hand may be random and
discrete- making it the main contributor in oscillation-noise.
In communication systems it is important for devices to stay in their defined operating frequency
band, thus phase noise plays an important role since the frequency stability of the total system is
Random phase variations
Discrete spurious signals
frequency
f0
Amplitude
Fig. 3.6 Output spectrum of RF oscillator
17
determined by the frequency stability of the oscillator. For instance a local oscillator may cause
channel interference due to its high phase noise when used in a down converter as shown in
Fig. 3.7.
Power
Frequency
Oscillator phase noise is defined as the ratio of the power of a side band to the power of the
carrier frequency at an offset frequency from the carrier. Typically the side band is
normalized by the unit bandwidth.
( ) ( )
0
⁄
3.4 Oscillator figure of merit
Figure of merit (FOM) is used to rank oscillators. The most common FOM is given by (3.8)
− +
−
(3.8)
Where is the phase noise in dBc/Hz , is the offset frequency , 0 is the oscillation
frequency and PDC is the transistor DC power consumption in milliwatt.
Interference wanted and adjacent channels
fIF fLO
Fig. 3.7 channel interference caused by noisy local oscillator
18
4 FBAR Oscillators
The small size and high quality factor of FBAR resonators make them to be of interest in
microwave oscillators.
Table 4.1 shows some publications of Oscillators based on FBAR.
Table 4.1: Summary of FBAR.based Oscillators
year Ref FBAR1
Resonator
integration
Technology f0
[GHz]
Pout
[dBm]
PDC
[mW]
PN
[dBc/Hz]
FOM
[dB]
1984 [17] ZnO,mm Mounted on
pcb
0.259 -24 -66@100 kHz
2001 [18] ZnO,mm Mounted on
pcb
AlGaAS
HBT
2.2 -108@100kHz
2003 [19] AlN,mm Mounted on
pcb
bipolar 1.985 10 -112@ 10 kHz 197
2005 [20] ZnO,mm Mounted on
pcb
1.1 -13.6 -115@ 10 kHz
2008 [21] sm Mounted on
pcb
130 nm
CMOS
2.2 6 -136@1 MHz 195
2008 [22] ALN,sm Mounted on
pcb
65 nm CMOS 22
0.062
-124@100kHz2
222
2008 [23] sm Mounted on
pcb
65 nm CMOS 22
0.92
-128@100kHz2
214
2009 [24] ALN,sm Mounted on
pcb
BiCMOS 2.11 21.6 -136@100 kHz 209
2011 [25] ALN,mm Mounted on
pcb
0.18 µm
CMOS
2 0.022 -121@100kHz 222.9
1mm=membrane mounted, sm=solidly mounted
2simulated
19
4.1 Fixed frequency FBAR Oscillator
The topology used in this work to design a fixed frequency FBAR oscillator is shown in Fig. 4.1.
A commercial BJT transistor (Infineon BFP-420) on common-base topology is used and the
circuit is designed as negative resistance Oscillator so that the transistor parasitic provides
necessary feedback required.
ZR Za
Fig.4.1 common base topology
For the circuit to oscillate the oscillation condition ZR+Za=0 must be satisfied. This can be
achieved by designing a proper output matching network.The resonator is placed at the emitter
port via a matching network. The resonator implemented in the circuit is of AlN solidly mounted
type presented in [26].
The equivalent MBVD model of the resonator is given in Fig 4.2, while Table 4.2 summarizes
corresponding parameters, - the series and parallel resonance frequencies, as well as the Q
factors.
Rm Cm Lm
Rs
R0 C0
Fig. 4.2 2 GHz FBAR MBVD model
Table 4.2: 2 GHz FBAR extracted parameters [26]
Rm= 1.4 Cm=72.2 fF Lm=83.9 nH Rs=2.8 R0=0.5 C0=3.7 pF
fs=2.045 GHz fp=2.065 GHz Qs=750 Qp=250
Resonator
output
matching
network
Resonator
matching
network
20
As it is discussed in chapter 2 FBAR shows much lower impedance at the series resonance than
of the parallel one making it easier to design the matching network for compensating the losses
of the resonator. Regarding this the Oscillator is designed functioning at the series resonance of
the resonator. The circuit is based on 0.6 mm Alumina substrate and microstrip transmission
lines are used as matching networks.
Simulation is done using Agilent ADS software. Large signal model provided by the
manufacturer is used to characterize the transistor, which is biased at Vcc=4 V and Ic=20 mA The
output matching is designed to create an impedance which cancels out the resonator losses so
there is no need for extra matching network at the resonator side.
Quarterwave transmission lines and capacitors are used as RF chokes, Harmonic balance is used
for large signal analysis of the steady state oscillation and finally the procedure is completed by
fine tuning the circuit. Fig. 4.3 shows the ADS schematic of the circuit.
Fig. 4.3 ADS schematic of the designed 2 GHz fixed frequency FBAR Oscillator
Fig. 4.4(a) shows the output spectrum of the circuit. It can be seen that the circuit oscillates at
2.044 GHz which agrees with the resonator series resonance frequency.
The Oscillator phase noise is given in Fig. 4.4(b). At 100 kHz offset from the fundamental, the
oscillator has a phase noise of -132 dBc/Hz which is better than the one reported in[22].
The output waveform is given Fig. 4.3(c) in two periods.
FBAR model
21
Fig. 4.3 2 GHZ FBAR Oscillator (a) output spectrum (b)phase noise plot
(c) output voltage in time domain
(a) (b)
(c)
22
4.2 Voltage controlled oscillator (VCO) Based on Non-tunable FBARs.
In LC oscillators frequency tuning can usually be achieved by directly connecting a control
voltage at the varactor part of the resonator. In AlN FBAR oscillators however this method
doesn‟t give frequency tuning range higher than 1 MHz. Some of the reported VCOs based on
non-tunable FBAR are listed in Table 4.3.
Table 4.3: comparisons of non-tunable FBAR VCOs
Reference f0(GHz) Power Tuning
range[MHz]
Best
PN@1MHz[dBc/Hz]
FOM[dB] Technology
[20] 1.1 - 0.2 -123 - Discrete
[19] 2 3.3V/35
mA
2.5 -150 -195 Bipolar
[27] 2.1 2.4V/24.
3mA
37 -144 -193 0.25µm
BiCMOS+
aboveIC
[28] 2 67µw 10 -149 -220 0.13µmCM
OS
Fig. 4.4 shows a colpitts based PCB oscillator, with the FBAR wire bonded to the circuit. This
topology is used in [20] and by applying a DC control voltage a very small frequency tuning
range is achieved.
Vtune Vdd
FBAR
Fig. 4.4.Colpitts based Oscillator [20]
23
The VCO in [19] is based on common-collector topology. The FBAR is wire bonded to the
Oscillator and a varactor is coupled to the FBAR to reach the tunability of 2.5 MHz at 2 GHz.
Fig. 4.5 FBAR VCO with tuning varactor [19]
The circuit diagrams in [27] and [28] are given Fig. 4.6 and Fig. 4.7.
Fig. 4.6 circuit diagram of the series resonance FBAR VCO core with a single ended output buffer [27]
24
The parasitic, non-tunable, reactive elements of the circuit always reduce the frequency
tunability. By comparing the works that have been done until now it can be understood that to
have more tuning range of the FBAR the circuit topology gets more complicated and yet the
tuning range is quite low compared to LC VCOs.
Due to limitations in fabrication process, in this work the purpose was to design a high tuning
range FBAR VCO keeping the circuit topology as simple as possible by using only one
transistor. Chapter 5 discusses the design of a voltage control oscillator based on intrinsically
tunable FBARs.
Fig. 4.7 FBAR-based differential Colpitts oscillator with gate-to-source feedback gain boosting [28]
25
5 Tunable FBAR VCO
As it is mentioned in section [2.3] tuning of an FBAR is possible by having ferroelectric material
like BSTO as the piezoelectric of the FBAR resonator. Oscillators based on tunable FBARs are
not studied previously and this work presents VCOs using tunable FBARS for the first time.
5.1 Design
In this section an integrated VCO based on Tunable FBAR is presented. The design and
simulation are done in Agilent ADS software and the final mask is prepared for fabrication
process.
A single –transistor topology is chosen in order to reduce the fabrication complexity. The tunable
FBAR is located on the emitter of the transistor as shown in Fig. 5.1. The oscillator frequency is
defined by the series resonance frequency of the device which is tunable according to the applied
bias voltage VR through the inductor L.
Fig. 5.1 Tunable FBAR Oscillator circuit Schematics
Decoupling capacitor C1 is used to isolate the resonator from the circuit in DC. An open stub
matching network at the output and inductive stub at the resonator ports make the compensation
for the resonator loss. Stubs S1 and S2 are quarter waves and AC shorted by capacitors C2 and
C3The output is taken from the collector by a 50 load. Coplanar wave guide which.is used for
the stubs and matching network and the transistor is the same as in section [4.1] (base, two
emitters and collector).One of the emitters connects to the FBAR via S3 and the other is used for
dc-bias through stub S1.
S3 Output
S1 S2
C1
TFBAR
R
VR
L
VEE Vcc C3 C2
26
5.1.1 ADS momentum design
To do electromagnetic simulation of the circuit, the design is done in ADS momentum.
Each part of the circuit is separately simulated and finally Co-simulation is done to analyze the
total circuit including DC analyses. The following section describes the design in momentum.
5.1.2 Substrate definition
The substrate used in this work was presented in [14] to achieve a tunable FBAR. Fig. 5.2 shows
the substrate layers with the corresponding thicknesses. The circuit is designed using 0.5µm
thick gold layer on top of the BSTO layer. For the resonator and circuit fabrication the metal
layers are patterned according the design as explained in sections [5.1.5-5.1.8].
Fig.5.2 Schematic view of the substrate cross section[14]
Ti/Al 10/100nm
Ba0.25Sr0.75TiO2 234nm
Silicon
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Ti/Tio2/Pt 20/25/100nm
27
5.1.3 Coplanar waveguide
The conventional coplanar waveguide (CPW) consists of conductors on top of a dielectric surface
[29]. The two ground planes are separated from the center strip by the gap as shown in Fig. 5.3.
W
The thickness and permittivity of the substrate, the dimensions of the center strip and the gap
width determine the characteristic impedance (Z0), the attenuation constant and the effective
dielectric constant (𝜀eff) of the CPW. [29].
Using CPW simplifies the fabrication, eliminates the need for via holes and reduces radiation
loss [30]. In CPW characteristic impedance is determined not only by the strip but also the slot
width, making possible to reduce the size without limit. The only drawback is higher losses[31].
To make low Z0 in conventional CPW, a very wide center strip conductor and a very narrow slot
width can be fabricated. This however shows high current density at the slot edges which
increases conductor losses; moreover a wide strip conductor can potentially couple power from
the dominant CPW mode to unwanted spurious propagation modes. Therefore it is not
recommended to have conventional CPW lines with Z0 less than 30 [29].
Fig. 5.3 shows the CPW used in this circuit design in ADS momentum.
Fig. 5.3 CPW line
S g
28
Fig. 5.3 CPW in ADS momentum
In this design, GSG ports are defined. Ports1 and 2 are defined as signal ports, ports 3 and 5 are
ground reference ports associated with port 1 and ports 4 and 6 are ground reference associated
with port 2.Table 5.1 gives the line parameters extracted from the software by keeping the center
strip width constant.
Table 5.1: Extracted CPW parameters from ADS momentum at 5.421 GHz
Z0(𝜴) 𝜀eff α (dB/mm) g(µm) S(µm)
41.677 8.216 0.146 5 100
47 7.422 0.101 25 100
48.85 7.273 0.091 35 100
49.4 7.217 0.087 40 100
49.8 7.169 0.083 45 100
50 7.144 0.081 48 100
51.7 7.095 0.078 55 100
53 7.037 0.073 65 100
54.1 6.99 0.069 75 100
29
It can be seen that the strip and gap widths of 100 and 48 µm respectively, determined the CPW
characteristic impedance of the 50 𝜴 which was later used in the circuit design.
Using this data in the general transmission line model, the Oscillator circuit was designed based
on the measured data of the tunable FBAR resonator introduced in [14].
The ADS schematic of the circuit is shown in Fig. 5.4.
Fig. 5.4 ADS schematic of designed tunable FBAR VCO circuit
Due to the relatively low Q factor of tunable FBAR resonator it was decided to integrate the
circuit to reduce the noise caused by parasitic effects as much as possible.
5.1.4 Grounding capacitors
In order to AC ground the quaterwave stubs in the design, the platinum layer forms bottom
electrode of the capacitor. As shown in Fig. 5.5.
Since there are conductive holes in BSTO a 100 nm silicon dioxide layer is considered over the
BSTO layer to isolate the top and bottom electrode in DC.
Tunable FBAR S1P model
30
Fig. 5.6 Reflection coefficient of the AC shorted quarterwave length stub
The reflection coefficient of the grounded quarterwave length stubs is shown in Fig. 5.6.The
stubs represent high impedance at the desired oscillation frequency range, suitable for isolating
RF signal from the DC bias voltage
Fig. 5.5 large size AC grounding capacitor
Pt+Au
SiO2
Au
Pt+Sio2+Au
31
5.1.5 Tunable FBAR resonator
The resonator was modeled and designed according to the measured data obtained in [14]. The
resonator is connected to the emitter by an integrated inductor made from the Al top electrode.
The active area of the resonator is 700µm2 Fig. 5.7 gives illustration of the resonator in
Momentum.
Fig. 5.7 Tunable FBAR resonator in ADS momentum
The geometry of the top-electrode in the active area is designed in way to suppress spurious
lateral acoustic resonances. The DC probe will be applied to the gold patch on the top of
aluminum inductor.To get the resonance frequency of the resonator Co-simulation is done by
adding the motional parameters from MBVD model as shown in Fig. 5.8.
Resonator
active area
Resonator
DC Bias
location
MBVD model
motional
parameters
Spiral inductor
made from Top
Al electrode
Fig. 5.8 Tunable FBAR Co-simulation in ADS
Au
Al
Pt
32
5.1.6 Decoupling capacitor
To isolate the DC bias of the resonator from other part of the circuit a series decoupling capacitor
near the emitter leg is used, as shown in Fig. 5.9. Here again the gold layer and bottom electrode
are isolated using silicon dioxide layer.
Fig. 5.9 Decoupling capacitor (C1 and C2 form a series capacitor letting through the RF and isolating DC of the resonator).
Capacitors C1 and C2 are formed between the platinum and gold layers with BSTO and SiO2 as
The dielectric material. The series capacitor
is large enough not to load the resonator
to affect the RF signal and the discontinuity at the gold layer prevents the DC bias signal
disturbing the rest parts of the circuit. The decoupling capacitor is illustrated in Fig. 5.10 which
shows the total designed circuit in momentum
Fig. 5.10 Total oscillator circuit in momentum
Transistor‟s Emitter
leg
Resonator
RF signal
Path
Pt
C1 BSTO C2
Au Au Sio2
33
5.1.7 Meandered circuit
Due to limitation in the fabrication and dimension of the sample (1 10mm) it was decided to
make the circuit as compact as possible so the transmission lines were meander in order to have
two circuits in one sample. The circuits with meandered lines are represented in Fig. 5.11
Fig. 5.11 meandered circuit
34
5.1.8 Co-Simulation
Some modification such as placing the resonators in the middle of the mask and adding some test
resonators needed to be done for the final mask to get the optimum layout for fabrication. To see
the oscillator performance Co-simulation is done which is shown in Fig. 5.12
Fig. 5.12 Co-simulation of final oscillator design
35
(c)
Co-simulation results for different bias voltages are given in Table 5.2.
Table 5.2: Oscillator Co-simulation results
Resonator bias voltage(V) Oscillation frequency(GHz) PN@100kHz(dBc/Hz)
5 5.655 -101
10 5.596 -106
15 5.561 -106.3
20 5.545 -106.3
25 5.526 -106.4
§
Output Spectrum
Fig. 5.13 Co-simulation results of tunable FBAR Oscillator (a)output voltage (b)phase noise (c) output spectrum
(a) (b)
(c)
Po
ut(
dB
m)
36
The Co-simulation results show that the Oscillation frequency tunability range of 129 MHz. The
oscillation frequency is slightly lower than the resonator series resonance frequency which is
expected for the loaded resonator. The phase noise is -106 dBc/Hz @ 100kHz frequency offset
from the carrier. The final mask for fabrication is given in figure 5.14
Fig. 5.14 final oscillator mask with the frame and alignment
Fig. 5.14 Final mask
37
6 Device Fabrication and Measurement
Fabrication of the device is done at Chalmers Cleanroom. A 20 nm thick layer of TiO2 for better
adhesion between SiO2 and Pt using magnetron sputtering . The device layer is sputtered on the
1 1cm sample containing the Bragg reflector. Bottom electrode pattern has been mapped from
the mask to the platinum bottom electrode by photolithography as shown in Fig. 6.1
Fig. 6.1 fabricated sample picture after bottom electrode pattering (the white areas show the platinum layer)
38
6.1 BSTO film growth by the PLD
The growth of the BSTO film has been performed using Pulsed Laser Deposition .(PLD) and
explained in following section.
6.1.1 PLD overview
PLD concept is basically simple and is shown in Fig. 6.2.
Fig. 6.2 Schematic of PLD device [32]
A short pulsed laser beam is focused onto a target (BSTO in this work). Plasma is formed
immediately on the target surface due to the pulse energy. The plasma then reaches the substrate
which had been mounted on a heater and heated to the defined temperature and causes the target
material to be deposited on the substrate.
There are numbers of parameters playing role in the deposited film quality. Some of those are
given in the following sections.
39
6.1.2 Laser-target interaction.
When the laser beam strikethrough the target surface, the material ablates out with same
stoichiometry as in the target. The vapor pressure, absorption of the material and pulse laser
wavelength determines the amount of ablated material [33].
6.1.3 The plume
The target forms a plume after ablation. This high energy plume tends to move towards the
substrate and presents forward peaking phenomenon [34].
Oxygen is often introduced into the chamber to keep constant the stoichiometry of the oxide
material and reduce the kinetic energy [33].
The pressure of the background gas and the distance between the substrate and target define the
shape of the plume.
6.1.4 Pulsed laser
The laser energy significantly affects the film quality. Higher energy increases the vapor pressure
and consequently the kinetic energy which could result in defects in the surface of the deposited
film due to re-sputtering.
Substrate temperature dramatically effects the deposited film quality in a way that higher
temperature results in better quality.
Table 6.1 summarizes the process parameters used by PLD system in BSTO film deposition.
Table 6.1: PLD system Setting
parameters comments
Laser source KrF
Laser wavelength 248 nm
Energy density 1.5 J.cm-2
Target BSTO Oxygen pressure 20 Pa
Repetition rate 10 Hz
Substrate Si/Hfo2/Sio2/…./Tio2/Pt
Substrate temperature 6200 -640
o C
Substrate-target distance 6 cm
40
Fig. 6.3 shows the sample after BSTO deposition
Fig. 6.3. 10 10 mm sample after BSTO deposition. The rain blow color shows the thickness difference of BSTO surface
Isolating SiO2, 100 nm aluminum top electrode and finally 500 nm gold layers were sputtered
and patterned on the sample. A step by step fabrication process is presented in Appendix 3.The
AFM picture of the BSTO film in the middle of the sample is shown in Fig. 6.4.
Fig. 6.4.AFM picture of the BSTO film (a) 2D (b) 3D view of the BSTO film surface
The sample is shown in Figs. 6.5 (a), (b) and (c) after pattering of each layer.
41
Fig. 6.5 fabricated sample after (a) Sio2 deposition & lift -off (b) Gold deposition & image reversal resist removal
(c) Al deposition & lift -off
(a) (b)
(c)
42
6.2 Measurements
6.2.1 Test resonator measurement
Test resonator one port measurement was done using Agilent PNA N5230A and probe station
with 150- µm GSG mounted microprobes. Figures 6.7, 6.8 and 6.9 show the fabricated test
resonator and measurement results.
Fig. 6.8 Test resonator reflection coefficient @ different DC bias voltages
Fig. 6.7 fabricated and measured Test resonator
2v
5v
10v
15v
20v
DC Bias
G S G
43
The test resonator measured results in comparison to the previously measured data which was
used in the oscillator circuit design show a shift of around 200 MHz downwards in the resonance
frequencies. It was also observed that maximum bias voltage for the resonator before the
breakdown is 20V.These effects are due to the integrated inductor and fabrication process
technology which differs slightly from the previously used resonator.
Fig. 6.9 measured series and parallel resonance frequencies
Test resonator series and parallel resonace frequencies
5.2
5.25
5.3
5.35
5.4
5.45
5.5
5.55
5.6
5.65
5.7
0 5 10 15 20 25
Voltage(V)
fs&
fp(G
Hz)
fp
fs
fs
fp
44
6.2.2 Oscillator measurements
To measure the oscillator, BJT transistors were mounted on the circuit by silver epoxy as shown
in Fig. 6.10.
Fig. 6.10 Final oscillator circuits including transistors
Prior to RF measurement of the oscillator output spectrum and the phase noise, a DC
measurement was done to verify the transistor is consuming the expected power as considered in
the simulation.
DC measurement showed that unfortunately the grounding capacitors were shorted to ground in
DC due to the pin holes in SiO2 layer. That could be because of FHR device which heats the
sample to several hundred degrees resulting a low quality SiO2 sputtered layer.
45
7 Conclusion and future work
BSTO intrinsically tunable FBAR and passive components have been monolithically integrated
on high resistivity silicon substrate. To demonstrate the optional of the technology, 5.5 GHz
voltage controlled Oscillator have been designed and fabricated on the substrate.
The simulated results showed high tunability with low phase noise compared to LC tank
oscillators.
Measured Test resonators represent tenability of 114MHz @ 5.5 GHz. DC measurement of the
Oscillator revealed short circuit in the integrated RF grounding capacitors due to pin holes in the
SiO2 and BSTO layer.
Future work will contain another fabrication round to prevent the pin holes by using E-beam
evaporation technology for SiO2 deposition. Depending on the Oscillator performance, the
fabrication process can be further optimized.
Oscillator circuit topology can be modified to improve the performance based on the
measurements of the latest presented tunable FBAR technology with higher Q factors.
46
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49
Appendix1: Tunable FBAR Resonator MBVD model Extracted
Parameters
Fig.1Tunable FBAR resonator MBVD model extracted parameters (a) series and parallel resonance frequencies
(b) series and parallel Q factors (c) effective coupling coefficient (d) impedance magnitude
(a) (b)
(c)
(d)
Imp
ed
an
ce m
agn
itu
de
,(O
hm
s)
50
Appendix2: Transmission line parameter extraction
Conversion S-parameter to ABCD
Conversion of S to ABCD-parameter is given in [1]
ABCD network for Transmission line
,Z0
Zo is a characteristic impedance of the transmission line.
is the length of the line.
Note that
+ Complex propagation constant
= attenuation constant NP/m
β = wave propagation constant
)cosh()sinh(
)sinh()cosh(
o
o
Z
Z
2112221121122211
2112221121122211
211111
1
1111
2
1
SSSSSSSSZ
SSSSZSSSS
SDC
BA
o
o
51
For a Lossless line
= 0
When the transmission line is lossless this reduces to
For TEM wave propagation the effective permittivity and Loss tangent can be obtained from [1]
𝜀 √
Where
Where is the attenuation constant due to dielectric in NP/m.
Extracted parameter for CPW from ADS momentum simulation are plotted from 0 to 20 GHz in
Fig. 1(a),(b),(c).
)sin()sinh( kjjk
)cos()sin(
)sin()cos(
kZ
kj
kjZk
o
o
)cos()cosh( kjk
52
Fig. 1 extracted parameters for a 1.4 mm length CPW with g=48µm and S=100 µm on the multilayer substrate (a) characteristic
impedance (b)effect permittivity (c)attenuation constant
Reference
[1] David M. Pozar, Microwave engineering Third edition. John Wiley & Sons, Inc., 2005
(a) (b)
(c)
53
Appendix3: Fabrication Steps and recipe
Note: the thicknesses are not to the same scale
Process Recipe Schematic 1.New sample including
Bragg reflector
Parameters:Hfo2 260nm/
SiO2 284nm(3 pairs) Size:10 10 mm
Thickness:501 m
2.Cleaning Tool: Wet bench
Parameters: Acetone, Ultrasonic bath for 3 min @ 100% power
Intention: To clean the photo resist used for protecting the wafer during cutting
3.TiO2 Deposition Tool: Sputter – NORDIKO 2000
Parameters: Ti and O2 for 8 min.
Intention: To deposit Tio2 on the sample for better adhesion between Sio2 and Pt.
4.Platinum Deposition Tool: Sputter – NORDIKO 2000
Parameters: Pt for 2 min @60 w
Intention: To deposit Pt on the sample as the bottom electrode
Piezo-electric material
Top-electrode
Silicon
bottom-electrode
Z1
Z2
Z2
Z1
Piezo-electric material
Top-electrode
Silicon
bottom-electrode
Z1
Z2
Z2
Z1
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
54
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
5.Cleaning Tool: Wet bench Parameters: Acetone, Ultrasonic bath for 3 min @
100% power Intention: To clean surface from impurities before
applying the photo resist
6.Resist applying and spinning
Tool: Hot plates and resist spinner
Parameters: photo resist S1813 @4000rpm for 30 sec. Hot Plates@ 90
0 C for 1 min.
Intention: To apply resist evenly on the sample for edge removal
7.Photo resist pattering-I
(edge removal)
Tool: Mask aligner –
KS MJB3-UV 400 Parameters: Soft contact, exposure time 1
min. Intention: To remove resist edges
8.Developing Tool: Wet bench
Parameters: Developer MF-319 for 1.5 min.
Intention: To develop exposed resist edges
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
55
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
9.Photo resist patterning -II Tool: Mask aligner – KS MJB3-UV 400 Parameters: soft contact,
exposure time 15 sec. Intention: To expose photo resist according to the bottom
electrode mask pattern
10.Developing Tool: Wet bench
Parameters: Developer MF-319 for 15 sec.
Intention: to develop the exposed photo resist
11.Etching Tool: Ion Beam Milling Oxford Chamber.
Parameters: Argon gas flow for 20 min.
Intention: To pattern the Pt bottom electrode layer
12.Resist removal
Tool: Wet bench, Ultra sonic
bath Parameter: Microposit remover @75
0C –Ultra sonic
bath @%100 for 10 min. Intention: to remove photoresist
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
56
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
BSTO 234nm
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
BSTO 234nm
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
13.Oxygen plasma strip Tool: Plasma Therm Batch Top Parameters:O2 plasma for 1
min @ 250 W. Intention: to remove organic residue from photo resists.
14.BSTO deposition
Tool: Pulsed Laser Deposition-(PLD) Parameters: Target BSTO
Temperature 620-640o C-
3100 laser pulses in 5 min. Intention: To deposit BSTO
film
15.Cleaning Tool: Wet bench
Parameters: Acetone Intention: to clean the sample
surface after BSTO deposition Ready to be taken to the main cleanroom
16.LOR Lift off- Resist Tool: Wet bench –Hot plates –Resist spinner Parameters: LOR 3A
@4000rpm for 1 min. Hot plates: 5 min@ 190
0 C.
Intention: Coat and prebake
LOR
BSTO 234nm
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
57
17.Coat and prebake imaging resist
Tool: Wet bench –Hot plates –Resist spinner Parameters: Photo resist
S1813@4000 rpm for 30 sec. Hot plates @110
0 C for 2 min.
Intention: Coat and prebake
resist making it ready for patterning.
18.Expose imaging resist Tool: Mask aligner –
KS MJB3-UV 400 Parameters: soft contact
exposure time 10 sec. Intention: To expose photo resist according to the SiO2
mask pattern
19.Develop the resist and LOR
Tool: Wet bench
Parameters: Developer MF-319 for 2 min Intention: To develop the
resist and LOR for SiO2
sputtering.
BSTO 234nm
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
BSTO 234nm
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
BSTO 234nm
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
58
BSTO 234nm
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
20.Sio2 layer sputtering Tool: FHR MS 150x4-L Sputter Deposition system. Parameters:O2 and Si
combination for 429 sec. Intention: To deposit SiO2 layer on the sample.
21.SiO2 Lift-off Tool: Wet bench-Ultra sonic
bath Parameters: Microsit Remover 1165@75
0C for 5 min.
Ultra Sonic bath @%20 for 1 min. Intention: To lift off SiO2 and have the pattern of it.
22.Resist applying for image
reversal work(Gold layer)
Tool: Wet bench–Resist
spinner Parameters: Photo resist
S1813@4000 rpm for 30 sec.
Intention: To apply photo resist for image reversal exposure.
BSTO 234nm
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
BSTO 234nm
pt 1 0 0 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
59
soluble
23.Exposure using inverted mask
Tool: Mask aligner – KS MJB3-UV 400
Parameters: exposure time 6 sec Intention: to expose the
sample for Gold deposition and patterning(the gold layer finally remains at exposed area)
24.Reversal bake Tool: Hot plates
Parameters: 1250C for 2 min.
Intention: To make the exposed area inert while the
unexposed area remains photo active.
25.Flood exposure without mask
Tool: Mask aligner – KS MJB3-UV 400 Parameters: flood exposure
for 60 sec. Intention: makes the resists, which was not exposed at
previous step, soluble in developer.
BSTO 234nm
pt 1 0 0 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
BSTO 234nm
pt 1 0 0 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
BSTO 234nm
pt 1 0 0 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
60
26.Developing
Tool: Wet bench Parameters: Developer AZ 351 B for 1 min
Intention: To develop the resist according to the mask pattern
27.Gold Deposition Tool: AVAC E-beam evaporator.
Parameters: 8.8 Å/ sec to reach 0.5 µm gold thickness.
Intention: To deposit the Gold layer on the sample.
28.Lift-off
Tool: Wet bench
Parameters: Acetone @750
C for 5 min.
Intention: To remove the extra Gold and have the pattern on the sample.
BSTO 234nm
pt 1 0 0 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
BSTO 234nm
pt 1 0 0 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
BSTO 234nm
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon
61
29.Deposition of Aluminum top electrode (same procedure as the Sio2 Layer)
Tool: FHR MS 150x4-L Parameter: Al deposition for 51 sec.
Intention: To deposit and pattern the Al layer on the sample.
BSTO 234 nm
pt 100 nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Sio2 284nm
Hfo2 260nm
Silicon