COMPARISON OF PLANAR AND WOUND TRANSFORMERS FOR FLYBACK

FORWARD AND HALF-BRIDGE SPACE POWER CONVERTERS

Thomas Björklund (1)

John Andreasen (2)

Finn Brøsen (3)

Erik Matthiesen (4)

Ole Poulsen (5)

(1) Magnetics Engineer, Flux A/S, Industrivangen 5-11, DK 4550 Asnaes (Denmark), tb@flux.dk

(2) Magnetics Engineer, Flux A/S, Industrivangen 5-11, DK 4550 Asnaes (Denmark), ja@flux.dk

(3) Power Engineer, Flux A/S, Industrivangen 5-11, DK 4550 Asnaes (Denmark), fb@flux.dk

(4) Power Manager, Flux A/S, Industrivangen 5-11, DK 4550 Asnaes (Denmark), em@flux.dk

(5) Technical Manager, Flux A/S, Industrivangen 5-11, DK 4550 Asnaes (Denmark), op@flux.dk

ABSTRACT

Planar technology has now entered the space domain.

The big advantages of planar technology are;

• Low profile

• Excellent repeatability

• Economical assembly

• Mechanical integrity

• Superior thermal characteristics

This is why the general power industries increasingly

are using planar magnetics in more and more

applications, and therefore also why we see a rising

demand for the usability of the planar technology

among space application developers.

The differences between wound and planar transformers

have been mapped with a detailed look on the various

parasitic component values, such as DC- and AC-

resistance, Leakage Inductance and stray capacitance,

and revealed the magnitude of the advantages of planar

technology.

This technical solution is proven in prototypes that have

been built in different combination of PCB’s and copper

foil, with more or less interleaving of windings.

Furthermore the transformers have been designed with

several outputs stacked together with a fairly high

number of primary turns, in order to have planar

transformers similar to the wound types that are

generally used for space applications.

INTRODUCTION

Obtaining the optimal design has been impossible when

engineers have developed printed circuit board (PCB)

windings using the frame of The European Cooperation

for Space Standardization (ECSS) standards for space

qualified processes. Where the requirements towards the

PCB manufacturers are driven in two main directions;

fine pitch technology for microprocessors and

impedance control for Radio Frequency (RF) domain

circuits, which is also reflected in the Interconnecting

and Packaging Electronic Circuits, now the Association

Connecting Electronics Industries (IPC) standards. Here

the Planar technology comes with greatly different

demands regarding the thickness of copper, laminate

tolerances and a strict control of etching the thick

copper foils and laminating with minimal resin, in order

to get a good fill factor in the final transformer.

Using PCB as a part of a component, is breaking with

the general thinking in the technologies of space

qualified PCB - and this divagation is necessary in order

to achieve functionality that can compete with

traditionally wounded transformers. Where high current

is occurring and thicker copper layers are needed,

folded copper foil windings makes a great alternative to

the PCB, and this pioneering mix of technology within

planar transformers, gives a whole new flexible design

frame for the space engineers.

Quality assurance is of utmost importance for the planar

transformer, where thermal drain, z-axis expansion and

via or Plated Through Hole (PTH) reliability, during

thermal cycling, are some of the most critical points. It

needs to be at the same low risk as traditionally wound

transformers, so tests and verification of functionality

out of the normal ECSS specified frame of use is

needed. This points to pioneering thermal tests that can

verify the life terms of the space planar transformer. I.e.

passive burn-in, power burn-in and thermal shock, but

also endurance testing to show the limits of this new

component technology.

The different topologies are selected to represent three

different usages of the ferrite shown in Fig. 1, Fig.2 and

Fig. 3

Fig. 1 B-H Curve of a flyback transformer

Fig. 2 B-H Curve of a forward transformer

Fig. 3 B-H Curve of a half-bridge transformer

1. THE TRUE COMPARISON

When comparing a standard transformer with a planar

transformer there are actual four different points of

comparison:

a) EELP vs. RM

Here the core shapes are different in standard sizes.

Designing the winding structure sets the criteria for the

needed window, which results in selecting a bigger core

than initially needed. RM cores enclose the windings

better than the EE cores, which gives a better EMI

performance.

b) PCB vs. Wire

Here the PCB is limited in winding-techniques and has

a thermal expansion factor, where the wire can be

wound in wrong number of turns from time to time, and

has pulling and bending issues.

c) Flat vs. Round

A thin flat conductor doesn’t suffer from skin effects in

high frequency alternating current (AC), like a round

conductor. But Flat conductors couples better to each

other hence generates a larger parasitic capacitance.

d) Tape & PCB vs. Coil former

Using a standardised bobbin or coil former, gives a high

reliability component since it is well tested with regards

to creepage and clearance distances. Tape and PCB

structures gives an individual design each time with

tolerances that needs verification testing dependent on

the tolerances.

Fig. 4 Planar EE18 transformer compared with RM5

(Left side), and Planar EE22 transformer compared

with RM10 (Right side)

2. THERMAL PERSPECTIVE

The optimal design for transformers is normally set to

be the balance of core and winding losses Fig.5, here

planar transformers can be designed towards a higher

core loss in order to gain less winding loss, because of

the better thermal drain possibility.

Fig. 5 Power loss balance

(1)

(2)

There is also the possibility of better thermal drain of

the winding structure, when using PCB for windings.

3. MECHANICAL INTEGRATION

One of the most obvious advantages of planar magnetics

is the different possibilities of integrating the trans-

former with the converter PCB. Stand-alone is the

common use of wound transformers, but sunken wound

transformers are also seen.

Fig. 6 PCB mounting options

Planar transformers are often integrated, but here the

tradeoffs between PCB layer build-up and transformer

performance becomes an issue. Even a small modi-

fication in the transformer, demands for a whole new

layout, where a stand-alone model easily can be

prototyped to the right fit, by removing or adding a

single turn.

A big advantage with stand-alone transformers is the

possibility of using several multilayer PCBs stacked

together, and even in combination with foil or wire

windings. Here the Hybrid model also has advantages,

when high secondary current can be placed in thick

copper foil combined with primary PCB windings, or

when multiple primary windings can be placed in thin

wire combined with the lower number of turns

secondary PCB windings in the converter board.

3.1. Fill Factor and Robustness

When looking at the RM wound transformer and the

EELP PCB in Fig.7 from a vibration point of view, it is

clear that the Centre of Mass Point is lower on the

planar core, as the core is lower. This gives a more

robust mechanical construction, despite of the windings.

Fig. 7 Centre of Mass Point

The winding construction of a PCB transformer can also

never reach as high a mass density as the wound trans-

former. This is due to the maximum possible fill factor

Fig.8 and the material density Fig.9, where polyimide

and plastics are lighter than copper.

Fig. 8 Typical Max Fill factors (Ku) for standard and

planar magnetics

Fig. 9 Density of transformer materials

This points on a more mechanical robust transformer,

however only if they are constructed with the same size

of core and same window area. It will then give a higher

current density in the planar transformer, and this is also

often the case in the design as the thermal transfer is

more efficient.

4. PCB TOLERANCES

When designing transformers, it is very important to

take thermal expansion into consideration due to the

fragile core that easily breaks in stressed conditions.

Introducing PCB is actually rather troublesome, as the

glass woven fibres are preventing x-y expansion and

instead transfer all the thermal expansion to the z-

direction. Which is exactly what is needed for a stable

Surface Mount Device (SMD) on a PCB, but absolutely

a bad idea for fitting into a transformer core.

The core dimensions comes with big tolerances of

window height shown in Fig.10 for EELP18 which is

±5%, but it is worse for the final PCB where the tole-

rances is ±10%.

Fig. 10 EELP18 Core tolerances

Examining the manufacturing process reveals that the

laminate and prepreg process is dependent on the resin

content, which depends on the heat and flow when

filling the bare woven fibre sheets, where the viscosity

also depends on the stock-age of the resin. For a 100µm

laminate the tolerance is as large as ±25% according to

IPC as shown in Fig.11

Fig. 11 Raw laminate tolerances

Additional the copper plating also has tolerances and

since copper is expensive, the manufacturer aims very

precise on the minimum tolerances Fig.12.

Fig. 12 Innerlayer copper thickness

When plating the drilled through holes, extra copper is

added to the entire surface. This gives an additional

thickness Fig.13

Fig. 13 Outerlayer copper thickness after plating (class

3 is for space and aerospace)

A final thickness of a 70µm winding is therefore in the

area of 55,7µm to 83,7µm depending on the PTH

structure.

4.1. Milling and registration

Other PCB tolerances, such as milling of the edge and

registration of the layers in-between are also the cause

for poor fill-factor, and this is a critical point regarding

clearance. In Fig.14 is shown the microsection of the

EELP18 transformer windings. The ECSS standards

states max ±100µm.

Fig. 14 Flyback transformer microsection

5. DIELECTRIC CONSTANT

When it is mentioned that the Planar transformer

repeatability is excellent, it is needed to take a deeper

look into predicting the parasitic capacitance according

to production tolerances.

The variations of dielectric constants for polyimide PCB

Fig.15 appears when the resin has a lower value than the

fibre glass, and the resin content varies.

Fig. 15 Dielectric constant and resin content of prepreg

Calculations due to laminate thickness tolerances and

dielectric constant variation results in huge differences.

Fig. 16 Capacitance variations of laminate thickness

Fig. 17 Capacitance variations of moist content

Polyimide is hygroscopic, and water has a dielectric

constant of 80,40, so within the max accepted value of

0,28% the variation is a few extra pF as in Fig.17

6. EMI AND EMC OF CORE SHAPES

Electromagnetic Interference (EMI) and Electromag-

netic Compatibility (EMC) are the unintentional gen-

eration, propagation and reception of electro-magnetic

energy of the transformer interacting with the surround-

ings. This is the almost impossible part to calculate

when designing the prototype, and it very much depends

on the core shape and usage.

The energy propagates in a combination of magnetic (B-

field) and electric (E-field) radiation as shown in Fig. 18

Fig. 18 Electromagnetic(EM) wave

The waves are only polarised in the Far-field area,

which is more than 2 wavelengths from the source, and

at 125kHz the wavelength is 2.4km long. In the Near-

field the EM wave is much more complex and can

therefor not be measured exactly on the two kinds of

transformers.

The E-core shapes in general are open and the windings

are wound outside the core structure. The Pot-core is

very closed and has the best EMI performance, however

the closed core makes it difficult to enter and exit the

windings. The RM core is therefore a fairly good

solution for EMI shielding of the windings. Fig. 19

shows the differences drawn with a Mean Length Turn

(MLT).

Fig. 19 Pot-, RM- and EELP cores with windings and

B-field lines indicated (Blue arrow)

Then of course there is the factor of relative ferrite

permeability µ r vs. the vacuum permeability, which is

also the permeability of air µ0. In N87 and 3F3 the

factor is around 2000.

6.1. Induction formulas

When looking at the standard textbook formulas for

transformers and how they are constructed it appears

that the core-shape differences are not a part of the

calculations. The induction formulas are derived from

two different systems, where the primary system is

based on a magnetic field from an endless long straight

lin shown in Eq.3-5 And Fig.20 and the self-inductance

is based on a toroidal solenoid Eq.9-10 And Fig.22

Fig. 20 Endless long line

(3)

(4)

(5)

Figure 21 reveals that the distance 2�x, where x is the

radius, is the same as the Magnetic Path Length (MPL),

when calculating the transformer.

Fig. 21 relationship of 2�r and MPL

In the Ferroxcube handbook, Eq.6 Is given for the

magnetic field strength based on the effective MPL

denoted le. (Square root 2 is used only for a sinus)

(6)

(7)

(8)

Where Eq.7 and Eq.8, From the Fundamentals of Power

Electronics theory book, shows that this is based on a

straight endless line.

The principle of a toroidal solenoid Fig.22 gives the

base for equations where the magnetic path length is a

circle.

Fig. 22 Toroidal solenoid

(9)

(10)

When 2�r equals the MPL, it is only true for toroidal

cores. Not for RM and EELP core structures. Hence the

differences in shape is ignored in the transformer

calculations Fig.23.

Fig. 23 Transformer equation (MPL is denoted lm)

When the shape of the magnetic path length is ignored,

the contribution, due to one or another shape, cannot be

defined.

7. TEST RESULTS

The prototypes in Fig. 4. are defined by the winding

turns (T) shown in Fig.24

Fig. 24 Flyback Transformer turns ratio

The mass budget is:

RM5: 5,12g of which the core and coil former is 3,61g

EELP18: 8,01g of which the core is 4,81g

Dimensions are:

RM5: H=10,52mm W=15,81mm L=15,95mm

EELP18: H= 8,09mm W=19,23mm L=22,52mm

Measurements have been performed on the prototypes

revealing the magnitude of the design.

Fig. 25 Flyback Transformer measurements

Comparison of the measurements Fig.25 shows that

they are overall the same, but the planar transformer has

lower leakage, which is around half of the standard

transformer, but also around four times as large capaci-

tance’s. These values are also the commonly expected.

31T 3T

7T

* *

*

16T 2T

5T

*

RM5 EELP18

The prototypes shown in Fig. 4. are defined by the

windings in Fig.26

Fig. 26 Forward Transformer turns ratio

The mass budget is:

RM10: 33,75g of which core and coil former is 27,97g

EELP22: 21,74g of which the core is 12,96g

Dimensions are:

RM10: H=18,61mm W=29,61mm L=39,50mm

EELP18: H=11,76mm W=24,15mm L=33,96mm

Measurements have been performed on the prototypes

to reveal the magnitude of the design.

Fig. 27 Forward Transformer measurements

Comparison of the measurements Fig.27 shows that the

two forward transformers over all are the same.

However here the wound transformer has lower leakage

and higher capacity than the planar transformer Fig.28.

This was anticipated, but not the most intuitive result.

Fig. 28 Forward Planar Transformer

The reason for having a better performance on the

wound transformer lies within the possibility of more

complex interleaving structure Fig.29 and Fig.30.

Fig. 29 Interleaving structure of Planar Transformer

Fig. 30 Interleaving structure of RM Transformer

(T is short for Turn)

7.1. Measuring tolerances

When measuring the leakage, it is ideal to short the

secondaries, so that only the none-coupled inductance is

measured Fig.31. Shorting with a wire adds inductance

to the circuit, hence the measurement has a large

tolerance relating from the inductance of the short.

Fig. 31 Simple and complex attempt to achieve a perfect

ac short

5T 3T

7T

* *

*

4T 3T

7T

*

RM10 EELP22

7T 7T5T 4T

8. DISCUSSION

For both traditional and planar transformers, it is

possible to have different structures of interleaving,

which is a choice of leakage vs. stray capacitance.

Prototyping the planar transformer is very costly, but

here simulation SW really comes in handy.

Fig. 32 Test converter mock-up

Test converters Fig.32 has been built so that the RM

transformer and the EELP transformer can be swapped.

This gives the possibility of testing load and efficiency

capabilities, and the general picture is as expected; the

one with lower leakage performs better. But as they

were designed on the same criteria they are overall the

same on performance.

Simulating the transformer makes it possible to predict

the leakage and stray capacitance by Finite Element

Analysis (FEA) but here tolerances are not included,

and Monte-Carlo plots is not possible due to standard

Central Processing Unit (CPU) performance.

Fig. 33 Capacitance model of a transformer

The EMC performance of the actual core can neither be

calculated nor measured precisely. The only way of

verifying the differences, due to core shapes, is to

design a PCB for each converter, with individually

optimised snubber circuits due to the different pin

positions on the RM and EELP transformer and the

layout. This part must be an issue that needs to wait for

a future study.

9. CONCLUSION

When looking on the core structure, the lower and plane

core has its mechanical and thermal advantages:

• Low building height

• Mechanical integrity

• Superior thermal characteristics

- The drawbacks are; EMC and winding area

When looking on introducing PCB windings, the thin

layers and photo-etching process has its advantages:

• Low building height

• Excellent repeatability

• Economical assembly

• Mechanical integrity

• High frequency performance (to some extend)

- The drawbacks are; prototype costs, reduces possible

number of turns. PTH or VIA reliability. Moist sen-

sitivity, thermal expansion and production tolerances.

10. REFERENCES

1. Robert W. Erickson & Dragan Maksimovic.

Fundamentals of Power Electronics, Springer; 2nd

edition (January 2001).

2. Colonel Wm. T. McLyman. Transformer and

Inductor design Handbook, Third Edition (2004)

3. Soft Ferrites and Accessories Data Handbook.

Ferroxcube (2008)

4. Hugh D. Young, Roger A. Freedman. University

Physics with Modern Physics with Mastering

Physics. Addison Wesley; 11 edition (August 8,

2003)

5. Generic Standard on Printed Board Design. IPC-

2221A, IPC (May 2003)

6. Chet Guiles, Rancho Cucamonga. Everything You

Ever Wanted to Know About Laminates, but Were

Afraid to Ask. Arlon-Materials for Electronics, 9th

Edition (Nov 2008)

7. Rafael Asensi, Roberto Prieto, José A. Cobos, and

Javier Uceda. Modeling High-Frequency

Multiwinding Magnetic Components Using Finite-

Element Analysis. UPM, División de Ingeniería

Electrónica, Madrid 2008

Thomas Björklund

Magnetics Engineer, Flux A/S

www.flux.dk

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