andreas brolund - uu.diva-portal.org

UPTEC E 17 011

Examensarbete 30 hpJanuari 2018

Evaluation of power quality and common design concept for AC-DC converters in aircraft

Andreas Brolund

Teknisk- naturvetenskaplig fakultet UTH-enheten Besöksadress: Ångströmlaboratoriet Lägerhyddsvägen 1 Hus 4, Plan 0 Postadress: Box 536 751 21 Uppsala Telefon: 018 – 471 30 03 Telefax: 018 – 471 30 00 Hemsida: http://www.teknat.uu.se/student

Abstract

Evaluation of power quality and common designconcept for AC-DC converters in aircraft

Andreas Brolund

This master thesis has been carried out in collaboration with Saab, AvionicsSystems in Jönköping, Sweden, during the spring of 2017. The thesis investigatesunidirectional rectifier topologies in aircraft and the focus has beenon evaluating the power quality requirements according to the aircraft standards,in the course of the More Electric Aircraft concept. Both passive andactive power factor correction topologies are considered, discussed and compared.Simulation models are designed in MATLAB/Simulink and the proceduresare presented. A modular concept regarding components is discussedwhere different power supplies and loads are considered. The simulationspresent both a passive 12-pulse auto-transformer rectifier unit and an activeDelta-switch rectifier fulfilling requirements for aircraft such as the total harmonicdistortion of the supply current. In addition, the input power factoris close to unity and an efficiency greater than 97% is obtained. Lastly, futureaspects of each topology are discussed and necessary improvements toobtain realistic simulation models are presented.

ISSN: 1654-7616, UPTEC E 17 011Examinator: Mikael BergkvistÄmnesgranskare: Markus GabryschHandledare: Ingemar Thörn

Popularvetenskaplig sammanfattning

Detta examensarbete ar utfort pa och i samarbete med Saab, Avionics Systems,Jonkoping, Sverige under varen 2017.

I examensarbetet utvarderades enkelriktade (kraftflodet kan bara ga i en rikt-ning, fran kalla till last) topologier for likriktning av vaxelstrom for anvandningi elektriska kraftsystem i flygplan. Likriktningen skedde fran 115-230 V vaxelsp-anning med en frekvens pa 360-800 Hz till 270-540 V likspanning med en re-sistiv last pa 2-10 kW. En passiv topologi (ej kontrollering av likriktningsproces-sen) baserad pa en 12-pulsig automatiserad transformator samt en aktiv topo-logi (kontrollering av likriktningsprocessen) baserad pa en delta-konfigureradhalvledare-anordning simulerades och utvarderades. Fran litteraturstudier ochsimuleringsresultat drogs slutsatsen att den passiva topologin i dagslaget arden basta topologin for likriktning da den erbjuder hog palitlighet, robusthetoch effektivitet. Den aktiva topologin visade pa goda resultat nar det kom tillel-kvalite. I och med pagaende framsteg och utvecklingar inom kraftelektro-nik och mikro-processorer, vilket ar en stor del inom den aktiva topologin, ardet berattigat att saga att den aktiva topologin aven kommer vara aktuell forlikriktning inom en snar framtid. Simuleringsmodellerna var forenklade upp-skattningar av hur det verkliga systemet sag ut och fungerade. Detta ledde tillatt det tyvarr var svart att dra nagon slutsats angaende huruvida ett modulartkoncept for likriktning givet olika in- och uteffekter kunde uppnas eller ej. Detmodulara konceptet skulle innebara att trots givet olika applikationer med oli-ka tekniska specifikationer skulle kunna anvanda en likriktnings-modul somfungerade for samtliga fall, vilket aven var en utav fragestallningarna. For attkunna fa mer realistiska simuleringsfall behovdes fler parametrar och variab-ler tas med i modellen. Exempel pa dessa parametrar var temperaturskillna-der, kosmisk stralning och lastprofiler som i samtliga simuleringsfall bortsagsifran. Framtida passiva och aktiva topologier som kommer vara av intresse ar18 och 30-pulsig automatiserad transformatorer och Vienna-likriktare baseradpa en konfiguration av sex transistor. Slutligen drogs slutsatsen att mjukvaran,MATLAB/Simulink, som anvandes for simuleringar ar ett utav de basta verk-tygen pa marknaden nar det kommer till analys, bedomning, validering ochforbattring av bade kraftelektronik och elektriska system.

Utifran utvarderingen av enkelriktade likriktnings-topologier i detta examens-arbete kan slutsatser dras om vilka topologier som kan vara intressanta att gavidare med samt vilka forbattringar som maste goras for att kunna anvanda deframtagna simuleringsmodellerna for mer realistiska fall. En overblick kan faspa framtida topologier varda att forska mer inom vid intresse.

Att forbattra denna typ av enkelriktad likriktning var av intresse for att mins-ka storningar och for att hoja el-kvaliten i det elektriska systemet pa flygplan.Detta da elektriska storningar och lag el-kvalite ger forsamrad prestanda ochsliter mer pa komponenterna vilket i sin tur ger lagre hallbarhet och forsamradpalitlighet. Slutligen leder detta till okade underhallskostnader och okad viktpa grund av att filtrering och blockering av storningar behovs. For att ta re-

da pa om en likriktare har bra prestanda finns det olika krav och riktlinjer attfolja for el-kvalite pa flygplan. Dessa ar sammanfattade i standard DO-160Foch MIL-STD-704F dar den forstnamnda oftast anvands for civilt flyg och densistnamnda oftast anvands for militart flyg. Dessa standarder ar sammanfattadei rapporten da de anvandes for validering och utvardering av simuleringsmo-dellerna.

For att minska vikt, underhallskostnader, forbattra palitligheten samt prestan-da av flygplan introducerades ett koncept benamnt More Electric Aircraft (merelektriska flygplan) pa tidigt nittiotal. Konceptet innebar att ersatta tunga me-kaniska, pneumatiska och hydrauliska system mot elektriska system. Detta led-de till okad efterfragan pa losningar av elektriska system som kan driva ochforsorja olika laster pa flygplan som tidigare varit drivna av mekaniska, pneu-matiska och hydrauliska system. Det ledde aven till utvidgning av det elekt-riska systemet pa flygplan, vilken i sin tur ledde till att god el-kvalite blev anviktigare.

Utover mojlig forbattring av el-kvalite ville Saab aven undersoka och utvarderaenkelriktade likriktare pa flygplan for att de olika in- och uteffekterna somanvandes i dagslaget i elkraftsystemet pa flygplan resulterade i olika losningarfor olika fall, vilket ledde till okade kostnader. Om ett modulart koncept kundefaststallas skulle det vara av intresse for Saab.

Tillvagagangssattet var att forst utfora en genomgaende litteraturstudie ochmarknadsundersokning av aktuella topologier for likriktning, men aven for attfa en overgripande forstaelse av den elektriska systemet pa flygplan. Informa-tion hamtades fran rapporter, artiklar, journaler och informativa webbplatsermen aven en del fran interna dokument pa Saab. Utifran detta presenteradesoch utvarderades de mest trovardiga och intressanta topologierna dar aven spe-cifikationerna, givna av Saab, och framtida aspekter for likriktare var med ochpaverkade. Utifran utvarderingen av de olika topologierna valdes tva vidare forytterligare utvardering i simuleringsmiljo. Modellerna simulerades for olika in-spanningar, utspanningar, frekvenser och laster i hopp om att nagon slutsatsangaende det modulara konceptet kunde dras samt for att validera modellernamed avseende pa el-kvalite. Slutligen utvarderades och diskuterades simule-ringsresultaten samt presenterades framtida koncept och slutsatser.

UPPSALA UNIVERSITY

MASTER THESIS

Evaluation of power quality andcommon design concept for AC-DC

converters in aircraft

Author:Andreas BROLUND

Examiner:Mikael BERGKVIST

Supervisor:Ingemar THÖRN

Subject reader:Markus GABRYSCH

A thesis submitted in fulfillment of the requirementsfor the degree of Master of Science at the

Department of Engineering Sciences

October 21, 2017

http://www.uu.se

http://teknik.uu.se/

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“Happiness only real when shared”

Christopher McCandless

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Uppsala University

AbstractDisciplinary Domain of Science and Technology

Department of Engineering Sciences

Master of Science

Evaluation of power quality and common design concept for AC-DCconverters in aircraft

by Andreas BROLUND

This master thesis has been carried out in collaboration with Saab, AvionicsSystems in Jönköping, Sweden, during the spring of 2017. The thesis inves-tigates unidirectional rectifier topologies in aircraft and the focus has beenon evaluating the power quality requirements according to the aircraft stan-dards, in the course of the More Electric Aircraft concept. Both passive andactive power factor correction topologies are considered, discussed and com-pared. Simulation models are designed in MATLAB/Simulink and the proce-dures are presented. A modular concept regarding components is discussedwhere different power supplies and loads are considered. The simulationspresent both a passive 12-pulse auto-transformer rectifier unit and an active∆-switch rectifier fulfilling requirements for aircraft such as the total har-monic distortion of the supply current. In addition, the input power factoris close to unity and an efficiency greater than 97% is obtained. Lastly, fu-ture aspects of each topology are discussed and necessary improvements toobtain realistic simulation models are presented.

http://www.uu.se

http://http://www.teknat.uu.se/

http://teknik.uu.se/

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AcknowledgementsI would like to thank my supervisor Ingemar Thörn for always taking thetime to listen when I had questions and thoughts. Markus Gabrysch, mysubject reader but also my second supervisor, for all the help with the simu-lation models. Jonas Dahlqvist for making sure I was integrated in into mynew environment, for always checking up on me so that everything was okayand for being a role model when it comes to being a leader. I would also liketo thank Mattias Johansson for all the help with troubleshooting simulationmodels and for the great discussion we had which made me think clearer.Last but not least, Aldin Avdic for making my time in Jönköping more enjoy-able by being my friend and training partner day in and day out.

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Contents

1 Introduction 11.1 Background . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11.2 Objective and Goal . . . . . . . . . . . . . . . . . . . . . . . . . 51.3 Methodology . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51.4 Delimitations . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51.5 Disposition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

2 Power Quality Requirements of Aircraft 72.1 Standard MIL-STD-704F and DO-160G . . . . . . . . . . . . . . 72.2 Voltage requirements . . . . . . . . . . . . . . . . . . . . . . . . 82.3 Power and Frequency requirements . . . . . . . . . . . . . . . 112.4 Current and EMC requirements . . . . . . . . . . . . . . . . . . 13

3 Rectifier Topologies in Aircraft 163.1 Auto-transformer rectifier unit . . . . . . . . . . . . . . . . . . 173.2 Active power factor correction . . . . . . . . . . . . . . . . . . 18

3.2.1 ∆-switch and Y-switch rectifier . . . . . . . . . . . . . . 183.2.2 Vienna rectifier . . . . . . . . . . . . . . . . . . . . . . . 20

3.3 Summary and Comparison . . . . . . . . . . . . . . . . . . . . 22

4 Design of Simulation Models 254.1 Power supply . . . . . . . . . . . . . . . . . . . . . . . . . . . . 254.2 Load . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 264.3 DC-link . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 264.4 12-pulse ATRU . . . . . . . . . . . . . . . . . . . . . . . . . . . 27

4.4.1 Auto-transformer . . . . . . . . . . . . . . . . . . . . . . 274.4.2 Diode bridge . . . . . . . . . . . . . . . . . . . . . . . . 304.4.3 Interphase transformer . . . . . . . . . . . . . . . . . . . 30

4.5 ∆-switch rectifier . . . . . . . . . . . . . . . . . . . . . . . . . . 334.5.1 Boost inductor . . . . . . . . . . . . . . . . . . . . . . . 334.5.2 Diode bridge and Switches . . . . . . . . . . . . . . . . 334.5.3 Control of switches . . . . . . . . . . . . . . . . . . . . . 33

5 Result and Discussion 395.1 12-pulse ATRU . . . . . . . . . . . . . . . . . . . . . . . . . . . 395.2 ∆-switch rectifier . . . . . . . . . . . . . . . . . . . . . . . . . . 44

6 Conclusion and Future Work 50

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A Appendix 53A.1 Defintions and Equations . . . . . . . . . . . . . . . . . . . . . 53

A.1.1 Average and Root Mean Square . . . . . . . . . . . . . . 53A.1.2 Efficiency . . . . . . . . . . . . . . . . . . . . . . . . . . 54A.1.3 Per-unit . . . . . . . . . . . . . . . . . . . . . . . . . . . 54A.1.4 Mutual inductance . . . . . . . . . . . . . . . . . . . . . 54

A.2 Power Quality Requirements . . . . . . . . . . . . . . . . . . . 55A.2.1 Conducted and radiated emission . . . . . . . . . . . . 55

A.3 Rectifier topologies . . . . . . . . . . . . . . . . . . . . . . . . . 57A.3.1 AT design topologies . . . . . . . . . . . . . . . . . . . . 57

A.4 Simulation parameters, values and models . . . . . . . . . . . 58

Bibliography 61

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List of Figures

1.1 Electrical system evolution in aircraft . . . . . . . . . . . . . . . 11.2 Power distribution of aircraft . . . . . . . . . . . . . . . . . . . 21.3 Electrical power distribution of B767 and B787 . . . . . . . . . 31.4 Electrical power distribution system in B787 and A380 . . . . 4

2.1 Phase sequence and phase shift of supply voltage . . . . . . . 82.2 Distortion spectrum for 270 VDC systems . . . . . . . . . . . . 102.3 Unity, lagging and leading power factor . . . . . . . . . . . . . 13

3.1 12-pulse ATRU . . . . . . . . . . . . . . . . . . . . . . . . . . . 173.2 ∆ and Y-switch rectifier . . . . . . . . . . . . . . . . . . . . . . 193.3 Switch-pair setup . . . . . . . . . . . . . . . . . . . . . . . . . . 193.4 The Vienna rectifiers . . . . . . . . . . . . . . . . . . . . . . . . 21

4.1 AT topology used in the design of simulation model . . . . . . 274.2 Simulation model of the AT with connections . . . . . . . . . . 284.3 Graphical representation and simulation model of the B6Us

and IPTs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 314.4 Block diagram of the control of the switches . . . . . . . . . . . 344.5 Sector detection with and without phase shift . . . . . . . . . . 37

5.1 12-pulse ATRU: Supply waveforms and FFT analysis of case 1 425.2 12-pulse ATRU: Supply waveforms and FFT analysis of case 1 435.3 ∆-switch: FFT analysis and over-modulation of case 4 . . . . . 475.4 ∆-switch: Supply waveforms and FFT analysis of case 6 . . . 485.5 ∆-switch: Supply waveforms and FFT analysis of case 8 . . . 49

A.1 Conducted emission limits . . . . . . . . . . . . . . . . . . . . . 55A.2 Ratiated emission limits . . . . . . . . . . . . . . . . . . . . . . 56A.3 CM and DM filter . . . . . . . . . . . . . . . . . . . . . . . . . . 56A.4 Different 18-pulse AT topologies . . . . . . . . . . . . . . . . . 57A.5 Zig-Zag AT topology . . . . . . . . . . . . . . . . . . . . . . . . 57A.6 Simulation model of voltage loop . . . . . . . . . . . . . . . . . 59A.7 Simulation model of carrier voltage generation . . . . . . . . . 59A.8 Simulation model of control loop . . . . . . . . . . . . . . . . . 60

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List of Tables

2.1 Voltage requirements for ACVF, ACCF and DC . . . . . . . . . 92.2 Power & frequency requirements for ACVF and ACCF . . . . 122.3 Definition of EMC ranges . . . . . . . . . . . . . . . . . . . . . 142.4 Current THD limits . . . . . . . . . . . . . . . . . . . . . . . . . 14

3.1 Comparison of ∆-switch, six-switch Vienna and 12-pulse ATRU 233.2 Comparison of B6U, 12-pulse ATRU and ∆-switch rectifier . . 23

4.1 Rating of coils in the AT . . . . . . . . . . . . . . . . . . . . . . 274.2 Rating of the IPT . . . . . . . . . . . . . . . . . . . . . . . . . . 324.3 Clamping action of the switches . . . . . . . . . . . . . . . . . . 36

5.1 12-pulse ATRU: Simulation cases . . . . . . . . . . . . . . . . . 395.2 12-pulse ATRU: Simulation result . . . . . . . . . . . . . . . . . 405.3 12-pulse ATRU: Simulation parameters . . . . . . . . . . . . . 415.4 ∆-switch: Simulation cases . . . . . . . . . . . . . . . . . . . . 445.5 ∆-switch: Simulation result . . . . . . . . . . . . . . . . . . . . 445.6 ∆-switch: Simulation parameters . . . . . . . . . . . . . . . . . 455.7 ∆-switch: Optimized system parameters . . . . . . . . . . . . 46

A.1 Rectifier bridge (B6U) parameters . . . . . . . . . . . . . . . . . 58A.2 Switch parameters . . . . . . . . . . . . . . . . . . . . . . . . . 58A.3 Default pu values . . . . . . . . . . . . . . . . . . . . . . . . . . 58

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List of Abbreviations

AC Alternating CurrentACVF Alternating Current Variable FrequencyACCF Alternating Current Constant FrequencyAPU Auxiliary Power UnitAPFC Active Power Factor Correction UnitAT Auto-TransformerATRU Auto-Transformer Rectifier UnitB6U Uncontrolled Three-phase Diode Bridge RectifierB6C Controlled Three-phase Thyristor/Transistor Bridge Rectifier circuitCE Conducted EmissionsCF Crest FactorCM Common ModeDC Direct CurrentDM Differential ModeEMC Electromagnetic CompatibilityEMI Electromagnetic InterferenceIPT Interphase TransformerIDG Internal Drive GeneratorL-L Line-to-LineL-N Line-to-NeutralMEA More Electric AircraftP ProportionalPF Power FactorPI Proportional IntegralPLL Phase Locked LoopP-P Peak-to-PeakP-V Peak-to-ValleyPWM Pulse Width ModulationRE Radiated EmissionsRF Ripple FactorRMS Root-Mean-SquareTHD Total Harmonic DistortionTRU Transformer Rectifier UnitVAC Volt Alternating CurrentVDC Volt Direct Current

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List of Symbols

Adist(f) distortion RMS value dB Vf supply frequency Hzfsw switching frequency HzFPI PI controllerCFV voltage crest factorCo DC-link capacitance FCs,D diode’s snubber capacitance FCs,S switch snubber capacitance FDF+ positive Vienna rectifier bridge diodeDF− negative Vienna rectifier bridge diodeDin negative ∆-switch bridge diodeDip positive ∆-switch bridge diodeDM+ positive Vienna rectifier midpoint switch diodeDM− negative Vienna rectifier midpoint switch diodeDN+ positive Vienna rectifier mains switch diodeDN− negative Vienna rectifier mains switch diodeg∗ reference conductance 1/Ωis,p measured L-N supply current phase p Ai∗s,p reference L-N supply current phase p Ais,A measured L-N supply current phase A Ais,B measured L-N supply current phase B Ais,C measured L-N supply current phase C Ai(t) instantaneous current A

I1 fundamental peak current A

Iipt peak current of IPT coil AIo output current AIRMS RMS current AIRMS,n RMS current of coil n AIRMS,ipt RMS current of IPT coil AIs RMS L-N supply current A

Is peak L-N supply current Aj imaginary unitk coefficient of couplingkboost ripple current factorkipt peak current of IPT related to output currentkn number of coils of type nKI,v integral coefficient of voltage controllerKP,i proportional coefficient of current controllerKP,v proportional coefficient of voltage controller

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La inductance of coil a HLb inductance of coil b HLbase base inductance HLc inductance of coil c HLd inductance of coil d HLe inductance of coil e HLf inductance of coil f HLipt inductance of IPT coil HLm,AT magnetizing inductance of AT HLm,pu magnetizing inductance pu valueLNi boost inductance HLon,D conduction inductance diode HLon,S conduction inductance switch HLpu inductance pu valuem modulation indexM midpoint of Vienna rectifierML mutual inductance HML,ipt mutual inductance of IPT HMR,ipt mutual resistance of IPT Hn type of coil in ATp notation of phase of voltage and currentP active power WPF power factorPin input active power WPo output active power WQ reactive power VArRa resistance of coil a ΩRb resistance of coil b ΩRbase base resistance ΩRc resistance of coil c ΩRd resistance of coil d ΩRipt resistance of IPT coil ΩRm,AT magnetizing resistance of AT ΩRm,pu magnetizing resistance pu valueRFV output voltage ripple factorRload load resistance ΩRon,D conduction resistance diode ΩRon,S conduction resistance switch ΩRpu resistance pu valueRs,D diode’s snubber resistance ΩRs,S switch snubber resistance ΩS complex power V A|S| apparent power V ASAT power rating of AT V ASbase base complex power V ASij switch ijSi switch i

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Si+ positive Vienna switch iSi− negative Vienna switch it time sT period (reciprocal value of f ) sTs sample time sTHDI current total harmonic distortionTHDV voltage total harmonic distortionvo measured output voltage Vv∗o reference output voltage Vvr P-P voltage ripple Vv∗rs,p reference L-N voltage at switch-pair phase p Vv∗r,ij reference L-L voltage over switch Sij Vv∗r,ji reference L-L voltage over switch Sji Vvs,p measured L-N supply voltage phase p Vvs,A measured L-N supply voltage phase A Vvs,B measured L-N supply voltage phase B Vvs,C measured L-N supply voltage phase C Vv(t) instantaneous voltage Vvtri triangular carrier voltage V

V peak voltage VV1 fundamental RMS voltage VVa(t) instantaneous voltage phase a VVb(t) instantaneous voltage phase b VVbase base voltage VVc(t) instantaneous voltage phase c VVf,D forward voltage diode VVf,S forward voltage switch VVn nth harmonic RMS voltage VVNi supply voltage APFCs VVs RMS L-N supply voltage V

Vs peak L-N supply voltage VVo output voltage VVo,AC output AC voltage VVAV G,o AVG output voltage VVRMS RMS voltage VVRMS,a RMS voltage of coil a VVRMS,b RMS voltage of coil b VVRMS,ipt RMS voltage of IPT coil VVRMS,n RMS voltage of coil n VVRMS,o,AC RMS output AC voltage VXbase base reactance Ωz complex variable of z-transformZbase base impedance ΩZbase,a base impedance of coil a ΩZbase,b base impedance of coil b ΩZbase,n base impedance of coil n ΩZbase,primary base impedance primary side of AT Ω

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Zbase,secondary base impedance secondary side of AT Ω

δa phase of voltage in phase a

δb phase of voltage in phase b

δc phase of voltage in phase c

φ phase shift between sinusoidal voltage and current

ω angular frequency rad/s

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Chapter 1

Introduction

1.1 Background

The power of an aircraft is obtained from the turbines (the engines of the air-craft) and can be divided in two separate parts where the first is the propul-sive part and the second is the non-propulsive part. The non-propulsivepart is used for different kinds of applications such as lights, cabin pressure,avionics, flight control and fuel pumps. The distribution of the power tothese applications is on a conventional aircraft, such as Airbus A330 and Boe-ing B767, carried out by a combination of hydraulic, electric, pneumatic andmechanical systems.

FIGURE 1.1: Evolution of the electrical power system in aircraft[1].

In recent years the development of the electrical power system have mademuch progress as seen in figure 1.1. A reason for this is the More Electric Air-craft (MEA) concept, introduced in the early nineties [2]. The first intention of

Chapter 1. Introduction 2

the concept was to reduce the overall weight, lower the maintenance costs,increase the reliability and to improve the performance of military aircraftby increasing the use of electronic equipment instead of heavy mechanical,pneumatic or hydraulic driven elements. With the increased capacity andrating of generators of civil aircraft the MEA concept also applied there. To-day, the objective is to completely replace the systems that in earlier aircraft(both military and civil) have been hydraulic, pneumatic or mechanical sys-tems with electrical systems [1]. An overview of the power distribution ofthe non-propulsive power of a conventional aircraft and a MEA is presentedin figure 1.2a and 1.2b, respectively. In conventional aircraft, bleed air (com-pressed air) is taken from the turbine and used for pneumatic applications(e.g. heating and pressurization of the cabin) and considerably reduces theefficiency of the turbine [2]. Hence, in the course of the MEA concept, a nobleed system is preferable which in turn result in a higher demand of newelectrical systems to supply power to these applications and loads.

(A)

(B)

FIGURE 1.2: Power distribution of the non-propulsive power of(A) a conventional aircraft and (B) a MEA [2].


The electrical power system in both civil aircraft and in military aircraft isoften based on a parallel connection of turbine-driven generators. In con-ventional (not MEA) and military aircraft the turbines are connected to inte-grated drive generators (IDG) via a mechanical gearbox resulting in a three-phase AC voltage with a constant frequency (ACCF) of 400 Hz and a rootmean square (RMS) line-to-normal (L-N) voltage of 115 V as output to themain AC bus. The latest electrical power distribution system based on theMEA concept is used in Airbus A380 and Boeing B787. An overview of a con-ventional (B767) and a MEA (B787) electrical power distribution system ispresented in figure 1.3 and in figure 1.4 a block representation of an electricalpower system used in a MEA is presented. The MEA electrical power systemincludes six generators, two per turbine and two per auxiliary power unit(APU). The main generators (the ones connected to the turbines) are eitherof permanent magnet synchronous machine (PMSM) or switched reluctancemachine (SRM) type. The aggregated rating of them is reaching up to 1 MVA.The main generators outputs (to the main AC bus), a three-phase AC voltagewith a variable frequency (ACVF) ranging between 360-800 Hz and a RMSL-N voltage of 115 V for the A380 and 230 V for the B787. The APU is alsoconnected to the main AC bus and is activated in case of an emergency andfor power supply on ground while the engines are not running. The variablefrequency of the main AC bus is a result of the elimination of the mechanicalgearbox between the turbine and the generators in accordance to the MEAconcept. This results in a more reliable generation with less maintenance andlower costs compared to the complex constant speed drive generation usedby the IDG in conventional aircraft [3]. The direct connection of the genera-tors to the turbine also allows the generator to act as a starter motor for theturbine, reducing the overall weight of the aircraft [2].

FIGURE 1.3: Electrical power distribution of the conventionalB767 (left) and the MEA B787 (right) [3].


FIGURE 1.4: Electrical power distribution system in MEA B787and A380 [1].

The APU and the redundancy of the system makes sure that the most impor-tant components of the aircraft have power even under emergencies and fail-ures. Still, the reliability and the performance of the elements in the system isof great importance. The generation and the conversion of power needs to berobust and deliver a high power quality. There are several aircraft standardsincluding standards created by aircraft companies which electrical powersystems in aircraft need to comply with. The two general industrial stan-dards are the DO-160G [4] used for civil aircraft and the MIL-STD-704F [5]used for military aircraft (also applicable on older/conventional civil air-craft). As seen in figure 1.4, the modern MEA electrical power distributionsystem consist of several voltage buses supplying several different loads.To obtain the voltage for each bus different kinds of converters are needed.Commonly, the 28 VDC bus is obtained by a transformer rectifier unit (TRU),the secondary AC bus at 115 VAC (not necessary in A380) is obtained by anauto-transformer unit (ATU) and finally the 270 VDC bus (or in the B787 a540 VDC bus) is obtained by a auto-transformer rectifier unit (ATRU). Theloads of each bus differ but common loads are actuators used for both pri-mary and secondary flight-control [2]. The actuators load demand varieswith the mission profile and are often in need of a low continuous powersupply during flight and a high peak power supply during starting and land-ing of the aircraft.

The last mentioned conversion, i.e. the rectification of the 115/230 VAC sup-ply to 270/540 VDC, is a part of a product that Saab [6] offers to their cus-tomers. In this thesis an investigation of this type of rectification on aircraft iscarried out and the focus will be on evaluating (in a simulation environment)the fulfillment of the power quality requirements according to the mentionedstandards. Both passive and active topologies are considered, discussed andcompared. Simulation models are designed and presented. Lastly, a modu-lar concept is discussed where different power supplies and loads are con-sidered.


1.2 Objective and Goal

The different types of power supplies used in aircraft leads to new design so-lutions for each converter case which in turn drives cost. Therefore, it wouldbe advantageous to keep certain design elements (e.g. inductors, capacitorsand ratings) common independently of input power and obtain a modularconcept. Further on, as the development of power systems and electronicsmoves forward in accordance to the MEA concept the topology used for rec-tification today has to be evaluated considering power quality, weight andreliability. Hence, the goal of the master thesis is to investigate topologiesfor rectification on-board aircraft and perform simulations in order to evalu-ate power quality and establish to what extent design elements can be keptcommon for a rectifier used for aircraft utility equipment, i.e if a modularconcept can be obtained. Both short duration (peak power) drive, continu-ous drive and different power inputs are simulated. Figures of merit suchas weight, reliability and cost should also be considered and finally futurerectifier topologies and aspects should be discussed.

1.3 Methodology

First, a literature study and a market research was done to get an insightin existing airborne power system and more specific in the topologies usedto rectify AC to DC in aircraft. The information was obtained from whitepapers, articles, journals and informative web pages but also from earlierprojects and internal documents at Saab. Three topologies for rectificationwere investigated further and compared regarding price, size, weight, relia-bility and performance.

From analysis and evaluation of the literature study and the market researchtwo of the three investigated topologies were chosen to be modeled for sim-ulation in MATLAB/Simulink. Simulations with different input voltage andfrequency were done to establish to what extent components could be keptcommon, independently of the input. To validate the performance of themodels, power quality requirements and characteristics defined in two stan-dards (one for civil and one for military aircraft) were used.

Finally, the simulation result was evaluated and summarized to conclude ifa common modular design concept could be obtained and how good thepower quality was. Future aspect and topologies were discussed and anoverall evaluation of the whole procedure of the master thesis was done.

1.4 Delimitations

The thesis only investigates the rectification for following input power sources

• Variable frequency 360-800 Hz, 230 VAC (L-N, RMS)


• Constant frequency 400 Hz, 115 VAC (L-N, RMS)

The output voltage and power range are limited to

• 270 VDC and 540 VDC

• Continuous power ≤ 2 kW

• Peak power ≤ 10 kW

Due to secrecy of the aircraft companies, the models are only evaluated usingthe following general standards

• DO-160G

• MIL-STD-704F

1.5 Disposition

The power quality requirements obtained from the aircraft standards aresummarized and presented in chapter 2. The outcome of the literature study,the market research and a summary of rectifier topologies in aircraft are pre-sented in chapter 3. The most promising rectification topologies from chap-ter 3 are designed for simulation and each step of the modeling are presentedin chapter 4. The result of the simulations, both during validation and withdifferent inputs, are discussed and presented in chapter 5. Finally, a conclu-sion is drawn considering the simulation result and the overall thesis out-come. Also future aspects and work are discussed in chapter 6.

7

Chapter 2

Power Quality Requirements ofAircraft

2.1 Standard MIL-STD-704F and DO-160G

The power quality requirements and characteristics used in today’s aircraftare described in MIL-STD-704F [5] and DO-160G [4] and are both widely usedfor verification of electrical applications. The MIL-STD-704F standard is formilitary and older civil aircraft and is more strict when it comes to voltagecharacteristics compared to the DO-160G standard but does not cover EMCand voltage spikes. The DO-160G standard is for newer civil aircraft and ismore comprehensive and covers test procedures as well. An extract from thescopes of the standards are presented as follows.

MIL-STD-704FThis standard establishes the requirements and characteristics of aircraft electricpower provided at the input terminals of electric utilization equipment. The pur-pose of this interface standard is to ensure compatibility between the aircraft electricsystem, external power, and airborne utilization equipment.

DO-160GThis document defines a series of minimum standard environmental test conditions(categories) and applicable test procedures for airborne equipment. The purpose ofthese tests is to provide a laboratory means of determining the performance charac-teristics of airborne equipment in environmental conditions representative of thosewhich may be encountered in airborne operation of the equipment.

In the next three sections, requirements stated in the standards concerningvoltage, frequency, power, current and quality in VAC (ACVF and ACCF)and VDC are summarized and the parameters and limits used for the devel-opment and validation of the simulation models are presented.

Chapter 2. Power Quality Requirements of Aircraft 8

2.2 Voltage requirements

The voltage supply should be a three-phase grounded neutral system with aphase sequence of A-B-C as illustrated in figure 2.1 and the supply voltageshould be a sine wave fulfilling the specified voltage quality requirementsin the standards. To get an overview of these requirements a summary anda comparison between the two standards for VAC and VDC are presentedin table 2.1. To obtain an estimation of the limits for 230 VAC the values intable 2.1 could be multiplied by two [4]. It is assumed that a multiplicationby two of the 270 VDC limits in table 2.1 is a valid estimation for the 540 VDClimits as well.

FIGURE 2.1: Phase sequence and phase shift of supply voltagewith a counterclockwise phase rotation [5].

Some of the main voltage quality requirements for VAC is the distortion fac-tor and crest factor. The distortion factor of VAC is equal to the total harmonicdistortion of the voltage (THDV ). The THD is defined as the ratio of the RMSvalue of the harmonics (i.e exclusive of fundamental) to the RMS value of thefundamental [4], [5]. The THDV is formulated in equation 2.1. For a puresine wave the THD is equal to zero since no harmonics are present.

THDV =

√∑∞n=2 V

2n

V1· 100% (2.1)

V1 = fundamental RMS voltage

Vn = nth harmonic RMS voltage

The crest factor of the voltage (CFV ) is defined as the ratio of the absolutepeak value to the RMS value. A pure voltage sine wave has a CF equal toVRMS

√2

VRMS=√

2 and a pure DC voltage (no peaks) has CF equal to 1. The CFVis formulated in equation 2.2.


TABLE 2.1: Steady state voltage requirements for ACVF, ACCFand DC [4], [5].

ACVF (115 V, 360-800 Hz) MIL-STD-704F DO-160GRMS L-N Supply voltage 108-118 V 100-122 VRMS voltage unbalance ≤3.0 V ≤8.0 VRMS voltage modulation ≤2.5 V ≤5.0 V, P-VVoltage phase difference 120±4 120±6

Distortion factor ≤5% ≤10%Crest factor 1.41±0.10 1.41±0.15DC component ±0.10 V ±0.20 V

ACCF (115 V, 400 Hz) MIL-STD-704F DO-160GRMS L-N Supply voltage 108-118 V 100-122 VRMS voltage unbalance ≤3.0 V ≤6.0 VRMS voltage modulation ≤2.5 V ≤5.0 V, P-VPhase difference 120±4 120±4

Distortion factor ≤5% ≤8%Crest factor 1.41±0.10 1.41±0.15DC component ±0.10 V ±0.20 V

DC (270 V) MIL-STD-704F DO-160GDC Voltage 250-280 V 220-320 VDistortion factor 0.015 N/ARipple amplitude ≤6.0 V ≤8.0 V

CF =|V |VRMS

(2.2)

V = peak voltageVRMS = RMS voltage

For VDC, ripple is the main topic when concerning voltage quality (in ad-dition to correct voltage amplitude). Ripple is defined as the variation ofvoltage about the steady state DC voltage during steady state operation [5].The requirements stated in the two standards covers distortion factor, dis-tortion spectrum and ripple amplitude. The distortion factor is equal to theripple factor of the output voltage of the rectifier (RFV ). The RF is definedas the ratio of the RMS value of the alternating voltage component on theDC voltage to the DC steady state voltage [4], [5]. The RFV is formulatedin equation 2.3. The ripple amplitude Vripple is the maximum absolute valueof the difference between the steady state and the alternating voltage com-ponent on the DC voltage [5]. The Vripple is formulated in equation 2.4. Theallowed ripple amplitude of a certain frequency component is defined in thedistortion spectrum illustrated in 2.2. With use of the spectrum the allowed


peak-to-peak voltage ripple (vr) at a switching frequency fsw can be calcu-lated as in equation 2.5. This is used later for a first approximation of theDC-link capacitance in chapter 4, section 4.3.

RFV =VRMS,o,AC

Vo(2.3)

Vripple = |Vo,AC | (2.4)

Vo,AC = Vo − VAV G,oVo,AC = output AC voltage

Vo = output voltageVAV G,o = AVG output voltage

VRMS,o,AC = RMS output AC voltage

FIGURE 2.2: Maximum distortion spectrum for 270 VDC sys-tems [5].

vr = 2√

2 · 10Adist(fsw)

20 (2.5)

Adist(fsw) = distortion RMS amplitude


2.3 Power and Frequency requirements

The power in an electrical system can be described using different quanti-ties. For an AC-system, the complex power is often considered. The complexpower (S) is the complex sum of the active and reactive power and is definedas in equation 2.6 [7].

S = P + jQ (2.6)

P = active powerQ = reactive powerj = imaginary unit

The apparent power (|S|) is for an AC system the magnitude of the complexpower but can also be expressed for systems with distortion via the RMSvoltage and the RMS current as in equation 2.7 [4].

|S| = VRMSIRMS (2.7)

VRMS = RMS voltageIRMS = RMS current

For AC-systems, the active power (P ) and the reactive power (Q) is the realand imaginary part of the complex power, respectively, as seen in equa-tion 2.6. The formulas defining these two are expressed in equation 2.8 andequation 2.9, respectively [7].

P = |S| cos(φ) (2.8)

Q = |S| sin(φ) (2.9)

|S| = apparent powerφ = phase shift between sinusoidal voltage and current

The active power is generally defined as the average of the product of theinstantaneous voltage and current, as expressed in equation 2.10

P =1

T

∫ T

0

v(t)i(t)dt (2.10)

T = period (reciprocal value of f )v(t) = instantaneous voltagei(t) = instantaneous currentt = time

The power factor (PF) is defined as the ratio of the active power (P ) flowingto the load to the apparent power (|S|) in the system, as expressed in equa-tion 2.11. It describes how efficiently the system transfer the active powerand since aircraft power system does not allow bidirectional power flow thePF can be a value between 0 and 1 (if bidirectional power flow is allowed thevalue is between -1 and 1) [4], [5]. How the PF affects the system is easier to


see if equation 2.7 is inserted in equation 2.11 and is rewritten with regardsto the current, as in equation 2.12.

PF =P

|S|(2.11)

IRMS =P

VRMSPF(2.12)

P = active power|S| = apparent power

VRMS = RMS voltageIRMS = RMS current

Equation 2.12 shows that a PF close to unity (1) will lead to a lower RMS-value of the current which in turn implies lower ohmic losses and that is whyunity power factor is desired in electrical systems [8]. Unity PF is achievedwhen there is no distortion and the phase shift between voltage and currentis equal to zero and this occurs when the system behaves essentially like aresistor. If the phase shift is negative the PF is said to be lagging and this oc-curs when the system is inductive (current is lagging the voltage). Similarly,the PF is said to be leading if the phase shift is positive and occurs when thesystem is capacitive (current is leading the voltage). A PF equal to unity, alagging PF and a leading PF are illustrated in figure 2.3.

A summary and a comparison between the two standards concerning powerand frequency requirements for VAC are presented in table 2.2. The PF valuefrom MIL-STD-704F is when operating at 50 percent or more of its rated loadcurrent and for a load greater than 500 VA. When operating at more than100 VA the system shall not have a leading PF [5]. The PF value from DO-160G is during full load and for a load greater than 150 VA [4].

TABLE 2.2: Steady state power and frequency requirements forACVF and ACCF [5][4].

ACVF (115 V, 360-800 Hz) MIL-STD-704F DO-160GFrequency 360-800 Hz 360-800 HzPower Factor (Lag) ≥0.85 ≥0.8Power Factor (Lead) 1 ≥0.968

ACCF (115 V, 400 Hz) MIL-STD-704F DO-160GFrequency ±7 Hz ±10 HzPower Factor (Lag) ≥0.85 ≥0.8Power Factor (Lead) 1 ≥0.968


FIGURE 2.3: Unity, lagging and leading power factor.

For a discretized system there is a sufficient condition to make sure that thediscrete sequence of samples capture all of the information from a continuous(time) signal of defined bandwidth. This condition is defined by the Nyquist-Shannon sampling theorem [9]. For a certain switching frequency the sampletime should be according to equation 2.13 to avoid aliasing.

Ts ≤1

2fsw(2.13)

Ts = sampling timefsw = switching frequency

2.4 Current and EMC requirements

There are different types of emission that have to be considered when deal-ing with power electronics on aircraft. The requirements concerning theseemissions are divided in three frequency ranges as presented in table 2.3.

The requirements regarding frequencies up to 150 kHz are covered by thecurrent harmonics limitations, presented in table 2.4. The limits are definedup to 40th harmonic which correspond to a frequency of 32 kHz considering


TABLE 2.3: Definition of frequency range and requirement con-cerning EMC [4].

Frequency range [MHz] Requirement

0 - 0.150 Current harmonics0.150 - 100 Conducted emission100 - 6000 Radiated emission

the maximum supply frequency of 800 Hz, hence limits of current harmon-ics above 32 kHz are not defined. The second frequency range deals withthe conducted emission (CE) and covers the common mode (CM) and differ-ential mode (DM) currents and the last range is the radiated emission (RE).Limits of the CE and RE are presented in appendix A.2, figure A.1 and A.2,respectively.

TABLE 2.4: Limits of harmonics in phase currents [4]. I1 is theamplitude of the fundamental peak current.

Harmonic Order Limit

3rd, 5th, 7th 0.02I1Odd Triplen Harmonics (n = 9, 15, 21, ..., 39) 0.1I1/n

11th 0.1I113th 0.08I1Odd Non Triplen Harmonics 17, 19 0.04I1Odd Non Triplen Harmonics 23, 25 0.03I1Odd Non Triplen Harmonics 29, 31, 35, 37 0.3I1/n

Even Harmonics 2 and 4 0.01I1/n

Even Harmonics > 4 (n = 6, 8, 10, ..., 40) 0.0025I1Harmonics > 40 N/A

The occurrence of emission disturbances in power equipment is inevitabledue to the fact of non-ideal components. To suppress and minimize the dis-turbance the design and placement of components and modules has to beconsidered. This includes the choice of using surface mounting or throughhole mounting, minimizing cross-section areas of sensitive nodes and loca-tion of components such as X and Y capacitors (described below), just tomention a few [10]. Apart from the design and placement, current harmon-ics are suppressed by harmonic injection and active/passive filter, conductedemission with the help of EMI filter and radiated emission by shielding.

The EMI filter often suppress both CM and DM currents. The part of theEMI filter suppressing CM currents (CM filter) often consists of a choke coiland a line bypass capacitor (Y) connected to the chassis ground and sup-presses disturbances which are conducted on all lines in the same direction,sometimes referred to as asymmetrical interference. The part of the EMI filter


suppressing DM currents (DM filter) often consists of a across-the-line capac-itor (X) and suppresses disturbances which are conducted on the signal lineand ground line in the opposite direction to each other, often referred to assymmetrical interference [11][10]. A graphical representation of a CM andDM filter are illustrated in appendix A.2, figure A.3.

16

Chapter 3

Rectifier Topologies in Aircraft

To meet the requirements of today’s aircraft standards basic rectifier topolo-gies such as the uncontrolled (passive) three-phase diode bridge rectifier(B6U) and the controlled (active) three-phase thyristor/transistor bridge rec-tifier (B6C) are not sufficient without any additional circuits [12]. The powerquality of the B6U is too low due to the high input current THD and low PFand the B6C is a bidirectional rectifier meaning it allows power flow in bothdirections which is prohibited in aircraft. Additionally, the reliability of theB6C is too low due to the risk of a shoot-through resulting in a short circuitof the DC-link [12]. These disadvantages and prohibitions can be avoided byimplementing additional circuits and complex rectification topologies. Foraircraft, the conventional topology to rectify AC to DC today is the passivetransformer rectifier unit (TRU) and in the latest aircraft such as B787 andA380 an auto-transformer rectifier unit (ATRU) is used [1]. In the future, ac-tive solutions such as the active power factor correction unit (APFC) mightbe an alternative as well.

The ATRU utilize the B6U together with an auto-transformer (AT), interphasetransformers (IPT) and a DC-link capacitor resulting in a robust system withimproved power quality compared to the basic B6U [12].

The APFC can be implemented using different design topologies: ∆-switch,Y-switch, Vienna, Buck, Boost and SWISS just to mention a few. The typicallyactive rectifiers are comprised of a B6C and a DC-link capacitor where theB6C is controlled to obtain a unity power factor at the supply. The APFCtopologies are often instead based on B6Us together with a switch-configura-tion. The slightly more complex design and control can still assure that noshoot-through occurs and that a constant DC-link voltage together with aunity power factor are obtained.

In the following sections the ATRU and the most promising APFC for air-craft rectification according to [2], [12]–[15] are investigated. Also, a sum-mary where the discussed topologies are compared and evaluated consider-ing power quality, weight, reliability and cost will be presented.

Chapter 3. Rectifier Topologies in Aircraft 17

3.1 Auto-transformer rectifier unit

The ATRU is a passive rectifier comprised of an AT, one or several B6Us de-pending on topology and two IPTs. According to [12] the employment of anAT instead of an isolated transformer reduces the weight and volume signif-icantly. As mentioned previously, there exist several design topologies of theAT, ranging from 6 to 30-pulses using winding configurations such as zig-zag, ∆ and hexagon just to mention a few. In this thesis the focus will be onthe 12-pulse topology and for that topology, the conventional winding con-figuration is the ∆-configuration. An overview of a full 12-pulse ATRU sys-tem based on this winding configuration is illustrated in figure 3.1. In addi-tion, three examples of 18-pulse AT topologies based on the ∆-configurationand a 12-pulse AT topology based on zig-zag configuration are presented inappendix A.2.1, figure A.4 and A.5, respectively. Further on, only the 12-pulse topology will be discussed.

FIGURE 3.1: 12-pulse ATRU based on a ∆-configured AT [16].

The AT divides the three-phase input A, B, C into two separate three-phasesystems 1,3,5 and 2,4,6, with a 30 phase shift. The two systems are rectifiedby the B6Us and then switched in parallel using IPT#1 and IPT#2 resultingin a DC-voltage with a 12-pulse ripple on the output [12]. The 5th and 7thharmonics of the input current can be significantly reduced in an ATRU. Howeffectively the reduction is depends on the design topology and how close torated power it is operating at [17]. Some of the challenges in the constructionand design procedure are

• Choosing winding material

• Selecting modular shape

• Defining and optimizing core geometry and aspect ratio

• Optimizing interactions between windings (leakage inductance, prox-imity effects)

To maximize the cancellation of the harmonics, fractional turns are requiredwhich in reality are approximated by full turns. Further, the ∆-configuration


can be used to circulate the triple harmonics produced in the rectificationprocess. Design and topology decision is explained and evaluated in [16]and [12] and therefore shall be omitted here for the sake of brevity.

The main advantages of the ATRU are the reliability and robustness due tothe lack of active electronic components. It also shows high overload capacitywhich is advantageous when it comes to applications requiring high peak-to-continuous power demand [12], [14], [18]. According to [16], industrial andaerospace power quality requirements, such as DO-160, can be satisfied witha 12-pulses ATRU. Additionally the ATRU does not need a common-modefilter at the input, due to the high common mode impedance [12], [17].

The main disadvantages of the ATRU are the weight and the unregulatedoutput voltage. The high weight is due to the auto-transformer which ac-counts for approximately 50% of the total weight [12]. The unregulated out-put voltage is due to the passive attribute, which in turn results in voltagedroops as the ATRU gets loaded. The droop is a function of the input sourceimpedance and the leakage reactance of the transformer and can result insubstantial voltage droops during full load [17]. The ATRU does neither of-fer soft start nor over current protection and an input differential-mode EMI-filter needs to be added [17], [18]. Additionally, the power factor of a passivesolution, such as the ATRU, cannot be controlled. Without EMI-filter, thepower factor is a function of loading and is typically lagging for an ATRU.With the EMI-filter the power factor can be made to range from leading tolagging depending on load, though leading power factor is not desired inpower systems [17].

3.2 Active power factor correction

3.2.1 ∆-switch and Y-switch rectifier

The ∆-switch and the Y-switch rectifier are both active three-phase two-levelunidirectional boost rectifiers. An circuit overview of the ∆-switch and theY-switch is presented in figure 3.2a and 3.2b, respectively. The topologiesare comprised of one inductor LNi per phase, a B6U Dip, Din with a bidi-rectional (current) bipolar (voltage) switch-configuration Sij (i, j ∈ 1, 2, 3)connected between the two diodes of the B6U on each phase-leg and finally aDC-link capacitor Co. The switch-configuration can be realized by two tran-sistors (a switch-pair) in series on each phase-leg, as illustrated in figure 3.3.The switch-pair on each phase-leg are connected in a ∆-coupling betweenthe phases for the ∆-switch and in a Y-coupling between the phases for theY-switch, hence the names. The individual switches are denominated as (topto bottom, left to right) S12, S21, S23, S32, S31 and S13. In [12] it is stated thatthe conduction losses of the ∆-switch topology are lower compared to theY-switch topology and therefore only the ∆-switch topology is consideredfrom now on.


(A)

(B)

FIGURE 3.2: APFC topologies (A) ∆-switch and (B) Y-switchboth with metal oxide semiconductor field effect transistors

(MOSFETs) as switching devices [2].

FIGURE 3.3: Switch-pair setup where i, j ∈ 1, 2, 3 denoteseach individual switch [2].

The three-phase input VNi is rectified by controlling the three switch-pairsSij and Sji in a manner that keeps the DC-link voltage constant and the in-put power factor as close to unity as possible. The diodes in the rectifier Dip

and Din make sure a short-circuit of the DC-link does not occur but on theother hand they have to commutate at switching frequency [2]. In order tominimize switching losses due to this, diodes with a low reverse-recoverycurrent are essential. In addition, choosing control strategies for the switcheshas to be done carefully. A hysteresis control may increase the effort of EMI-filtering and a constant switching frequency control may result in too highsupply current THD if the switching frequency is too low [15]. A survey cov-ering different control strategies is presented in [19] and therefore shall beomitted here for the sake of brevity.


The presence of the inductors on the AC side has an impact on several con-verter properties. It gives a continuous input current and the attribute of aboost rectifier which implies an output voltage Vo with a relation to the sup-ply voltage VNi as expressed in equation 3.1 [2]. The right hand side is thepeak value of the line-to-line supply voltage. Having an output voltage of atleast this peak-line-to-line supply voltage will guarantee that the system doesnot get over-modulated. If over-modulation occur, it could lead to distortionin the supply current [2].

Vo ≥√

3√

2VNi (3.1)

Hence, the inductors are often referred to as boost inductors. The total weightof the ∆-switch rectifier is mainly determined by them, since they are approx-imately 50% of the total weight [12]. However, the capacitor needed in theEMI-filter is small due to the presence of the boost inductors [15].

The main advantages of the ∆-switch rectifier are the active attribute en-abling a power factor at unity and a constant DC-link voltage enhancing theoverall power quality of the rectifier [12]. Additionally, the ∆-switch rectifierhas a high reliability and low complexity compared to other APFC topolo-gies and can handle a phase loss without changing controller structure [15].The main disadvantage of the ∆-switch rectifier, or APFC topologies in gen-eral, is the EMI topic [12]. They need both DM and CM-filtering, which isa disadvantage concerning weight and volume. However, it is thought thatthe development and improvement of power electronics and computationalpower of digital signal processors will enable a reduction of filter compo-nents [17]. In addition, boost solutions require additional circuitry for softstart-up to minimize the inrush current [13].

3.2.2 Vienna rectifier

The Vienna rectifier is an active three-phase three-level unidirectional boostrectifier [20]. A circuit overview is presented in figure 3.4a. The topology iscomprised of one inductor LNi per phase with the same purpose as the onesused in the ∆-switch rectifier, a B6U DF+, DF− with an advantageously inte-grated bidirectional switch-configuration Si (i ∈ 1, 2, 3) on each phase-legand finally two DC-link capacitors Co at the output creating the three-levelproperty.

The switch-configuration is comprised of a transistor as switch and a diodebridge. The diode bridge is comprised of two diodes DN+, DN− connected tothe supply and two diodesDM+, DM− connected to the midpointM betweenthe output capacitors Co. During operation, the current flows through twodiodes for every switching state which implies conduction losses. In order toreduce the losses a topology called six-switch Vienna rectifier can be used [2].A circuit overview of this topology is presented in figure 3.4b. Compared tothe original Vienna rectifier the diodesDM+,DM− in the switch-configurationconnected to the midpoint M and the switch Si are replaced by a switch-pair


Si+, Si−. Considering all three phases, it sums up to a total of six switches,hence the name six-switch Vienna rectifier. Considering the reduced conduc-tion losses the focus will be on the six-switch topology from now on.

(A)

(B)

FIGURE 3.4: APFC topologies (A) original Vienna and (B) six-switch Vienna with MOSFET as switching device [2].


The operation of the six-switch Vienna rectifier uses the same approach as the∆-switch rectifier, i.e the rectification, the constant DC-link voltage and theunity power factor are achieved by controlling the switch-pairs. The diodesconnected to the supply DN+ and DN− are commutated with supply fre-quency meanwhile the diodes in the B6U DF+ and DF− commutates withswitching frequency. Similarly to the ∆-switch rectifier, diodes with a lowreverse-recovery current are needed for the B6U, whereas for all other diodesstandard rectifier diodes can be applied [2]. Due to the boost attribute, theminimal output voltage Vo in relation to the supply voltage VNi is also for thistopology as stated in equation 3.1, according to [2]. Finally, the main advan-tages and disadvantages of the six-switch Vienna rectifier are also similarly tothe ones for the ∆-switch topology. Though, according to [15] the efficiencyof the six-switch Vienna rectifier is slightly lower compared to the ∆-switchrectifier for a certain power supply and load as presented in table 3.1. The ef-ficiency here and further on in the report refers to the efficiency of a electricalsystem as defined in appendix A.1.

3.3 Summary and Comparison

In [15] a comparison between a constructed ∆-switch and six-switch Viennarectifier is carried out and the outcome is presented in table 3.1. In the sametable, values obtained for a 12-pulse ATRU from [18] are also presented. Aslightly higher efficiency for the ∆-switch topology is achieved comparedto the Vienna topology given the specified supply and load. Though, [15]also state that the six-switch Vienna rectifier might be more efficient for ahigher supply voltage (e.g. 230 VAC). The ATRU shows the highest efficiency,though the given value is for different supply voltage and output voltagethan for the active topologies. The biggest different can be seen on the sup-ply current THD between the active and passive topology. This is also veri-fied in a comparison carried out in simulation environment of a conventionalB6U, a 12-pulse ATRU and a ∆-switch rectifier done in [12]. The results arepresented in table 3.2. Apart from the power quality which supply currentTHD is a major part of, there is a close race between the passive topology(ATRU) and the active ∆-switch topology. Considering the development inpower electronics which will improve the reliability and lower the cost, theactive solutions will get even better. Meanwhile, a possibility to improvethe ATRU by development of the magnetic material technology is unlikelyin the near future [17]. But one possible way to improve it is to increase theweight and implement an 18-pulse ATRU instead, which would especiallyimprove current THD [18]. One can also see that a B6U is the overall bestsolution of the three when disregarding the power quality. However, powerquality is of big importance. As mentioned in chapter 2, a low THD and apower factor close to unity results in higher efficiency. The power quality isespecially essential when considering the MEA concept which will lead to anincrease in electric power systems which in turn implies that a good powerquality is important for the overall system performance. Further on, com-plexity of a topology generally determines the cost and failure rate. That is


why the passive solutions are a better choice cost-wise and reliability-wisesince active solutions is in need of semiconductor devices such as transistors,additional control and gate drive circuitry. That active solutions at the mo-ment are more expensive than passive ones is also verified in [17]. Finally, in[13] a comparison between a six-switch Vienna rectifier and a 12-pulse ATRUis carried out and the result shows a higher efficiency for the ATRU but witha higher weight for a supply voltage of 115 V and an output power of 5000 W.

One of the reasons for the ATRU being the commonly used rectifier topologytoday is (in addition to the high reliability and efficiency) that it has, as de-scribed earlier, a capability to provide current in excess of its nominal value.This is used in applications where the peak-to-continuous demand is highwhich is the case of actuators on aircraft. The active solution is in this caselimited by the durability of the switching devices and the power rating ofthe rectifier has to increase for application where overloading capability isneeded [17].

TABLE 3.1: Comparison of ∆-switch rectifier, six-switch Vi-enna rectifier and 12-pulse ATRU. Values are obtained from[2][15][18], and values that were not available in the reports are

marked with "-".

∆-switch Vienna 12-pulse ATRU

Supply L-N RMS voltage [V] 115 115 230Supply frequency [Hz] 400 400 400Switching frequency [kHz] 72 72 passiveOutput voltage [V] 400 400 270Output power [kW] 5.0 5.0 4.5Supply current THD 2-3% - 11%Power factor 0.999 - 0.986Efficiency 94.9% 94.4% 95.0%Input EMI-filter Yes Yes NoPower density [kW/dm3] 1.91 1.91 0.995Power weight ratio [kW/kg] 1.32 1.32 1.82

TABLE 3.2: Comparison of B6U, 12-pulse ATRU and ∆-switchrectifier [12]. The topologies are evaluated from very good (++)

to very bad (- -).

B6U 12-pulse ATRU ∆-switch

Power quality - - + ++Weight ++ - - -Reliability ++ + - -Cost ++ o -Efficiency ++ + -


In summary, the active topologies ∆-switch and Vienna offer today betterpower quality but a slightly lower efficiency compared to the ATRU. Thecost, the reliability and the ability to vary the output power with ease is infavor of the ATRU, but in the future the active topologies might replace theATRU. Considering these comparisons, the models to simulate is chosen andthe design of them is presented in the next section.

25

Chapter 4

Design of Simulation Models

Due to time limitations not all the topologies presented in chapter 2 weremodeled and simulated. The 12-pulse ATRU and the ∆-switch rectifier werechosen to study in more detail. The reason for this choice was that the ATRUwas the topology that Saab uses at the moment and that the ∆-switch recti-fier has a slightly higher efficiency for an input voltage of 115 V accordingto [15]. In all simulations EMC, net inductance and resistance, transient be-havior, saturation of the AT and start-up circuitry to minimize inrush currentare neglected due to time limitation and secrecy. The modeling and all simu-lations are carried out in MATLAB/Simulink using the Simscape Power System- Specialized technology toolbox.

4.1 Power supply

The power supply was in all simulations modeled as a balanced three-phaseYg-connected sinusoidal voltage source using the Three-Phase Source block.The phase voltages (Va(t), Vb(t), Vc(t)) generated from the block can mathe-matically be expressed as in equation 4.1.

Va(t) = Vs sin(ωt+ δa)

Vb(t) = Vs sin(ωt+ δb)

Vc(t) = Vs sin(ωt+ δc)

(4.1)

Vs =√

2Vs

Vs = peak L-N supply voltageVs = RMS L-N supply voltageω = angular frequency = 2πf

f = supply frequencyδa = phase of voltage in phase aδb = phase of voltage in phase bδc = phase of voltage in phase c

For each simulation the supply voltage (amplitude and phase) and the fre-quency were set according to the requirements specified in chapter 2, sec-tion 2.2 and 2.3 respectively.

Chapter 4. Design of Simulation Models 26

4.2 Load

The rectifier generally has an inverter driven AC motor as load which in turndrives an actuator. The characteristics of varying switches, torque and powerof the inverter and motor was disregarded due to time limitation and com-plexity. Instead, the load was in all simulations modeled as a purely resistiveload by using the Series RLC Branch block. The simplified load characteris-tics of the motor were set (by Saab [6]) to be 2000 W continuously and 10000W peak with an output voltage of 270 VDC or 540 VDC. In order to achievethe right active output power, a first approximation of the load resistanceRload was estimated as in equation 4.2 using the reference output voltage andpower.

Rload =V 2o

Po(4.2)

Vo = output voltagePo = output power

4.3 DC-link

The DC-link was comprised of a capacitor (known as the DC-link capacitor)and was modeled using the Series RLC Branch block. The value of the ca-pacitor Co depended on the topology and was for the passive case (ATRU)set to a value in accordance to [21] and for the active case (∆-switch) a firstapproximation was estimated using equation 4.3 [12].

Co =Pom

vrfswVo(4.3)

m =

√3√

2VsVo

m = modulation indexVs = RMS L-N supply voltageVo = output voltagePo = output powervr = P-P voltage ripplefsw = switching frequency

The switching frequency is the frequency of the power electronic switchesused in the ∆-switch and are discussed more thoroughly in section 4.5.3. Thevalue of the voltage ripple was estimated using figure 2.2 and equation 2.4.Further on, the supply voltage (Vs) was set according to the requirementsin 2.2 and the output voltage (Vo) and power (Po) was set to values accordingto the specifications of the load (see section 4.2). In addition, for the ∆-switch,since no start-up circuitry is implemented, the initial DC-link voltage was set


to a value of 95% of the desired output voltage in order to minimize theinrush current and to get a more stable system.

4.4 12-pulse ATRU

4.4.1 Auto-transformer

The AT was modeled according to a topology presented in [16]. A graphicalrepresentation of that AT topology is illustrated in figure 4.1 and the ratingof each coil type (denoted a, b, c, d) is presented in table 4.1. The graphicalrepresentation helps to understand the connections inside the AT more easily.Coils that are connected have an electrical connection and coils that are inparallel to each other are magnetically connected.

FIGURE 4.1: AT topology used in the design of simulationmodel. A,B,C denotes the phases, a, b, c, d the coils and

1, 2, 3, 4, 5, 6 the outputs.

TABLE 4.1: Rating of coils in the AT at 230 VAC input,270 VDC/5 kW output [16].

Coil (n) Voltage (VRMS,n) Current (IRMS,n) Number Of Coils (kn)[V] [A]

a 197.9 4.9 3b 31.9 7.6 3c 31.9 7.6 3d 123.7 8.3 3

With the knowledge of the electrically and magnetically connections of theAT, it was modeled using three Multi-Winding Transformer blocks (one foreach phase). Each block had four input ports (representing coil a and d) onthe primary side and four output ports (representing coil b and c) on thesecondary side. Using this set up, the connection of the AT could be carriedthrough as follows:


• Phase A was connected to coil d of the A-block. Coil a of the A-blockrepresents the coil between phase A and C. Coil c and b represents thetwo shorter coils at phase B (output 3 and 4, respectively).

• Phase B was connected to coil d of the B-block. Coil a of the B-blockrepresents the coil between phase B and A. Coil c and b represents thetwo shorter coils at phase C (output 5 and 6, respectively).

• Phase C was connected to coil d of the C-block. Coil a of the C-blockrepresents the coil between phase C and B. Coil c and b represents thetwo shorter coils at phase A (output 1 and 2, respectively).

The final simulation model with all the connections is illustrated in figure 4.2.

FIGURE 4.2: Simulation model of the AT with the three multi-winding transformer representing each phase, the connection

between them and the six outputs.

To determine the parameters for the blocks the apparent power rating of theAT SAT was needed. It was obtained by summarizing the products of theRMS voltage and the RMS current of each coil in the AT and dividing bytwo [22]. The equation is expressed in 4.4 and the voltage and current ratingsfor each coil were obtained from table 4.1.

SAT =1

2

∑n

knVRMS,nIRMS,n (4.4)

n = type of coil = a, b, c, dkn = number of coils of each type

VRMS,n = RMS voltage of coil nIRMS,n = RMS current of coil n


With the rating of the AT a base impedance Zbase,n for the primary side andthe secondary side of the AT could be calculated according to equation 4.5.

Zbase,n =V 2RMS,n

SAT(4.5)

The base impedance of the primary side Zbase,primary was calculated with re-spect to the voltage rating of coil a and the base impedance of the secondaryside Zbase,secondary was calculated with respect to the voltage rating of coil b,as in equation 4.6 and 4.7, respectively.

Zbase,primary = Zbase,a =V 2RMS,a

SAT(4.6)

Zbase,secondary = Zbase,b =V 2RMS,b

SAT(4.7)

Finally, using the base impedances the resistance and the inductance of eachcoil was estimated using the default per-unit (pu) value of the block Rpu, Lpuas in equation 4.8, 4.9, 4.10 and 4.11.

Rprimary = Ra = Rd = Zbase,primaryRpu =V 2RMS,a

SATRpu (4.8)

Rsecondary = Rb = Rc = Zbase,secondaryRpu =V 2RMS,b

SATRpu (4.9)

Lprimary = La = Ld = Zbase,primaryLpuω

=V 2RMS,a

SAT

Lpu2πf

(4.10)

Lsecondary = Lb = Lc = Zbase,secondaryLpuω

=V 2RMS,b

SAT

Lpu2πf

(4.11)

ω = angular frequencyf = supply frequency

The losses of the AT were realized by a magnetizing resistance and induc-tance parameter Rm,AT , Lm,AT in the Multi-Winding Transformer block. Thevalue was estimated using the default p.u value of the block Rm,pu, Lm,putogether with the impedance base of the primary side as in equation 4.12and4.13. Since three blocks were used to model the AT the values were mul-tiplied by three.

Rm,AT = 3Zbase,primaryRm,pu (4.12)

Lm,AT = 3Zbase,primaryLm,puω

(4.13)


4.4.2 Diode bridge

The six outputs (two three-phase systems with a 30 phase shift) of the ATwere directly connected to and rectified by two B6U. The two B6Us weremodeled using the Universal Bridge block and by selecting diodes as the powerelectronic device. The forward voltage parameter of the block was obtainedfrom an ultrafast Si-diode (APT30D60BHB) [23] (which was used for the ∆-switch simulation as well) and the other diode parameters were kept at thedefault values. The final parameter values used for the B6U in simulationsare presented in appendix A.4.

4.4.3 Interphase transformer

In order to reduce harmonics the four outputs of the B6Us were connected inparallel [24]. And to avoid voltage differences which can lead to circulatingcurrent this was done by implementing two IPTs [12]. A graphical repre-sentation of the connections of the B6Us and the IPTs together with the finalsimulation model are illustrated in figure 4.3a and 4.3b, respectively. The twoIPTs were modeled using the Mutual Inductance block and as type of mutualinductance two windings with equal mutual terms was selected. The ratings ofthe coils in the IPT are presented in table 4.2.

To estimate the main parameter of the block, the winding inductance Lipt,equation 4.14 was used [12]. The parameter kipt is the relation between thepeak current seen by the interphase transformer and the output current. Afirst approximation should be around 0.4-0.5 [12].

Lipt = 0.1V 2s

fkiptPo(4.14)

kipt =IiptIo

Iipt =√

2IRMS,ipt

kipt = peak current of IPT related to output current

Iipt = peak current of IPT coilIRMS,ipt = RMS current of IPT coil

Io = output currentVs = RMS L-N supply voltagef = supply frequencyPo = output power


(A)

(B)

FIGURE 4.3: Graphical representation (A) of the connection be-tween the B6Us and the IPTs and (B) simulation model of thefinal set up. 1, 2, 3, 4, 5, 6 denotes the outputs from the AT and

e, f the coils of the IPT.


TABLE 4.2: Rating of the IPT at 230 VAC input, 270 VDC/5 kWoutput [16].

Coil Voltage (VRMS,ipt) Current (IRMS,ipt)[V] [A]

e 22 9.1f 22 9.1

Since the rating presented in table 4.2 was for a certain case those valueswere used for determination of the inductance. Further on, the mutual in-ductance ML between coil e and f was estimated according to equation 4.15.

ML = k√LeLf (4.15)

k = coefficient of couplingLe = inductance of coil eLf = inductance of coil f

Considering that coil e and f had the same voltage and current rating (ta-ble 4.2) which implied equal inductance in accordance to equation 4.14, themutual inductance of the interphase transformer ML,ipt was estimated asstated in equation 4.16. The winding resistance Ript and the mutual resis-tance MR,ipt were neglected.

ML,ipt = kLipt (4.16)


4.5 ∆-switch rectifier

4.5.1 Boost inductor

The boost inductors LN were modeled using the Series RLC Branch block.A first approximation of the inductance value was estimated using equa-tion 4.17 [2].

LN =Vo

kboostfswIs

m√3

(1−√

3

2m

)(4.17)

m =

√3√

2VsVo

Is =√

2Is

m = modulation indexVs = RMS L-N supply voltageVo = output voltage

Is = peak L-N supply currentIs = RMS L-N supply current

kboost = ripple current factorfsw = switching frequency

The ripple current factor kboost is typically chosen to a value of 0.1-0.2 in orderto set the P-P ripple current to 10-20% of the peak supply current [2].

4.5.2 Diode bridge and Switches

After the boost inductors each phase is connected to its respective phase legin the diode bridge and its respective phase leg in the ∆-configured switchset up. The diodes were modeled using the Diode block and the same pa-rameter values as for the B6U in the ATRU model (see section 4.4.2) wereused. The switches were modeled as MOSFETs according to [15] using theMOSFET block. The internal diode forward voltage parameter value andthe drain-source conduction resistance parameter value were obtained froma Cool-MOS (IPP60R045CP) [25]. The rest of the parameters was set to de-fault values. The final parameter values used for the diode bridge and theswitches in simulations are presented in appendix A.4.

4.5.3 Control of switches

The aim of the control of the rectifier was to achieve a constant DC-linkvoltage and a unity input power factor. This was done by implementing aconstant switching frequency control using pulse width modulation (PWM)based on [2]. The ∆-connected switches affect the L-L voltage since eachswitch-pair (Sij, Sji) is connected between two phases. This gave rise tothe idea of controlling the currents through the ∆-configured switches. Toavoid inconvenient caused by redundant switching states and thus increased


clamping action the L-N current of the rectifier was controlled [2]. This re-sulted in that the modulation voltage needed for the PWM had to be a L-Lquantity.

FIGURE 4.4: Block diagram of the control of the switches.

A block representation of the implemented control loop is presented in fig-ure 4.4. An inner current loop (to achieve the sinusoidal supply current) andan outer voltage control loop (to achieve the constant output voltage) wasimplemented. Further on, in [2] a thorough explanation of the whole controlstrategy is presented and discussed and therefor omitted here for the sake ofbrevity. Instead, a brief summary of the steps taken in order to obtain themodulation voltage and finally generate PWM signals is given:

• The L-N supply voltage (vs,p), the L-N supply current (is,p) and the out-put voltage (vo) were measured, where p denotes the phases (A, B, C).

• The reference supply current (i∗s,p) was obtained by multiplying the sup-ply voltage with a reference conductance (g∗) in order to achieve ohmicinput behavior.

i∗s,p = g∗vs,p (4.18)

• The reference conductance was obtained by a PI controller (FPI) withthe difference between the measured output voltage and the referenceoutput voltage (v∗o) as input.

g∗ = FPI(v∗o − vo) (4.19)

• The reference L-N voltage at the switch-pair (v∗rs,p) was obtained by sub-tracting the difference between the reference supply current and the


measured supply current multiplied by a P controller of gain KP,i fromthe supply voltage.

v∗rs,p = vs,p −KP,i(i∗s,p − is,p) (4.20)

• The reference L-L voltage over the switch Sij (vr,ij) was obtained by aY -∆ transformation with the reference L-N voltage over the switchesas input, where ij denotes each switch in accordance to figure 3.3.

v∗r,ij = v∗rs,p=i − v∗rs,p=j (4.21)

The reference L-L voltages over the switches was used as the modulationvoltage for the PWM in order to create the control pulses for the switches.This was done as follows

• The reference L-L voltage over the switches (v∗r,ij) was divided in twoindependent but equal parts where one of them was multiplied by afactor of -1 in order to obtain v∗r,ji.

v∗r,ji = (−1)v∗r,ij (4.22)

• A unipolar constant frequency triangular carrier voltage (vtri) were sub-tracted from the modulation voltages v∗r,ij and v∗r,ji, respectively.

• Depending on the difference between the modulation voltage and thecarrier voltage the signal was saturated to either 0 or 1, i.e either low(OFF) or high (ON).

vtri − v∗r,ij < 0⇒ saturation = 0

vtri − v∗r,ij > 0⇒ saturation = 1

vtri − v∗r,ji < 0⇒ saturation = 0

vtri − v∗r,ji > 0⇒ saturation = 1

(4.23)

• The outputs from the saturation were the PWM signals (pwmij) andbefore the signals was transferred to the switches a clamping actionwas implemented in order to obtain the correct switching states.

• The clamping action was controlled by measuring the sector (0− 360)of phase A of the L-N supply voltage (vs,A). The clamping action forthe switches was divided in six equally sized sectors of 60 and theresulting switching states for all switches for each sector are presentedin table 4.3

The L-N supply voltage and the L-N supply current were measured with theThree-Phase VI measurement block at the supply and after the boost inductors,respectively. The output voltage was measured with the Voltage measurement


TABLE 4.3: Required clamping action for all the switches. 0indicates that the switch is OFF, 1 indicates that the switch isON and the denomination pwmij indicates that the switch is

modulated by the PWM signals [2].

Sector S12 S21 S23 S32 S31 S13

330 − 30 pwm12 1 0 0 1 pwm13

30 − 90 0 0 pwm23 1 1 pwm13

90 − 150 1 pwm21 pwm23 1 0 0150 − 210 1 pwm21 0 0 pwm31 1210 − 270 0 0 1 pwm32 pwm31 1270 − 330 pwm12 1 1 pwm32 0 0

block at the DC-link and the output reference voltage was set according to thespecifications in section 4.2. A unipolar triangular wave was used as carriervoltage to yield an optimized switching sequence in accordance to [2]. Thecarrier waveform was generated using the Pulse Generator block, the Constantblock, the Sum block and the Integrator block as can be seen in appendix A.4,figure A.7. The block parameters of the pulse generator and the constantwas set in order to obtain a switching frequency in accordance to [2] and anamplitude equal to the output voltage. The summation and the integrationblock was modeled using the basic blocks. Further on, the sector detectionwas obtained with a phase locked loop (PLL) measurement by using the PLLblock. The L-N supply voltage, phase A, vs,A (obtained by the supply voltagemeasurement) was used as input, giving the phase angle and the frequencyof the input as outputs. By multiplying the phase angle by 360

2πthe sector in

degrees was obtained. The clamping action presented in table 4.3 was basedon [2], where it was stated that the supply current for sector 330−30 shouldbe positive in phase A and negative in B and C, which implied a voltage asstated in equation 4.24. Since vs,A had a sinusoidal behavior the signs of thevoltages in sector 330 − 30 were as illustrated in figure 4.5a and as seen,they did not cohere with the required values stated in equation 4.24. In orderto obtain correct signs, the phase angle obtained from the PLL block wasphase shifted by −90 in order to detect the correct sector as illustrated in4.5b. Finally, the frequency and sample time parameters for the PLL blockwas set according to chapter 2, section 2.3 and the rest was set to default.

330 − 30 :

is,A > 0

is,B < 0

is,C < 0

⇒

vs,A > 0

vs,B < 0

vs,C < 0

(4.24)

The PI controller FPI used to obtain the reference conductance was mod-eled using the PID controller block. The controller was set to be a discretePI controller with a sample time Ts in accordance to chapter 2, section 2.3,and the rest as default. An overview of the simulation model is presentedin appendix A.4, figure A.6. In equation 4.25 the PI controller’s compensator


(A)

(B)

FIGURE 4.5: Sector detection with no phase shift (A) resultingin wrong sign of the voltage and sector detection with a −90

phase shift (B) resulting in correct signs of voltage and hencecurrent.

formula is presented. The P controller (KP,i) used in the current loop to ob-tain the reference L-N voltage at the switch-pair was modeled similarly. Forboth the PI and the P controller, the respectively coefficients KP,v, KI,v, KP,i

were tuned in order to obtain a stable system and output and a unity powerfactor. The final coefficients values used for the PI and the P controller insimulations are presented in chapter 5, section 5.2.

FPI(z) = KP,v +KI,vTsz

z − 1(4.25)

KP,i = Proportional coefficient of voltage controlKP,i = Integral coefficient of voltage controlTs = sample timez = complex variable of z-transform


With these measurements, detections, generations and controls the neces-sary quantities for the control loop were obtained. An illustration of thesimulation model of the control loop for the three phases is presented in ap-pendix A.4.

39

Chapter 5

Result and Discussion

5.1 12-pulse ATRU

The 12-pulse ATRU was simulated in four different cases as presented intable 5.1. The supply current distortion and the input power factor of therectifier (THDI and PF ), the efficiency (η) and finally the ripple factor of theoutput voltage (RFV ) are presented for each case in table 5.2. The valuesof the primary and secondary resistance and inductance (Rprimary, Lprimary,Rsecondary, Lsecondary), magnetizing resistance and inductance of the AT (Rm,AT ,Lm,AT ), winding and mutual resistance and inductance of the IPTs (Ript, Lipt,MR,ipt, ML,ipt), load resistance (Rload), DC-link capacitor (Co) and finally sam-ple time (Ts) used for each case are presented in table 5.3. The reason forrestricting the simulations to a supply voltage of 230 V was because the onlyavailable rating of the AT and the coils was for that supply voltage. A bigeffort was made when trying to re-scale the AT to make it suitable for dif-ferent supply voltages. Different ratings of each coil and different values ofthe resistance and inductance were tested in order to at least be able to sim-ulate for a supply voltage of 115 V, but unfortunately without any success.As mentioned before, in all simulations EMC, net inductance and resistance,transient behavior (inrush current, charging of DC-link) and saturation of theAT were neglected. In all four simulations the inductance of primary coil d,determined using the pu value of the transformer block (see section 4.4.1),was increased by a factor of ten in order to minimize current THD.

TABLE 5.1: Different cases that were simulated for the 12-pulseATRU. Supply voltage is given in L-N RMS.

Case Supply voltage Supply frequency Output voltage Output power[V] [Hz] [V] [W]

1 230 400 270 20002 230 400 270 100003 230 800 270 20004 230 800 270 10000

The only case not resulting in a THDI fulfilling the requirements in table 2.4was case 1. The elimination of the third harmonic was successful but un-fortunately, the fifth harmonic was slightly too large which can be seen infigure 5.1c where the FFT analysis of case 1 is presented. The supply voltage

Chapter 5. Result and Discussion 40

and current for case 1 is presented in figure 5.1a and 5.1b, respectively. Infigure 5.2a and 5.2b the supply voltage and current are presented for case 4which showed the best result of all simulated cases. One can clearly see inboth case 1 and 4, that the supply current has 12 pulses per period due to the12-pulse rectification. The THD of case 4 is 3.73% and the triple harmoniccomponents were not as successfully eliminated as in case 1 but still withinthe limits, which can be seen in the FFT analysis presented in figure 5.2c.One possible reason why the fifth harmonic is too large in case 1 and whythe triple harmonics is not as successfully eliminated in case 4 may be thatthe simulation model was just a rough estimation of a real 12-pulse ATRU.In addition, as seen in table 5.3, the winding and mutual resistance of theIPT was set to zero. Further on, in the design of the whole simulation modelseveral block parameters were set to default values. This could also have af-fected the simulation result. The reason for using default values was lack ofinput parameters, which in turn was because the secrecy of companies (theydid not publish specific values). In general, there exist only few publicationsin this topic, especially concerning simulation models. In the papers pub-lished, the design of simulation models and justification of parameters usedwas neither thoroughly explained nor presented.

TABLE 5.2: Simulation results of the 12-pulse ATRU for eachsimulation case

Case THDI PF RFV η

1 10.3% 0.974 (lag) 0.00000647 0.9722 6.15% 0.956 (lag) 0.00000883 0.9803 9.02% 0.974 (lag) 0.00000177 0.9724 3.73% 0.924 (lag) 0.00000627 0.979


TAB

LE

5.3:

Sim

ulat

ion

para

met

ers

ofth

e12

-pul

seA

TRU

for

each

case

.*de

note

sth

atth

epr

imar

yco

ild

(see

figur

e4.

1fo

rde

finit

ion)

iste

nti

mes

the

stat

edva

lue

Cas

eRprimary

[mΩ

]Rsecondary

[mΩ

]Lprimary

[µH

]Lsecondary

[µH

]Rm,AT

[kΩ

]Lm,AT

[mH

]Ript

[mΩ

]Lipt

[mH

]MR,ipt

[mΩ

]ML,ipt

[mH

]Rload

[Ω]

Co

[µF]

Ts

[µs]

152

.61.

3783

7*2.

181.

5862

80

3.81

00.

381

36.5

940

52

52.6

1.37

837*

2.18

1.58

628

03.

810

0.38

17.

2994

05

352

.61.

3783

7*2.

181.

5862

80

3.81

00.

381

36.5

940

54

52.6

1.37

837*

2.18

1.58

628

03.

810

0.38

17.

2994

05


(A)

(B)

(C)

FIGURE 5.1: Supply voltage (A), supply current (B) and FFTanalysis of supply current (C) of simulation case 1.


(A)

(B)

(C)

FIGURE 5.2: Supply voltage (A), supply current (B) and FFTanalysis of supply current (C) of simulation case 4.


5.2 ∆-switch rectifier

The ∆-switch rectifier was simulated for eight different cases as presented intable 5.4. The distortion and the power factor of the supply (THDI and PF ),the efficiency (η) and finally the ripple factor of the output voltage (RFV ) arepresented for each case in table 5.5. The value of the boost inductance (LN ),load resistance (Rload), DC-link capacitor (Co), coefficients of the PI and Pcontroller (KP,v, KI,v and KP,i), switching frequency (fsw) and finally sampletime (Ts) used for each case are presented in table 5.6. As mentioned before,in all simulation cases EMC, net inductance and resistance, transient behav-ior and start-up circuitry to minimize inrush current were neglected.

TABLE 5.4: Different cases that were simulated for the ∆-switchrectifier. Supply voltage is given in L-N RMS.

Case Supply voltage Supply frequency Output voltage Output power[V] [Hz] [V] [W]

1 115 400 270 20002 115 400 270 100003 115 800 270 20004 115 800 270 10000

5 115 400 540 20006 115 400 540 100007 115 800 540 2000

8 230 400 1080 10000

TABLE 5.5: Simulation results of the ∆-switch rectifier for eachsimulation case. * denotes that the system is over-modulated.

Case THDI PF RFV η

1* 12.89% 0.987 (lag) 0.000169 0.9862* 3.31% 0.995 (lag) 0.000202 0.9873* 8.54% 0.986 (lag) 0.0000434 0.9864* 2.77% 0.984 (lag) 0.000103 0.987

5 5.93% 0.998 (lag) 0.0000102 0.9876 1.18% 0.999 (lag) 0.0000451 0.9897 5.93% 0.998 (lag) 0.00000953 0.986

8 4.75% 0.999 (lag) 0.00000896 0.991

The THDi for case 1-4 looked good at a first glance, but in the FFT analysisthe fifth harmonic turned out to be too large in all cases considering the cur-rent requirements presented in table 2.4. The FFT analysis of case 4, whichhad the best THDI of case 1-4, is illustrated in figure 5.3a. The reason for the


TABLE 5.6: Simulation parameters of the ∆-switch rectifier foreach case.

Case LN [µH] Rload [Ω] Co [mF] KP,v KI,v KP,i fsw [kHz] Ts [µs]

1 330 36.5 1.47 0.01 10 10 72 12 330 7.29 1.47 0.01 10 10 72 13 330 36.5 1.47 0.01 10 10 72 14 330 7.29 1.47 0.01 10 10 72 1

5 720 146 1.47 0.01 10 100 72 16 720 29.2 1.47 0.01 10 100 72 17 720 146 1.47 0.01 10 100 72 1

8 720 117 1.47 0.01 10 100 72 1

distortion could be that the system was over-modulated. This occurs in ac-cordance to equation 3.1, which states that a minimum output voltage has tobe around 280 V. In [2] it is stated that over-modulation will lead to distortionin the supply current which was confirmed by the simulations. In figure 5.3b,the modulation voltage and the carrier voltage for case 4 is presented and asseen in the graph, the modulation voltage has a larger amplitude comparedto the carrier wave, which results in over-modulation. The best simulationresult for the ∆-switch rectifier was obtained in case 6. In this case, the out-put voltage was 540 V which is far above to prevent over-modulation andthe output power was 10 kW. The supply voltage and current are presentedin figure 5.4a and 5.4b, respectively. One can see that the current is followingthe voltage, which was obtained by the control of the switches, resulting ina PF of 0.999. The current is still slightly distorted but from the FFT anal-ysis, presented in figure 5.4c, one can see that all individual harmonics arewithin the limits stated in table2.4 and a THDI of 1.18% is obtained. Sim-ulation case 6 resulted in an efficiency of 0.989 and a RF fulfilling the limitspresented in table 2.1 with a large margin. In case 8 a supply voltage of 230V, an output voltage of 1080 V and a output power of 10 kW was simulated,which is not used in any civil or military aircraft today but it was of interestfor future applications. To be able to compare, the boost inductance for thiscase was kept at the same value as for case 5-7, even though the theoreticalvalue should be higher. The supply voltage, supply current and FFT analysisof case 8 is presented in figure 5.5a, 5.5b and 5.5c, respectively. As one cansee, the ∆-switch rectifier shows promising results for an increased outputvoltage with a low THDI , high efficiency and a power factor close to unity.One big difference, compared to e.g. case 6, is the increase of even harmonics.Even harmonics means that the waveform does not have half-wave symme-try any more. By comparing the supply current of case 6 with the one in case8, one can see an increase of distortion in case 8, which is due to this increaseof even harmonics/DC-component. Why an increase of even harmonics oc-cur is hard to say, but it might be improved by implementing EMI-filters atthe input and output.


Since the control strategy was based on [2] and due to time limitation, valuesfor the boost inductance, DC-link capacitance and switching frequency wereset directly according to that paper. The equation of the boost inductanceand the DC-link capacitance presented in equation 4.17 and 4.3, respectively,were used as a first approximation but if they differed too much with theones stated in [2] the values were not used since it resulted in an unstablesystem. The reason for this was probably that the control strategy used wasoptimized for a system with specific parameters. The ones used in [2] arepresented in table 5.7.

TABLE 5.7: Parameters used in [2] which the control strategywas optimized for. The supply voltage is given in L-N RMS.

Vs [V] f [Hz] Vo [V] Po [kW] fsw [kHz] LN [µH] Co [mF]

115 360-800 400 5 72 330 1.47230 360-800 650 10 72 720 1.47

Further on, the optimized control based on the current and voltage controlparameters (KP,i, KP,v, KI,v) in [2] was set in order to stabilize and maximizethe performance of the system for these values presented in table 5.7. There-for, to be able to simulate the cases presented in table 5.4 without ending upwith an unstable system, the control parameters had to be modified. The bestway, performance wise, to do this would have been to go through each con-trol block and calculate new coefficients by considering the phase margin,gain margin and stability criteria. Unfortunately, this would have been tootime consuming. Instead, the modification of the control parameters was ob-tained in a trail and error fashion and it ended up with the values presentedin table 5.6. Considering this, the result for some simulations case, especiallycase 5 and 7 which showed a higher THDI compared to case 6 even thoughthe system is not over-modulated, could be explained.


(A)

(B)

FIGURE 5.3: FFT analysis of supply current (A) of simulationcase 4 with the fifth harmonic reaching above 2%. The reason ofthe distortion is (B) the modulation voltages vr,12∗, vr,21∗ reach-ing above the carrier voltage vtri, resulting in over-modulation.


(A)

(B)

(C)

FIGURE 5.4: Supply voltage (A), supply current (B) and FFTanalysis of supply current (C) of simulation case 6. The currentfollows the voltage resulting in almost a unity power factor and

a low THD is obtained.


(A)

(B)

(C)

FIGURE 5.5: Supply voltage (A), supply current (B) and FFTanalysis of supply current (C) of simulation case 8. The pres-ence of even harmonics result in a more distorted supply cur-

rent compared to case 6.

50

Chapter 6

Conclusion and Future Work

The goal of this master thesis was to evaluate the power quality of AC/DCrectifier on aircraft and to establish if a common design concept could beobtained. Based on a literature and market study on AC/DC rectifier on air-craft a passive ATRU and two APFCs were selected and analyzed for aircraftapplications. Figures of merit such as weight, reliability and cost are dis-cussed and compared and the aircraft standards considering power qualityare summarized for voltage, current and frequency requirements. The pas-sive 12-pulse ATRU and the active ∆-switch rectifier were chosen for evalu-ation by simulation in MATLAB/Simulink. The evaluation considered a sup-ply of 115 VAC/230 VAC, 400-800 Hz and an output voltage and power of270/540 VDC and 2-10 kW, respectively. The design of simulation models ispresented with a thoroughly step-by-step explanation for the 12-pulse ATRUand a more brief explanation for the ∆-switch rectifier. The two rectifiers aresimulated in different cases and evaluated considering total harmonic distor-tion, power factor and ripple factor. The efficiency of the simulated converteris given and waveforms of the supply current and voltage together with aFFT analysis of the supply current are presented and discussed.

In section 5.1, the simulation result of the 12-pulse ATRU model showed anoverall good THDI ranging from 3 to 10 % and a markable damping of tripleharmonics was achieved. The presence of the 5th harmonic was in one offour cases too large considering the requirements. Finally, considering thatthe model was simulated for only one supply voltage and output voltagetogether with the fact that the simulation model was not validated made itdifficult to draw any conclusion regarding common design concept. The coilsof the AT are designed and rated for specific supply and output values, thesame holds for the DC-link capacitor. For the model to become more realistic,rating and values of coils, inductance and resistance in the AT and IPT arenecessary. Since companies and manufactures are very stringent and care-ful when it comes to this, it perhaps could be obtained from sophisticatedmeasurements and tests on a physical model instead. In addition, if a morerealistic simulation model should be obtained, implementation of EMI-filtersand saturation of the AT are necessary as well. Future passive topologiesthat should be considered are the 18-pulse and the 30-pulse ATRU. The bothimplies an increase in weight but an improved current THD together withsimilarly reliability and robustness as the 12-pulse topology.

Chapter 6. Conclusion and Future Work 51

In section 5.2, the simulation of the ∆-switch rectifier model showed for thefirst four cases that an output voltage of 270 V was too low for a supply volt-age of 115 VAC. This was due to the boost property of the ∆-switch rectifierresulting in over-modulation if a too low output voltage was used, whichin turned led to distortion in the supply current. With an output voltage540 VDC instead the APFC showed promising result with a THD as low as1.18%, almost unity power factor and an efficiency of 0.989. Further on, alsowith an increased supply voltage of 230 VAC and an output of 1080 VDC thetopology showed interesting results which all were within the requirements.For all simulated cases, the output capacitor could be kept common togetherwith the switching frequency. If this could be realized in modular conceptis hard to say. The simulation model was unfortunately very unstable andsensitive to even small changes in parameter values. The conclusion drawnfrom this, is that the ∆-rectifier is not sufficient for rectification of 115 VAC to270 VDC nor 230 VAC to 540 VDC but instead is an interesting topology forfuture aspects in the course of the MEA concept. For a more stable and realis-tic simulation model, a more thoroughly design of the control of switches forthe desired supply and output together with implementation of a start up-circuit has to be done. In addition, EMI-filters need to be implemented sincethey are one of the bigger drawbacks of active solutions. For an increase ofthe supply to 230 VAC, the component stress has to be considered more care-fully. The recently released SiC (Silicon Carbide) MOSFET with high blockingvoltage could come to use in applications like this. Finally, credits has to begiven to the author of [2] where exactly this is done and it is presented in agood way with a thoroughly explanation of each step. Unfortunately, it wasdone for a different output power and voltage that was of concern in thisthesis. But it is a good guideline if one want to simulate an APFC for aircraftapplications.

Lastly, it has to be mentioned, that in this thesis neither environmental as-pects such as temperature, altitude and restriction of electrical componentssuch as the MOSFET due to cosmic radiation are considered. This is of courseof concern in real aircraft applications. Also, the load profile and the missionprofile are neglected and the load are instead set to be either 2 kW or 10 kW.This could also be improved for future simulations.

My conclusion is that the active solutions, both the simulated ∆-switch recti-fier and the earlier discussed six-switch Vienna is promising for future air-craft applications. The reliability and cost has earlier been the reason toprefer passive solution before active. But in the course of the MEA con-cept and development of power electronics and computational power theactive solutions will be the preferable since they can offer better power qual-ity which is crucial for a power system in need of long life time and highreliability. Also, the Simscape Power System - Specialized technology toolbox inMATLAB/Simulink is shown to be a powerful tool for analyze and simula-tions of power electronics and power systems. The library includes most ofthe necessary blocks and components for simulation of any type of electronic

Chapter 6. Conclusion and Future Work 52

circuit and more. A forum for explanations, examples and Q & A are avail-able at the web which makes it is easy to get going and to find help andguidance if needed. There are great opportunities for improvements and de-velopments at Saab by implementing this simulation tool. Especially in thearea of power electronics and power systems, e.g. to make estimations, vali-dations and performance improvements.

53

Appendix A

Appendix

A.1 Defintions and Equations

A.1.1 Average and Root Mean Square

The average (AVG) of an periodic voltage or current waveform x(t) is equalto the arithmetic mean of the waveform. The formula defining AVG in thisthesis is

XAV G =1

T

∫ T

0

x(t)dt (A.1)

XAV G = average value of voltage or currentx(t) = instantaneous voltage or currentT = period (reciprocal value of f )

The root mean square (RMS) of a periodic waveform x(t) is equal to thesquare-root of the arithmetic mean of the square of the waveform. The for-mula defining RMS in this thesis is

XRMS =

√1

T

∫ T

0

x2(t)dt (A.2)

XRMS = RMS value of voltage or currentx(t) = instantaneous voltage or currentT = period (reciprocal value of f)

If the waveform x(t) is expressed by RMS values of harmonics, the total RMSvalue can be written as

XRMS =

√√√√ ∞∑n=1

X2n,RMS +X2

AV G (A.3)

XRMS = RMS value of voltage or current

Xn,RMS = nth harmonic RMS value of voltage or current

Appendix A. Appendix 54

A.1.2 Efficiency

The efficiency η of a system is defined as

η =PoPin

(A.4)

Po = output powerPin = input power

A.1.3 Per-unit

Per-unit (pu) system is often used in power systems to simplify calculationsand circuit representations. Usually a line-to-neutral base voltage (Vbase) anda one phase base complex power (Sbase) are selected and then the remainingquantities can be determined with regards to the chosen bases [7]. Followingrelations are used for calculations with per-unit

pu quantity (dimensionless) =actual quantity (SI unit)

base value of quantity (SI unit)(A.5)

Zbase =V 2base

Sbase(A.6)

Rbase = Zbase (A.7)

Xbase = ωLbase = Zbase (A.8)

A.1.4 Mutual inductance

The mutual inductance (M ) between two coils is expressed as

M = k√L1L2

k = coefficient of couplingL1 = inductance of coil 1L2 = inductance of coil 2

(A.9)

where the coefficient of coupling k describes how well magnetically coupledthe two coils are, ranging from 0 to 1. If k = 1 the two coils are perfectlycoupled, if k > 0.5 the two coils are said to be tightly coupled and if k < 0.5the two coils are said to be loosely coupled [26].


A.2 Power Quality Requirements

A.2.1 Conducted and radiated emission

FIGURE A.1: Conducted emission limits according to DO-160G[4].


FIGURE A.2: Radiated emission limits according to DO-160G[4]

FIGURE A.3: Graphical representation of a CM and DM filter[11].


A.3 Rectifier topologies

A.3.1 AT design topologies

FIGURE A.4: Three different 18-pulse AT topologies based onthe ∆-configuration [18].

FIGURE A.5: 12-pulse Zig-Zag AT topology [22].


A.4 Simulation parameters, values and models

In table A.1 the parameters values used for the diode bridge (B6U) in sim-ulations are presented. The forward voltage drop value Vf was obtainedfrom data sheet [23], the rest were set as default. In the 12-pulse ATRU theB6U was modeled with a complete block meanwhile in the ∆-switch rectifiermodel the B6U was implemented with individual diodes. The default valueof the snubber capacitance was for the complete block Cs = ∞ and for theindividual diode Cs = 250 pF . The parameter values for the switches usedin the ∆-switch rectifier are presented in A.2. The internal diode forwardvoltage drop value Vf and the drain-source conduction resistance Ron wasobtained from data sheet [25], the rest were set as default. In table A.3 the de-fault pu values used for calculation of AT resistance, inductance and lossesare presented.

TABLE A.1: Parameter values used in the rectifier bridge (B6U)simulation model for both 12-pulse ATRU and ∆-switch recti-

fier.

Vf,D [V ] Ron,D [Ω] Lon,D [H] Rs,D [Ω] Cs,D [pF ]

1.6 0.001 0 105 ∞/250

TABLE A.2: Parameter values used in the switch Sij simulationmodel for the ∆-switch rectifier.

Vf,S [V ] Ron,S [Ω] Lon,S [H] Rd,S [Ω] Rs,S [Ω] Cs,S [pF ]

0.9 0.045 0 0.01 105 ∞

TABLE A.3: Default pu values used for calculation of resistance,inductance and losses of the AT for the 12-pulse ATRU.

Rpu Lpu Rm,pu Lm,pu

0.005 0.02 50 50

In figure A.6 the voltage control loop to obtain the reference conductanceg∗ is presented, for values of the PI control see section 5.2. In figure A.7the simulation model used to generate the carrier voltage is presented. Theconstant block was set with a value of 2A/P with A = Vo and P = 1/fsw,further on the pulse generator block was set with an amplitude of 4A/P , aperiod of P , a pulse width of 50% and a phase delay of 0.


FIGURE A.6: Simulation model of the voltage loop used to ob-tain the reference conductance g∗.

FIGURE A.7: Simulation model of the generation of the unipo-lar triangular carrier voltage


FIGURE A.8: Simulation model of the full control loop includ-ing all three phases.

61

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