a family of efficient complex halfband filters

8/3/2019 A Family of Efficient Complex Halfband Filters

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A Family of Efficient Complex Halfband Filters

Heinz G. Gckler and Sanja Damjanovic

Digital Signal Processing Group

Ruhr-Universitt Bochum, D-44780 Bochum, Germany

Mihajlo Pupin Institute, Belgrade, Serbia and Montenegro{goeckler, damjanovic}@nt.rub.de

Abstract Halfband lowpass and bandpass filters, versatilebuilding blocks in a multitude of DSP applications, are revisited.Linear-phase FIR and minimum-phase IIR halfband filters forsample rate alteration are recalled, their computational loads arecompared, and optimum signal flow graphs are presented. Thesame comparisons are carried out for corresponding complexhalfband filters (Hilbert-transformers) with and without anadditional frequency offset by = /4. As expected, thesmaller the transition width between passband and stopband the

higher the relative advantage of the IIR approach in terms ofcomputation. Yet, there exist ranges of specifications where linear-phase FIR halfband filters outperform their minimum-phase IIRcounterparts.

I. INTRODUCTION

A digital halfband filter (HBF) is, in its basic form with

real-valued coefficients, a lowpass filter with one passband

and one stopband region, where both specified bands have

the same bandwidth. The zero-phase frequency response of a

nonrecursive (FIR) halfband filter with its symmetric impulse

response exhibits an odd symmetry about the quarter samplerate ( = 2 ) and half magnitude (

12

) point [33], where =2f/fn represents the normalised (radian) frequency and fn =1/T the sampling rate. The same symmetry holds true forthe squared magnitude frequency response of minimum-phase

(MP) recursive (IIR) halfband filters [22], [34]. As a result

of this symmetry property, the implementation of a real HBF

requires only a low computational load since, roughly, every

other filter coefficient is identical to zero [3], [28], [34].

Due to their high efficiency, digital halfband filters are

widely used as versatile building blocks in digital signal

processing applications. They are, for instance, encountered

in front ends of digital receivers and back ends of digital

transmitters (software defined radio, modems, CATV-systems,

etc. [13], [15], [16], [32]), in decimators and interpolators for

sample rate alteration by a factor of two [2], [3], [4], [11],

[40], in efficient multirate implementations of digital filters

[4], [10], [15] (cf. Fig. 1), in tree-structured filter banks for

FDM de- and remultiplexing (e.g. in satellite communications)

according to Fig. 2 and [7], [13], [14], [15], etc. A frequency-

shifted (complex) halfband filter (CHBF), generally known

as Hilbert-Transformer (HT, cf. Fig. 3), is frequently used

to derive an analytical bandpass signal from its real-valued

counterpart [18], [19], [22], [25], [33], [34], [35]. Finally, real

IIR HBF or spectral factors of real FIR HBF, respectively, are

used in perfectly reconstructing sub-band coder (cf. Fig. 4)

Fig. 1. Multirate fi ltering applying dyadic HBF decimators, a basic fi lter,and (transposed) HBF interpolators

Fig. 2. FDM demultiplexer fi lter bank; LP/HP: lowpass/highpass directionalfi lter block based on HBF

and transmultiplexer filter banks [10], [15], [28], [38], whichmay apply the discrete wavelet transform [5], [6], [10], [36].

Digital linear-phase (LP) FIR and MP IIR HBF have thor-

oughly been investigated during the last three decades starting

in 1974 [4] and 1969 [17], respectively. An excellent survey

of this evolution is presented in [33]. However, the majority

of these investigations deals with the properties and the design

of HBF by applying allpass pairs [31], [37], also comprising

IIR HBF with approximately linear-phase response [33], [34],

[35]. Hence, only few publications on efficient structures

Fig. 3. Decimating Hilbert-Transformer (a) and its transpose (b)

Fig. 4. Two-channel sub-band coder (SBC) fi lter bank


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e.g. [3], [4], [22], [23], [40], present elementary signal flow

graphs (SFG) with minimum computational load. Moreover,

only real-valued HBF and complex Hilbert-Transformers (HT)

with a centre frequency of fc = fn/4 (c =2

) have been

considered in the past.

The major goal of this contribution is to show the existence

of complex HBF with their passbands (stopbands) centred at

fc = c fn8

, c = 1, 2, 3, 5, 6, 7 (1)

that require roughly the same amount of computation as their

real HBF prototype (fc = f0 = 0), and to present the mostefficient elementary SFG for sample rate alteration. These will

be given for LP FIR [12] as well as for MP IIR HBF for real-

and complex-valued input and/or output signals, respectively.

Detailed comparison of expenditure is included.

The organisation of the paper is as follows: First, we recall

the properties of both classes of the afore-mentioned real

HBF. The efficient multirate implementations are based on the

polyphase decomposition of the transfer function [3], [15],

[27], [38]. Next, we present the corresponding results oncomplex HBF (CHBF), the classical HT, by shifting a real

HBF to a centre frequency according to (1) with c {2,6}.Finally, complex offset HBF (COHBF) are derived by applying

frequency shifts according to (1) with c {1,3,5,7}, and theirproperties are investigated. Illustrative design examples and

implementations thereof will be given.

II. REA L HALFBAND FILTERS (HBF)

In this section we recall the essentials of LP FIR and

MP IIR lowpass HBF with real-valued impulse responses

h(k) = hk H(z). From such a lowpass (prototype) HBF

a corresponding real highpass HBF is readily derived by using

the modulation property of the z-transform [29]

zkc h(k) H(z

zc) (2)

by setting in accordance with (1)

zc = z4 = ej2f4/fn = ej = 1 (3)

resulting in a frequency shift by f4 = fn/2 (4 = ).

A. Linear-Phase (LP) FIR Filters

Throughout this paper we describe a real LP FIR (lowpass)filter by its non-causal impulse response with its centre of

symmetry located at k = 0 according to

hk = hk k (4)where the associated frequency response H(ej) R is zero-phase [28].

Specification and Properties: A real zero-phase (LP) low-

pass HBF, also called Nyquist(2)filter [28], is specified in the

frequency domain as shown in Fig. 5, for instance, for an

equiripple or constrained least squares design, respectively,

allowing for a dont care transition band between passband and

stopband [26], [28], [33]. Passband and stopband constraints

Fig. 5. A possible specifi cation of a zero-phase FIR HBF

p = s = are identical, and for the cut-off frequencies wehave:

p + s = . (5)

As a result, the zero-phase desired function D(ej) R aswell as the frequency response H(ej) R are centrosym-metric about D(ej/2) = H(ej/2) = 1

2. From this frequency

domain symmetry property immediately follows

H(ej) + H(ej()) = 1, (6)

indicating that this type of halfband filter is strictly comple-

mentary [33].

According to (4), a real zero-phase FIR HBF has a sym-

metric impulse response of odd length N = n + 1 (denoted astype I filter [28]), where n represents the even filter order. Incase of a minimal (canonic) monorate filter implementation,

n is identical to the minimum number nmc of delay elementsrequired for realisation, where nmc is known as the McMillandegree [38]. Due to the odd symmetry of the HBF zero-phase

frequency response about the transition region (dont care band

according to Fig. 5), roughly every other coefficient of the

impulse response is zero [26], [33], resulting in the additional

filter length constraint:

N = n + 1 = 4m 1, m N. (7)Hence, the non-causal impulse response of a real zero-phase

FIR HBF is characterised by [4], [15], [26], [33]:

hk = hk =

12

k = 00 k = 2l l = 1, 2, . . . , (n 2)/4

h(k) k = 2l 1 l = 1, 2, . . . , (n + 2)/4(8)

giving rise to efficient implementations. Note that the name

Nyquist(2)filter is justified by the zero coefficients of the

impulse response (8). Moreover, if an HBF is used as an anti-imaging filter of an interpolator for upsampling by two, the

coefficients (8) are scaled by the upsampling factor of two

replacing the central coefficient with h0 = 1 [10], [15], [27].As a result, independent of the application, this coefficient

does never contribute to the computational burden of the filter.

Design Outline: Assuming an ideal lowpass desired func-

tion consistent with the specification of Fig. 5 with a cut-off

frequency of t = (p + s)/2 = /2 and zero transitionbandwidth, and minimising the integral squared error, yields

the coefficients [15], [30] in compliance with (8):

hk =

t

sin(kt)

kt =

1

2

sin(k 2 )

k 2 , |k| = 1, 2, . . . ,n

2 .(9)


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This least squares design is widely optimal for multirate HBF

in conjunction with spectrally white input signals since, e.g

in case of decimation, the overall residual power aliased by

downsampling onto the usable signal spectrum is minimum

[15]. To master the Gibbs phenomenon connected with (9), a

centrosymmetric smooth desired function can be introduced in

the transition region [30]. Requiring, for instance, a transition

band of width = s p > 0 and using spline transitionfunctions for D(ej), the above coefficients (9) are modifiedas follows [15], [30]:

hk =1

2

sin(k 2

)

k 2

sin(k

2)

k2

, |k| = 1, 2, . . . , n

2, R.

(10)

Least squares design can also be subjected to constraints that

confine the maximum deviation from the desired function:

The Constrained Least Squares (CLS) design [8], [15]. This

approach is also efficiently applicable to the design of high-

order LP FIR filters with quantised coefficients [9].

Subsequently, all comparisons are based on equiripple de-signs obtained by minimisation of the maximum deviation

maxH(ej) D(ej) on the region of support ac-

cording to [24]. To this end, we briefly recall the clever

use of this minimax design procedure in order to obtain

the exact values of the predetermined coefficients of (8), as

proposed in [39]: To design a two-band HBF of even order

n = N 1 = 4m 2, as specified in Fig. 5, start withdesigning i) a single-band zero-phase FIR filter g(k) of oddorder n/2 = 2m 1 for a passband cut-off frequency of 2pwhich, as a type II filter [28], has a centrosymmetric zero-

phase frequency response about G(ej) = 0, ii) upsample

g(k) by two without changing the sample rate, which yieldsan interim filter h(k) of the desired odd length N with acentrosymmetric frequency response about H(ej/2) = 0,iii) lift the passband (stopband) of H(ej) to 2 (0) byreplacing the centre zero coefficient with 2h(0) = 1, and iv)scale this coefficients of the impulse response by 1

2.

Efficient Implementations: Monorate FIR filters are com-

monly realised by using one of the direct forms [27]. In our

case of an LP HBF, minimum expenditure is obtained by

exploiting coefficient symmetry, as it is well known [28], [29].

The count of operations or hardware required, respectively, is

included below in Table I (column MoR). Note that the mul-

tiplication by the central coefficient h0 does not contribute tothe overall expenditure.

The minimal implementation of an LP HBF decimator

(interpolator) for twofold down(up)sampling is based on the

decomposition of the HBF transfer function into two (type 1)

polyphase components [3], [15], [38]:

H(z) = E0(z2) + z1E1(z2). (11)

In the case of decimation, downsampling of the output signal

(cf. upper branch of Fig. 1) is shifted to the system input

by exploiting the noble identities [15], [38], as shown in

Fig. 6(a). As a result, all operations (including delay and its

control) can be performed at the reduced output sample rate

fo = fn/2: Ei(z2) := Ei(zo), i = 0, 1. In Fig. 6(b), the

Fig. 6. Polyphase representation of a decimator (a,b) and an interpolator (c)

for sample rate alteration by two; shimming delay: z1/2o := z

1

input demultiplexer of Fig. 6(a) is replaced with a commutator

where, for consistency, the shimming delay z1/2o := z1

must be introduced [15].

As an example, in Fig. 7(a) an optimum, causal real LP

FIR HBF decimator of n = 10th order and for twofolddownsampling is recalled [4]. Here, the odd-numbered coeffi-

cients of (8) are assigned to the zeroth polyphase component

E0(zo) of Fig. 6(b), whereas the only non-zero even-numberedcoefficient h0 belongs to E1(zo).

For implementation we assume a digital signal processor as

a hardware platform. Hence, the overall computational load of

its arithmetic unit is given by the total number of operations

(times the operational clock frequency [15]) NOp = NM+NAcomprising multiplication (M) and addition (A). All contribu-

tions to the expenditure are listed in Table I as a function

of the filter order n, where the McMillan degree includes

the shimming delays. Obviously, both coefficient symmetry(NM < n/2) and the minimum memory property (nmc < n[3], [10], [15]) can concurrently be exploited.

The application of the multirate transposition rules on the

optimum decimator according to Fig. 7(a), as detailed in [15],

yields the optimum LP FIR HBF interpolator, as depicted in

Fig. 6(c) and Fig. 7(b), respectively. Table I shows that the

interpolator obtained by transposition requires less memory

than that published in [3], [4].

B. Minimum-Phase (MP) IIR Filters

In contrast to FIR HBF, we describe MP IIR HBF always

by its transfer function H(z) in the z-domain.


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Fig. 7. Optimum SFG of LP FIR HBF decimator (a) and interpolator (b)

MoR: fOp = fn Dec: fOp = fn/2 Int: fOp = fn/2

nmc n n/2 + 1NM (n + 2)/4NA n/2 + 1 n/2

NOp 3n/4 + 3/2 3n/4 + 1/2

TABLE I

EXPENDITURE OF REAL LINEAR-P H A S E FIR HBF; n:ORDER, nmc:

MCMILLAN DEGREE, NM(NA): NUM BER OF M ULTIPLIERS (ADDERS)

Specification and Properties: The magnitude response of

an MP IIR lowpass HBF is specified in the frequency domain

by D(ej), as shown in Fig. 8, again for a minimax orequiripple design. The constraints of the designed magnituderesponse

H(ej) are characterized by the passband andstopband deviations, p and s, related [22], [33] by

(1 p)2 + 2s = 1. (12)The cut-off frequencies of the IIR HBF satisfy the symmetry

condition (5), and the squared magnitude responseH(ej)2

is centrosymmetric aboutD(ej/2)2 = H(ej/2)2 = 1

2.

We consider real MP IIR lowpass HBF of odd order n. Thefamily of the MP IIR HBF comprises Butterworth, Chebyshev,

elliptic (Cauer-lowpass) and intermediate designs [37], [41].

The MP IIR HBF is doubly-complementary [28], [31], [37],

satisfying the power-complementarityH(ej)2 + H(ej())2 = 1 (13)and the allpass-complementarity conditionsH(ej) + H(ej() = 1. (14)H(z) has a single pole at the origin of the z-plane, and(n 1)/2 complex-conjugated pole pairs on the imaginaryaxis within the unit circle, and all zeros on the unit circle

[34]. Hence, the odd order MP IIR HBF is suitably realized

by a parallel connection of two allpass polyphase sections as

H(z) =

1

2 A0(z2) + z1A1(z2) , (15)

Fig. 8. Magnitude specifi cation of minimum-phase IIR lowpass HBF

where the allpass polyphase components can be derived by

alternating assignment of adjacent complex-conjugated pole

pairs of the IIR HBF to the polyphase components. The

polyphase components Al(z2), l = 0, 1 consist of cascadeconnections of second order allpass sections:

H(z) =1

2

n121

i=0,2,...

ai + z2

1 + aiz2 A0(z2)

+z1n121

i=1,3,...

ai + z2

1 + aiz2 A1(z2)

,

(16)

where the constants ai, i = 0, 1, ..., (n12

1), with ai < ai+1,denote the squared moduli of the HBF complex-conjugated

pole pairs in ascending order; the complete set of n poles is

given by

0, ja0, ja1, ..., ja n121

[27].

Design Outline: In order to compare MP IIR and LP

FIR HBF, we subsequently consider elliptic filter designs.

Since an elliptic (minimax) HBF transfer function satisfies

the conditions (5) and (12), the design result is uniquely

determined by specifying the passband p (stopband s) cut-off frequency and one of the three remaining parameters:

the odd filter order n, allowed minimal stopband attenuationAs = 20log(s) or allowed maximal passband attenuationAp = 20log(1 p).

There are two most common approaches to elliptic HBF

design. The first group of methods is performed in the analog

frequency domain and is based on classical analog filter design

techniques: The elliptic HBF transfer function H(z) is mappedonto an analog elliptic filter function by applying the bilinear

transformation [27], [29]. The magnitude response of the

analog elliptic filter is approximated by appropriate iterative

procedures to satisfy the design requirements [1], [33], [34],

[40]. The other group of algorithms starts from an elliptic HBF

transfer function (16) and also iteratively optimizes the filter

coefficients by nonlinear optimisation techniques minimising

the peak stopband deviation. For a given transition bandwidth,

the maximum deviation is minimized e.g. by the Remez

exchange algorithm or by Gauss-Newton methods [40], [41].

For the class of elliptic HBF with minimal Q-factor, closed-

form equations for calculating the exact values of stopband and

passband attenuation are known, allowing for straightforward

designs if the cut-off frequencies and the filter order are given

[22].

Efficient Implementation: In case of a monorate filter im-

plementation, the McMillan degree nmc is equal to the filterorder n. Having the same hardware prerequisites as in the

previous section for the FIR HBF, the computational load of


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Fig. 9. Optimum minimum-phase IIR HBF decimator block structure (a)and SFG of the 1st (2nd) order allpass sections (b)

MoR: fOp = fn Dec: fOp = fn/2 Int:fOp = fn/2

nmc n (n + 1)/2NM (n 1)/2NA 3(n 1)/2 + 1 3(n 1)/2

NOp 2n 1 2n 2

TABLE II

EXPENDITURE OF REAL M INIM UM-PHASE IIR HBF; n: ORDER, nmc:

MCMILLAN DEGREE, NM (NA): NUM BER OF M ULTIPLIERS (ADDERS)

hardware operations per output sample is given in Table II

(column MoR). Note that multiplication by the constant of

0.5 does not contribute to the overall expenditure.

In the general decimating structure, as shown in Fig. 9(a),

decimation is performed by an input commutator in con-

junction with a shimming delay according to Fig. 6(b). By

the underlying exploitation of the noble identities [15], [38],

the cascaded second order allpass sections of the transferfunction (16) are transformed to first order allpass sections:ai+z

2

1+aiz2:=

ai+z1o

1+aiz1o

, i = 0, 1, ..., n12

1, as illustrated in Fig.9(b). Hence, the polyphase components Al(z2) := Al(zo), l =0, 1 of Fig. 9(a) operate at the reduced output sampling ratefo = fn/2, and the McMillan degree nmc is almost halved.The optimum interpolating structure is readily derived from

the decimator by transposition. Computational complexity is

presented in Table II, also indicating the respective operational

rates fOp for the NOp arithmetical operations.Elliptic filters also allow for multiplierless implementations

with small quantization error, or implementations with a

reduced number of shift-and-add operations in multipliers [20],[21].

C. Comparison of real FIR and IIR HBF

The comparison of the Tables I and II shows that NFIROp 0, (20)as it is expected from a Hilbert-Transformer. Note that thecentre coefficient h0 is still real, whilst all other coefficientsare purely imaginary rather than complex-valued.

Specification and Properties: All properties of the real HBF

are basically retained except of those which are subjected to

the frequency shift operation of (17). This applies to the filter

specification depicted in Fig. 5 and, hence, (5) modifies to

p +

2+ s +

2= p+ + s = 2. (21)

Obviously, strict complementarity (6) is retained as follows

H(ej(

2)) + H(ej(

2)) = 1, (22)

where (2) is applied in the frequency domain.

Efficient Implementations: The optimum implementation of

an n = 10th order LP FIR CHBF for twofold downsampling isagain based on the polyphase decomposition of (19) according

to (11). Its SFG is depicted in Fig. 12(a) that exploits the odd

symmetry of the HT part of the system. Note that all imaginary

units are included deliberately. Hence, the optimal FIR CHBF

interpolator according to Fig. 12(b), which is derived from

the original decimator of Fig. 12(a) by applying the multirate

transposition rules [15], performs the dual operation with

respect to the underlying decimator. Since, however, an LP

FIR CHBF is strictly rather than power complementary (cf.

(22)), the inverse functionality is only approximated [15].

Fig. 12. Optimum SFG of decimating LP FIR HT (a) and its transpose (b)

Fig. 13. Optimum SFG of decimating linear-phase FIR CHBF

In addition, Fig. 13 shows the optimum SFG of an LP FIR

CHBF for decimation of a complex signal by a factor of two.

In essence, it represents a doubling of the SFG of Fig. 12(a).

Again, the dual interpolator is readily derived by transposition.The expenditure of the half- (R C) and the full-complex

(C C) CHBF decimators and their transposes is listed inTable III. A comparison of Tables I and III shows that the

overall numbers of operations NCFIROp of the half-complexCHBF sample rate converters (cf. Fig. 12) are almost the same

as those of the real FIR HBF systems depicted in Fig. 7. Only

the number of delays is, for obvious reasons, higher in the

case of CHBF.


In the IIR CHBF case the frequency shift operation (2) is

again applied in the z-domain. Using (17), this is achieved by


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the impulse response (8) of any real LP FIR HBF yields the

complex-valued COHBF impulse response:

hk = ejc

4 h(k)zkc = h(k)ej(k+1)c

4 = h(k)jc(k+1)/2, (28)

where n2 k n2 and c = 1, 3, 5, 7. By directly equating(28) for c = 1, and relating the result to (8), we get:

hk =

121+j2 k = 0

0 k = 2l l = 1, 2, . . . , (n 2)/4j(k+1)/2h(k) k = 2l 1 l = 1, 2, . . . , (n + 2)/4

(29)

where, in contrast to (20), the impulse response exhibits the

symmetry property:

hk = jckhk k > 0. (30)Note that the centre coefficient h0 is the only truly complex-valued coefficient where, fortunately, its real and imaginary

parts are identical. All other coefficients are again either purely

imaginary or real-valued. Hence, the symmetry of the impulse

response can still be exploited, and the implementation of anLP FIR COHBF requires just one multiplication more than

that of a real or complex HBF [12].

Specification and Properties: All properties of the real HBF

are basically retained except of those which are subjected to

the frequency shift operation according to (27). This applies

to the filter specification depicted in Fig. 5 and, hence, (5)

modifies to

p + c

4+ s + c

4= p+ + s = + c

2. (31)

Obviously, strict complementarity (6) reads as follows

H(ej(c

4)) + H(ej((1+c/4))) = 1. (32)

Efficient Implementations: The optimum implementation of

an n = 10th order LP FIR COHBF for twofold downsamplingis again based on the polyphase decomposition of (29). Its SFG

is depicted in Fig. 16(a) that exploits the coefficient symmetry

as given by (30).

The optimum FIR COHBF interpolator according to Fig.

16(b) is readily derived from the original decimator of Fig.

16(a) by applying the multirate transposition rules. As a result,

the overall expenditure is again retained (Invariant property of

transposition [15]).

In addition, Fig. 17 shows the optimum SFG of an LP FIR

COHBF for decimation of a complex signal by a factor of

two. It represents essentially a doubling of the SFG of Fig.

16(a). The dual interpolator can be derived by transposition.

The expenditure of the half- (R C) and the full-complex

(C C) LP COHBF decimators and their transposes is listedin Table V in terms of the filter order n. A comparison ofTables III and V shows that the implementation of any type of

COHBF requires just two or four extra operations over that of

a classical HT (CHBF), respectively (cf. Figs. 12 and 13). This

is due to the fact that, as a result of the transition from CHBF

to COHBF, only the centre coefficient changes from trivially

real (h0 =12

) to complex (h0 =1+j

22

) calling for only one

multiplication. The number nmc of delays is, however, of the

order of n, since a (nearly) full delay line is needed both for

Fig. 16. Optimum SFG of decimating LP FIR COHBF (a) and its transpose(b)

Fig. 17. Optimum SFG of decimating linear-phase FIR COHBF

the real and imaginary parts of the respective signals. Note

that the shimming delays are always included in the delay

count. (The number of delays required for a monorate COHBF

corresponding to Fig. 17 is 2n.)


In the IIR COHBF case the frequency shift operation (2) is

applied in the z-domain. This is achieved by substituting the

complex z-domain variable in the respective transfer functions


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LP FIR MP IIR

NOp nmc Fig. NOp nmc Fig.

HBF Decimator 12 8 7 9 3 9

CHBF: R C 11 11 12(a) 8 3 14

CHBF: C C 24 16 13 18 6 15

COHBF: R C 13 14 16(a) 17 5 18

COHBF: C C 28 16 17 36 10 19

TABLE VII

EX P E N D I TU R E S O F R E A L A N D C O M P L E X HB F DECIM ATORS BASED ON

THE DESIGN EXAM PLES OF FIG . 11; NOp: NUM BER OF OPERATIONS

In the case of the odd-numbered HBF centre frequencies of

(1), c {1, 3, 5, 7}, there exist specification domains, wherethe computational loads of complex FIR HBF with frequency

offset range below those of their IIR counterparts. This is

confirmed by the two bottom rows of Table VII, where this

table lists the expenditure of a twofold decimator based on

the design examples given in Fig. 11 for all centre frequencies

and all applications investigated in this contribution. This sec-

toral computational advantage of LP FIR COHBF is, despitenIIR < nFIR, due to the fact that these FIR filters still allowfor memory sharing in conjunction with the exploitation of

coefficient symmetry [12]. However, the amount of storage

nmc required for IIR HBF is always below that of their FIRcounterparts.

To improve the basis of comparison, in a forthcoming

extension of this contribution we will include approximately

LP IIR HBF [33], [34] in our comparative investigations of

optimum filter implementations.

REFERENCES

[1] R. Ansari. Elliptic fi lter design for a class of generalized halfband fi lters.

IEEE Trans. Acoust., Speech, Sign. Proc., ASSP-33(4):11461150, 1985.[2] R. Ansari and B. Liu. Effi cient sampling rate alternation using recursive

IIR digital fi lters. IEEE Trans. Acoustics, Speech, and Signal Process-ing, ASSP-31(6):13661373, Dec. 1983.

[3] M. Bellanger. Digital Processing of Signals - Theory and Practice. JohnWiley & Sons, New York, 2nd edition, 1989.

[4] M. G. Bellanger, J. L. Daguet, and G. P. Lepagnol. Interpolation,extrapolation, and reduction of computation speed in digital fi lters. IEEETrans. Acoust., Speech, and Sign. Process., ASSP-22(4):231235, 1974.

[5] S. Damjanovic and L. Milic. Examples of orthonormal wavelet transformimplementad with II R fi lter pairs. In Proc. SMMSP 2005, Riga, Latvia,ICSP Series No.30, pages 1927, Jun. 2005.

[6] S. Damjanovic, L. Milic, and T. Saramki. Frequency transformations intwo-band wavelet I IR fi lter banks. In Proc. EUROCON 2005, Belgrade,Serbia and Montenegro, pages 8790, Nov. 2005.

[7] G. R. Danesfahani, T. G. Jeans, and B. G. Evans. Low-delay distortion

recursive (IIR) transmultiplexer. Electron. Lett., 30(7):542543, 1994.[8] G. Evangelista. Zum Entwurf digitaler Systeme zur asynchronen Abtas-tratenumsetzung. PhD thesis, Ruhr-Universitt Bochum, 2001.

[9] G. Evangelista. Design of optimum high-order fi nite-wordlength digitalFIR fi lters with linear phase. EURASIP Sign. Process., 82(2):187194,2002.

[10] N. Fliege. Multiraten-Signalverarbeitung: Theorie und Anwendungen.B. G. Teubner, Stuttgart, 1993.

[11] L. Gazsi. Quasi-bireciprocal and multirate wave digital lattice fi lters.Frequenz, 40(11/12):289296, 1986.

[12] H. G. Gckler. German patent application P 19627787, 1996.

[13] H. G. Gckler and H. Eyssele. Study of on-board digital FDM-demultiplexing for mobile SCP C satellite communications. Europ.Trans. on Telecommunications, 3(1):730, Jan.-Feb. 1992.

[14] H. G. Gckler and B. Felbecker. Digital on-board F DM-demultiplexing

without restrictions on channel allocation and bandwidth. In Proc. 7th Int. Workshop on Dig. Sign. Proc. Techn. for Space Communications,

Sesimbra, Portugal, 2001.

[15] H. G. Gckler and A. Groth. Multiratensysteme - Abtastratenumsetzungund digitale Filterbnke. Schlembach Fachverlag, Wilburgstetten, 2004.

[16] H. G. Gckler and K. Grotz. DIA MAN T: All digital frequency divisionmultiplexing for 10 Gbit/s fi bre-optic C ATV distribution system. InProc. EUSIPCO94, Edinburgh, UK, pages 9991002, Sep. 1994.

[17] B. Gold and C. M. Rader. Digital Processing of Signals. McGraw-Hill,New York, 1969.

[18] I. Kollar, R. Pintelon, and J. Schoukens. Optimal FIR and IIRHilbert Transformer design via LS and minimax fi tting. IEEE Trans.

Instrumentation and Measurement, 39(6):847852, Dec. 1990.[19] B. Kumar, S. C. Dutta Roy, and S. Sabharwal. Interrelations between

the coeffi cients of FIR digital differentiators and other FIR fi lters anda versatile multifunction confi guration. Signal Processing, 39(1-2):247262, 1994.

[20] M. D. Lutovac and L. D. Milic. Design of computationally effi cientelliptic II R fi lters with a reduced number of shift-and-add opperationsin multipliers. IEEE Trans. Sign. Process., 45(10):24222430, 1997.

[21] M. D. Lutovac and L. D. Milic. Approximate linear phase multiplierlessIIR halfband fi lter. IEEE Trans. Sign. Process. Lett., 7(3):5253, 2000.

[22] M. D. Lutovac, D. V. Tosic, and B. L. Evans. Filter Design for SignalProcessing Using MATLAB and Mathematica. Prentice Hall, NJ, 2001.

[23] E. De Man and U. Kleine. Linear phase decimation and interpolationfi lters for high-speed application. Electron. Lett., 24(12):757759, 1988.

[24] H. J. McClellan, T. W. Parks, and L. R. Rabiner. A computer programfor designing optimum FIR linear phase digital fi lters. IEEE Trans.

Audio and Electroacoustics, AU(21):506526, Dec. 1973.

[25] K. Meerktter and K. Ochs. A new digital equalizer based on complexsignal processing. In Z. Ghassemlooy and R. Saatchi, editors, Proc.CSDSP98, volume 1, pages 113116, Apr. 1998.

[26] F. Mintzer. On half-band, third-band, and Nth-band FI R-fi lters and theirdesign. IEEE Trans. Acoustics, Speech, and Signal Processing, ASSP-30(5):734738, Oct. 1982.

[27] S. K. Mitra. Digital Signal Processing: A Computer Based Approach.The McGraw-Hill Co., 1998.

[28] S. K. Mitra and J. F. Kaiser, editors. Handbook for Digital SignalProcessing. John Wiley & Sons, New York, 1993.

[29] A. V. Oppenheim and R. W. Schafer. Discrete-Time Signal Processing.Signal Processing Series. Prentice Hall, NJ, 1989.

[30] T. W. Parks and C. S. Burrus. Digital Filter Design. John Wiley &Sons, New York, 1987.

[31] P. A. Regalia, S. K. Mitra, and P. P. Vaidyanathan. The digital all-passfi lter: A versatile signal processing building block. Proc. of the IEEE,76(1):1937, Jan. 1988.

[32] M. Renfors and T. Kupianen. Versatile building blocks for multirateprocessing of bandpass signals. In Proc. EUSPICO 98, Rhodos, Greece,pages 273276, Sep. 1998.

[33] H. W. Schler and P. Steffen. Halfband fi lters and HilbertTransformers. Circuits Systems Signal Processing, 17(2):137164, 1998.

[34] H. W. Schler and P. Steffen. Recursive halfband-fi lters. AE,55(6):377388, 2001.

[35] H. W. Schssler and J. Weith. On the design of recursive Hilbert-transformers. In Proc. ICASSP 87, Dallas, TX, pages 876879, 1987.

[36] G. Strang and T. Nguyen. Wavelets and Filter Banks. Wellesly-Cambridge Press, Wellesley, MA, 1996.

[37] P. P. Vaidyananthan, P. A. Regalia, and S. K. Mitra. Design of doubly-complementary IIR digital fi lters using a single complex allpass fi lter,with multirate applications. IEEE Trans. Circuits and Systems, CAS-34(4):378389, Apr. 1987.

[38] P. P. Vaidyanathan. Multirate Systems and Filter Banks. Signal

Processing Series. Prentice Hall, New Jersey, 1993.[39] P. P. Vaidyanathan and T. Q. Nguyen. A trick for the design of FIRhalf-band fi lters. IEEE Trans. Circuits and Systems, CAS-34:297300,Mar. 1987.

[40] R. A. Valenzuela and A. G. Constantinides. Digital signal process-ing schemes for effi cient interpolation and decimation. IEE Proc.,130(6):225235, Dec. 1983.

[41] X. Zhang and T. Yoshikawa. Design of orthonormal II R wavelet fi lterbanks using allpass fi lters. Signal Processing, 78(1):91100, 1999.

a family of efficient complex halfband filters

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