6report

Wideband Antenna Technology in Military Applications

1. INTRODUCTION

Wideband antennas are essential for military systems requiring spectrum-agility,

novel radar modes or multi-functionality. However, the characteristics that are

‘wideband’ and the physical constraints differ in different systems.

High-performance radar or communication systems, particularly ‘shared aperture’

architectures, require wideband VSWR and radiation pattern characteristics. Some

proposed multifunction phased arrays require multi-octave radiating elements capable

of wide angle (±45-60º) beam scanning. The performance is critically dependent, not

just on the radiating aperture, but the choice of technology used for signal distribution

behind the aperture. Design of the host platform (e.g. aircraft / naval ship) is closely

linked to the requirement to accommodate the sensor system.

In contrast, low / medium gain communication systems, particularly for mobile

applications, require wideband VSWR but can tolerate some radiation pattern

degradation. For communications applications ‘wideband’ might be ‘only’ 20-50%.

Physical constraints are very different from those for phased arrays, with the focus on

low-cost, lightweight solutions that operate in unpredictable environments. Body

worn antennas are gaining prominence in a wide range of communications

applications. When integrated into clothing antenna size is constrained by the

dimensions of the human body.

Here we describe wideband antenna technologies for two military applications

– phased array sensors and body-worn communications systems.

Dept. of Electronics & Communication 1 SJCET Palai


2. WHAT IS A WIDEBAND ANTENNA?

In communications, wideband is a relative term used to describe a wide range of

frequencies in a spectrum. A system is typically described as wideband if the message

bandwidth significantly exceeds the channel's coherence bandwidth. Some

communication links have such a high data rate that they are forced to use a wideband

bandwidth; others links may have relatively low data rates, but deliberately use a

wider bandwidth than "necessary" for that data rate in order to gain other advantages.

A wideband antenna is one with approximately or exactly the same operating

characteristics over a very wide passband. Distinguished from broadband antennas,

where the passband is large, but the gain and/or pattern need not stay the same over

the passband.

The term Wideband Audio or (also termed HD Voice or Wideband Voice) denotes a

telephone conversation using a wideband codec, which uses a greater frequency range

of the audio spectrum than conventional telephone calls, resulting in a clearer sound.

According to the United States Patent and Trademark Office, WIDEBAND is a

registered trademark of Wideband Corporation, a USA based manufacturer of Gigabit

Ethernet managed switches, adapters, and networking equipment.

In some contexts wideband is distinguished from broadband in being broader.



3. WIDEBAND PHASED ARRAYS

In wave theory, a phased array is a group of antennas in which the relative phases of

the respective signals feeding the antennas are varied in such a way that the effective

radiation pattern of the array is reinforced in a desired direction and suppressed in

undesired directions. Phased array transmission was originally developed in 1905 by

Nobel Laureate Karl Ferdinand Braun who demonstrated enhanced transmission of

radio waves in one direction During World War II, Nobel Laureate Luis Alvarez used

phased array transmission in a rapidly-steerable radar system for "ground-controlled

approach", a system to aid in the landing of aero planes in England. At the same time

GEMA in Germany built the PESA Mammut 1. It was later adapted for radio

astronomy leading to Nobel Prizes for Physics for Antony Hewish and Martin Ryle

after several large phased arrays were developed at the University of Cambridge. The

design is also used in radar, and is generalized in interferometric radio antennas.

DARPA researchers recently announced a 16 element phased array integrated with all

necessary circuits to send at 30–50 GHz on a single silicon chip for military purposes.

An antenna array is a multiple of active antennas coupled to a common source or load

to produce a directive radiation pattern. Usually the spatial relationship also

contributes to the directivity of the antenna. Use of the term "active antennas" is

intended to describe elements whose energy output is modified due to the presence of

a source of energy in the element (other than the mere signal energy which passes

through the circuit) or an element in which the energy output from a source of energy

is controlled by the signal input.

Wideband phased arrays offer the advantages of novel radar modes (multi-band, high

resolution) and ‘shared apertures’ performing multiple functions (radar,

communications, EW, satcom) from a common aperture. These reduce the number of

antennas on a platform, simplifying accommodation of multiple systems on small

vehicles (e.g. UAVs) and potentially reducing RCS. The initial / through-life cost,



robustness and serviceability are also important considerations. Vivaldi, Balanced

Antipodal Vivaldi Antenna (BAVA) and Highly Coupled Dipole (HCD) elements

offer multi-octave bandwidths but mutual coupling must be predicted accurately as

part of their design.

3.1. Vivaldi antenna elements:

A Vivaldi-antenna is a co-planar broadband-antenna, which is made from a on both sides

metalized dielectric plate.

Fig 3.1.1: Pattern of a Vivaldi Antenna, made from double sided printed circuit board material

The feeding line excites a circular space via a microstrip line, terminated with a

sector-shaped area. From the circular resonant area the energy reaches an exponential

pattern via a symmetric slot line.

Vivaldi antennas can be made for linear polarized waves or - using two devices

arranged in orthogonal direction - for transmitting / receiving both polarization

orientations.

If fed with 90deg phase shifted signals orthogonal devices can transmit / receive

circular oriented electromagnetic waves.

Vivaldi antennas are useful for microwave frequencies exceeding 3GHz.


http://en.wikipedia.org/wiki/File:Vivaldiantenna.jpg


Advantages of Vivaldi antennas are their broadband characteristics (suitable for ultra-

wideband signals), their easy manufacturing process using common methods for PCB

production, and their easy impedance matching to the feeding line using microstrip

line modeling methods.

The MWEE collection of EM simulation benchmarks includes a Vivaldi antenna

Orthogonal Vivaldi elements (Fig. 1), with feed and slotline regions along a common

axis, have almost coincident phase centres for perpendicular polarisations but slotline

impedance is constrained by the substrate thickness of the orthogonal board.

Alternatively, orthogonal elements can be manufactured in an ‘egg-box’ configuration

with phase centres of orthogonal elements offset along the sides of the array ‘unit cell’

However, wide bandwidth requires current continuity in the ‘corners’ of the unit cell,

complicating manufacture and replacement of failed elements in an operational

environment (a complete board may need to be disconnected and replaced).

Fig 3.1.2: Dual-polar Vivaldi element Fig3.1.3: Offset Vivaldi array

Vivaldi antenna arrays can be designed for operation over 10:1 bandwidths. Simulated

results for an infinite array of stripline-fed Vivaldi elements on Fr=6 substrate

performs well for scan to 450 and frequency 1.8-18 GHz. Vivaldi antenna elements

are typically 2-3 wavelengths long at the highest frequency of the array. The addition

of a properly designed cover to an array of relatively short Vivaldi antennas can

improve its performance over bandwidths on the order of 2:1. Broadside and scanned

performance are improved by the cover. Gaps between elements of the array would be



desirable so the array can be assembled and repaired by inserting independent

modules. However, gaps between the elements cause resonances that destroy the

array’s wideband performance.

Single Vivaldi antennas, also known as "tapered slot" or "notch" antennas, can

operate as end-fire, traveling wave antennas producing somewhat directive radiation

patterns with 8-10 dBi gain and sidelobe levels of-10 to -15 dB when the antenna is a

few wavelengths long. As a traveling wave antenna, the impedance and pattern

bandwidths are moderately wide. Printed Vivaldi antennas are easy to fabricate,

having no highly sensitive dimensional tolerances. The dielectric substrate of the

printed antenna slows the traveling wave in the slot, producing enhanced endfire gain

for a band of frequencies, but also causing pattern degradation when the dielectric

loading becomes too great.

When operating in a scanning array, Vivaldi antennas must be spaced no more than

0.5 wavelengths at the highest frequency of operation. In this configuration, the

Vivaldi antennas are too small to operate effectively in the travelling wave mode.

Rather, they work with neighboring elements to radiate a collimated beam of radiation

with good impedance match to the guided power of the feed lines. As in all phased

arrays, mutual coupling between the Vivaldi elements plays an important role in

performance. By properly designing the Vivaldi elements to use mutual coupling,

antenna arrays can be designed to operate over a decade of frequency and to scan

more than 450 from broadside. The excellent performance of Vivaldi antenna arrays

is achieved when the elements are electrically connected to their neighbors. It would

be preferable to construct the array by using individual elements or small sub arrays

that can be easily inserted or removed from the aperture. The array could then be

manufactured, maintained, reconfigured or scaled by placing elements to fill the

desired aperture area. Unfortunately, separation between the Vivaldi elements causes

anomalies that disrupt the wideband performance of the array. The Doubly Mirrored

Balanced Antipodal Vivaldi Antenna (DmBAVA), a variation of the BAVA, is not

connected to its neighbors and it has been successfully designed to operate over two

octaves of bandwidth. The Vivaldi elements of a phased array can be as short as about

0.25 wavelengths at the lowest operating frequency. This makes the array reasonably

compact, but some applications require even shorter elements. Recent work shows



that a properly designed dielectric radome can enhance the performance of short

elements, especially when the required bandwidth is less than about 3:1. Because

Vivaldi antenna arrays can operate over very wide bandwidths, their use in ultra

wideband (UWB) systems is feasible. In such applications, signal integrity becomes

important. Simulations of large (infinite array approximation) Vivaldi arrays show

that these arrays transmit signals with nearly constant amplitude and low delay

dispersion over their band of operation, resulting in very little distortion of the

transmitter pulse. Compared to dielectric-free Vivaldi antennas, arrays with a

dielectric substrate have slightly better amplitude bandwidth, but slightly higher

dispersion of the group delay.

3.1.1 Wideband Vivaldi Arrays

Vivaldi antenna arrays are particularly suited to applications requiring operation over

multiple octaves of frequency and electronic scanning to 450 or more. The VSWR

simulated for an infinite, dual-polarized array is shown in Fig. 1. The predicted

VSWR for this array with substrate permittivity Er=6 is mostly less than 2 and less

than 2.7 for all scan angles to 450 and over the frequency range 1.8-18 GHz. Similar

antennas have been designed with VSWR < 2 for these frequency and scan ranges.

3.1.2 Principle of operation

As stated earlier, the Vivaldi antenna is a type of a traveling-wave antenna of the

“Surface-type”. The waves travel down the curved path of the flare along the antenna.

In the region where the separation between the conductors is small when compared to

the free-space wavelength, the waves are tightly bound and as the separation

increases, the bond becomes progressively weaker and the waves get radiated away

from the antenna. This happens when the edge separation becomes greater than half-

Radiation from high-dielectric substrates is very low and hence for antenna

applications significantly low dielectric constant materials are chosen.

The wideband performance of Vivaldi arrays is achieved only if the array elements

are electrically connected to adjacent elements. This makes it difficult to manufacture

and repair the Vivaldi arrays. If the elements are individually fabricated and inserted



into an array without insuring electrical contact to neighboring elements, severe

anomalies limit the array bandwidth to approximately 3:1 see Figure.

Figure 3.1.4: VSWR of typical Vivaldi arrays with complete electrical contact and with gaps between

elements. The gaps cause anomalous resonances.

3.1.3 Vivaldi Arrays with Radome Matching

In many applications, it is desirable to make the Vivaldi elements as short as possible.

However, the achievable bandwidth of the array decreases when the length of the

Vivaldi elements become too short. It has long been recognized that single or multiple

layers of dielectric can be used in close proximity to an array aperture to improve the

impedance match versus frequency and or scan. Initial studies of Vivaldi arrays show

that a properly designed cover improves bandwidth and impedance match even for

relatively short elements.

The basic configuration of a dielectric cover at the array aperture is shown in Figure,

where a single sheet of thickness t and permittivity Er is placed flush against the array

surface.



Figure3.1.5: Single-polarized Vivaldi array with dielectric cover.

A Vivaldi element with length equal to width typically is not well matched over an

octave of bandwidth. Such an element is shown in Figure 4 along with its VSWR for

several cover thicknesses. For this cover, Er = 4, the best VSWR is obtained for cover

thickness in the range 0.5cm < t < l cm. The upper frequency limit of the plot is 1.875

GHz, where the element spacing (and length) is Xo/2. The scan performance of the

array with a somewhat optimized cover, Er = 4 and t = 0.6 cm, is shown in Figure .

The cover improves match across the band, resulting in 2.5:1 bandwidth for 2:1

VSWR. Performance improvement is obtained for 45° scan in E and H planes as well,

except for 450 H-plane at the highest frequencies.

The dielectric cover can degrade array performance if its electrical thickness becomes

too great. Blindness can occur when the cover supports a guided wave with

propagation constant in the aperture plane that approximately matches the transverse

wave number of a higher order Floquet mode for the array. For thicknesses greater

than about 1 cm, blindness occurs at 450 for higher frequencies within the operating

band. The blindness moves to lower frequencies as the cover thickness increases.

3.1.4 Dispersion Characteristics of Vivaldi Arrays

The availability of very wide bandwidth antenna arrays and radar systems creates a

need to understand and characterize the time-domain behavior of antenna arrays for

very short pulses. By using infinite array analyses like those typically used for

wideband array design and Fourier transforms; we have characterized the



performance of some (infinite) wideband arrays in terms of the normalized transfer

function and associated group delay.

Vivaldi antenna arrays of the type we studied extensively have been characterized and

compared to arrays of long slots with periodic excitation, TEM horns, and

capacitively coupled dipoles. The magnitude of the normalized transfer function and

the normalized group delay deviation of a Vivaldi array are shown in Fig. 9. This

particular Vivaldi array has VSWR < 3 for 0.2 <ff0 < 1 and VSWR < 2 for0.25 < ffo

< 1. The array has low dispersion over its operating band. Previously, we have found

that including a dielectric substrate of moderately high permittivity, Er - 4-10, can

improve array VSWR performance. However, the dispersion studies indicate that

higher permittivity substrates cause greater group delay deviation.

3.1.5 Bandwidth characteristics

At different frequencies, different parts of the antenna radiate, while the radiating part

is constant in wavelength. Thus the antenna theoretically has an infinite bandwidth of

operation and can thus be termed frequency independent. As the wavelength varies,

radiation occurs from a different section which is scaled in size in proportion to the

wavelength and has the same relative shape. This translates into an antenna with very

large bandwidth. Again referring to Figure 2.1 it can be seen that the Vivaldi antenna

is divided into two areas:

a propagating area defined by WE < W < WA

a radiating region defined by WA < W < WO

Where,

W - Slot width

WE - Input width

WA - Slot width at radiating area

WO - Output width

The original Vivaldi antenna proposed earlier employed a taper that opened up

real fast thus providing an almost constant beamwidth over the entire frequency range

of about two octaves when plotted with frequency or normalized length. For antennas

with smaller opening angles, the beam width becomes dependent upon the frequency.

Theoretically, the TSA is capable of having an operating bandwidth within a frequenc

range of 2 GHz to 90 GHz while practically the operating bandwidth is limited by the



transition from the feeding transmission line to the slot line of the antenna and by the

finite dimensions of the antenna. Thus to achieve a wider bandwidth, it is imperative

for the designer to have in mind the following two aspects:

The transition from the main input transmission line to the slot line for feeding

the antenna. This is designed for a low reflection coefficient to match the

potential of the antenna.

The dimensions and shape of the antenna, to obtain the required beam width,

side

lobes and back lobes, over the operating range of frequencies

3.1.6 Dual-Polar Vivaldi Antennas

Linearly polarized Vivaldi radiating elements were introduced by Gibson and have

received wide attention due to their broad bandwidths and relative ease of

manufacture. Development of Vivaldi elements with a dual-polar capability is more

complex because of the non-planar antenna geometry and feed configuration. Also, in

array applications, the element design must take account of the complex mutual

coupling environment at the array aperture. Prediction of the coupling environment,

and in particular the variation of the antenna match with array scan angle, requires use

of full-wave electromagnetic modeling techniques. In this paper, we describe the

design and evaluation of dual polar Vivaldi elements optimised for operation in the

array environment with frequency bandwidths of up to 3: l and scan angles of up to

60”. The wide-angle scan performance of elements immersed within infinite arrays

has been modelled using FDTD techniques. Comparison of FDTD predictions with

the measured performance of isolated elements is presented. – The wide scan range

and bandwidth requirements required the use of customised “blind-mate” connectors.

3.2 BALANCED ANTIPODAL VIVALDI ELEMENTS

The Vivaldi antenna, a form of tapered slot radiator, has been shown to produce well

Performance over a wide bandwidth, limited only by the traditionally used slot line to

microstrip feed transition. The authors present a new antenna, the balanced antipodal

Vivaldi, which incorporates an ultra-wide bandwidth transition and overcomes the



poor polarisation performance of the antipodal form. Good performance over a 1 to 40

frequency range has been obtained. The use of the antenna in a linear phased array has

also been investigated using elements constructed on high permittivity substrate.

Wideband wide angle scanning with good cross-polarisation levels is obtained.

Fig3.2.1: Single BAVA element

Multi-octave performance phased arrays are important for a number of applications,

including elctronic warfare and multiple mode radar systems. Wide bandwidth array

action is obtained primarily through the use of wide bandwidth array elements,

although arrays incorporating clusters of elements covering sections of the desired

bandwidth have been reported. Such elements should have, in addition to wide

bandwidth, symmetrical beamwidths to optimize scanning and should be compact to

allow sufficiently small element spacing to prevent grating lobe formation at the

maximum operating frequency. An additional requirement is that the element should

allow integration with transmit receive modules constructed using a printed circuit

transmission medium such as microstrip. There are several ways of creating a wide

bandwidth array element. The ridged horn exhibits bandwidths of up to two octaves

with a highly symmetric beam and good power handling capability. The spiral

antenna has bandwidths in excess of four octaves but requires a wide bandwidth

balun. Log periodic antennas have been used in HF or VHF sky wave radar to give

wide angle scanning. However, none of these elements are of a suitable form for

circuit integration. The tapered slot antenna, however, can be fabricated using printed



circuit techniques and is thus ideal for circuit integration. The slot antenna can be

fabricated in either triplate stripline or microstrip. The stripline version, known as the

tapered notch, is generally fabricated with an exponential taper. All other types of

tapered slot are fabricated on micro strip and include the Vivaldi, with an exponential

taper, the linear taper, broken linear taper and constant width slot antennas . All these

antennas exhibit low cross-polarisation characteristics in the principle planes,

however in the diagonal plane the CO- to cross-polarisation ratio decreases rapidly

away from bore sight . This group of antennas is now widely used not only in phased

arrays but also in radio astronomy, remote sensing, multiple beam satellite

communications and spatial power combining techniques. In this study both the

tapered stripline notch and the Vivaldi antennas have been tested using identical

elliptical tapers, these antennas being fed by stripline and micro strip respectively.

Nearly identical performance is noted in our studies, in terms of gain, beamwidths and

cross-polarisation, while references and suggest differences in operation. However

one difference which is pertinent to phased array operation is that the Vivaldi antenna

has an open feed line which can radiate and perturb the radiation pattern. Although

both these elements can have equal beamwidths and can in principle be directly

connected to an integrated circuit, the slot line to feed line transition limits the

bandwidth and requires considerable ingenuity to give broadband performance. This

paper describes the development and performance of a new tapered slot antenna

element that overcomes the transition problem to produce an ultra-wideband element

for circuit integration. Performance in a small linear phased array is also presented

and performance discussed. If the feed transition is made a collinear extension of the

slot, then the band limiting effect is removed giving very wide bandwidth operation.

In the antipodal Vivaldi, a smooth transition between twin line and microstrip is used.

The metallization on either side of the substrate is flared in opposite directions to form

the tapered slot It is clearly seen that band limitation caused by the Vivaldi transition

is removed and wideband action is indeed obtained. The lower frequency limit is now

determined by the cut-off mechanism of the flare, namely that at the lowest operating

frequency the aperture is half a wavelength wide. However the antipodal nature of the

antenna gives rise to very high levels of cross polarisation particularly at high

frequencies, due to the skew in the slot fields close to the throat of the flare.



To overcome this high cross-polarisation we have added a further layer of

metallisation to form a balanced antipodal Vivaldi. The resultant electric field in the

slot region is now oriented parallel to the metallisation whilst the output transmission

medium is triplate stripline. Elliptical radiating tapers were chosen for this antenna

because previous work with the Vivaldi and tapered notch antennas showed that this

particular taper gave similar E and H beamwidths. The notch length-to-width ratio

was set at 2:l to give gain of between 5 and l0dBi as found in our previous Vivaldi

studies

The input return loss, is similar to that obtained for the antipodal Vivaldi while the

cross polarisation, Fig. 3, is improved and is typically below -20dB for this particular

E~ = 2.32 substrate antenna. Fig. 6 shows that E and H beamwidths are approximately

equal and constant over a 6 to l8GHz bandwidth.

It can be seen from Fig. 7 that there is a squint of about 15" in the E-plane radiation

pattern. This squint appears to be independent of both frequency and permittivity.

It is believed that this squint is due primarily to the unequal propagation velocity

experienced by the currents on each side of the slot due to their different

geometries. Measurements of the antenna aperture fields confirmed the presence of

phase asymmetries together with some amplitude asymmetry. Various methods were

tried to reduce the squint including cutting away some of the substrate, reducing the

distance between the radiating flares and the transition flares, the addition of

balancing flares and shorting pins between the two outer flares. However none of

these reduced the squint but the use of asymmetric flares allowed the introduction of

some asymmetry into the patterns which was found to offset the squint as shown

in Fig. 7. The inset shows the shape of the flares used in this example. The tapers in

the transition are of a similar elliptical form with length and spacing between tapers

chosen to be greater than half a wavelength at the lower operating frequency. We

have not performed optimisation on these transitions and it may well be that shorter

antennas could be developed.



3.2.1 Arrays of balanced antipodal Vivaldi antennas

z y

X

Fig 3.2.2: Balanced antipodal Vivaldi array

Although the balanced antipodal Vivaldi antenna is intended to be used in a dual

polarised array, only its performance in E-plane arrays has been demonstrated. To

incorporate this wideband element into a scanning array, the elements must be placed

at &I2 at the highest frequency, where ho is the free space wavelength. Thus at the

bottom end of a 3:l band these elements will be spaced by h016. It is well known that

tapered slot antennas exhibit a low frequency cut-off which occurs when the

maximum flare width at the aperture is hJ2, where hs is the wavelength in the slot.

Although this may not occur in very large phased arrays due to high mutual coupling

at the lowest frequency, it will occur in the small arrays considered here and the

following design method is therefore appropriate. Assuming an effective dielectric

constant of cS in the slot these two conditions are met when (1) A0 - As - A0 6 2 &2

giving E, = 9. The substrate dielectric constant to achieve this could be approximately

derived from uniform slot theory but its value is constrained both by the materials

available and the likely materials to be used in the microwave integrated circuit. The

value chosen for the initial demonstrator was 10.5, with an additional array being

made on E, = 6 material while using separation of alternative elements in the H-plane



to prevent grating lobes. The silhouette of the E, = 10.5 array is shown in Figs. 10 and

11 with that for the E~ = 6 array being similar. The seven element array was produced

using one manufacturing process, to work over a 3 to 9GHz range. Initial results for a

single element using a network analyser in both frequency and time domain mode

revealed a substantial reflection from the dielectric edge at the flare aperture. Shaping

of a dielectric extension beyond this aperture was found to reduce this reflection, with

a semicircular extension as indicated in Figs. 10 and 11, giving optimum

performance. This was then used on all array elements, for both the 10.5 and 6

permittivity substrates. The beamwidths for the E, = 10.5 elements are in general very

large and therefore the E-plane asymmetry noted in the low permittivity elements is

not observed. This is due to surface wave interaction and the fact that the radiation

occurs from the front of the dielectric extension. In the E, = 6 case the trapped

waves are reduced and some element asymmetry is observed in the E-plane. The

principle plane cross polarisation levels in these higher permittivity elements

are similar to those found in the low permittivity versions (5 -20dB). However the

cross-polarization in the diagonal planes is generally better in the higher dielectric

constant medium. This is due to the fact that the physically smaller elements have

shorter longitudinal current paths and therefore an increase in effective cancellation

occurs. 0 -10 dB -20-30 The limitations on the bandwidth of the Vivaldi antenna due

to the slot line to microstrip transition have been overcome whilst preserving low

cross-polarisation by the development of the balanced antipodal Vivaldi antenna. The

new antenna allows simple integration with microwave integrated circuit transmit/

receive modules using an additional stripline to microstrip transition which on E, =

10.5 substrate has been shown to have a loss of less than I dB An antenna on E, =

2.32 substrate has been shown to have a bandwidth in excess of 40:1, whilst over a

3:1 bandwidth, cross-polarisation below -20dB is obtained. Radiation patterns are in

general well controlled but an E-plane squint is noted which can in principle be

compensated for using asymmetrical flares. Performance on E, = 10.5 substrate and to

some extent on E, = 6 is affected by the dielectric-air interface at the flare aperture

and this mismatch has been reduced with the introduction of a semicircular substrate

extension. Two 7-element E-plane arrays of these balanced antipodal Vivaldi

elements have been constructed, one on E, = 10.5 and the other on E, = 6.0 with a



triangular lattice structure to avoid the formation of grating lobes at the high

frequency end of the band. Wideband wide angle scanning has been achieved with

these arrays while maintaining suitable cross-polarisation levels.

Phased arrays for use in future multifunction systems require wideband elements

capable of wide angle scanning. Vivaldi elements are widely used but their

performance is sensitive to current continuity with neighbouring elements. This

complicates manufacture and element replacement in operational use, especially for

dual-polar versions. Here we describe the design of a dual-mirrored, dual-polar

BAVA element (DmDpBAVA) for use in large phased array antenna apertures over

the 6-18GHz frequency range. FDTD simulation, validated against a Finite Element

code, is used to predict the array performance over a 3:1 frequency range for beam

scanning out to ±50°. BAVA elements do not require connection with adjacent

elements and are therefore attractive for use in phased arrays with modular

construction Vivaldi antennas are widely established as broadband radiating antenna

elements and have been used in a wide variety of military and civil applications.

Isolated Vivaldis can achieve bandwidths of many octaves. However, in a wideband

array the element spacing is small at the bottom of the band and mutual coupling is an

important factor in determining array performance, especially at wide scan angles.

Here we describe design of Balance Antipodal Vivaldi Antenna (BAVA) radiating

elements for use in wide scanning array applications at 6-18GHz. The use of BAVAs

simplifies the design, manufacture and maintenance in operational use of the radiating

elements

3.2.2 BAVAS for modular array construction

In some applications it is desirable to use dual-linear or circular polarisation for

maximum versatility in array apertures. Examples include novel radar modes, satellite

data terminals and multifunction shared aperture array antennas. Two approaches are

available for wideband dual-polar Vivaldi operation. Orthogonal Vivaldi elements

(Figure 3.1.2) can be constructed in which the feed and slot line regions lie along a

common axis so that phase centers for perpendicular polarisations (e.g. ‘H’ and ‘V’)

are almost coincident. However, a wideband hybrid-coupler arrangement is required



in the feed region and the slotline width (and impedance) is constrained by the

substrate thickness of the orthogonal board. Alternatively orthogonal elements can be

manufactured in an ‘egg-box’ configuration (Figure 3.2.2) with phase centers of

orthogonal elements separated along the sides of the ‘unit cell’ of the array. Arrays of

such elements have been demonstrated for wideband applications. However, the

operational bandwidth, defined in terms of acceptable VSWR, is sensitive to the

continuity of currents in the ‘corners’ of the unit cell. The practical implications of

this are,

complex manufacture that must achieve adequate control of currents in the

‘corners’ of the unit-cell

Maintenance or replacement of individual array elements is difficult, if not

impossible; in an operational environment (a complete board must be

disconnected and replaced).

The manufacturing process must therefore ensure continuity of currents in the

corners. Alternatively, designs have been investigated having no electrical contact

in the array corners with the conducting ground plane surfaces terminated a short

distance from the corner of the unit cell. However, these geometries are narrow

band and can exhibit VSWR resonances due to additional modes propagating

between the substrate boards. Elsallal and Schaubert [5], [6] have investigated a

development of the Vivaldi element described as the Balanced Antipodal Vivaldi

Antenna (BAVA). An advantage of this class of element is that adjacent elements

need not be in physical contact. Arrays of orthogonal BAVA elements can

therefore be manufactured from individual elements without the need for current

continuity in the corners of the unit cell. Individual elements are then easier to

maintain or replace in an operational environment and a modular approach can be

used in the manufacture and maintenance of the arrays. The BAVA element could

be manufactured as an integral part of the transmit / receive module (TRM) in the

array. Failed radiating elements / TRMs can then be replaced as single integrated

items during the lifetime of the equipment. Arrays of BAVA elements exhibit a

number of resonance phenomena that limit the useable bandwidth of the array.

However, Elsallal and Schaubert discovered that a simple rotation of alternate

elements, forming what has become known as Dual-Mirrored Dual-Polar BAVA

(DmDpBAVA) elements, can suppress some resonances. In this paper we describe



design of a DmDpBAVA element for use in large phased array antennas with

octave bandwidth and wide scan capability (>±50º) in any plane. Both Finite

Difference Time Domain (FDTD) and Finite Element (FE) simulation techniques

were used to predict the active reflection coefficient. In particular we show,

DmDpBAVA elements are capable of achieving wide scan angle (>±50º)

performance across at least an octave bandwidth.

Use of DmDpBAVA elements suppresses some resonance phenomena.

Confirmation, using FDTD simulations, of the BAVA resonance phenomena

reported in reference [6].

Unit-cell aperture field distributions for BAVAs at resonance and non-

resonance frequencies.

A possible explanation for a resonance condition in BAVA arrays which

explains why resonances are suppressed in dual-mirrored BAVA geometries.

3.2.3 BAVA radiating elements

A typical linearly polarised BAVA element is shown in Figure 3.3.1. The BAVA is

driven by a stripline feed and is manufactured using conventional PCB techniques.

The outer arms of the element are extensions of the stripline ground planes (not

shown in the diagram), and the inner arm is a continuation of the stripline inner

conductor. The BAVA has superior polarisation characteristics to the antipodal

Vivaldi array [3], as the outer arms are symmetrically placed relative to the inner

conductor. The BAVA has an asymmetric structure leading to high cross-polar. All

modelling presented here assumes that the BAVA elements work against a continuous

ground plane that is perpendicular to the base of the stem (the z=0 plane in Figure

3.2.1). Periodic boundary conditions are used to simulate the performance of the



element immersed in the mutual coupling environment of an infinite array.

Fig 3.2.3: SpBAVA Fig 3.2.3: mSpBAVA

Figure shows unit cells for a linearly (singly) polarized BAVA (SpBAVA), and a

mirrored version (mSpBAVA). The latter shows that alternate elements in the array

are mirrored about the y axis. The active match of the two variants predicted using FE

modelling is shown in Figure 5 and is in excellent agreement with Figure 4 of [6].

This confirms that mirroring is needed to eliminate the unwanted resonance at

7.2GHz, which appears to occur when the diagonal spacing between elements is

approximately half a free-space wavelength.

3.2.4 Dual polarised array

Figure 8 shows unit cells for 3-9GHz dual-polar, unmirrored (DpBAVA) and

mirrored (DmDpBAVA) radiating elements based on the linearly polarised element

geometry in [6]. As for the single polarised array, mirroring is used to remove the

mid-band resonance. Mirroring is required for both polarisations, resulting in the

DmDpBAVA unit cell containing eight elements, due to the changed symmetry

planes of the array. Modelling using our FDTD code Agate is preferred to use of a FE

code because of the reduced memory requirement and because the performance at

multiple frequencies can be obtained from a single simulation.



Fig3.2.4: DmDpBAVA unit cell Fig3.2.5: DpBAVA unit cell

Design of a BAVA element has demonstrated a scan capability of ±50º over an octave

bandwidth for a 2:1 VSWR. Wider bandwidth can be obtained by a trade-off with

maximum scan angle. Simulations show good agreement between FDTD, FE and

results presented in the literature. Calculations of field distributions in the aperture

plane show rotation of the E-field at one of the resonance conditions which appears to

occur when the diagonal element spacing is close to half the free-space wavelength. In

common with other wideband element designs such as the Vivaldi, best performance

is achieved in the E-plane. A disadvantage of ‘egg-box’ configurations of Vivaldi

antennas as candidates for dual polar radiating elements is that the performance is

sensitive to current continuity in the ‘corners’ of the unit-cell, complicating

manufacture and maintenance. BAVA element geometries do not require current

continuity or even physical contact between neighboring elements. This simplifies

manufacture and replacement of elements during operational use and allows a

modular construction of sensor arrays of different aperture sizes.

3.3 HIGHLY COUPLED DIPOLE ELEMENTS

An array of dipole antenna elements is formed on the substrate with each dipole

antenna element positioned on a respective one of the array tiles. Each dipole antenna

element includes a medial feed portion and a pair of legs extending outwardly

therefrom. Adjacent legs of adjacent dipole antenna elements include respective

spaced apart end portions forming a gap between the respective end portions and

defined by separate tiles. A capacitor coupler is positioned at each respective spaced

apart end portion of adjacent legs and bridging a gap for capacitive coupling



respective spaced apart end portions of respective adjacent dipole antenna elements

together.

Fig 3.1: typical HCD element

With space at a premium and a need for increasing functionality at lower cost, multi-

octave phased arrays will enable shared aperture systems to be designed incorporating

multiple RF functions. The Highly Coupled Dipole (HCD) elements based on

geometries described below have coincident phase centers for orthogonal

polarizations. Calculations predict that such arrays are capable of two-octaves

bandwidth. However, manufacture requires accurate tolerance control in multiple

antenna and feed components across the array. This shows a dual polarised

implementation using coaxial cables. However, initial investigations prior to the

design optimisation indicated that a stripline feed is easier to fabricate.

Fig 3.3.1: HCD Elements Fig3.3.2: HCD Demonstrator Array

]Lightweight phased array antennas having a wide frequency bandwidth and a wide

scan angle can be economically manufactured and conformally mounted on a surface,



such as a nose cone of an aircraft. Examples of such antenna include a current sheet

array (CSA) formed of at least one dipole layer and using coupling capacitors

between antenna dipole elements. The capacitors often are formed as interdigitated

"fingers." The coupling capacitance between dipole elements can be increased by

lengthening the capacitor "digits" or "fingers," which results in additional bandwidth

for the antenna

Often these types of phased array antennas are formed as large arrays, often with sub

arrays, and operable in the 2.0 through 18.0 GHz range. They can be constructed from

different modules with separate array panels, for example, each about 12×18 inches

and forming an antenna aperture. They can be constructed with an interdigitated

assembly of various beam former components, sub array beam formers,

transmit/receive modules and associated components, with connections that are ribbon

bonded to antenna feed portions and associated legs extending outward therefrom.

The antenna elements form the dipoles. As a result, these phased array antenna

structures have an array of tightly packed and closely spaced dipole elements

connected to neighboring dipole elements through capacitor coupling.

The antenna can have dual polarization by using horizontal and vertical dipole

elements and solder connections at feed points. The capacitor coupling imparts a

broadband performance, and can be formed using interdigitated or in some cases end-

coupled capacitor elements. The interdigitated capacitor elements have lengthened

"fingers" so as to increase capacitance The current sheet array (CSA) or dipole layer

has typically closely-coupled, dipole elements embedded in dielectric layers above a

ground plane. Inter-element coupling in these prior art examples is achieved with

interdigital capacitors. Coupling can be increased by lengthening the capacitor

"fingers". The additional coupling provides more bandwidth. It is believed that the

capacitors tend to act as a bank of quarter-wave (λ/4) couplers. Coupling can be

maintained to extend the bandwidth of a particular design. In this prior art example,

the necessary degree of inter-element coupling can be maintained by placing coupling

plates on separate layers around or adjacent to the interdigital capacitors. Shortening

the capacitor "digits" or "fingers" moves the gain dropout out-of-band, but reduces

coupling and bandwidth. Adding the coupling plates the capacitive coupling to

maintain or improve bandwidth on these separate layers increases.



4. SIGNAL DISTRIBUTION AND BEAM FORMING

Technologies behind the antenna aperture must support wideband operation if the

benefits of the element bandwidth are to be realised. In many cases there are mass and

volume constraints on the antenna system itself (e.g. on a ship’s mast) and there is a

requirement to move wideband signals over significant distances. Optical fiber is

suitable for distributing RF signals, and is now sufficiently mature for use in radar and

EW systems. Optical fiber is inherently wideband and has low mass, low loss and

good EMC properties. However, it has been difficult to support the high dynamic

ranges required for radar systems because of the modulation techniques used. Recent

improvements in device technology have allowed much larger dynamic ranges to be

supported, and the technology is now more attractive to system designers. Phased

arrays implemented with phase weights are intrinsically narrow band as the weights

are approximations to time delays. Operation over wide bandwidths then leads to

beam squint and loss of gain. For wide band operation the use of True Time Delay is

required, where time delays rather than phase shifts are applied to the element signals.

For large arrays time delays are bulky and lossy if implemented in conventional

microwave technologies, and the use of optical fiber is attractive. Systems have been

demonstrated that provide a single scanned beam and a set of staring beams. Optical

technology has also been used for very wideband signal sampling, providing the

antenna system with a wide instantaneous bandwidth. A number of sampling

strategies can be used, such as using fast optical switches to interleave a number of

conventional analogue to digital convertors. Although waveform generation is not

normally required to have the same bandwidth as a receiver, a similar approach in

reverse can be used for wideband waveform generation. Advances in digital

technology, such as the increasing speed and capacity of FPGA technologies, allow

very high bandwidths to be handled using highly parallelized processing architectures.

It is now possible to construct wideband RF systems that support many multi-function

operations.



Beamforming is a signal processing technique used in sensor arrays for directional

signal transmission or reception. This spatial selectivity is achieved by using adaptive

or fixed receive/transmit beam patterns. The improvement compared with an

omnidirectional reception/transmission is known as the receive/transmit gain (or loss).

Beamforming can be used for both radio or sound waves. It has found numerous

applications in radar, sonar, seismology, wireless communications, radio astronomy,

speech, acoustics, and biomedicine. Adaptive beamforming is used to detect and

estimate the signal-of-interest at the output of a sensor array by means of data-

adaptive spatial filtering and interference rejection. Beamforming takes advantage of

interference to change the directionality of the array. When transmitting, a

beamformer controls the phase and relative amplitude of the signal at each

transmitter, in order to create a pattern of constructive and destructive interference in

the wavefront. When receiving, information from different sensors is combined in

such a way that the expected pattern of radiation is preferentially observed.

5. BODY WORN ANTENNAS

In the military domain, communications is of paramount importance and increasingly

every military asset forms a part of a large, ad-hoc network. Military personnel are

expected to carry a large amount of equipment and are likely to require the addition of

sensors around the body for situational awareness and medical monitoring. To avoid

an increase in the already significant burden suffered by military personnel, new

technology must remain as unobtrusive as possible. Flexible, lightweight electronics

and antennas are, therefore, of interest in the military sector. This paper presents the

development and assessment of flexible, body wearable antennas Much of the open

literature has concentrated on WLAN, GSM or UWB frequencies [2-4] but this work

considers the frequency range from 100MHz to 1GHz. This presents a challenge as

antennas which operate at this frequency are typically of comparable size to the body.

Flexible antennas which conform to the body are essential at these frequencies to

provide unobtrusive solutions. An added challenge is that a requirement of this work

was to develop wideband solutions. Three antenna designs are considered, a spiral, a



bowtie and a broadband wire dipole. The broadband wire dipole covers the lowest

frequencies, from 100MHz to 250MHz. This element consists of a wire dipole with

lumped loads placed in each arm to produce a broadband antenna. The spiral and

bowtie are both capable of covering the band from around 250MHz to 800MHz.

Scaled versions of the spiral and bowtie can cover higher bands as required. The

antenna designs have previously been reported [5] so will not be repeated here in

detail for reasons of brevity. Although the work was specifically aimed at body

wearable antennas, the manufacturing techniques are applicable to the general

development of conformal antennas and electronics. The majority of the techniques

considered are low cost processes requiring no specialized equipment and utilizing

commercially available materials. The antennas have been designed using full-wave

Computational Electromagnetics (CEM) software. The antennas were modelled

placed onto the body and the effect of proximity to the body is discussed in Section II.

A review of novel candidate manufacturing techniques has been conducted

to provide methods to allow the integration of flexible antennas into typical garments.

Several manufacturing techniques were identified and trialled. These include

conducting nylon, copper coated fabric, weaving conducting thread, conducting nylon

sheets, thin conducting mesh, screen printing conducting inks and spraying

conducting paints. Each manufacturing technique was utilised to fabricate one or

more of the antennas designs and to integrate the antenna into clothing. The materials

and manufacturing techniques are presented in Section III. Each technique was

assessed by performing input impedance measurements and calibrated gain

measurements with the antennas integrated in clothing and worn on a real human.

5.1 Electromagnetic modelling

The three antenna types were designed using an in-house Finite Difference Time

Domain (FDTD) tool. The antennas were modelled in free-space and worn on the

body with various antenna/body spacing. The material parameters for the body were

taken from the FCC website [6]. Several computer models were trialled to ascertain

the fidelity of the model required at these frequencies. The first body model used a

homogenous model with the ‘Average Muscle’ dielectric properties (see Table 1). The

separation between the body and antenna is a critical parameter affecting the antenna



performance. Even at the relatively long wavelengths at the lower frequency range of

interest movement of the antennas within a few cm of the body has a noticeable

effect. When placed directly on the skin the high loss of the average muscle dielectric

caused the currents on the antenna to short and the antenna barely radiated. Studying a

typical cross-section of a human body it is evident that 5-10mm consists of fatty

tissue, which dielectric constant has much lower real and imaginary dielectric

constants than the average muscle value (see Table 1). A second body model was

constructed which included this thin layer of fatty tissue around a core of average

muscle. An effect of this change is that when the antenna is placed on the body the

conducting elements are not in direct contact with the highly lossy muscle material.

Although there is a degradation in the antenna performance when placed on the body,

it is not as dramatic as when the body is modelled as a single ‘average muscle’

homogenous dielectric. It was found that the model with the thin fatty tissue layer

produces excellent results compared to the measured results of the antenna on a real

human. A further body model was created which included the average abdominal

dielectric constant within the body cavity but this made little difference to the

results. This was anticipated as the average abdominal values have similar loss to the

average muscle. To demonstrate the difference between the first two models, the

bowtie antenna was modelled placed on a slab of dielectric. If the slab was

homogenous with the average muscle dielectric value the boresight gain was -27dBi

at 300MHz. If a 10mm layer of fatty tissue is included the boresight gain is -8dBi.

The results of the modelling of the difference antenna types are shown in

Section IV, along with measured data.

Fig 5.1: COMPARISON OF TISSUE PERMITTIVITY AT 500MHZ



5.2 Materials and manufacturing

The manufacturing techniques for the wire dipole antenna were different than for the

bowtie and spiral. The bowtie and spiral antennas required similar manufacturing

techniques. The aim was to integrate the antennas into conventional clothing and a

‘polo shirt’ style t-shirt and ‘combat’ style trousers were used. The broadband wire

dipole was integrated into the trousers and t-shirt and the spiral and bowtie were

integrated into the back of the t-shirt. A wide range of materials were considered and

a summary is provided below.

1) Conducting Ribbon

This is a commercially available product consisting of typically 3-6 tracks of

conductive thread woven into a nonconductive backing ribbon fabric. The material

comes in a reel so long lengths can be used. The material is 15mm wide and

extremely flexible. It was not suitable for the bowtie as it comes in thin strips. It was

also not suitable for the spiral as it kinks when wound around corners. It appeared

ideal for the broadband dipole. A disadvantage is that the material cannot be soldered

to so crimp type connectors were used for the antenna terminals and to connect the

lumped loads.

2) Insulated Wire

Standard flexible insulated wire is an ideal candidate for the broadband wire dipole.

This is a proprietary high voltage, multi-strand, single conductor wire with a thick

silicone rubber insulating cladding. It is flexible and can be soldered to directly. The

wire is easily available and can be integrated into the seam of conventional clothing.

3) Conducting Paint

The conductive paint contains a high silver content to produces a low electrical

resistance. The conducting paint antenna was low-profile and unobtrusive. The paint

could not be soldered to so a conducting epoxy was used which maintained a strong

bond throughout the measurements, see Figure 3.



4) Conducting Nylon

Conducting nylon is available from a variety of manufacturers. The conducting nylon

was adhesive backed and then cut into the bowtie and spiral patterns. The pattern was

then adhered to the t-shirt. The adhesive formed a strong bond to the t-shirt. The

conducting nylon could not be soldered to so a conducting epoxy was used, see Figure

5) Phosphor Bronze Mesh

Phosphor bronze mesh is typically used for EMC shielding in clothing. This material

was again adhesive backed and cut into the spiral and bowtie shapes. The thermal

conductivity was good enough to solder directly to. It was noticed that once folded the

mesh would retain a bend which made the antenna more obtrusive when worn as the

kinks were clearly visible through the fabric.

6) Conducting Thread

Conducting thread is suitable for the spiral antenna but not for the wire dipole or

bowtie. A local firm with a computer controlled embroidery machine was used to

create a spiral antenna. A satin stitch was used to try and ensure a good conducting

path along the arms of the antenna. The balun was connected using conducting epoxy,

see Figure 2.

7) Screen Print

An obvious method to write a pattern onto a t-shirt is to use a traditional screen print

technique. Conducting ink suitable for screen printing was procured and trialled. The

printed ink could not be soldered to directly so a conducting epoxy was used to

connect the balun.

8) LCP

Liquid Crystal Polymer (LCP) is an alternative substrate material to polyimide for the

manufacture of flexible PCBs. Typically the substrate and copper foil are covered

with an outer Kapton layer to provide electrical insulation and some moisture

resistance. To enable easy design changes without the need for PCB layout and



processing, initial experiments were conducted on LCP with ½ ounce copper but not

coated with Kapton. Components can be directly soldered to the LCP material.

9) Copper coated fabric

There are a variety of commercial processes available to coat fabric with copper. The

antenna shape is defined by masking the fabric and the final product provides a

flexible, low resistivity surface which forms an integral part of the fabric without any

additional bonding operations. Electrical connection to the copper may be achieved by

soldering. The fabric substrate was attached to the garment using iron-on adhesive,

see Figure 4. The adhesive was clearly visible and made the antenna rather obtrusive

so different adhesive methods should be used.

Fig 5.2.1: Conducting nylon spiral Fig5.2.2: Embroidered conducting thread spiral

Fig5.2.3: Conducting paint Fig 5.2.4: Copper coated fabric bowtie



5.3 Antenna Feed

All the antennas considered required a balanced feed. The feed point of the dipole was

around the waist-band of the wearer so the balun is unobtrusive. The spiral and bowtie

antenna required a balun at the center of the element which is at the center of the torso

and therefore both the balun and the connecting coaxial cable had to be low profile.

Thin coaxial cable was procured which had suitable performance across the required

frequency band and has a diameter of 1.8mm. As well as being unobtrusive, the balun

was required to be wideband (since the antennas were wideband) and handle a typical

portable transceiver power level of a few watts. A suitable balun was procured which

can handle 3W, has a bandwidth of 0.5-1000MHz and had a height of 4mm.

The antennas were modelled placed close and in direct

contact with the body. When close to the body it was clear that capturing the body

composition was important as it has a significant effect on the body. Including a thin

layer of fatty tissue modified the results, particularly when the antenna was placed on

the body. There is excellent agreement between modelled and measured patterns. A

variety of manufacturing techniques were investigated to ascertain methods to

integrate antennas into clothing. The phosphor bronze mesh, LCP and copper coated

fabric have the advantage that the antennas can be directly soldered to. The nylon,

painted and screen printed antennas used conducting epoxy. For the nylon and painted

antennas the bond held securely during measurements but on the screen print material

the bond disconnected. The use of epoxy could be of concern if a robust antenna

solution is required.

5.4 Antenna Selection

A comprehensive literature review was conducted to identify candidate antennas. For

reasons of brevity, only a summary of relevant work is described in this section. One

of the earliest relevant papers found was published in 1968 and was concerned with

an experimental study of the effect of human proximity to portable communications

devices operating from 30MIHz - 150Nffz [I]. It was noted that an antenna tuned to

work in proximity to the human body had increased range when moved away from the



body, despite the introduction of a mismatch at the feed. The reason given is that the

human body is strongly absorbing at these frequencies and significantly reduces the

radiation efficiency. The earliest body-worn antenna is a multi-turn loop developed by

King [2]. This is a shoulder mounted antenna which can be tuned to operate over

150-170M4Ez, with a 3:1 VSWR bandwidth of 1.4MH~z. The antenna was designed

to have a low visual signature and operate under light clothing but is not low profile

enough for this work. A recent alternative to the loop antenna is a Planar Inverted F

antenna (PIFA) mounted on the shoulder [3]. PIFA antennas are commonly use in

mobile telephones (operating over GSM frequencies) and papers optimising, the

antenna for mobile handsets are too numerous to mention. The PIFA described in

reference [3], however, is designed to operate at 350MI~z. A stated advantage of the

PIFA is that it can receive both horizontal and vertical polarisation (the results show

HP has around 10-12dB lower gain to VP). Since the PIFA works above a ground-

plane (the size of which is made small in [3]) the effect of proximity to the human

body remains small. Although this antenna is small, it is not low profile enough for

this work. Candidate solutions for wearable FM (50-140Mffz) antennas have been

proposed. Prototype antennas have been designed, fabricated and measured [4]. The

designs are variations of asymmetric dipoles, including wide dipole and meandered

dipoles. As the designs are narrow band and have no ground-plane between the

antenna and the dipoles, variation of body position has a significant impact on VSWR

and patterns. An alternative method to create unobtrusive antennas is to camouflage

them within conventional clothing construction. For example, a standard metal button

from a pair of jeans has been modified into a dual-band WLAN antenna. This is

unlikely to be possible at the low frequencies of interest in this work. A number of

papers concentrate on the design of antennas incorporating textiles/fabrics but they

generally operate around WLAN frequencies (2.4GHz). A textile Electronic Band gap

(EBG) structure has been inserted in the middle of a fabric patch antenna. A solid

copper tape ground-plane is attached to a layer of felt. The felt is covered with square

patches of copper tape which is covered with a second layer of felt. A copper tape

patch is then placed on the top felt layer. The EBG antenna has 30% increase in

bandwidth with 50% decrease in size compared to a conventional patch, but is still

relatively narrowband. The use of the human body as a counterpoise for an antenna

has been reported. An RF source is placed one side of a conducting surface (which



represents a portable handset) with the other side of the surface touching a human

hand. The other side of the source is attached to a 'ground-plane'. The length of the

'ground-plane' is varied to produce a resonance as it is driven against the hand.

Although there is a large number of published articles concentrating on body wearable

antennas, few were of direct relevance to this work, particularly due to the low

frequency and wide bandwidth requirements.

At the frequencies of interest, a wavelength become comparable to the dimensions of

a typical human body. The body, or section of body used to mount the antenna,

determines the maximum size of the antenna. Assuming the lowest frequency of

operation of an antenna is when its length is around half a wavelength, the body

sections which can be used at different frequencies are shown in Table below.

Fig 5.4.1 Body section against frequency of body mounted antenna

The table shows that to achieve the lowest frequency of interest, it is necessary to

have an antenna the length of the torso and leg. Such an antenna would have to be

physically narrow to fit unobtrusively along the leg. The obvious candidate is a

dipole, but this is narrowband. It is possible, however, to load the dipole with passive

components to increase the bandwidth of the antenna. This antenna type would not

have the bandwidth to cover the entire band of interest and it is unlikely passive

components can be procured which operate correctly at higher frequencies. Placing an

antenna around the torso or leg may provide an antenna which operates at the lower

frequencies. It is unlikely that wrapping the antenna around a material as lossy as the

body will provide good performance. Simple elements like dipoles/bowties would

also have the wrong polarisation. Mounting an antenna on the chest or back is an

attractive option. The frequencies in Table 1 assume free-space but of course

proximity to the body will have an effect and could allow the antennas to operate at a

lower frequency than specified. Two wideband elements were chosen to be placed on



the chest, the bowtie and spiral. Although the spiral is circularly polarised, not vertical

as desired, this may provide benefits for certain body positions.

6. CONCLUSION

Military systems are now exploiting a very broad range of wide band antenna, feed

and beamformer technologies. Multi-octave phased array antennas have stringent

performance requirements, and are manufactured in relatively low numbers. Body

Worn Antennas (BWAs) are generally low gain systems where radiation pattern

performance is less critical, due to the uncertainties in the propagation environment.

Body Worn Antennas (BWAs) are low cost and have the potential for manufacture in

large quantities (thousands). Advances in EM simulation underpin exploitation of

wideband phased arrays (calculation of mutual coupling, tolerance and materials

effects) and BWAs (prediction of interactions with the human body).



7. REFERENCES

1. Rob A. Lewis, “Wideband Antenna Technology in Military Applications”,

International Workshop on Antenna Technology (iWAT), 2010.[ Base paper]

2. D. H. Schaubert, S. Kasturi, A. O. Boryssenko and W. M. Elsallal, “Vivaldi

Antenna Arrays for Wide Bandwidth and Electronic Scanning”, European

Conf. on Antennas and Propagation (EuCAP), Edinburgh, UK, Nov 2007.

3. J. C. G. Matthews and G. Pettitt, “Development of Flexible, Wearable

Antennas”, European Conf. on Antennas and Propagation (EuCAP), Berlin,

April 2009.

4. J. Langley, P. S. Hall and P. Newham, “Balanced Antipodal Vivaldi Antenna

for Wide Bandwidth Phased Arrays”, IEE Proc. Microwaves, Antennas and

Propagation, Vol 143, No. 2, April 1996.

5. http://www.antenna-theory.com/arrays/main.php.

6. http://www.q-par.com/products/patch-antennas-and-arrays/vivaldi-antenna.

7. http://en.wikipedia.org/wiki/Wideband.

8. http://en.wikipedia.org/wiki/Phased_array.

9. http://en.wikipedia.org/wiki/Beamforming.

10. http://www.freepatentsonline.com/7463210.html.


6report

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