wide band linearly tapered slot

7/30/2019 Wide Band Linearly Tapered Slot

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THE UNIVERSITY OF QUEENSLAND

School of Information Technology and Electrical Engineering

Submitted for the degree of Bachelor of Engineering (Honours)

in the division of Electrical Engineering

May 2003

Wide Band Linearly Tapered SlotAntenna

ByJ ustin J oseph Paul


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Justin Joseph Paul

8/260 Sir Fred Schonell Drive

St. Lucia, QLD 4067

Australia

Tel. (07) 32177851

23rd May 2003

The Dean

School of Information Technology and Electrical Engineering

University of Queensland

St. Lucia, QLD 4067

Dear Sir,

In accordance with the requirements of the degree of Bachelor in Engineering (Honours) in

the division of Electrical Engineering at the University of Queensland, I present the

following thesis entitled A Wide Band Linearly Tapered Slot Antenna. This work was

performed under the supervision of Associate Professor Nicholas Shuley.

I declare that all the work submitted in this thesis is my own, except as acknowledged in

the text, and have not been submitted for a degree at the University of Queensland or any

other institution.

Yours sincerely,

Justin Joseph Paul


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Abstract

i

Abstract

A Wide Band Linearly Tapered Slot Antenna (LTSA) acting as a receiving antenna used

for identification of objects through measurement of pulses was experimentally designedand constructed. Since a simulated design along with an identical prototype was completed,

results obtained for the S11 return loss, bandwidth and radiation patterns of both the

simulated and measured results were evaluated.

The simulation of the Wide Band LTSA was done on a program called FEKO. A return

loss of 48dB and a usable bandwidth of 66% was obtained for the simulated design

whereas the prototype obtained a return loss of 39dB and a usable bandwidth of only 15%.

There seem to be an alarming difference in the usable bandwidth and thus, we can conclude

that a wide bandwidth was apparent only for the simulated design and not the prototype.

The radiation patterns, however, produce almost similar results for both the simulated and

measured designs. This proved that the signal is being transmitted in the same directions in

both cases.

At the end of the thesis, a simulated design of a Wide Band LTSA was successfully

constructed into a prototype and tested. The overall results obtained agreed, to a certain

extent, with the research and theoretical background found in several journals on this thesis

topic.


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Acknowledgements

ii

Acknowledgements

I wish to express sincere thanks to my supervisor, Associate Professor Nicholas Shuley, for

his patience and guidance throughout the course of this project. This thesis would not havebeen completed without his invaluable time and advice given during the design process of

this thesis.

I would like to thank my parents, who gave me unlimited love and support throughout these

years, even though they are all the way back home in Singapore. This would not have been

possible without them.

Finally, many thanks go out to all those who have helped me throughout the course of the

project. I would like to thank Keith and Denis from the UQ Electronics Laboratory, for

rendering their professional skills to the completion of the prototype. And also to Russell

Clark from the Microwave Laboratory, who provide assistance during the setting up and

testing of the antenna in the Anechoic Chamber.


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Contents

iii

Contents

Abstract i

Acknowledgments ii

L ist of Figures vi

List of Tables ix

1 Introduction 1

1.1 Aims and Objectives of the Thesis 1

1.2 Overview of Thesis 2

1.3 Simulation Program: FEKO 3

2 The Tapered Slot Antenna 5

2.1 Characteristics of a Tapered Slot Antenna 5

2.1.1 Radiation Characteristics 6

2.1.2 Bandwidth Characteristics 6

2.2 Design Considerations 6

2.3 Taper Profiles 7

1.3.1 Effect of Curvature on Taper Profile 9

2.4 Feeding Techniques 10

2.4.1 Coaxial Line Feed 11

2.4.2 Microstrip Line Feed 13

2.5 Summary 15

3 Microstrip Transmission Line 16

3.1 Microstrip Principles 16


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Contents

iv

3.2 Substrate Materials 18

3.3 Microstrip Design Formulas 19

3.3.1 Effective Dielectric Constant 20

3.3.2 Wavelength 21

3.3.3 Characteristic Impedance 22

3.4 Quarter-wave Microstrip Transformer 22

3.4.1 Design on FEKO 23

3.4.2 Simulated Results Using FEKO 25

3.5 Discussion of Results 26

3.6 Summary 27

4 Microstrip to Slot Transition 28

4.1 Microstrip to Slot Transition 28

4.1.1 Microstrip to Slot Transition 28

Using a Double Y Balun

4.2 A Back-to-Back Microstrip to Slot Transition 30

4.2.1 Design on FEKO 30

4.2.2 Simulated Results 33

4.2.3 Sketch of Prototype 34

4.2.4 Measured Results 35


4.4 Summary 38

5 Design and Simulated Results of the Wide Band LTSA 39

5.1 Features of LTSA 39

5.2 Design Considerations 39

5.3 Design on FEKO 40

5.4 Simulated Results 41

5.4.1 S11 Return Loss and Bandwidth 42

5.4.2 Radiation Patterns 42


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Contents

v


5.6 Summary 47

6 Prototype and Measured Results of the Wide Band LTSA 48

6.1 Sketch of Prototype 48

6.2 Measured Results 48

6.2.1 S11 Return Loss and Bandwidth 50

6.2.2 Radiation Patterns 50


6.4 Summary 54

7 Evaluation 55

7.1 Evaluation of Simulated and Measured 55

S11 Return Loss and Bandwidth

7.2 Evaluation of Simulated and Measured 56

Radiation Patterns

7.3 Summary 57

8 Conclusion 58

8.1 Future Work 59

Appendix A FEKO Programs 61

Appendix B Sketches 69

References 71


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List of Figures

vi

List of Figures

Chapter 2:

Figure 2.1: Top view of Taper Slot Antenna 7

Figure 2.2: Cross-sectional View of Slotline 7

Figure 2.3: Different Taper Profiles of a TSA: (a) exponential, 8

(b) tangential, (c) parabolic, (d) linear, (e) linear-constant,

(f) exponential-constant, (g) step-constant, (h) broken linear

Figure 2.4: Schematic of TSA Taper Profiles 10

Figure 2.5: Different Feeding Techniques of a TSA: (a) coaxial line, 11

(b) microstrip line, (c) CPW, (d) air-bridge/GCPW,

(e) FCPW/centre-strip, (f) FCPW/notch

Figure 2.6: Model of Coaxial Line to Slot Transition 12

Figure 2.7: Equivalent Circuit of Coaxial Line to Slot Transition 12

Figure 2.8: Model of Microstrip to Slotline Transition 14

Figure 2.9: Equivalent circuit of a Microstrip to Slotline Transition 15


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List of Figures

vii

Chapter 3:

Figure 3.1: Structure of Microstrip Transmission Line 17

Figure 3.2: Wide and Narrow Microstrip Line 20

Figure 3.3: Equivalent Circuit of a Quarter Wave Microstrip Transformer 23

Figure 3.4: Top View of Design in WinFEKO 24

Figure 3.5: Return Loss obtained from GraphFEKO 25

Figure 3.6: Voltage Standing Wave Ratio obtained from GraphFEKO 26

Chapter 4:

Figure 4.1: Microstrip to Slot Transition Using a Double Y-balun 29

Figure 4.2: Equivalent Circuit of Back-to-Back Microstrip to Slot Transition 31

Figure 4.3: Top View of Back-to-Back Microstrip to Slot 32

Transition in WinFEKO

Figure 4.4: Simulated Return Loss, S11, of a Back-to-Back 33

Microstrip to Slot Transition

Figure 4.5: Top View of Finished Prototype 35

Figure 4.6: Bottom View of Finished Prototype 36


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List of Figures

viii

Figure 4.7: Measured Return Loss, S11, of a Back-to-Back 37

Microstrip to Slot Transition

Chapter 5:

Figure 5.1: Top View of Wide Band LTSA in WinFEKO 40

Figure 5.2: Simulated Return Loss, S11, of the Wide Band LTSA 42

Figure 5.3: Radiation Patterns for E-plane 44

Figure 5.4: Radiation Patterns for H-plane 45

Chapter 6:

Figure 6.1: Top View of Finished Prototype 49

Figure 6.2: Bottom View of Finished Prototype 49

Figure 6.3: Measured Return Loss, S11, of the Wide Band LTSA 50

Figure 6.4: General Orientation of Wide Band LTSA for E-plane and H-plane 51

Figure 6.5: Radiation Patterns for E-plane 52

Figure 6.6: Radiation Patterns for H-plane 52


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List of Tables

ix

List of Tables

Table 3.1: Properties of Microwave Dielectric Substrates 19

Table 4.1: Parameters of Back-to-Back Microstrip to Slot Transition 31

Table 5.1: Parameters for the Wide Band LTSA 41

Table 7.1: Compiled Simulated and Measured Results 55


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Chapter 1: Introduction

1


This chapter highlights the importance of the Wide Band Linearly Tapered Slot Antenna

(LTSA). After defining the aims and objectives of the thesis, this chapter closes with an

overview of the thesis. The overview briefly summarizes the topics discussed in every of

the following chapters in this thesis.

1.1 Aims and Objectives of the Thesis

The aim of this thesis is to successfully design and construct a wide band LTSA that is able

to identify objects through measurement of pulse responses. Aiming to complete the design

and construction of this project on schedule is given the highest priority. Also, obtaining

similar simulated and measured results is another objective to fulfil. Throughout the course

of the thesis, gaining a comprehensive understanding of travelling wave antennas, matching

capabilities as well as microstrip and slotline characteristics will definitely provide useful

knowledge for the future.

The antenna and all the designs carried out in this thesis were carried out using ROGERSdielectric substrate with relative permittivity of 2.0 and thickness of 0.5mm. All the

simulated and measured results were done over a frequency range of 1 GHz to 8 GHz.

FEKO is a new program introduced due to the need to design and simulate the wide band

LTSA. Being able to gain knowledge on FEKO will prove to be a useful tool for the

present and future. This, together with learning and understanding how basic antennas

generally work, are also aims of this project.


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1.2 Overview of Thesis

Chapter 1begins with some aims and objectives set at the commencement of the thesis. An

overview of the thesis will also be given at the end of the chapter.

Chapter 2begins with an introduction of a basicTapered Slot Antenna. Characteristics and

design considerations will give a better insight into the operations of the antenna. Most

importantly, the effects of the taper profile and two feeding techniques will be discussed.

Chapter 3 will provide an in depth look into the principles of a Microstrip Transmission

Line. Different substrate materials that can be used for antennas are mentioned along with a

table highlighting their properties. Microstrip design formulas are also included in this

chapter to provide useful equations for designing. Finally, all the above are implemented in

a Quarter Wave Microstrip Transformer design that is done using FEKO.

Chapter 4 begins with an introduction of the most important factor leading to the

completion of the wide band linearly tapered slot antenna, the Microstrip to Slot Transition.

An extremely useful implementation of this transition is through a Double Y-Balun. The

first prototype designed and built during this thesis was a Back-to-Back Microstrip to Slot

Transition. Steps that were taken while approaching this design as well as simulated results

obtained are also covered in this chapter. These simulated results are then compared with

measured results obtained from the laboratory.

Chapter 5will cover theDesign and Simulated Results of theWide Band Linearly Tapered

Slot Antenna. A detailed look into the features and the design considerations of the antenna

is available in this chapter. Simulated results obtained for the antenna design will be

discussed at the end of the chapter.


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Chapter 6 will focus mainly on the Prototype and Measured Results of the Wide Band

Linearly Tapered Slot Antenna. In this chapter, information through sketches of the

prototype and measured results will be provided. The chapter will conclude with

discussions done on the measured results.

Chapter 7 will contain the Evaluation of the results obtained for both the simulated and

measured results of the wide band linearly tapered slot antenna. Comparisons will be made

between both the results and explanations of similarities and differences will be done as

well.

Chapter 8 will give an overall Conclusion by summarising all the work done during the

course of it and future prospects for the wide band linearly tapered slot antenna.

1.3 Simulation Program: FEKO

The program FEKO is based on the Method of Moments (MOM). Electromagnetic fields

are obtained by first calculating the electric surface currents on conducting surfaces and

equivalent magnetic and electric surface currents on the surface of a dielectric solid. The

currents are calculated using a linear combination of basis functions, where the coefficients

are obtained by solving a system of linear equations. Once the current distribution is

known, further parameters such as near and far field, radar cross sections, directivity of

input impedance can be found. Only time domain harmonic sources are supported in the

current version and calculation is done in the frequency domain. FEKO uses ejt time

conversion.


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WinFEKO is the main user interface module and is used to control the solution of a

problem. The geometry is defined in terms of high level commands in the *pre input file

which also sets the solution parameters. The customised text editor, EditFEKO, assists the

user in creating and editing the input file. The processor/mesher, PREFEKO, processes this

file and prepares the input file, *fek, for the program FEKO which is actually the field

calculation code. The PREFEKO enables the user to create complex geometries with a

single command. The output file, *out, of FEKO contains all the solution information. The

resulting fields and/or currents can be displayed in 3D in WinFEKO or as 2D plots in

GraphFEKO.

The above description of the simulation program FEKO is a summarised version obtained

from the User Manual of FEKO. For more detailed information on FEKO, this manual is

recommended. To put the above in laymans terms, EditFEKO is used to write the code for

the simulation program. Upon running PREFEKO, we are then able to obtain a

visualisation of the outlook of the design in WinFEKO. Finally, all the 2D plots for the

simulation can be obtained through GraphFEKO.


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Chapter 2: The Tapered Slot Antenna

5


The chapter begins with a review on some characteristics of the Tapered Slot Antenna. This

is followed by design aspects to be taken into consideration in the process of constructing

the antenna. In general, designs differ only in the taper profile of the slot and the feeding

technique. Some of the common taper profiles and feeding techniques will be presented in

this chapter.

2.1 Characteristics of a Tapered Slot Antenna

The tapered slot antenna (TSA) belong to the general class of end-fire travelling wave

antennas and consist of a tapered slot etched onto a thin film of metal. This is done either

with or without a dielectric substrate on one side of the film. Besides being efficient and

lightweight, the more attractive features of TSAs are that they can work over a large

frequency bandwidth and produce a symmetrical end-fire beam with appreciable gain and

low side lobes [1]. An important step in the design of the antenna is to find suitable feeding

techniques for a slotline excited TSA.

Understanding the characteristics of the TSA is fundamental and would help a great deal in

designing the antenna. From research journals on the TSA, we can confirm that TSAs

generally have wider bandwidth, higher directivity and are able to produce symmetrical

radiation patterns [2].


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2.1.1 Radiation Characteristics

As the TSA is a travelling wave antenna, the phase velocity and the guide wavelength, g,

varies with the change in thickness, dielectric constant and taper shape. Having the gain

proportional to L/g, parameters such as length, width and taper profiles also have direct

impact on the radiation patterns, directivity and cross-polarisation level of the antenna. The

radiation characteristics of the antenna are also affected by the substrate thickness and

ground plane. [4]

2.1.2 Bandwidth Characteristics

The TSA is capable of having an operating bandwidth within a frequency range of 2 GHz

to 90 GHz. To achieve a wider bandwidth, it is ideal for the TSA to have a perfect

impedance match at both the feed transition and the slot termination. Different methods for

bandwidth broadening depend on the feed methods chosen. This will be described further

in section 2.4. The bandwidth is normally proportional to the change in frequency,f. [4]

2.2 Design Considerations

A TSA is formed by slowly increasing the width of a slot from the point of its feed to an

open end of width generally greater than O/2 [3]. This is illustrated in figure 2.1.

Experimental results done in various journals have confirmed that the impedance,

bandwidth and radiation patterns are greatly affected by parameters such as length, width

and taper profile of a TSA. The dielectric substrates thickness and relative permittivity are

also important as they contribute to the efficiency of the antenna. Figure 2.1 and 2.2 show

the top view and cross-sectional view of a slotline on a dielectric substrate with its

important parameters illustrated. The shaded area in both the figures represents the

remaining copper on the dielectric substrate after etching is done.


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Figure 2.1: Top view of Taper Slot Antenna

Figure 2.2: Cross-sectional View of Slotline

2.3 Taper Profiles

Many taper profiles exist for a normal TSA. Figure 2.3 shows different planar designs and

we can observe that each antenna differs from one another only in the taper profile of the

slot. Of all the designs illustrated in figure 2.3 [4], only the Vivaldi [5] and linearly tapered

profile [3] have been thoroughly studied over the past few years.


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Planar tapered slot antennas have two common features. The radiating slot acts as the

ground plane for the antenna and the antenna is fed by a balanced slotline. However,

drawbacks for a planar TSA come in the form of using a low dielectric constant substrate

and obtaining an impedance match for the slotline. By fabricating on a low dielectric

constant substrate, relatively high impedance is obtained for the slotline. If a microstrip

feed is chosen, it makes matching very difficult. Thus, the microstrip to slot transition will

limit the operating bandwidth of the TSA.

(a) (e)

(b) (f)

(c) (g)


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(d) (h)

Figure 2.3: Different Taper Profiles of a TSA: (a) exponential, (b) tangential, (c)

parabolic, (d) linear, (e) linear-constant, (f) exponential-constant, (g) step-constant,

(h) broken linear

2.3.1 Effect of Curvature on Taper Profile

Tapered slot antennas with linear, exponential or constant taper profile are commonly

reported and their journals can be easily found. However, information on the effects of the

curvature on a taper profile is not readily available. From the authors of [6], we are able to

obtain experimental investigation and results on the effects. The important points will be

briefly summarized in this sub-section. For more detailed explanation and illustrations, it is

recommended that the particular journal be referred to.

Figure 2.4 shows the schematic of linear (1) and exponential (2), (3) and (4) taper profiles

of a TSA. As seen in the figure, four TSAs of same length and terminating slot width, but

with different taper profiles, were fabricated and tested. Fabrication was done on the same

type of substrate with the same relative permittivity.


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Figure 2.4: Schematic of TSA Taper Profiles

The cross polarization is generally improved with the decrease in the radius of the curvature

except for the E-plane, which will not show any improvement. More importantly, thedecrease on the radius of the curvature also reduces the bandwidth of the antenna. This is

not ideal as the later part of the thesis focuses on designing an antenna with a wide

bandwidth.

2.4 Feeding Techniques

A slot generally always excites a TSA. In order to test and design slotline circuits, it is

necessary to have a transition between a slot and another transmission medium [7]. These

transitions should be very compact and have low loss. Some feeding techniques and their

transitions are shown in the figure 2.5 [4]. The commonly used methods are the coaxial line

feed and the microstrip line feed. These will be illustrated and discussed in the next two

sub-sections.


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(a) (b)

(c) (d)

(e) (f)

Figure 2.5: Different Feeding Techniques of a TSA: (a) coaxial line, (b) microstrip

line, (c) CPW, (d) air-bridge/GCPW, (e) FCPW/centre-strip, (f) FCPW/notch

2.4.1 Coaxial Line Feed

A coaxial line feed provides a direct path for coupling of fields across the slot [4]. A

commonly used coaxial line to slot transition is shown in figure 2.6. The transition consists

of a coaxial line placed perpendicular at the end of an open circuited slot. The outer

conductor of the cable is electrically connected to the ground plane on one side of the slot

while the inner conductor of the coaxial line forms a semicircular shape over the slot as

shown in figure 2.6. This transition has been analysed in [8]. An equivalent circuit, also

from [8], is shown in figure 2.7.


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Figure 2.6: Model of Coaxial Line to Slot Transition

Figure 2.7: Equivalent Circuit of Coaxial Line to Slot Transition

From the equivalent circuit, we can predict that the slot impedance will be transform to a

lesser value, by a factor of n, so as to match a 50 coaxial cable. To do this, a slot

impedance of around 75 is needed. However, in practice, it is difficult to obtain a slot

impedance of around 75 because a slotline impedance of less than 100 will have a

very small width and this makes fabrication with etching difficult and inaccurate.


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2.4.2 Microstrip Line Feed

A microstrip to slot transition consists of a slot, etched on one side of the substrate,

crossing an open circuited microstrip line, located on the opposite side, at a right angle. The

slot extends to one quarter of a wavelength beyond the microstrip and the microstrip

extends one quarter of a wavelength beyond the slot as shown in figure 2.8. The latters

wavelength has to minus a length extension ofL. The length extension is due to fringing

at the end of the open circuited line, which makes the line appear electrically longer [4].

The length extension can be approximated using the following expression [9]:

( )

( )

+

++=

8.0258.0

264.03.0412.0

,

,

h

wh

wh

L

effr

effr

(2.1)

An equivalent circuit of the microstrip to slot transition is shown in figure 2.9 [4]. An

impedance match between the microstrip and slotline can be obtained at a given frequency

by applying equation (2.2). The equation can also be applied to the coaxial to slot

transition.

ms ZnZ2= (2.2)

where ( )

=

oo

utqutn

2sincot2cos (2.2.1)

)vu

out

q

1

tan2

+=

(2.2.2)


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,

2

=

s

oru

1

2

=

s

ov

(2.2.3)

where Zs = characteristic impedance of the slot, Zm = characteristic impedance of themicrostrip, o =free space wavelength ands =slotline wavelength

To achieve proper impedance match, multi-step quarter wave transformers are sometimes

used. By terminating the microstrip with a radial stub and the slot with an elliptical shaped

cavity, the bandwidth can be broadened. Also, when terminating the slot with the elliptical

cavity, the operating bandwidth of the transition tends to shift down in frequency [4].

Figure 2.8: Model of Microstrip to Slotline Transition


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Figure 2.9: Equivalent circuit of a Microstrip to Slotline Transition

2.5 Summary

From this chapter, a better understanding on the characteristics and design considerations

inevitably helps in the designing and constructing of a tapered slot antenna. Various taper

profiles and feeding techniques were described and illustrated to give the reader different

options while designing a TSA. The effects the angle of the taper profile has on the antenna

were also highlighted. Finally, the transitions of the two more common feeding techniques,

coaxial line and microstrip line, were explained.

The overall design of the wide band LTSA was closely modelled after some of the figures

presented in this chapter. Figures 2.1 and 2.2 were taken into consideration when designing

the linearly taper profile of the slot. Figure 2.8 illustrates the most important considerations

that the microstrip to slot transition was designed after. However, a good transition at one

particular frequency does not work well over all the frequencies as the wavelength changes

with the frequency.


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Chapter 3: Microstrip Transmission Line

16


This chapter will provide an overview on the basic principles and operation of a microstrip

transmission line. By providing definitions on important parameters and a detailed

description on its operation, one is able to gain a better knowledge and understanding of the

design specifications of a microstrip transmission line. This is essential in the latter

chapters as the microstrip transmission line plays an important part in this thesis.

3.1 Microstrip Principles

The microstrip transmission line is the most commonly used Microwave Integrated Circuit

(MIC) transmission medium and is also one of the most popular type of planar transmission

line. A planar configuration implies that the dimensions in a single plane can determine the

characteristics of the element. For example, the width, w, of a microstrip line on a dielectric

substrate can be adjusted to control its impedance.

The microstrip transmission line is popular due to the fact that the mode of propagation on

microstrip is almost TEM. This allows easy approximate analysis and yields wide bandcircuits [7].

The structure of a microstrip transmission line is shown in the figure 3.1. The most

important dimension parameters of a microstrip circuit design are the width, w, of the

microstrip line and the height, h, which is equivalent to the thickness of the dielectric

substrate [10]. The relative permittivity, r, of the substrate is also another important

parameter. The fabrication of a microstrip transmission line is often done through etching

on a microwave substrate material.


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Figure 3.1: Structure of Microstrip Transmission Line

There are generally two types of dielectric substrates, soft and hard. Soft substrates are

normally used as they are flexible, cheap and can be easily fabricated. Hard substrates have

better reliability and lower thermal expansion coefficients. However, they are more

expensive and not flexible. Substrates materials will be mentioned in the next section.

From the above, we can conclude that the microstrip line has many advantages, such as low

cost, small in size and use of photolithographic method for fabrication that leads to good

repeatability, reproducibility and ease of mass production. However, the microstrip line

does have its disadvantages that include higher loss, lower power-handling capability and

greater temperature instability. The thickness of the strip, t, and the conductivity, , are not

important parameters and are often neglected.


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3.2 Substrate Materials

The choice of dielectric substrate plays an important role in the design and simulation of

the microstrip transmission line as well as any other antennas. Some important dimensions

of the dielectric substrate are:

The dielectric constant.

The dielectric loss tangent that sets the dielectric loss.

The thermal expansion and conductivity.

The cost.

The manufacturability.

The thickness of the copper surface.

There are numerous types of substrates that can be used for the design of antennas. They

often have different characteristics and their dielectric constants normally range from 2.2

r 12. Thick substrates with low relative dielectric constants are often used as they

provide better efficiency and a wider bandwidth. However, using thin substrates with high

dielectric constant would result in smaller antenna size. But this also results negatively on

the efficiency and bandwidth. Therefore, there must be a design trade-off between antenna

size and good antenna performance [11].

Material Relative

Dielectric

Constant

Loss Tangent at

10 GHz (tan )

Thermal

Conductivity, K

(W/cm/C)

Dielectric

Strength

(kV/cm)

Sapphire 11.7 10-4 0.4 4 x 103

Alumina 9.7 2 x 10-4 0.3 4 x 103

Quartz (fused) 3.8 10-4 0.01 10 x 103

Polystyrene 2.53 4.7 x 10-4 0.0015 280


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Beryllium Oxide

(BeO)

6.6 10-4 2.5 -

GaAs 12.3 16 x 10-4 0.3 350

Si 11.7 50 x 10-4 0.9 300

3M 250 type GX 2.5 19 x 10-4 0.0026 200

Keene DI-clad

527

2.5 19 x 10-4 0.0026 200

RT Duriod 5870 2.35 12 x 10-4 0.0026 200

3M Cu-clad 233 2.33 12 x 10-4 0.0026 200

Keene DI-clad

870

2.33 12 x 10-4 0.0026 200

RT Duriod 5880 2.20 9 x 10-4 0.0026 200

3M Cu-clad 217 2.17 9 x 10-4 0.0026 200

Keene DI-clad

880

2.20 9 x 10-4 0.0026 200

RT Duriod 6010 10.5 15 x 10-4 0.004 160

3M epsilon IU 10.2 15 x 10-4 0.004 160

Keene DI-clad

810

10.2 15 x 10-4 0.004 160

Air 1.0 0 0.00024 30

Table 3.1: Properties of Microwave Dielectric Substrates

3.3 Microstrip Design Formulas

To design a basic microstrip transmission line, one must be able to obtain dimensions such

as effective dielectric constant, wavelength and characteristic impedance. This can be

calculated through some simple equations that will be shown in the next few sub-sections.


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3.3.1 Effective Dielectric Constant

One might think that the effective dielectric constant, r,eff, is the same as the dielectric

constant, r, of the substrate. This appears to be true only for a homogeneous structure and

not for a non-homogeneous structure. For microstrip structures, we are able to calculate the

effective dielectric constant that comes in two different cases. These two cases are

illustrated in figure 3.2 whereby the top diagram shows a microstrip with width, w, greater

than the thickness, h, of the substrate (wh). The opposite can be said about the bottom

diagram [10].

Figure 3.2: Wide and Narrow Microstrip Line

By looking at the diagram with wh, we can conclude that the circuit performs similar to

having two parallel planes as most of the fields as kept under the wide microstrip width.

Thus, reff is approximately equivalent to r. When wh, half of the fields will be in air with

r =1, while the other half of the fields will be confined to the substrate with r,eff=(r +

1). Therefore, the range of a dielectric constant can be said to be:


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( ) reffrr + ,12

1(3.1)

The following equations can be used to obtain a precise value ofr,eff. Equations (3.2) and

(3.3) take into consideration negligible thickness of the microstrip.

1104.012

12

1

2

122

1

,

+

+

++=

h

wfor

h

w

hw

rreffr

(3.2)

112

12

1

2

12

1

,

+++=

h

wfor

hw

rreffr

(3.3)

3.3.2 Wavelength

For a propagating wave in free space, the wavelength of that medium is equal to the speed

of light divided by its operating frequency. To obtain the wavelength of a given wave-guide

or antenna, the free space wavelength is simply divided by the square root of the effective

dielectric constant of the wave-guide. These are shown in equations (3.4) and (3.5).

o

of

c= (3.4)

effr

og

,

= (3.5)


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22

where c =speed of light, fo =operating frequency, o =free space wavelength andg=the

guide wavelength.

3.3.3 Characteristic Impedance

The characteristic impedance, Zo, of any line is the function of its geometry and dielectric

constant. For a microstrip transmission line, the characteristic impedance is defined as the

ratio of voltage and current of a travelling wave. For a microstrip line with width, w, we are

able to calculate the characteristic impedance through the following two equations [12]:

125.08

ln60

,

+=

h

wfor

h

w

hw

Zeffr

o

(3.6)

( )1

444.1ln667.0393.1

120

, +++

=h

wfor

hw

hw

Zeffr

o

(3.7)

Note: Negligible microstrip thickness is taken into consideration

3.4 Quarter Wave Microstrip Transformer

A quarter wave microstrip transformer was modelled and designed in FEKO. Figure 3.3

shows an equivalent circuit that is simple and useful for matching a real load impedance to

a transmission line. A step in width will exist at the junction of two microstrip lines due to

both lines having different impedance. This is commonly encountered when designing

transitions. Designing a single section quarter wave microstrip transformer will prove to be

useful in the latter part of this thesis involving the transition from microstrip to slotline.


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23

Figure 3.3: Equivalent Circuit of a Quarter Wave Microstrip Transformer

One drawback of the quarter wave transformer is that it can only match a real load

impedance. However, by using an appropriate length of transmission line between the load

and the transformer, a complex load can always be changed into a real impedance.

3.4.1 Design on FEKO

A microstrip line with a characteristic impedance, Zo, of 50 is matched to a real load

impedance, ZL, of 100 by a single section quarter wave transformer. The characteristicimpedance of the quarter wave matching section can be obtained by equation (3.8).

LoZZZ =1 (3.8)

where Zo, Z1 and ZL represent the given characteristic impedances as seen in figure 3.3

Programs such as PUFF and PICAARD can easily determine the length, l, of the singlesection quarter wave transformer.

Z Z1 ZL

l


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24

From equation (3.8), we are able to calculate the characteristic impedance, Z1, of the

quarter wave matching section to be 70.71. By setting a design frequency, fo, we are also

able to obtain the electrical length of the quarter wave section, g/4, of the matching section

through the wavelength equations given previously. However, the electrical length of the

matching section will definitely differ at other frequencies and thus, a perfect match can no

longer be achieved.

Figure 3.4 shows the top view of the microstrip design as seen in WinFEKO. The pink area

surrounding the design represents the substrate with a dielectric constant of 2.0 and

thickness of 0.5mm. From the figure, we can clearly see a length of microstrip transmission

line, in orange and yellow, stepping down to a single section quarter wave transformer, in

green. This looks similar to the one seen in figure 3.3. The FEKO code for this design is

attached in the Appendix A.

Figure 3.4: Top View of Design in WinFEKO

/4

ZoZ1

Terminated

withZL


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25

3.4.2 Simulated Results Using FEKO

Figure 3.5 and 3.6 are results of the return loss, S11, and voltage standing wave ratio,

VSWR, of the quarter wave microstrip transformer respectively. Figure 3.5 shows the

transformer to have a very good return loss over a frequency range of 1 GHz to 8 GHz and

figure 3.6 shows a VSWR of approximately less than 2 for the same frequency range.

Figure 3.5: Return Loss obtained from GraphFEKO


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Figure 3.6: Voltage Standing Wave Ratio obtained from GraphFEKO

3.5 Discussion of Results

From the results obtained for the return loss and standing wave ratio of the quarter wave

microstrip transformer, we can conclude that the 50 transmission line matches well with

the terminated load impedance of 100 through the 70.71 single section quarter wave

transformer. Normally, antennas generally require a return loss of at least 10dB or a

VSWR of less than 2 for it to work effectively. Both the graphs confirm that these criteria

are met and that the design works excellently over a wide frequency range of 1 GHz to 8

GHz.


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3.6 Summary

Understanding the principles and operation of a quarter wave microstrip transformer helped

during the course of this thesis project. This is because the basic knowledge of how to

match impedance was learnt through design and simulation of this transformer. Matching

of impedances will play an important part in the latter part of the thesis because the wide

band linearly tapered slot antenna requires a microstrip to slot transition.

Many problems were faced during the first design stage of the thesis because learning the

use of FEKO proved to be difficult task. As the quarter wave microstrip transformer was

the first design done on FEKO, a better understanding of how to operate FEKO and the

various geometrical and control options was obtained.


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Chapter 4: Microstrip to Slot Transition

28


The microstrip to slot transition is the most important factor when it comes to the wide

band linearly tapered slot antenna. The point at which the microstrip crosses the slot has to

be almost exact for an antenna to produce a good performance. This chapter will further

describe the implementation of the transition through a double Y-balun and a back-to-back

transition.

4.1 Microstrip to Slot Transition

By feeding a slot with a microstrip line, we are creating a transition between the two. This

transition has been described and illustrated in chapter 2 under section 2.4. An improved

microstrip to slot transition has been proposed using a double Y-balun. Theoretically, this

would prove to be extremely useful for the microstrip to slot transition. However, due to

time constrains of this thesis, the double Y balun was not implemented during the design of

the wide band linearly tapered slot antenna. Hence, only a brief description of the balun

will be given in the next sub-section to give the reader an outlook as to how the transition is

done using a double Y balun.

4.1.1 Microstrip to Slot Transition Using a Double Y Balun

The double Y-balun is an extremely effective method while doing a microstrip to slot

transition. Double Y-baluns are based on the 6 port double Y junction, which consists of 3

balanced and 3 unbalanced lines placed alternately around the centre of the structure.

Figure 4.1 illustrates how a double Y-balun looks like when used in a microstrip to slot

transition [13].


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.

Figure 4.1: Microstrip to Slot Transition Using a Double Y-balun

In order to make a structure work as a balun with perfect transmission between an opposite

balanced and unbalanced ports, opposite pairs of lines should have reflection coefficients

with opposite phases. This means that one pair of lines should be short circuit and the other,

open circuit. The electrical lengths of the lines from the open or short circuits to the centre

of the junction should be equal:

ssmm ll = (4.1)

Wheremands are the phase constants and lmand ls are the line lengths for microstrip and

slot, respectively. In experimental realization, the values of length, l, are made small. To

avoid radiation effects, the following should be observed:

42

l (4.2)


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4.2 Back-to-Back Microstrip to Slot Transition

By designing a back-to-back microstrip to slot transition, we are basically creating a two-

port network. A microstrip line, following the same guidelines stated in the microstrip to

slot transition, feeds the slot antenna that is terminated by an elliptical shaped cavity. This

part will be used for the wide band linearly tapered slot antenna as well. A back-to-back

microstrip to slot transition is created by simply duplicating this design symmetrically at

the end of the slot.

4.2.1 Design on FEKO

The back-to-back microstrip to slot transition was again designed using FEKO. Figure 4.2

shows a equivalent circuit of the transition and figure 4.3 shows the top view of it in

WinFEKO. The design incorporates a microstrip transmission line with a single section

quarter wave transformer that was described in Chapter 3. The design was created using the

measurements of a ROGERS dielectric substrate with a dielectric constant of 2.0 and

thickness of 0.5mm. This particular substrate was use because of its mass availability in the

laboratory. Other substrates of different dielectric constants were available but not in

quantity. This might prove to be a problem in repeated fabrication as the design might have

to be redesigned when the substrate runs out.

The FEKO code for this design can be found in the Appendix A. Included in this code are

the parameters for the back-to-back microstrip to slot transition. They are as follows:


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Length of Substrate 132mm

Width of Substrate 61mm

Width of Slot, Ws 3mm

Length of Slot, Ls 50mm

Width of Microstrip Transmission Line, Wm 2.4mm

Length of Microstrip Transmission Line, Lm 20mm

Width of Quarter Wave Section, Wq 1.2mm

Length of Quarter Wave Section, L q 21.5mm

Radius of Elliptical Cavity 20mm

Table 4.1: Parameters of Back-to-Back Microstrip to Slot Transition

The dimensions given in the previous page were a combination of analysis of designs used

in journals as well as through trial and error. Through the dimensions used, we were able to

obtain the best available results.

Figure 4.2: Equivalent Circuit of Back-to-Back Microstrip to Slot Transition

Slotline

O/C

Line of Symmetry

50 Load

Excitation

MicrostripMicrostrip

Slotline

O/C


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Figure 4.3: Top View of Back-to-Back Microstrip to Slot Transition in WinFEKO

Similar to the microstrip design in Chapter 3, the pink surface represents the dielectric

substrate. On the surface, we are able to see many triangles that represent the metallic

surface of the substrate. The slot is terminated by an elliptical cavity on both sides. There

are also two strips crossing the slot that represents the location of the microstrips located at

the bottom of the substrate. One side of the transition is excited by a 50 microstrip line

while the microstrip line on the other half is terminated by a 50 real load impedance.

This can be interchanged due to the design being symmetrical.

Line ofSymmetry

EllipticalCavity

MicrostripLine(bottom ofsubstrate)

Slot


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33

4.2.2 Simulated Results

When simulating the back-to-back microstrip to slot transtion in FEKO, only the result for

return loss, S11, was really looked at and taken into consideration. This is because the

design is done just to prove the matching capabilities of a symmetrical microstrip to slot

transition through a two-port network. If it matches well, we are then able to use half of the

design and attach it to a linear taper profile.

Figure 4.4: Simulated Return Loss, S11, of a Back-to-Back Microstrip to Slot

Transition

Figure 4.4 shows the return loss of the transition obtained from GraphFEKO after running

the simulation in FEKO. From it, we can observe that the results of the return loss were

taken over a frequency range of 1 GHz to around 8 GHz.

UsableBandwidth


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34

By observing the result of the simulated return loss, S11, for the back-to-back microstrip to

slot transition, it can be seen that the return loss is about 30dB at the resonant frequency of

3.4 GHz. As for the usable bandwidth of this transition, it is defined over the frequency

range at which the S11 is at 10dB. Figure 4.4 shows the user bandwidth to be

approximately from 3 GHz to 7.25 GHz. Thus, the bandwidth can be calculated through the

equation:

%100)( 12 =

of

ffBW (4.3)

where fo is the frequency at which S11 is minimum

f1 and f2 are the frequencies at which S11 is at 10dB

By applying the above equation, the usable bandwidth for the back-to-back microstrip to

slot transition is approximately 125%. This result is considered to be satisfactory as the

objective of obtaining a wide bandwidth and impedance match of this transition is

achieved.

4.2.3 Sketch of Prototype

The sketch of the prototype, with all its dimensions, can be found in the Appendix B and it

shows how the antenna would look like physically. This sketch looks similar to the design

in FEKO as illustrated in figure 4.3.


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4.2.4 Measured Results

A physical version of the back-to-back transition was needed in order to verify the

simulated results obtained from FEKO. The program Protel was used to draw out an exact

design of both the top and bottom side of the transition with precise dimensions. Upon

completion, the Protel design was submitted to the Electronics Laboratory located in S309,

Hawken Engineering Building. The fabrication took only two hours and the etching came

out according to the given dimensions. The top and bottom view of the finished product can

be seen in figure 4.5 and 4.6 on the next page. As seen from the figures, a SMA connector

is soldered onto each of the microstrip lines. Soldering them to the side with the slot also

grounds the connectors.

Figure 4.5: Top View of Finished Prototype


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36

Figure 4.6: Bottom View of Finished Prototype

The result for the return loss of the back-to-back microstrip to slot transition was done

through a Network Analyser stationed in S507, Hawken Engineering Building. Since the

design is symmetrical, either connector can be connected to a coaxial cable which

calculates the return loss while the other is terminated by a 50 load. This results in the

design working as a one-port network.

From figure 4.7, the plot for the measured results, though not a clear one, can be seen. The

measured return loss for the transition was also taken over a frequency range of 1GHz to

8GHz on the x-axis, scaled at 0.7 GHz per division. The return loss on the y-axis was

scaled at 10dB per division. The same results were obtained when the load and excitation

were switched around.


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1 2.7 3.8 5.6 6.6 8 (GHz)

Figure 4.7: Measured Return Loss, S11, of a Back-to-Back Microstrip to Slot

Transition

From the plot, we can observe that the usable bandwidths for the measured results are

obtained over two sets of approximate frequency ranges: 2.7 GHz to 3.8 GHz and 5.6 GHz

to 6.6 GHz. This works out to a calculated bandwidth of around 34% for the first range and

17% for the second range. This shows that the transition to have only a narrow bandwidth

over two resonant frequencies of 3.2 GHz and 5.9 GHz. The return losses at these two

frequencies are observed to be at 37dB and 30 dB respectively.


It is normally difficult to obtain similar simulated and measured results when it comes to

testing of antennas. In the case of the back-to-back microstrip to slot transition, this proves

likewise.

S11ReturnL

oss(dB) UsableBandwidths

0

3.2 5.9

-10

-30-37


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38

For the simulated results obtained from FEKO, one has to realise that the program

considers the transition to be in an optimised environment unless stated. FEKO will only

take into consideration the dimensions and control parameters that the user includes in the

code. Other parameters, such as losses, mutual inductance and capacitance, if not included,

will be assumed to be at an optimised level. Therefore, because the transition is simulated

in an ideal case, the plot in figure 4.4 can be said to produce a good result.

By looking at figure 4.7, we can conclude that the measured result of the transition did not

produce a wide bandwidth as obtained for the simulated result. This could be due to the

factors as mentioned in the previous paragraph. However, the two peaks for the simulated

and measured return losses are located at around the same frequency point with a slightly

better return loss for the measured result. This proves that the transition works at the same

simulated and measured frequency. By testing the back-to-back transition physically, the

product tends to be subjected to other physical factors affecting it. For example, there are

bound to be losses in the cables used for testing and even a slight bend in a cable tends to

shift the result a little.

4.4 Summary

At the start of this project, the major concern was that the simulated results would not tally

with the measured results. Designing the back-to-back microstrip to slot transition allowed

the comparison of these results to be done. As expected, both the results were not similar

except for the resonant frequency. The reasons behind this have been discussed in the

previous section.


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Chapter 5: Design and Simulated Results of the Wide Band LTSA

39


A wide band linearly tapered slot antenna incorporates a slot antenna with a linear tapered

profile and is fed by a microstrip line. Both taper profiles and feeding techniques have

already been described in chapter 2 and further description of the linear taper profile and

microstrip feed can also be found in that chapter. This chapter will describe the design and

simulation processes of the wide band LTSA.

5.1 Features of LTSA

The LTSA generally has a lot of features including narrow beam width, high element gain

and wide bandwidth. A major disadvantage of the LTSA is that it requires either a

microstrip to slot transition or a coplanar waveguide to slot transition as part of its feeding

network. Due to this, the antenna design complexity increases and there is also a limit to

the wideness of the bandwidth of which the antenna can achieve.

5.2 Design Considerations

To design a wide band LTSA, it is important to obtain the required free space wavelength

and guide wavelength of the antenna. The effective dielectric constant is also required for

the guide wavelength. To obtain these parameters, it is recommended that section 3.3 of

Chapter 3 be referred to. In that section, equations are given to calculate these parameters.

Also, programs like PCAARD and slotline or microstrip calculators available on the

Internet can be use to obtain these parameters. The results from these programs tend to be

not as accurate. Thus, calculations through equations are still strongly recommended.


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5.3 Design on FEKO

Similar to the back-to-back microstrip to slot transition, the design for the wide band LTSA

was also done using FEKO. From the results observed in section 4.4 of chapter 4, we can

conclude that the simulated back-to-back transition works well. Thus, as mentioned before,

half of the transition is used for the wide band LTSA. A linearly taper profile is simply

attached to the end of the slot. Figure 5.1 shows how the top view of the wide band LTSA

would look like using the program WinFEKO.

Figure 5.1: Top View of Wide Band LTSA in WinFEKO

Since the measurements concerning the microstrip to slot transition of the antenna has

already been calculated in Chapter 4, all we need to do is alter some of the measurements

and include the dimensions of the linear taper profile. The overall dimensions for the wide

band LTSA are tabulated in the next page.

EllipticalCavity

MicrostripLine(bottom ofsubstrate)

14

Slot

TaperAngle


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Length of Antenna 133.5mm

Width of Antenna 61mm

Width of Slot, Ws 3mm

Length of Slot, Ls 25mm

Width of Microstrip Transmission Line, Wm 2.4mm

Length of Microstrip Transmission Line, Lm 20mm

Width of Quarter Wave Section, Wq 1.2mm

Length of Quarter Wave Section, L q 21.5mm

Radius of Elliptical Cavity 20mm

Taper Angle, 14

Table 5.1: Parameters for the Wide Band LTSA

Again, although the dimensions of the antenna can be calculated through given equations,

some of them are implemented into the design through the trial and error process. The

dimensions of the design should only be adjusted a little at a time when testing because

every little change matters when it comes to designing of antennas. The code written using

FEKO can be found in Appendix A.

5.4 Simulated Results

The wide band LTSA was simulated using FEKO and the results were plotted out using

GraphFEKO. For this particular antenna, the results taken into considerations were the

return loss, S11, and the radiation patterns of the E-plane and H-plane. The E-plane

represents radiation with respect to the vertical plane and the H-plane represents radiation

with respect to the horizontal plane. Similar to the back-to-back microstrip to slot

transition, the wide band LTSA was design to operate over a frequency of 1 GHz to 8 GHz.


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5.4.1 S11 Return Loss and Bandwidth

Figure 5.2: Simulated Return Loss, S11, of the Wide Band LTSA

Figure 5.2 shows the plot of the return loss of the wide band linearly tapered slot antenna as

obtained from GraphFEKO.

5.4.2 Radiation Patterns

The plots shown in figure 5.3 and 5.4 illustrate the E-plane and H-plane radiation patterns

obtained for four frequencies, 2.4 GHz, 3.38 GHz, 5.34 GHz and 6.46 GHz. Thesefrequencies were chosen because the return losses peak at that particular point. The peaks

can be observed from figure 5.2.

UsableBandwidth


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43

At 2.4 GHz

At 3.38 GHz

At 5.34 GHz


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44

At 6.46 GHz

Figure 5.3: Radiation Patterns for E-plane

At 2.4 GHz

At 3.38 GHz


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At 5.34 GHz

6.46 GHz

Figure 5.4: Radiation Patterns for H-plane


From the simulated return loss, S11, for the antenna, we are able to obtain a minimum

return loss of about 48dB at the resonant frequency of 6.46 GHz. A return loss of 10dB

can be obtained over the frequency of 3.25 GHz to 7.5 GHz. By referring to figure 5.2 and

using the equation (4.3), the usable bandwidth of the antenna is calculated to be

approximately 66%. Thus, the antenna can be considered to be wide band.


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As mentioned previously, the radiation patterns were taken at four different frequency

points of 2.4 GHz, 3.38 GHz, 5.34 GHz and 6.46 GHz with the bore sight of the antenna at

90. We will first discuss the radiation patterns taken with respect to the E-plane. At 2.4

GHz, a main lobe can be seen at the angle of 135. A couple of side lobes at the angles of

45 and 195 exist. The side lobes are negligible as they prove to be only very small. At

3.38 GHz, the main lobe can again be seen at the angle of 135. However, there are

considerable back lobes existing at 225 and 315 and a side lobe at 30. As the frequency

increases to 5.34 GHz, the main lobe at the bore sight is reduced significantly and is shifted

to the angle of 120. There are also more back lobes with a large back lobe appearing at the

angle of 240. At 6.46 GHz, we basically get the same radiation pattern as for 5.34 GHz

except there seems to be the existence of more back lobes but at a smaller scale. Judging

from the radiation patterns obtained at all four frequencies, we can conclude that more back

lobes are evident as the frequency increases. These extra lobes are undesirable because they

represent energy wasted from the antenna.

Looking at the radiation patterns on the H-plane, we can conclude the antenna eludes

almost symmetrical patterns. At a lower frequency of 2.4 GHz, we can observe that there is

a sufficiently large main lobe at the bore sight with no side lobes, although a couple of back

lobes do exist. As the frequency increases, one can observe the main lobe at the bore sightto have narrowed. However, there is an appearance of side and back lobes with

considerable gain appearing at the angles of 225 and 315 approximately. These are again

considered to be undesirable as there will be a loss of power from the antenna.

Judging from the discussion of the radiation patterns, the wide band linearly tapered slot

antenna tends to perform better at lower frequencies because there is less power loss due to

side and back lobes and thus the antenna can transmit a stronger signal. Normally, both the

free space and guide wavelengths are affected by frequency. An increase in frequency

would lead to a decrease in wavelength. Since antennas are design to operate better over a

given frequency, operating it at a different frequency requires a change in dimension as

well.


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5.6 Summary

This chapter provides a detailed design and simulation process of the wide band linearly

tapered slot antenna. The exact parameters used for the designing of the antenna in FEKO

were given with illustrations along the way.

From the results for return loss and bandwidth, the criteria set at the start of this project of

obtaining a wide band antenna was relatively achieved. By taking the vertical and

horizontal radiation patterns over four points from 1 GHz to 8 GHz, we were able to

observe the performance of the antenna and thus, can conclude that the antenna performs

better at lower frequencies.


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Chapter 6: Prototype and Measured Results of the Wide Band LTSA

48


A prototype of the wide band LTSA was built with the help of the Electronics Laboratory

located at S309, Hawken Engineering Building. The design for the prototype was done in

Protel before submitting to the laboratory for fabrication. This chapter will provide a sketch

of the prototype but primarily focuses on the measured results obtained from the Network

Analyser located in room S507, Hawken Engineering Building.

6.1 Sketch of Prototype

A sketch of the prototype was done in order to have a visualisation on how the antenna

would look like if built. It also gives us an idea on how big the antenna would be and if the

design will come out as planned. The sketch of the prototype can be found in Appendix B

and looks similar to figure 5.1 of chapter 5.

6.2 Measured Results

Similar to the back-to-back microstrip to slot transition, the same process was taken to

design and fabricate the antenna on the same ROGERS dielectric substrate of 2.0 dielectric

constant and thickness of 0.5mm. This process proved to be as efficient as before with the

help of the personnel at the Electronics Laboratory. The top and bottom view of the

finished product can be seen in figure 6.1 and 6.2. From the figures, we can clearly see a

slot terminated by an elliptical cavity on one side while the other side opens up to a linear

taper with an angle of 14. A microstrip line can also be seen crossing the slot at the bottom

of the substrate. A SMA connector is soldered to the microstrip in order to excite it.


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Figure 6.1: Top View of Finished Prototype

Figure 6.2: Bottom View of Finished Prototype


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6.2.1 S11 Return Loss and Bandwidth

1 5 5.8 8 (GHz)

Figure 6.3: Measured Return Loss, S11, of the Wide Band LTSA

Figure 6.3 shows the measured return loss obtained through the network analyser. The x-

axis is scaled at 0.7 GHz per division and the y-axis is scaled at 10 dB per division.

6.2.2 Radiation Patterns

The radiation patterns, E-plane and H-plane, for the wide band linearly tapered slot antenna

was taken through testing in the Anechoic Chamber. A mount for the antenna had to be

built in order to securely attach the antenna to the rotational stand in the chamber. This

mount was built with the help of the Electrical Engineering Workshop. During testing, the

antenna radiates a signal sent from a source horn antenna. The different radiation planes

can be changed by rotating the LTSA or the source antenna 90 accordingly.

S11ReturnLoss(dB

)

UsableBandwidth0

-39

5.3

-10


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Figure 6.4 illustrates the general orientation of the antenna from which the E-plane and H-

plane are taken. The orientations for the radiation patterns at all the frequencies are not

illustrated like in chapter 5 because the Network Analyser, unlike FEKO, is unable to

produce the sketches. The measured results for the radiation patterns were taken at the same

frequency points as the simulated results. Figures 6.5 and 6.6 show the measured results for

the radiation patterns for the E-plane and H-plane respectively.

Figure 6.4: General Orientation of Wide Band LTSA for E-plane and H-plane

90 90

270

180

270

0 180


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Figure 6.5: Radiation Patterns for E-plane

Figure 6.6: Radiation Patterns for H-plane

9090

180 180

270270

00

00

9090

180180

270 270

0 0

270 270

180 180

90 90


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By observing the measured return loss, S11, in figure 6.3, we are able to obtain a minimum

return loss of 39dB at the resonant frequency of 5.3 GHz. A return loss of 10dB can be

obtain over a frequency of 5 GHz to 5.8 GHz. Referring to figure 6.3 and applying equation

(4.3), the usable bandwidth can be calculated to be approximately 15%. From this, we can

conclude that the antenna can only work over a very narrow band.

Figures 6.4 and 6.5 show the radiation patterns for the E-plane and H-plane taken at the

frequencies of 2.4 GHz, 3.38 GHz, 5.34 GHz and 6.46 GHz. The bore sight of the antenna

was taken at 90. Again, the plots were taken from the Network Analyser.

Looking at the radiation pattern for the E-plane at 2.4 GHz, we notice a main lobe at the

angle of 45. There are minor small side and back lobes that may be neglected due to the

size of them. At 3.38 GHz, the main lobe has shifted towards the bore sight and there are

still the same side and back lobes. At the frequencies of 5.34 GHz and 6.46 GHz, a larger

main lobe is seen at the bore sight. However, more side and back lobes do exist.

For the radiation patterns obtained for the H-plane, we are able to notice almost

symmetrical patterns at all the frequencies. At all the frequencies, we are able to see a main

lobe at the bore sight that narrows as the frequency increases. However, as the frequency

increases, there are also side and back lobes appearing. These are apparent at the higher

frequencies of 5.34 GHz and 6.46 GHz with obvious side lobes around the angles of 0 and

180.

Judging from the plots for both the E-plane and H-plane taken at the four frequencies, we

can conclude that the wide band linearly tapered slot antenna works better at lower

frequencies. This is because the antenna is able to transmit a stronger signal, as there is less

power loss due to the appearance of side and back lobes.


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6.4 Summary

From this chapter, we are able to draw a conclusion to the physical performance of the wide

band linearly tapered slot antenna. Although the measured return loss was not obtained as

what was expected, the radiation patterns were almost identical to what was predicted.

Experience was also gain in the form of hands on experience when it comes to the setting

up and operation of the Network Analyser and Anechoic Chamber. This might prove to be

useful in the working environment.


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Chapter 7: Evaluation

55


This chapter will compare and evaluate all the simulated and measured results obtained for

the wide band LTSA. Judging from chapter 4, the simulated results of the back-to-back

microstrip to slot transition tend to differ a lot from the measured ones. Thus, this chapter

will discuss the similarities and differences of both sets of results.

7.1 Evaluation of Simulated and Measured S11 Return Loss and Bandwidth

The simulated and measured results of the S11 return loss and bandwidth can be seen in

figures 5.2 and 6.3 in chapters 5 and 6 respectively. Both results were taken over the same

frequency and were supposedly taken under the same design and physical considerations as

well. The comparisons will be made on the minimum return loss, resonant frequency and

usable bandwidth. These are tabulated below.

Simulated Measured

Minimum Return Loss -48dB -39dB

Resonant Frequency 6.46 GHz 5.3 GHzUsable Bandwidth 66% 15%

Table 7.1: Compiled Simulated and Measured Results

Judging from the tabulated results, we can conclude that the antenna designed in FEKO has

a better return loss of 48dB and operates at a higher resonant frequency of 6.46 GHz. The

physical design has its resonant frequency shifted down to 5.3 GHz and a slightly worse

return loss of 39dB. The change in the return loss and resonant frequency for the physical

design could be due to the fabrication and testing process.


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56

As mentioned before, the fabrication of antenna was done by submitting a Protel design to

the Electronics Laboratory. Although the design done through Protel should produce an

antenna with exact parameters, it is still bound to have a slight percentage of error. For

example, the Keep Out Layer in Protel is formed by having a border of certain thickness.

This thickness is bound to overlap into the actual dimensions of the antenna, thus making

the length or the width slightly off the required measurements. A slight change in

dimensions, be it 0.1mm, will make a difference in the output of the antenna. This, together

with reasons mentioned in section 4.3 of chapter 4 concerning the testing of the antenna,

tends to have at least some effect on performance of the antenna.

7.2 Evaluation of Simulated and Measured Radiation Patterns

Radiation patterns of the wide band LTSA helps us understand how much signal is being

transmitted and in which direction. Comparisons will be made on the E-plane and H-plane

of both the simulated and measured results to see if the antenna designed in FEKO

performs similarly as to the one built.

By observing figure 5.3 of chapter 5 and 6.5 of chapter 6, we are able to draw a comparison

between the simulated and measured radiation patterns for E-plane. Unlike the difference in

the return losses, the radiation patterns for both the simulated and measured results seem to

have almost identical patterns at the bore sight of 90. The main lobe, taken at all the

frequencies, are all facing the same direction although there seems to be the existence of

more side and back lobes for the measured results. Also, there seems to be a lot of noise in

the measured results but this is due to the reflections off objects such as the rotational stand

or the source antenna in the Anechoic Chamber. Having absorbers located around this

objects can reduce this noise but still, not all the noise can be subdued.

Referring to figures 5.4 and 6.6 of chapter 5 and 6 respectively, we are also able to

conclude that the simulated and measured radiation patterns for the H-plane produce almost


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57

similar outcomes. All the main lobes for both sets of results are pointing at the bore sight of

90.The only difference is that the main lobe of the simulated results tends to be wider and

larger. There are generally the same side and back lobes located in the same direction.

There is again noise similar to those obtain for the E-plane.

7.3 Summary

This chapter enables us to compare both sets of radiation patterns obtained for the wide

band linearly tapered slot antenna. From the comparisons made, we can conclude that the

actual antenna performs according to the design specification made in FEKO. However, a

great difference in bandwidth is apparent where the simulated results obtained a 66%

bandwidth whereas the measured ones only had a 15% bandwidth. This shows the actual

antenna can operated well only over a narrow bandwidth and thus, not meet the criteria of

the antenna being wide band. Steps that can be taken to improve this will be mentioned in

section 8.1 of chapter 8.


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Chapter 8: Conclusion

58


The main focus of this thesis was to successfully design and construct a Wide Band

Linearly Tapered Slot Antenna and at the same time obtain almost satisfactory results. An

important objective was to ensure that the simulated results of the antenna done in FEKO

and the measured results obtain from the actual testing of the antenna produce almost

similar outcomes.

On the whole, the antenna was completed on time. The simulated usable bandwidth of the

antenna was calculated to be 66% while the measured usable bandwidth was calculated to

be only 15%. Therefore, the simulated results met the criteria of obtaining a wide band for

the antenna although the measured results did not. The radiation patterns for both thesimulated and measured results were almost the same to prove that the signal of the antenna

in FEKO and actual antenna radiates the signal in similar directions.

Upon the completion of this thesis, all the aims and objectives made at the start of the

project were met. The wide band linearly tapered slot antenna was completed on time and a

great deal was learnt about the program FEKO. Finally, knowledge on the microwave field

and travelling wave antenna was gain and this will be an invaluable asset for the future.


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8.1 Future Work

Although this thesis was considered to be a minor success, several factors of the wide band

linearly tapered slot antenna can still be improved. As we have seen during the course of

this project, the simulated and measured results were not similar at times and it was also

difficult to obtain a wide bandwidth for the antenna. Also, due to time constrains, different

designs of the antenna were researched on but further work was not carried out.

Listed below are three factors that should be research on and implemented to ensure better

performance and results if future work was to be done on this thesis.

Implementation of the Double Y-balun

When designing the antenna, one of the major problems was the microstrip to slot

transition that is crucial in obtaining a wide bandwidth for the antenna.

Implementing a double y-balun by following closely to the steps mentioned in

section 4.1 of chapter 4 would result in the bandwidth of the antenna improving a

great deal.

Changing of parameters to observe differences in results

Design and construction of the wide band linearly tapered slot antenna was done on

a fixed set of parameters. Although satisfactory results were obtained, one has to

wonder how different the results would have been if some of the parameters were

changed. Therefore, comparisons of results should be made on the same type of

antenna with a series of different parameters.


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60

Measurement of discontinuities of the antenna

When comparing the simulated and measured return loss of the antenna, the

simulated result obtained a much better return loss and bandwidth than the

measured result. The reason behind this can be found out by measuring the

discontinuities of the antenna in the time domain through the Network Analyser.

By experimenting and implementing the recommended future work, this thesis project can

be use as a base for the next student to build a solid foundation on. Hence, this will ensure

that an improved wide band linearly tapered slot antenna will perform better and exceed

more requirements.


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Appendix A

61

Appendix A

FEKO code for Quarter Wave Microstrip Transformer

** Scaling factor since all dimensions below in mm

SF 1 0.001

#h = 1.0

#epsr = 2.0 ** Relative permittivity

#effepsr = 1.983 ** Effective permittivity

#freq = 8.0e9

#lam = 1000 * #c0 / #freq / sqrt(#effepsr)

#x = 20

#lam1 = #lam/4 + #x

#a = 1.20

#b = 2.40

** Segmentation parameters#tri_len = #lam / 12

#fine_tri = #lam / 16

IP 0 #tri_len

** Define Points

DP A 0 0 #h

DP B 0 #x #h

DP C #a #x #h

DP D #a 0 #h

DP E #a 0 #h

DP F #a #x #h

DP G #b #x #h

DP H #b 0 #h

DP 1 0 #x #h

DP 2 0 #lam1 #h

DP 3 #a #lam1 #h

DP 4 #a #x #h

** Microstrip Feed

LA 1

BP A B C D #fine_tri #fine_tri

LA 2

BP E F G H #fine_tri #fine_tri

LA 3

BP 1 2 3 4 #fine_tri #fine_tri

** End of geometry input

EG 1 0 0 0 0 1 1GF 10 1 0 1 1 0

#h

1.0 #epsr 1 0.001

LE 2 3 3 100 0


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Appendix A

62

** Frequency

FR 51 0 1.0e9 #freq

** Voltage source at the wire centre with impressed power

PW 3 0 1.0 50 0

AE 0 A H 3 1.0 0 0

** Far-field pattern

FF 1 181 1 1 0 0 2


FF 1 181 1 1 0 90 2

** End

EN

FEKO code for Back-to-Back Microstrip to Slot Transition


SF 1 0.001

#h = 0

#t = -0.5

#epsr = 2.0

#freq = 8.0e9

#lam = 1000 * #c0 / #freq / sqrt(#epsr)

#x = -45

#y = -65

#z = -66

#xx = 45

#yy = 65#zz = 66

#a = 78

#b = 81

#c = 90.5

#d = 110

#1 = -20.5

#2 = -18.1

#3 = 20.5

#4 = 18.1

#5 = -19.3

#6 = 19.3

#L = 49

#tl = #lam/4

#tl1 = min(#tl, 2) ** Comparable to slot width

#tl2 = #tl/2

IP 0 #tl


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Appendix A

63

** Define Points of Slotline

DP A -25 #L #h

DP B -25 #a #h

DP C 0 #a #h

DP D 0 #L #h

DP E -25 #b #h

DP F -25 #d #h

DP G 0 #d #h

DP H 0 #b #h

** Define Points of Microstrip

DP 1 #1 #l #t

DP 2 #1 69 #t

DP 3 #5 69 #t

DP 4 #5 #l #t

DP 1.1 #5 #l #t

DP 2.1 #5 69 #t

DP 3.1 #2 69 #t

DP 4.1 #2 #l #t

DP 1.2 #1 69 #t

DP 2.2 #1 #c #t

DP 3.2 #5 #c #t

DP 4.2 #5 69 #t

DP 11 #1 #L #h

DP 12 #1 #d #h

DP 13 #2 #d #h

DP 14 #2 #L #h

DP 22 #1 #a #h

DP 23 #2 #a #h

DP 32 #1 #b #h

DP 33 #2 #b #h

** Elliptical Open Circuit

DP I #x #L #h

DP J #x #a #h

DP K #y #L #h

DP L #y #a #h

DP M #x #b #h

DP N #x #d #h

DP O #y #b #h

DP P #y #d #h

DP Q #z #L #h

DP R #z #a #h

DP S #z #b #h

DP T #z #d #h

DP M1 #x 101 #h

** Define Points of Slotline2

DP AA 0 #L #h

DP BB 0 #a #h


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Appendix A

64

DP CC 25 #a #h

DP DD 25 #L #h

DP EE 0 #b #h

DP FF 0 #d #h

DP GG 25 #d #h

DP HH 25 #b #h

** Define Points of Microstrip2

DP 1a #3 #l #t

DP 2a #3 69 #t

DP 3a #6 69 #t

DP 4a #6 #l #t

DP 1a1 #6 #l #t

DP 2a1 #6 69 #t

DP 3a1 #4 69 #t

DP 4a1 #4 #l #t

DP 1a2 #3 69 #t

DP 2a2 #3 #c #t

DP 3a2 #6 #c #t

DP 4a2 #6 69 #t

DP 11a #3 #L #h

DP 12a #3 #d #h

DP 13a #4 #d #h

DP 14a #4 #L #h

DP 22a #3 #a #h

DP 23a #4 #a #h

DP 32a #3 #b #h

DP 33a #4 #b #h

** Elliptical Open Circuit2

DP Ia #xx #L #h

DP Ja #xx #a #h

DP Ka #yy #L #h

DP La #yy #a #h

DP Ma #xx #b #h

DP Na #xx #d #h

DP Oa #yy #b #h

DP Pa #yy #d #h

DP Qa #zz #L #h

DP Ra #zz #a #h

DP Sa #zz #b #h

DP Ta #zz #d #h

DP M1a #xx 101 #h

LA 0

PH J L K I L #tl1

PH M N P O M1 #tl1

PH J B A I B #tl1

PH M E F N E #tl1

BP L R Q K


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Appendix A

65

BP P T S O

BP O S R L #tl1

PM A 11 22 B

#tl2 #tl1

BP 22 11 14 23 #tl2 #tl1

PM 23 14 D C

#tl2 #tl1

BP 32 33 13 12 #tl1 #tl2

PM 32 12 F E

#tl2 #tl1

PM 13 33 H G

#tl2 #tl1

LA 1

BP 1 2 3 4 #tl2 #tl1

BP 1.1 2.1 3.1 4.1 #tl2 #tl1

BP 1.2 2.2 3.2 4.2 #tl2 #tl1

** Add feed strip

BP 1 4.1 14 11 #tl1 #tl1

LA 2

PH Ja La Ka Ia La #tl1

PH Ma Na Pa Oa M1a #tl1

PH Ja CC DD Ia CC #tl1

PH Ma HH GG Na HH #tl1

BP La Ra Qa Ka

BP Pa Ta Sa Oa

BP Oa Sa Ra La #tl1

PM AA 14a 23a BB

#tl2 #tl1

BP 22a 11a 14a 23a #tl2 #tl1

PM 22a 11a DD CC

#tl2 #tl1

BP 32a 33a 13a 12a #tl1 #tl2

PM 33a 13a FF EE

#tl2 #tl1

PM 12a 32a HH GG

#tl2 #tl1

** Add feed strip

BP 1a 4a1 14a 11a #tl1 #tl1

LA 3

BP 1a 2a 3a 4a #tl2 #tl1

BP 1a1 2a1 3a1 4a1 #tl2 #tl1

BP 1a2 2a2 3a2 4a2 #tl2 #tl1

** End of geometry input

EG 1 0 0 0 0

GF 11 2 0 1 1

#h

0.5 #epsr 1 0.001

1 1


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Appendix A

66

** Frequency

FR 51 0 1.0e9 #freq

** Voltage source at the wire centre with impressed power

PW 1 1.0

AE 0 0 1 0 1 0

LE 2 3 0 50

OS 1 1


FF 1 181 1 1 0 0 2


FF 1 181 1 1 0 90 2

** End

EN

FEKO code for Wide Band Linearly Tapered Slot Antenna


SF 1 0.001

#h = 0

#t = -0.5

wide band linearly tapered slot

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