transverse-flux-type cylindrical linear synchronous motor ... · fig. 1 [7]. for conventional...

4346 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 61, NO. 8, AUGUST 2014

Transverse-Flux-Type Cylindrical LinearSynchronous Motor Using Generic Armature

Cores for Rotary MachineryJung-Seob Shin, Member, IEEE, Ryuji Watanabe, Student Member, IEEE, Takafumi Koseki, Member, IEEE,

and Houng-Joong Kim

Abstract—This paper presents the design and analysis of atransverse-flux-type cylindrical linear synchronous motor usinggeneric armature cores for rotary machinery that can addressthe problem of complex structures in conventional transverse-flux-type topologies. First, the operational principle and structural ad-vantages of the proposed model are explained. The thrust densityand cogging force are investigated during the initial design stageusing an application in which large thrust density and low coggingforce are required. The proposed model is both theoretically andnumerically designed by using a magnetic-circuit method and a3-D finite-element method, respectively. Finally, the results andefficacy of our structural concept are experimentally validated.

Index Terms—Cogging force reduction, finite-element (FE)method (FEM), permanent-magnet linear synchronous motor(PMLSM), thrust design, transverse-flux-type machine.

NOMENCLATURE

lg Air-gap length.lgc Air-gap length considering the Carter coefficient.τp Pole pitch.a Half length of the field magnet.z Distance in the moving direction.Bg Air-gap flux density under no-load conditions.Br Remanent flux density of the field magnet.Am Dimension of a magnet.Ag Dimension of the air-gap area.μrm Relative recoil permeability of magnet.lm Magnet length in the magnetization direction.N Number of winding turns on a salient pole.Erms RMS value of the back electromotive force (EMF).p Number of pole pairs.I Armature current.Hc Magnetic coercive force.da Half slot length in the moving direction.lc Height where the armature coil is wound.Kcp Packing factor of coil.Sc Dimension of the armature coil.

Manuscript received January 31, 2013; revised May 1, 2013 and June 25,2013; accepted July 6, 2013. Date of publication July 24, 2013; date of currentversion February 7, 2014.

J.-S. Shin, R. Watanabe, and T. Koseki are with the Department ofElectrical Engineering, School of Engineering, The University of Tokyo,Tokyo 113-0033, Japan (e-mail: [email protected];[email protected]; [email protected]).

H.-J. Kim is with KOVERY Company, Ltd., Suwon 440-301, Korea (e-mail:[email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TIE.2013.2274426

I. INTRODUCTION

L INEAR MOTION is an essential requirement in manyapplications in industrial fields such as manufacturing

processes [1], transportation [2], and robotics [3].Two methods are generally used to achieve linear motion.

One method involves the use of rotary motors with rotary-to-linear converters. Because both rotary and linear motorshave been extensively studied, such converters can be designedat low costs and with large thrust densities. However, suchmechanical conversion creates large noise and friction, whicheventually leads to the deterioration of positioning accuracy.The second method is to employ linear motors. Here, a directlinear drive is possible, which can yield low noise, easy mainte-nance, and high positioning accuracy. Several types of linearmotors are available. In particular, because of the advent ofrare-earth permanent magnets [4], a permanent-magnet linearsynchronous motor (PMLSM) has been used in applicationsthat require large thrust densities and high positioning accu-racies. However, one of the drawbacks of the PMLSM is thatthe thrust density is comparatively lower than that of rotarymotors equipped with rotary-to-linear converters; this drawbackhas severely limited the high-performance applications of thePMLSM.

Hence, increasing the thrust density is one of the mostimportant research topics with regard to the PMLSM today. Atransverse-flux-type topology is an ideal alternative in whichthe flux is carried in the iron back in a plane transverse(perpendicular) to the direction of motion and current flow.Therefore, the magnetic and electric loads in the machine arealong different planes. This allows for an inverse relationshipto exist between the capability of the machine to produceforce and the pole pitch, which results in a higher thrust den-sity [5].

However, the process for manufacturing conventionaltransverse-flux-type topologies is generally difficult because ofthe complex structure resulting from the presence of a 3-Dmagnetic circuit [6]. In conventional manufacturing processes,a large number of segmented components are needed to forma magnetic circuit, and the use of lamination is difficult; thislimits its application in industrial fields.

In earlier works, a transverse-flux-type cylindrical linearsynchronous motor using generic armature cores for rotary ma-chinery has been proposed to address the problem of complexstructures in conventional transverse-flux-type topologies [7].

0278-0046 © 2013 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.

SHIN et al.: LINEAR SYNCHRONOUS MOTOR USING GENERIC ARMATURE CORES FOR ROTARY MACHINERY 4347

Fig. 1. Fundamental principle of the proposed model. (a) Cross section ofa general rotary synchronous motor. (b) Cross section of the proposed linearmotor.

However, in these works, the structural advantages of the pro-posed configuration have not been investigated. Furthermore,the design results have not been experimentally validated.

This paper mainly focuses on three things. First, the op-erational principle and structural advantages of the proposedmodel are investigated. Second, the thrust design and coggingforce are evaluated by using an application in which largethrust density and small cogging force are required. Third, theprototype model is experimentally verified. In Section II, theoperational principle and structural advantages are introduced.In Section III, we explain a method that can reduce the coggingforce of the proposed model. In Sections IV and V, the proposedmodel is both theoretically and numerically designed by using amagnetic-circuit method and a 3-D finite-element (FE) method(FEM), respectively. Finally, in Section VI, the results andefficacy of the structural concept of our initial prototype modelare experimentally validated.

II. OPERATIONAL PRINCIPLE AND

STRUCTURAL ADVANTAGES

A. Fundamental Principle of the Proposed Model

The fundamental principle of the proposed model is shown inFig. 1 [7]. For conventional rotational synchronous machinery,as shown in Fig. 1(a), the number of armature poles is differentfrom that of the field magnets. Therefore, the rotor rotatesafter the application of an armature current that has a phasedifference of 120◦ with each armature coil.

However, if the number of armature poles is equal to thatof the field magnets, as shown in Fig. 1(b), the number ofmagnetic circuits that can be generated is the same as thenumber of field magnets, and the rotor remains stationary since

the magnetic balance between the armature and field sides ismaintained.

The fundamental principle of the proposed model is thegeneration of a longitudinal force that can drive the armatureside toward the z-direction.

B. Structure and Operational Principle

Fig. 2 shows the configuration of the three-phase unit ofthe proposed model. In the overall configuration, the armatureside is the mover, and the field side is the stator, as shown inFig. 2(a). The armature side consists of armature units in anonmagnetic material box. The field side consists of field unitsinside a stainless steel pipe fixed at both ends.

A basic armature unit has an even number of salient poles anda concentrated winding structure along with a four-salient-poleconfiguration, as shown in Fig. 2. Every armature coil is woundin series with a phase difference of 180◦. By applying a currentto these coils, each of them is excited with a phase difference of180◦. In Fig. 2(b), −U denotes the current component shiftedby 180◦ from U . A field unit consists of an even number of fieldmagnets (equal to the number of salient poles in the armatureunit) and an iron core, as shown in Fig. 2(b). The field magnetsare magnetized along the radial direction, similar to that ina conventional cylindrical structure. In this configuration, notonly the armature core of three-phase machines but also thearmature core of the stepping motor can be used as the armaturecore. Furthermore, the field side is designed according to theshape of the armature core.

When a field unit is accurately located in the center ofthe armature unit, a magnetic circuit is formed, as shown inFig. 2(c). The main flux flows transversely from the north poleto the south pole along the armature core. In this manner,the cross-sectional symmetrical magnetic form of the balancedmagnetic circuits, which is the same number of salient poles inarmature unit, emerges.

The armature cores are arranged along the direction ofmovement (i.e., the z-direction), as shown in Fig. 2(d). Eachcore is spatially separated by a difference of 120◦. The fieldmagnets are arranged along the z-direction, and each magnetis electrically separated by 180◦. The nonmagnetic materialspacer isolates the magnetic paths between the iron cores. Inthis structure, each unit in the armature and field sides is mag-netically separated. The flux flow is transverse to the directionof movement, and the longitudinal flux flow (similar to that in aconventional cylindrical structure) back to the common core isabsent [8], [9]. Therefore, any core–pole combination, includ-ing not only the fundamental three-core–two-pole combinationbut also the nine-core–eight-pole combination, can be easilyachieved by arranging each unit in the armature and field sidesalong the direction of movement.

Fig. 3 shows the principle of generating thrust when a three-phase ac current is applied to the armature coil in each phasein the fundamental three-core–two-pole combination. The prin-ciple of generating thrust in the proposed model is basicallythe same as that in a conventional PMLSM. When the U -phasecore is between the north and south poles where the electricangle of the U -phase core is equal to 90◦, the armature current


Fig. 2. Configuration of the three-phase unit. (a) Entire configuration.(b) Armature and field units. (c) Magnetic circuits. (d) Configuration alongthe direction of movement. (In (d), the iron cores in the field side have beenremoved for better understanding and clarity.)

with magnitude I is applied to the armature coil in the U -phase,and the armature current with magnitude 0.5I is applied to thearmature coils in the V - and W -phase coils in the directionopposite to the U -phase, as shown in Fig. 3(a). Under thiscondition, the north pole is generated at the salient pole in theU -phase, and the south pole is generated at the salient pole in

Fig. 3. Principle of generating thrust. (a) Three-phase ac current. (b) Inter-action between the armature and field sides. (In (b), only a salient pole in thearmature core for each phase is considered because of its symmetrical structure.Furthermore, the iron cores in the field side have been removed for betterunderstanding and clarity.)

the V - and W -phases. After interaction with the field magnets,the armature side moves along the z-direction, as shown inFig. 3(b). In this manner, the three-phase ac current applied toall armature coils in each phase generates the moving magneticfield through all salient poles; thereafter, this moving magneticfield interacts with the field magnets. As a result, a longitudinalforce is generated that drives the armature side along thez-direction.

C. Structural Advantages

With regard to the PMLSM research, the structural aspecthas to be considered because it considerably affects the per-formance and stiffness of the linear drive system and is alsorelated to the versatility of its applications in industrial fields.The proposed model has structural advantages because of thefollowing reasons.

1) Simple structure: The magnetic structure comprises onlyone cylindrical field unit and one unsegmented armaturecore. Because of the unsegmented armature core, themechanical stiffness is high.

2) Easy assembly: Since these magnets can be easily placedinto the holes of the iron core, there is no need for using astrong adhesive, and specific equipment is not needed tofix these magnets.

3) Use of lamination: Unlike the configuration in con-ventional transverse-flux-type topologies, the proposedmodel has a 2-D magnetic circuit in which the main flux


flows transversely from the north pole to the south polealong the armature core, as shown in Fig. 2(c). In thismagnetic circuit, the iron cores in the armature and fieldunits can be easily fabricated using laminated steel platesthat are arranged along the z-direction.

4) Cancellation of the strong normal attractive force be-tween the armature and field sides: The strong normalattractive force between the armature and field sides isone of the most important structural problems in con-ventional single-sided PMLSM; furthermore, it consid-erably affects the mechanical stiffness of a linear drivesystem. Therefore, the burden placed on the supportingmechanism is large, thereby complicating the manufac-turing process [4]. However, the proposed model is in-herently compensated by the cross-sectional symmetricalmagnetic form of the balanced magnetic circuits. As aresult, mechanical support can be easily achieved. There-fore, the burden placed on the supporting mechanism issmall, thereby simplifying the manufacturing process.

III. CONSIDERATION OF COGGING FORCE REDUCTION

A cogging force arises from the interaction between the fieldmagnet and the armature core. Large cogging forces resultin thrust ripples and noise, which result in poor positioningaccuracy.

In applications such as the driving source in liquid-crystal-display color filter inspection systems in which positioningaccuracy of less than 5 μm is required, cogging force reductionis one of the most important factors to be considered in thePMLSM design. Many methods have been proposed regardingthe reduction of the cogging force in PMLSMs, e.g., skewing,semiclosed slots, and optimization of the magnet length [10],[11]. However, these methods can be a burden during the man-ufacturing stage and can occasionally affect the manufacturingcost.

The magnitude of the cogging force is inversely proportionalto the least common multiple (LCM) of the number of coresand poles. Therefore, the cogging force can be significantlyreduced by properly selecting a core–pole combination. Thenine-core–eight-pole combination (in which the LCM is fourtimes that of the nine-core–six-pole combination based onthe fundamental three-core–two-pole combination) has beenverified as a useful method for cogging force reduction; usingthis method, the cogging force is reduced to over 50% ascompared to the other combinations based on the fundamentalthree-core–two-pole combination [12].

Fig. 4 shows the characteristics of nine-core–six-pole andnine-core–eight-pole combinations. The former consists of onlythree sets of fundamental three-core–two-pole combinations, asshown in Fig. 4(a). In this configuration, the coil connectionis U , V , and W because the V - and W -phase cores in eachset have electric phase differences of 120◦ and 240◦ with theU -phase core in each set, respectively.

In the nine-core–eight-pole combination, the position rela-tionship between the armature cores and field magnets is thesame as that in the nine-core–six-pole combination. However,the sequence of placement of the cores and magnets is different.

Fig. 4. Characteristics of two types of core–pole combinations. (a) Configura-tion of nine-core–six-pole combination. (b) Configuration of nine-core–eight-pole combination. (c) Coil connection of the proposed model. (d) Averageeffective flux in the U -phase. (−U , −V , and −W are the current componentsshifted by 180◦ from their U , V , and W counterparts, respectively. Further-more, in (a) and (b), only a salient pole in the armature core for each phase isconsidered because of its symmetrical structure.)

As shown in Fig. 4(b), nine armature cores are placed alongsideeight magnets for a total distance of T . If the pole pitch is 9 mm,the total distance is 72 mm, and the slot pitch is 8 mm. For thisposition relationship, the fourth and seventh cores have electricphase differences of 120◦ and 240◦ with the first core, respec-tively. Furthermore, the fifth, sixth, eighth, and ninth cores haveelectric phase differences of 120◦ and 240◦ with the secondand third cores, respectively. Therefore, if the first and third


cores are U -phase cores, the fourth and sixth cores are V -phasecores, and the seventh and ninth cores are W -phase cores; thisimplies that three continuous cores have the same phase.

In the proposed model, a Y-connection is used in order toapply a three-phase ac current to the armature coil, as shownin Fig. 4(c). Every armature coil in each core in each phaseis wound to all the salient poles in the series with a phasedifference of 180◦. Furthermore, from the position relationshipbetween the armature cores and field magnets in which a southpole is present below the U2 core, the armature coils in U2, V 2,and W2 have to be wound in the opposite direction to the othercores in the same phase. The armature coil in each core for eachphase is connected in parallel.

On the other hand, the disadvantage of applying a nine-core–eight-pole combination is a decrease in the thrust density.This is a result of the asymmetric characteristics between thearmature cores and field magnets, i.e., U1 and U3 cores have anelectrical phase difference of 20◦, as shown in Fig. 4(b). Hence,the average effective flux in the U -phase decreases by 4%, asshown in Fig. 4(d). As a result, the thrust in a nine-core–eight-pole combination is approximately 4% less than that in a nine-core–six-pole combination. However, we have decided to usea nine-core–eight-pole combination in the proposed model forcogging force reduction because the decrease in the coggingforce is much larger than that in the thrust density.

IV. PRELIMINARY DESIGN OF THRUST

We have employed the magnetic-circuit method for theoreti-cal modeling of thrust in the preliminary design stage and madethe following assumptions [13], [14].

1) The field unit is accurately located in the center of thearmature unit.

2) The permeability of iron cores is infinitely large. Mag-netic saturation is neglected.

3) The slot effect is compensated by the Carter coefficient.4) The flux leakage is neglected. All flux from the field

magnet flows into the armature teeth through an air gap.5) Behavior in a one-phase configuration is considered be-

cause behavior in a three-phase configuration can beestimated from the results in the one-phase configuration.

A. Air-Gap Flux Density

If the slot effect is compensated by the Carter coefficient, theair-gap flux under no-load condition by moving an armatureunit φg(z) is distributed, as shown in Fig. 5. In Fig. 5, z isthe distance in the moving direction and can be expressed byvelocity v multiplied by time t. Under these conditions, φg(z)can be expressed by a Fourier series, as shown in

φg(z)=∑k=1

4φg

(2k − 1)πsin

((2k−1)πa

2τp

)cos

((2k − 1)πz

τp

).

(1)

From (1), the fundamental component of φg(z) and itsdensity Bg(z) are expressed in (2) and (3). In (3), Bg has been

Fig. 5. Air-gap flux distribution by moving an armature unit.

derived on the basis of the magnetic-circuit method [13] and isexpressed in (4)

φg(z) =4φg

πsin

(πa

2τp

)cos

(πz

τp

)(2)

Bg(z) =φg(z)

Ag=

4Bg

πsin

(πa

2τp

)cos

(πz

τp

)(3)

where

Bg∼= Br(

Ag

Am+

μrmlgclm

) . (4)

If the dimensions of the air gap and air-gap length areconstant and the flux leakage is neglected, the air-gap flux underno-load conditions is affected by Am and lm. The effectivedesign for large air-gap flux is to select Am and lm as large aspossible. This affects the design for large magnetomotive force(MMF) and relatively small magnetic reluctance of the fieldmagnet, which results in a large flux in the magnetic circuit.However, Am and lm cannot be infinitely large, owing to factorssuch as limited space in the field side, manufacturing cost, andmechanical strength. Therefore, proper selection of Am and lmin the designated volume is important in the actual design stage.

B. Maximum Thrust

If the flux leakage is negligible, all flux from the fieldmagnets is linked with armature coils, and the back EMF canbe expressed as

E(z) = −Ndφg(z)

dt. (5)

In the control, when the d-axis current is maintained atzero, the maximum thrust per armature unit Ft_1unit can beexpressed as shown in (6). The maximum thrust is generatedin a position in which the electrical phase difference betweenthe armature core and north pole is 90◦

Ft_1unit = pErmsI

v=

2√2p πφgNI

τpsin

(πa

2τp

). (6)

From (6), thrust is generally determined by electrical loadNI multiplied by the effective flux flowing to the armature corefrom the magnetic load Hclm if other conditions are constant.

The maximum thrust per three-phase Ft_3phase, consideringseveral core–pole combinations, is estimated from (7). In (7),


Fig. 6. Three-dimensional mesh FE model.

1.5 specifies the total amount of thrust for a general three-phasemachine, and m is the number of armature units per phase.

Also, α is a coefficient that denotes the ratio of the averageeffective flux in the U -phase in core–pole combinations whichhave the asymmetric characteristics between the armature coresand field magnets to the average effective flux in the U -phasein core–pole combinations that are based on fundamental three-core–two-pole combinations. In the case of a nine-core–eight-pole combination, α is 0.96, as shown in Fig. 4(d). This is theresult of the asymmetric characteristics between the armaturecores and field magnets in nine-core–eight-pole combination,i.e., U1 and U3 cores in nine-core–eight-pole combination havean electrical phase difference of 20◦, as shown in Fig. 4(b),compared with that in nine-core–six-pole combination

Ft_3phase = 1.5 Ft_1phase = 1.5 m α Ft_1unit. (7)

V. THRUST DESIGN USING 3-D FEM

In spite of our theoretical modeling, all flux from the fieldmagnet may, in reality, not flow into the armature teeth throughthe air gap. Therefore, a design in the initial design stage em-ploying numerical tools as part of the design process providesuseful information regarding flux leakage, flux distribution, andmagnetic saturation. In the numerical design and analysis using3-D FEM, the JMAG-Designer 10.4.3h commercial package isused [15].

The 3-D mesh of the proposed model is shown in Fig. 6. Inthe 3-D FEM analysis, a four-salient-pole model is considered,and three armature cores per phase and a periodic boundarycondition have been applied to save the computation time.

A. Specification and Materials in the Proposed Design

Table I and Fig. 7 show the primary design specifications andmaterials used in the proposed model.

A total volume of 60 mm × 60 mm × 108 mm and a nine-core–eight-pole combination were selected. In this volume, the

TABLE IMAIN DESIGN SPECIFICATIONS AND MATERIALS [16], [17]

Fig. 7. Main design specifications of parts. (a) x–y plane. (b) x–z plane.

pole pitch τp and slot pitch τs are 13.5 and 12 mm, respectively.All dimensions in the proposed model, except for da, are fixedand selected on the basis of spatial limitations and mechanicalstrength. The core of the 50JN230 in which the maximummagnetic flux density in the linear operation region is closed to1.5 T and the thickness of laminated piece is 0.5 mm has beenused as iron core in the armature and field sides [16]. Also, thepermanent magnet of N50M in which Br is about 1.4 T hasbeen used as the field magnet [17]. The velocity v is 1 m/s, andthe drive frequency f is 37.03 Hz.


B. Key Design Variables for Thrust Design

When only the structure and total volume are known, it isimportant to find the point at which the maximum thrust isgenerated.

As shown in (6), thrust is proportional to the multiplication ofmagnetic and electric loads. Magnetic load Hclm represents theMMF of the field magnet. Electric load NI represents the MMFof the armature side. When the structure and total volume aredetermined, both loads depend on the space of the total volume.

For the design of maximum thrust in the proposed model,we have determined lm to be 2.9 mm on the basis of spatiallimitations and mechanical strength in the field unit.

The key design variable in thrust design is da. da denotes thehalf slot length in the moving direction, and an armature coil iswound in da. It is related to the electric load, as expressed as (8).In (8), N(da, lc) is the number of windings per armature pole,and Aw(da, lc) is the cross-sectional area where the armaturecoil is wound. Geometrically, the larger da is, the higher thewinding turn per armature pole can be achieved. This representsthe increase of electrical load by increasing MMF

N(da, lc)I =kcpAw(da, lc)I

Sc(8)

where

Aw(da, lc) = da lc. (9)

However, the term da affects the width lz and cross sectionAa of the armature core, as expressed in (10). In (10), the largerthe value of da under the same τs, the smaller Aa(da, lt) is,which means a decrease in the amount of effective flux flowingto the armature core. Therefore, the design for maximum thrustin the designated volume should find the point where themultiplication of spatially determined effective flux and electricloads reaches the maximum value [18]

Aa(da, lt) = lz(da) lt (10)

where

lz(da) = τs − 2da. (11)

C. Total Longitudinal Force

Total longitudinal force per phase acting in moving directionconsists of cogging force and thrust [7].

Here, let us define that the maximum thrust per phase at acertain design point Ft_1phase is obtained from the differencebetween the total longitudinal force Ftotal_1phase and the cog-ging force Fc_1phase in the mover position where the maximumthrust is generated, as expressed in (12). Cogging force has beencalculated by applying the virtual displacement principle, asexpressed in (13). In (13), W is the magnetic energy stored in aphase, which is calculated as the consequence of the numericalelectromagnetic field calculation

Ft_1phase =Ttotal_1phase − Fc_1phase (12)

Fc_1phase = −dW

dz

∣∣∣∣φ=const.

(13)

Fig. 8 and Table II show the 3-D FEM results of thrust perphase under the rated condition where I = 4 A and the winding

Fig. 8. Three-dimensional FEM results of thrust.

TABLE IIWINDING TURN PER ARMATURE POLE

Fig. 9. Ratio of cogging force to thrust.

turn per armature pole. The maximum thrust is at da = 3.5 mmand approximately 27 N. Thrust decreases from da = 3.5 mm,which means decreasing flux linkage to the armature coil,regardless of spatially increased MMF on the armature side.

However, in selecting the design of a PMLSM, cogging forceis also an important aspect because large cogging force causesthrust ripple, which results in poor positioning accuracy. Thus,it is important to find the point at which large thrust and lowcogging force can be achieved. We have evaluated this aspectusing the ratio of cogging force to thrust. Fig. 9 shows the ratioof cogging force to thrust based on 3-D FEM values.

From the results, da = 2.5 mm is a good value when con-sidering a design for large thrust and low cogging force withina designated dimension. For that reason, we have decided thatda = 2.5 mm is the best point in the initial design.

VI. FUNDAMENTAL EXPERIMENT FOR THE VALIDATION

OF RESULTS IN THE DESIGN POINT

On the basis of the design point in the initial design stage, wehave fabricated the initial prototype model. In this section, theefficacy of our structural concept is experimentally validated. Inthe experiment, resistance, inductance, back EMF, thrust, andcogging force are measured and analyzed.


TABLE IIIRESISTANCE AND INDUCTANCE

Fig. 10. Prototype model and measuring equipment for back EMF.

Fig. 11. Three-dimensional FEM and empirical results of back EMF.

A. Resistance and Inductance

Table III shows the measurement results of resistance andinductance. They were measured using a digital multifunctionalmeter and LRC meter at room temperature (22.7 ◦C).

B. Back EMF

The prototype model and the equipment used to measureback EMF are shown in Fig. 10.

The back EMF is measured from the open-circuit voltage.The three-phase voltage waveforms were recorded using adigital oscilloscope (TECTONICS-TDS 3034C). Fig. 11 showsthe 3-D FEM and empirical back EMF with the mover positionat v = 1 m/s. The 3-D FEM data were calculated from fluxlinking with the armature core at no-load conditions. Thereis relatively good agreement between 3-D FEM and empiricalvalues, and both waveforms are close to sinusoidal.

C. Thrust

The equipment used to measure thrust basically consists of aload cell, a potential meter, a dc amplifier, and a data collector,

Fig. 12. Measuring equipment for thrust.

Fig. 13. Three-dimensional FEM and empirical results of thrust. (a) Thrustat I = 4 A. (b) Thrust–armature-current characteristics. (In (b), the data inparentheses in the x-axis denote the current density.)

as shown in Fig. 12. The placement of the prototype was care-fully planned to prevent measuring error. Thrust was measuredusing a load cell at different mover positions when dc currentwas applied to the armature coil. The displacement of moverwas measured using a potential meter, and the data from theload cell and potential meter were recorded in the data collector.

Fig. 13 shows the thrust under the rated condition whereI = 4 A and thrust–armature-current characteristics. There isa good agreement between 3-D FEM and empirical values.Empirical values are approximately equal to 3-D FEM values.These results prove the validity of the design point that wasobtained from the design developed using 3-D FEM.


Fig. 14. Three-dimensional FEM and empirical results of cogging force.

D. Cogging Force

In this research, the method used to measure cogging forcewas basically the same as that used for thrust, except for thearmature current. Cogging force was measured by the loadcell at different mover positions under no-load conditions.The displacement of the mover was also measured using thepotential meter, and the data from load cell and potential meterwere recorded in the data collector.

Fig. 14 shows the 3-D FEM and empirical results of coggingforce. Large cogging force per phase is significantly reduced byapplying the nine-core–eight-pole combination. The maximumvalue of total cogging force was approximately 5 N, which wasapproximately 6.4% of the thrust in the rated region.

However, considering the performance requirement for ap-plications in which a positioning accuracy of less than 5 μm isrequired, further improvement of cogging force remains to bedesired.

E. Thrust Density

Thrust density is an important aspect to evaluate. We havecalculated thrust density on the basis of the following to com-pare it with other proposals for commercial products [19], [20].To compare thrust density, rated thrust has been used. Table IVshows the thrust density in the proposed model and four othercommercial products.

1) Fvolume: thrust per total volume Vt in which the armatureside is faced with the field side.

2) Fdimension: thrust per total dimension St of active air-gaparea.

3) Fweight: thrust per total mover weight Wt.It is difficult to evaluate the superiority or inferiority of not

only the proposed model but also other proposed models onaccount of insufficient data about the measuring methods, driveconditions, current density, cooling methods, etc.

However, results from the initial prototype model, which isbased on the proposed configuration, are not in the top tiereven though they are in the range of the latest linear motorperformance. The small thrust density results from low slot fillfactor in the initial prototype model. The number of windingturns in a salient pole in the proposed model was determinedby the slot in the moving direction and space around a salientpole in the cross section in the x–y plane, as shown in Fig. 2.

TABLE IVCOMPARISON OF THRUST DENSITY

The slot in the moving direction was determined by slot pitch,and the space around a salient pole in the cross section inthe x–y plane was determined by the number of salient polesin the armature core. Compared with the slot fill factor in aslot with a moving direction of 0.75 in the initial prototypemodel, the slot fill factor in the cross section of the x–y plane isapproximately 0.16, which indicates that there is a large amountof wasted space. Therefore, for large thrust density, the optimalratio between pole pitch and the number of salient poles in thearmature core should be considered. From this result, our futurework, which will focus on structural and design improvementsfor large thrust density, has been clarified.

VII. CONCLUSION

In this paper, a transverse-flux-type cylindrical linear syn-chronous motor using generic armature cores for rotary ma-chinery to address the problem of the complex structure intransverse-flux-type topologies has been proposed. The advan-tages of the proposed configuration are the following:

1) simple structure;2) easy assembly;3) the use of lamination;4) cancellation of the strong normal attractive force between

the armature and field sides.

These advantages are helpful to facilitate the manufacturingof PMLSMs with transverse-flux-type topology.

In the initial design stage, the design for thrust and cog-ging force in an application in which large thrust and smallcogging force are required was undertaken using a magnetic-circuit method and 3-D FEM. Thrust is proportional to the


multiplication of electric and magnetic loads; therefore, a de-sign that optimizes maximum thrust in the designated volumerequires finding the point at which the multiplication of spa-tially determined effective flux and electric loads reaches themaximum value. However, in selecting the design of a PMLSM,cogging force should also be considered because large coggingforce causes thrust ripple, which results in a poor positioningaccuracy. Therefore, the point at which large thrust and smallcogging force could be achieved was selected as the criticaldesign point in the initial design stage.

The initial prototype model was fabricated to validate theresults for the selected design point and our structural concept,and a fundamental experiment was carefully conducted. A goodagreement between 3-D FEM and empirical values proved thevalidity of the design point and the effectiveness of our struc-tural concept. In addition, it was verified that a nine-core–eight-pole combination was useful for significant reduction in theproposed model.

Although it was verified that the results from the initialprototype model were close to the latest linear motor technol-ogy, improvements of thrust and cogging force remain to beachieved. Therefore, in the future, we will propose structuraland design improvements for high performance, including largethrust density and small cogging force.

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Jung-Seob Shin (M’11) received the M.S. degree inelectrical engineering from The University of Tokyo,Tokyo, Japan, in 2011, in the laboratory of Prof.T. Koseki, where he is currently working toward thePh.D. degree in electrical engineering.

His interests include linear synchronous motors,electrical machinery design, and motor control.

Mr. Shin is a member of the IEEE IndustrialElectronics Society.

Ryuji Watanabe (S’13) received the B.S. degree inelectrical engineering from The University of Tokyo,Tokyo, Japan, in 2012, where he is currently workingtoward the M.S. degree in electrical engineering.

His research interests include linear synchronousmotors, synchronous generators, and electrical ma-chinery design.

Takafumi Koseki (S’87–M’92) received the Ph.D.degree in electrical engineering from The Universityof Tokyo, Tokyo, Japan, in 1992.

He is currently an Associate Professor with theDepartment of Electrical Engineering, School ofEngineering, The University of Tokyo. His currentresearch interests include public transport systems,particularly linear drives, and the analysis and con-trol of traction systems.

Prof. Koseki is a member of the Institute of Elec-trical Engineers of Japan, Japan Society of Mechan-

ical Engineering, Japan Society of Applied Electromagnetics and Mechanics,Japan Society of Precision Engineering, and Japan Railway Electrical Engi-neering Association.

Houng-Joong Kim received the Ph.D. degreein electrical engineering from Musashi Instituteof Technology, (currently Tokyo City University),Tokyo, Japan, in 1997.

He was a Researcher with the Hitachi ResearchLaboratory, Hitachi Ltd., Hitachi, Japan, until 2009,where he was involved in linear drive research andinvented the tunnel actuator (TA). This design is usedto create the globally fastest drilling machine forprinted circuit boards. He is currently the CEO ofKOVERY Company, Ltd., Suwon, Korea. His cur-

rent research interests include linear motor designs and precise drive controlsfor industrial applications.

Dr. Kim is a member of the Korean Institute of Electrical Engineers. He wasa recipient of an R&D 100 Award for the TA in 2007.

transverse-flux-type cylindrical linear synchronous motor ... · fig. 1 [7]. for conventional...

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