soft single switched dc-dc converters with coupled inductors

8/19/2019 Soft Single Switched DC-DC Converters With Coupled Inductors

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Implementation of PWM Soft single switched DC-DC converters with coupled inductors

Page1

CHAPTER 1

INTRODUCTION

In order to reduce the size and weight of switching converters and increase power

density, a high switching frequency is required. However, in hard-switching

converters, as the switching frequency increases, switching losses and

electromagnetic interference increase. To solve this problem, soft-switching

converters are indispensable. In recent years, great amount of research is done to

develop soft-switching techniques in dc-dc converters. In these converters, it is

desirable to control the output voltage by pulse width modulation (PWM) because of

its simplicity and constant frequency. A low number of components, particularly

active components, is also desirable. Quasi-resonant converters do not have any extraswitch to provide soft-switching conditions; however, they must be controlled by the

variation of switching frequency. Furthermore, zero-voltage transition, zero-current

transition, and active clamped converters are PWM controlled but require at least

two switches, which increases the complexity of power and control circuits. PWM

soft-single-switched (SSS) converters and lossless passive snubbers, enjoy all the

mentioned advantages, usually at the cost of additional current and voltage stresses.

However, they usually have a large number of passive elements, which makes the

converter implementation difficult. The lossless passive snubber circuit is simple and

easy to implement. However, in this converter, a soft-switching condition is not

achieved for the switch turnoff instant. Furthermore, an additional diode is added in

the main power path, which would further increase the converter conduction losses. In

this project, a family of PWM SSS converters without any substantial increase in

voltage and current stresses is presented. Furthermore, in this converter family,

the number of additional components is not high. The switch in all proposed

converters is turned on under zero-current-switching (ZCS) condition and is turned

off at almost zero-voltage-switching (ZVS) condition. The converter main diode turns

on under ZVS condition and turns off under zero-voltage zero-current switching

(ZVZCS) conditions. Furthermore, an auxiliary diode turns on under ZVS condition

and turns off under ZCS condition. The proposed method can be easily applied to

single-switch converters such as buck, boost, and buck-boost. Cuk, SFP1C, and Zeta.

Furthermore, it can be applied to isolated single-switch converters such as forward,

Flyback, isolated CUK, isolated SFP1C, and isolated Zeta.


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1.1 ELECTROMAGNETIC INTERFERENCE:

As the name suggests, it is related with the disturbance caused

due to electromagnetic waves to the operation of the dc-dc. converter or any otherelectronic circuit Because of the rapid changes in voltages and currents within a

switching converter, power electronic equipment is a source of electromagnetic

interference with other equipment as well as with its own proper operation. The

electromagnetic or more devices to work together without interfering with one

another. Susceptibility is a measure of a device's sensitivity to external

interference. Because a good receiving antenna is a good transmitting antenna,

devices with susceptibility are usually devices that generate electromagnetic

interference. Government regulation is monitored through regulatory agencies. The

regulation dealing with computer generated electromagnetic interference not only

specifies the amount of electromagnetic interference that is acceptable but also the

testing methods used to measure electromagnetic interference. Testing and

approval is usually done by private companies authorized to perform testing. For there

to be an electromagnetic interference problem there must be: (1) a source of RF

power, such as a digital device, (2) coupling interference is transmitted in two forms:

radiated and conducted. The switching converters supplied by the power lines

generate conducted noise into the power lines that is usually several orders of

magnitude higher than the radiated noise into free space. Metal cabinets used for

housing power converters reduce the radiated component of the electromagnetic

interference. Conducted noise consists of two categories commonly known as the

differential mode and the common mode. The differential mode noise is a current or a

voltage measured between the lines of the source, that is, a line-to-line voltage or the

line current 1 dm. the common mode noise is a voltage or current measured between

the power lines are present in general on both the input lines and the output lines. Any

filter design has to take into account both of these modes of noise. Governments

regulate the amount of radio frequency interference that any equipment can produce

so that computers and televisions, for example, do not interfere with each other

Several terms are important in dealing with this subject. Electromagnetic interference

is, actually, the production of unwanted RF energy. Electromagnetic compatibility is

the ability of two via es and ground, such as I cm. Both differential mode and


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common mode noise a transmission line to (3) a radiating structure or antenna. Two

examples illustrate the problem:

Desktop PC:-

A new desktop personal computer is placed on a turntable in an

electromagnetic compatibility chamber which is a shielded room lined with absorbing

pyramids. Three meters away there is a broadband antenna connected to a spectrum

analyzer. As the personal computer rotates on the turntable, a technician scans across

the frequency spectrum. The tests reveal an unwanted emission at one half the clock

speeds of the Pentium microprocessor, and it appears to be worse when the back of

the personal computer faces the antenna- The personal computer fails its test. A small

probe, called a sniffer, is used to localize the source of emission. Not only does the

Pentium radiate but other devices connected to it radiate. A graphics card operating at

the Pentium clock speed also radiates and it is on the back side of the personal

computer from which radiation is strongest. The problem is resolved by rearranging

the positions of the cards in the personal computer and adding more screws to the

cover lid of the metal card box to prevent leakage.

Compact disc player on aircraft: -

It happened that the controls of a jet aircraft went haywire the reason of which

the pilot could not find out. After talking to the flight staff it was learnt that a

passenger had turned on his compact disc player. Whenever the player was on, the

display controls of the plane showed dubious signals. The compact disc player and

others like it were subjected to tests in an electromagnetic compatibility

and found to have radiation at the frequency of the air navigation system. The

players were several months old

and the aluminum case had oxidized, forming an insulating layer. Aluminum

oxide, or alumina, is a good insulator and radiation was leaking out. Although

emitting little energy, the compact disc player was at a seat next to a window and only

a couple of meters from the jet's navigation antenna. On the other hand, the navigation

beacon could he 100 km away, giving the compact disc player a

(100000/2)*(100000/2) = 25*100000000 or 94 dB advantage as an interfering source.

A better box design is required.


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1.2 GENERATION OF ELECTROMAGNETIC INTERFERENCE:

Switching waveforms are inherent in all switching converters. Because of short rise

and fall times, these waveforms contain significant energy levels at harmonic

frequencies in the radio frequency region, several orders above the fundamental

frequency. The transmission of the differential mode noise is through the input line to

the utility system and through the dc-side network to the load on the power converter.

Moreover, conduction paths through stray capacitances between components and due

to magnetic coupling between circuits must also be considered.

Fig.1.1. Block diagram of overhead line interference

Fig.1.2. Waveforms of interference pulses

The transmission of the common mode noise is entirely through parasitic or stray

capacitors and stray electric and magnetic fields. These stray capacitances exist

between various system components and between components and ground. For safety

reasons, most power electronic equipment has a grounded cabinet. The noise

appearing on the ground line contributes significantly to the electromagnetic

interference.


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CHAPTER 2SOFT SWITCHING

2.1 Introduction:

The high operating efficiency of switched mode power supplies (SMPS)

resulted in widespread acceptance in all power processing applications. The design

demand is forever moving towards higher power densities. This requirement leads to

processing of power at higher switching frequencies. In the conventional SMPS,

higher frequency of operation leads to high switching losses and associated problems.

The concept of soft switching addresses several of these problems. This chapter gives

an overview of soft switching converters.

Historically, the linear power supplies were common during the late 1950's

and 1960's. The controlling devices in these power supplies operate in the active

region. Consequently, power dissipation is large; this power loss is transformed into

heat. The efficiency is therefore poor. The power density is low. Search for higher

efficiency and power density lead to Switched Mode Power Supplies. Here, thedevices are switched either ON or OFF. The control is by pulse width modulation Soft

Switching Converters (PWM). In conventional PWM SMPS, the switches turn-on to

full current from full voltage and turn-of to full voltage from full current as shown in

fig 2.1. Such switching is referred as hard switching. The switching losses during hard

switching are considerable.

Fig.2.1. Hard Switching

An ideal switch takes a finite time to switch-on (tr) and switch-off (tf ). During the

switch-on time the device voltage is defined. During switch-off the device current is


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defined. The second quantity during switching (device current during turn-on and

device voltage during turn-off) is decided by the external circuit to the switch. By

making use of the above characteristics of the devices, attempts are made to minimize

the switching losses. Snubber circuits were used in conventional SMPS to reduce the

switching losses. Snubber circuits are effective, to a limited extent, in reducing the

device stress during the switching transitions. However, they do not reduce the

switching losses appreciably; they only shift the power loss from the switches to the

snubber resistors. Soft switching converters address these issues in a more efficient

way as explained below.

2.2 Soft Switching Converters:

Most soft switching converters rely on the process of resonance. Resonant

switching techniques reduce the switching losses to practically zero; the switching

frequency then may be increased to hundreds of kHz to achieve higher power density.

Such converters in general are classified as Soft switching converters. The

distinguishing feature of the soft switched converters is that, they switch ON and OFF

at zero current and/or zero voltage.

Soft Switching Converters have the following advantages

1. Circuit operation is possible at much higher frequencies, giving scope for reducing

the size of energy storage elements in the converter.

2. di/dt and dv/dt stresses on the switching devices are reduced; noise and interference

are reduced.

3. The parasitic elements of the circuit such as leakage inductance of transformer,

device junction capacitances etc., contribute to well defined desirable switching

transitions leading to low or no switching losses.

The switching techniques in the resonant converter employ zero voltage

switching and/or zero current switching. In zero current switching, the device turns-on

with zero current and turns off after the current drops to zero. In zero voltage

switching, the switch turns-off at zero voltage and turns-on after the device voltage

drops to zero. Fig. 2.2 illustrates the ZVS and ZCS switching trajectory.


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Fig.2.2.Switching strategy

In the ZVS converters, the voltage across the device is brought to zero by

external means, just prior to turn-on, thus eliminating the turn-on losses. During the

turn-off process, the rate of voltage raise is limited by external means (by a capacitor

in shunt with the main switch), so that the device current falls to zero before the

voltage rises substantially i.e. Assisted turn-off.

In the ZCS converters, the current through the device is brought to zero by

external means, just prior to turn-off. Thus the turn-off losses as well as the voltage

spikes due to stray inductance are totally eliminated. During the turn-on process, the

current raise is slowed down by external means (by an inductor in series with themain switch), so that the device voltage falls to zero before the current becomes

appreciable i.e. assisted turn-on.

2.3 Hard Switching and Soft Switching Techniques:

In the 1970’s, conventional PWM power converters were operated in a

switched mode operation. Power switches have to cut off the load current within the

turn-on and turn-off times under the hard switching conditions. Hard switching refers

to the stressful switching behavior of the power electronic devices. The switching

trajectory of a hard-switched power device is shown in Fig.1. During the turn-on and

turn-off processes, the power device has to withstand high voltage and current

simultaneously, resulting in high switching losses and stress. Dissipative passive

snubbers are usually added to the power circuits so that the dv/dt and di/dt of the

power devices could be reduced, and the switching loss and stress be diverted to the

passive snubber circuits. However, the switching loss is proportional to the switching

frequency, thus limiting the maximum switching frequency of the power converters.Typical converter switching frequency was limited to a few tens of kilo-Hertz


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(typically 20kHz to 50kHz) in early 1980’s. The stray inductive and capacitive

components in the power circuits and power devices still cause considerable transient

effects, which in turn give rise to electromagnetic interference (EMI) problems. Fig.2

shows ideal switching waveforms and typical practical waveforms of the switch

voltage. The transient ringing effects are major causes of EMI.

In the 1980’s, lots of research efforts were diverted towards the use of

resonant converters. The concept was to incorporate resonant tanks in the converters

to create oscillatory (usually sinusoidal) voltage and/or current waveforms so that

zero voltage switching (ZVS) or zero current switching (ZCS) conditions can be

created for the power switches. The reduction of switching loss and the continual

improvement of power switches allow the switching frequency of the resonant

converters to reach hundreds of kilo-Hertz (typically 100kHz to 500kHz).

Consequently, magnetic sizes can be reduced and the power density of the converters

increased. Various forms of resonant converters have been proposed and developed.

However, most of the resonant converters suffer several problems. When compared

with the conventional PWM converters, the resonant current and voltage of resonant

converters have high peak values, leading to higher conduction loss and higher V and

I ratings requirements for the power devices. Also, many resonant converters require

frequency modulation (FM) for output regulation. Variable switching frequency

operation makes the filter design and control more complicated.

In late 1980’s and throughout 1990’s, further improvements have been made in

converter technology. New generations of soft-switched converters that combine the

advantages of conventional PWM converters and resonant converters have been

developed. These soft-switched converters have switching waveforms similar to those

of conventional PWM converters except that the rising and falling edges of the

waveforms are smoothed with no transient spikes. Unlike the resonant converters,

new soft-switched converters usually utilize the resonance in a controlled manner.

Resonance is allowed to occur just before and during the turn-on and turn-off

processes so as to create ZVS and ZCS conditions. Other than that, they behave just

like conventional PWM converters. With simple modifications, many customized

control integrated control (IC) circuits designed for conventional converters can be

employed for soft-switched converters. Because the switching loss and stress have

been reduced, soft-switched converter can be operated at the very high frequency

(typically 500kHz to a few Mega-Hertz). Soft- switching converters also provide an


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effective solution to suppress EMI and have been applied to DC-DC, AC-DC and DC-

AC converters. This chapter covers the basic technology of resonant and soft-

switching converters. Various forms of soft-switching techniques such as ZVS, ZCS,

voltage clamping, zero transition methods etc. are addressed. The emphasis is placed

on the basic operating principle and practicality of the converters without using much

mathematical analysis.

I

VOff

On

Soft-switching

Hard-switching

Safe Operating Area

snubbered

Fig.2.3.Typical switching trajectories of power switches

Fig.2.4.Typical switching waveforms of (a) hard-switched and (b) soft-switched

devices


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2.4 Resonant Switch

Prior to the availability of fully controllable power switches, thyristors were

the major power devices used in power electronic circuits. Each thyristor requires a

commutation circuit, which usually consists of a LC resonant circuit, for forcing the

current to zero in the turn-off process. This mechanism is in fact a type of zero-current

turn-off process. With the recent advancement in semiconductor technology, the

voltage and current handling capability, and the switching speed of fully controllable

switches have significantly been improved. In many high power applications,

controllable switches such as GTOs and IGBTs have replaced thyristors. However,

the use of resonant circuit for achieving zero-current-switching (ZCS) and/or zero-

voltage-switching (ZVS) has also emerged as a new technology for power converters.

The concept of resonant switch that replaces conventional power switch is introduced

in this section.

A resonant switch is a sub-circuit comprising a semiconductor switch S and

resonant elements, Lr and C r . The switch S can be implemented by a unidirectional or

bidirectional switch, which determines the operation mode of the resonant switch.

Two types of resonant switches, including zero-current (ZC) resonant switch and

zero-voltage (ZV) resonant switches, are shown in Fig.3 and Fig.4, respectively.

Lr

CrS

(a)

Lr

CrS

(b)

Fig.2.5.Zero-current (ZC) resonant switch.

Lr

S

(a)

Cr

Lr

CrS

(b)

Fig.2.6.Zero-voltage (ZV) resonant switch.


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2.5 ZC resonant switch

In a ZC resonant switch, an inductor Lr is connected in series with a power

switch S in order to achieve zero-current-switching (ZCS). If the switch S is a

unidirectional switch, the switch current is allowed to resonate in the positive half

cycle only. The resonant switch is said to operate in half-wave mode. If a diode is

connected in anti- parallel with the unidirectional switch, the switch current can flow

in both directions. In this case, the resonant switch can operate in full-wave mode. At

turn-on, the switch current will rise slowly from zero. It will then oscillate, because of

the resonance between Lr and C r . Finally, the switch can be commutated at the next

zero current duration. The objective of this type of switch is to shape the switch

current waveform during conduction time in order to create a zero-current condition

for the switch to turn off.

2.5.1 ZV resonant switch

In a ZV resonant switch, a capacitor Cr is connected in parallel with the switch

S for achieving zero-voltage-switching (ZVS). If the switch S is a unidirectional

switch, the voltage across the capacitor Cr can oscillate freely in both positive and

negative half-cycle. Thus, the resonant switch can operate in full-wave mode. If a

diode is connected in anti-parallel with the unidirectional switch, the resonant

capacitor voltage is clamped by the diode to zero during the negative half-cycle. The

resonant switch will then operate in half-wave mode. The objective of a ZV switch is

to use the resonant circuit to shape the switch voltage waveform during the off time in

order to create a zero-voltage condition for the switch to turn on.

2.5.2 ZCS-QRCs

A ZCS-QRC designed for half-wave operation is illustrated with a buck type

dc-dc converter. The schematic is shown in Fig.5(a). It is formed by replacing the

power switch in conventional PWM buck converter with the ZC resonant switch in

Fig.3(a). The circuit waveforms in steady state are shown in Fig.5(b). The output filter

inductor L f is sufficiently large so that its current is approximately constant. Prior to

turning the switch on, the output current I o freewheels through the output diode D f .

The resonant capacitor voltage V Cr equals zero. At t 0, the switch is turned on with

ZCS. A quasi-sinusoidal current I S flows through Lr and C r , the output filter, and the

load. S is then softly commutated at t 2 with ZCS again. During and after the gate

pulse, the resonant capacitor voltage V Cr rises and then decays at a rate depending on

the output current. Output voltage regulation is achieved by controlling the switching


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frequency. Operation and characteristics of the converter depend mainly on the design

of the resonant circuit Lr - C r . The following parameters are defined: voltage

conversion ratio M , characteristic impedance Z r , resonant frequency f r , normalized

load resistance r , normalized switching frequency γ .

i

o

V

V M = (1a)

r

r r

C

L Z = (1b)

r r

r

C L f

π=

2

1 (1c)

r

L

Z

Rr = (1d)

r

s

f

f =γ (1e)

It can be seen from the waveforms that if I o > V i / Z r , I S will not come back to zero

naturally and the switch will have to be forced off, thus resulting in turn-off losses.

The relationships between M and γ at different r are shown in Fig.5(c). It can

be seen that M is sensitive to the load variation. At light load conditions, the unused

energy is stored in C r , leading to an increase in the output voltage. Thus, the switching

frequency has to be controlled, in order to regulate the output voltage.

Vi

S CR1

Df

Lr Lf

Cr Cf RLVoV

Cr

Io

iLr

Fig.2.7a.Schematic diagram.


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gate signal

to S

ILr

VDS

VCr

Tt1

t0

Vi /Z

r

IO

Vi

Vi

Fig.2.7b.Circuits waveforms.

0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1

0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1

γ

M0.5

1

r =2510

Fig.2.7c.Relationship between M and .

Fig.2.7c.Half-wave, quasi-resonant buck converter with ZCS

If an anti-parallel diode is connected across the switch, the converter will be operating

in full-wave mode. The circuit schematic is shown in Fig.6(a). The circuit waveforms

in steady state are shown in Fig.6(b). The operation is similar to the one in half-wave

mode. However, the inductor current is allowed to reverse through the anti-parallel

diode and the duration for the resonant stage is lengthened. This permits excess

energy in the resonant circuit at light loads to be transferred back to the voltage source

Vi. This significantly reduces the dependence of Vo on the output load. The


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relationships between M and γ at different r are shown in Fig.6(c). It can be seen that

M is insensitive to load variation.

Vi

S

Df

Lr Lf

Cr Cf RL

iLr

VoVCr

Io

Fig.2.8a.Schematic diagram.

VDS

Tt0

gate signal

to S

ILr

VCr

t1

Vi /Z

r

IO

Fig.2.8b.Circuit waveforms.

0

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1

0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1

γ

M

r =1-10

Fig.2.8c.Relationship between M and .

Fig.2.8.Full-wave, quasi-resonant buck converter with ZCS

By replacing the switch in the conventional converters, a family of QRC with ZCS is

shown in Fig.7.


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2.6 Comparisons between ZCS and ZVS

ZCS can eliminate the switching losses at turn-off and reduce the switching losses at

turn-on. As a relatively large capacitor is connected across the output diode during

resonance, the converter operation becomes insensitive to the diode’s junction

capacitance. The major limitations associated with ZCS when power mosfets are used

are the capacitive turn-on losses. Thus, the switching loss is proportional to the

switching frequency. During turn-on, considerable rate of change of voltage can be

coupled to the gate drive circuit through the Miller capacitor, thus increasing

switching loss and noise. Another limitation is that the switches are under high current

stress, resulting in high conduction loss.

BUCK

S1

D1 L1

C1

S1

D1 L1

C1

BOOST

C1

L1

D1

BUCK/

BOOST

L1

C1D1

S1

D1 L1

C1

S1

D1 L1

C1

CUK

C1

L1

D1

L1

C1D1

SEPIC

C1

L1

D1

L1

C1D1


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FLYBACK

C1

L1

D1

L1

D1

C1

FORWARD L1

D1

C1

C1

L1

D1

Fig.2.9.A family of quasi-resonant converter with ZCS

capacitance. The major limitations associated with ZCS when power mosfets

are used are the capacitive turn-on losses. Thus, the switching loss is proportional tothe switching frequency. During turn-on, considerable rate of change of voltage can

be coupled to the gate drive circuit through the Miller capacitor, thus increasing

switching.

It should be noted that ZCS is particularly effective in reducing switching loss for

power devices (such as IGBT) with large tail current in the turn-off process.

ZVS eliminates the capacitive turn-on loss. It is suitable for high-frequency

operation. For single-ended configuration, the switches could suffer from excessivevoltage stress, which is proportional to the load. It will be shown in Section 15.5 that

the maximum voltage across switches in half-bridge and full-bridge configurations is

clamped to the input voltage.

For both ZCS and ZVS, output regulation of the resonant converters can be

achieved by variable frequency control. ZCS operates with constant on-time control,

while ZVS operates with constant off-time control. With a wide input and load range,

both techniques have to operate with a wide switching frequency range, making it not

easy to design resonant converters optimally.


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Fig.2.10.Block Diagram

DC source:

It is the first stage of this project. So it is give the DC supply to Inverter. The DC

source may be Battery or fuel cell or rectified from AC source

Soft single switched Boost Converter:

Boost converter is used to convert low voltage dc to high voltage dc .switching

operate in high frequency. High switching frequency generates EMI and high voltage

stress. This voltage stress should be reduced using soft switch technique .the auxiliary

circuit is used to achieve soft switching

Filter.

Rectifier converts AC to DC. This output has ripples. It is filtered with a help of LC

filters.

Load:

The output has DC output voltage. It is used to run the motor, battery charging, and

telecommunication applications.

Micro controller:

Micro controller is used to generate triggering pulse for mosfets. It is used to control

the outputs. Micro controller have more advantage compare then analog circuits and

micro processor such as fast response, low cost, small size and etc.

Driver:

It is also called as power amplifier because it is used to amplify- the pulse output from

micro controller. It is also called as opto coupler IC. It provides isolation between

microcontroller and power circuits.

Regulated Power supply (RPS):

RPS give 5V supply for micro controller and 12V supply for driver. It is converted

from AC supply. AC supply is step down using step down transformer


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2.7 MODES OF OPERATION:

The proposed converter has six distinct operating modes in a switching cycle.

Mode 1 [t0—t1]:

Before the first mode, it is assumed that the main switch is off and I in flows

through the main diode Do , and thus, the Cr voltage is equal to Vo. At t0, the switch

is turned on under ZCS condition due to series inductor Lrl.

ILr1 (t) = (Vo/Lr1) (t-to) … (2.1)

∆t1=t1-to = (Lr1 Iin)/Vo ... (2.2)

Fig.2.11.Mode 1

Mode2 [t1-t2]:

At tl, the Lr1 current has reached Iin, and the diode Do current has reached

zero. Thus, the diode do turnoff is under ZCS. In addition, due to the existence of Cr

and based on (3), the Do voltage rises slowly and is considered ZVS. Consequently,

the Do turnoff is ZVZCS. In this mode, Lrl starts to resonate with Cr. This mode ends

when the Cr voltage reaches zero.

))1(cos()( t t r Vot Vcr −= ω … (2.3)

))1(sin()(1 t t r Zr

Vo Iint I

lr −+= ω

Cr Lr

fr r

1

12 =Π=ω

Cr Lr Zr / 1=

r t t t ω 2 / 122 Π=−=∆


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Fig.2.12.Mode 2

Mode 3 [t2-t3]:

When Vcr reaches zero, diode D1 starts to conduct under ZVS condition, and

the Cr voltage is clamped at zero. Since the total ampere turns of Lr2 and Lrl should

stay constant. Furthermore, the Lr1 current is equal to the sum of the input current and

the Lrl current. The interval of this mode and the previous mode is effectively the

converter duty cycle. This mode ends when the switch is turned off.

211)( 21 N I N I N Zr

Vo Iin

Lr Lr +=+

Zr n

Vo Iin Ilr

)1(1 ++=

Zr n

Vo Ilr

)1(2

+=

)21(3 t t DTst ∆+∆−=∆

Fig2.13.Mode 3

Mode 4 [t3 — t4]:

By turning the switch off, the ampere turn of Lr1 is transferred to Lr2 , and now,

the Lr2 ampere turn is the sum of its previous ampere turn plus the Lr1 ampere turn as


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described by

2121 21 N I N I N I Lr Lr =+

where I Lr1 and I Lr2 are the values of the coupled inductor currents in the previous modeand I 1 is the Lr 2 current at t 3. Thus, by substituting (8) and (9) in (11), the following is

obtained

)( / 1)3(21 Zr

Vo Iinnt Ilr I +==

The resonant capacitor charges by Lm plus the Lr2 current until its voltage reaches V 0.

The switch voltage, C r voltage, and Lr2 current during this mode are

))3(1

sin()1()1()( t t r

n

I Iin Zr nt Vsw −++= ω

))3(1

sin()1()( t t r n

I IinnZr t Vcr −+= ω

Iint t r n

I Iint ILr −−+= ))3(1

cos()1()(2 ω

It can be observed from (13) that the switch is turned off under ZVS condition at the

beginning of this mode. However, in practice, due to the small leakage inductance of

the coupled inductors, a small voltage spike appears across the switch, and then, the

switch voltage rises slowly to its final value. Thus, actually, the switch is turned off

under almost ZVS condition even though the spike peak is usually much smaller than

the switch maximum voltage.

At t 4, V cr reaches V o; thus, the duration of this mode and the maximum voltage stress of

the switch are

))1(

(sin344 1

ZrIinnVo

Vo

r

nt t t

++=−=∆ −

ω

Von

t VswVsw )1

1()4(max +==


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Fig.2.14.Mode 4

Mode 5 [t4-t5]:

This mode begins when the Cr voltage reaches Vo and the diode Do turns on

under ZVS condition. In this mode, the Lrl current decreases linearly. At t5, the Lr2

current reaches zero, and the diode D1 turns off under ZCS condition.

Iin Iin Zr n

Von Iin

n

n I −

++

+=

2

22 )1(2)1

(2

)4(2

2)(2 t t Lr

Vo I t ILr −−=

Vo

I Lr t t t

22455 =−=∆

Fig.2.15. Mode 5

Mode 6 [t5-t6]:

In this mode, /in freewheels through the diode Do, and the current through the

resonant inductors remains zero and the voltage across the resonant capacitor stays at

Vo. The converter operations in continuous conduction mode (CCM) and

discontinuous conduction mode (DCM) are similar. However, CCM is preferred since


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the energy stored in the leakage inductance of the coupled inductors in this condition

is less than that in DCM.

)54()1(6 t t Ts Dt ∆+∆−−=∆

Fig.2.16.Mode 6

Conventional Circuit Drawbacks:

• More switching loss

• More EMI

• High voltage stress and current stress

• Required more components for achieving soft switching

• Complex control circuit.

Advantages:

• Less switching losses due to soft switching

• 1ess voltage and current stress

• No need for additional switches for soft switching

• High step up voltage

• Simple control method

• Better voltage control due to PWM technology

Applications:

• Speed control of DC motor

• Power supply for DC drives

• Battery charging

• Back up for server system.

2.8 PWM DC-DC CONVERTERS

2.8.1 POWER STAGES OF THE PWM DC-DC CONVERTERS

The full-bridge and half-bridge converters is shown in fig.2.9 and fig.2.10 are


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mostly used in high power applications.

Fig.2.17. Full-Bridge Converter

Fig. 2.18. Half-Bridge Converter

Pulses of opposite polarity are produced on the primary and secondary windings of

the transformer by switching of the transistor.

In a connection with the half–bridge inverter, the capacitors Cd1 and Cd2 establish a

voltage midpoint between zero and the input dc voltage. The input voltage is equally

divided between the capacitors. The relationship between the input and output voltage

for the half–bridge is

2

1

.o

d

V N D

V N =

And for the full bridge is

2

1

2 .o

d

V N D

V N =

Where duty cycle ison

D T T = and 0


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Comparison of the full–bridge (FB) converter with the half–bridge (HB) converter for

identical input and output voltages and power ratings requires the following turn’s

ratio:

2 2

1 1

2 HB FB

N N N N

=

Neglecting the ripple in the current through the filter inductor at the output and

assuming the transformer magnetizing current to be negligible in both circuits, the

transistor currents IC are given by

( ) 2( )C HB C FB I I =

2.8.2 PWM STRATEGIES FOR FULL-BRIDGE CONVERTER

The conventional control diagram used for hard driven converters is shown in

Fig.2.11. The transistors (T1, T2) and (T3, T4) are switched as pairs alternatively at

the selected switching frequency, which alternately places the transformer primary

across the input supply U for same interval ton. The maximum duty cycle is

50%(D=0.5).

A disadvantage of this switching mode is that when all four switches are turned off,

the energy stored in the leakage inductance of the power transformer causes severe

ringing with junction capacitance of switching devices.

Fig.2.19. Waveforms Of Hard Switching Converter With Conventional PWM

The control scheme in Fig. 2.12 is almost the same as previous except that the duty

cycle in one leg (transistors T1, T4) is constant (D=0.5) and in second leg (transistors

T2, T3) is variable in a range between zero and 50%.


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Fig.2.20. waveforms of converter with modified PWM control

Both legs (transistors T1, T4 and T2, T3) of the bridge operate with a 50% duty cycle,

and the phase shift between the legs is controlled. When the two legs operate in phase,

the differential voltage applied to the transformer is zero, and zero DC output voltage

is obtained. When the two legs of the bridge are in opposite phase, the differential

voltage applied to the transformer, and also the output voltage is maximal.

Fig.2.21. waveforms of converter with phase-shifted PWM control

The phase shift pulse width modulation (PS-PWM) leads to asymmetrical switching

waveforms. The leading leg consists of transistors T1, T4 and the lagging leg consists

of transistors T2, T3. The transistor currents in these legs are not symmetrical. The

PS-PWM control strategy leads to zero-voltage turn-on of the transistors in both of

legs, as it is evident from oscillograms as shown in fig. 2.14 and 2.15


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.

Fig.2.22. Oscillogram

Transistor voltage uCE1 and current iC1 in the leading leg at turn–on and turn–off

Fig.2.23. Oscillogram

Transistor voltage uCE2 and current iC2 in the lagging leg at turn–on and turn–off

2.9. Soft Switching PWM Converters

The soft switching PWM converter is defined here as the combination of converter

topologies and switching strategies that result in zero–voltage and/or zero–current

switching. They are called also pseudo–resonant, quasi-resonant, resonant transition,

clamped voltage topologies and other. In these converters the resonant transition is

employed only during a short switching interval. The output voltage is usually

controlled by PWM with constant switching frequency.

Soft switching PWM converters can be classified as follows:

1 ZVS PWM converters

2 ZCS PWM converters

3 ZVS ZCS PWM converters


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2.9.1 ZVS PWM Converters

The simplest ZVS PWM full bridge converter is shown in Fig. 2.16. The converter

snubbers consist of capacitances C1–C4 and inductance LR, which are represented by

transistor and diode output capacitances and transformer leakage inductance

respectively.

Fig.2.24. Full-Bridge ZVS PWM converter

The transistors (MOSFETs or IGBTs) in leading or lagging leg are turned–on while

their respective anti-parallel diodes conduct. Since the transistor voltage is zero during

the entire turn-on transition, switching loss does not occur at turn–on.

Fig.2.25. Switch (transistor MOSFET T1 and its body diode D1) voltage Uds1 and

current Ids1 during turn–on and

turn–off (leading leg)


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Fig.2.26.Switch (transistor MOSFET T2 and its body diode D2) voltage Uds2 and

currentI2 during turn–on and turn–off (lagging leg)

By utilising small snubber capacitors C1 – C4 the turn–off losses are sufficiently

reduced. If the transistor turn–off time is sufficiently fast, then the transistor is

switched fully off before the collector voltage rises significantly above zero, and thus

negligible turn–off switching loss is incurred. The ZVS converter exhibits low

primary–side switching loss and generated EMI. However, conduction losses are

increased with respect to an ideal hard switching PWM full bridge topology.

At light load, the leakage inductance energy is not sufficient to ensure zero–

voltage switching in the lagging leg of the converter. This critical load condition is

also a function of the line condition. The worst case is high input voltage when more

capacitive energy is required. Another consideration is the delay time from the turn–

off T4 until the turn–on of T1 and vice versa. If the delay time td is too short, then the

device capacitance may not be fully discharged. However, if the delay time td2 is too

long, the capacitor voltage will peak, continue to resonate and drop. Fortunately, the

time of peak charge is relative independent of the input voltage and load condition

and is equal to one quarter of LRC time constant


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Fig.2.27.Transistor voltage Uce1 and current Ic1 at turn–off –detail

The secondary–side diodes switch at zero current. This leads to switching losses and

ringing as a result of interaction of diodes capacitance with the leakage inductance of

the transformer. Additional snubber circuitry is usually required, for prevention of

excessive diode voltage stress. To remove the above-mentioned disadvantages a lot of

derivations of the ZVS PWM converters were developed. The penalty for the

improvement is usually higher complexity of the converter topology.

2.10 ZCS PWM Converters

The ZCS PWM converters can be derived from the ZVS PWM converters by

applying the duality principle.

Fig.2.28. Basic circuit diagram of the FB ZCS-PWM converter

The transformer leakage inductance, the rectifier’s junction capacitances, and the

transformer winding capacitances can be utilised in this circuit.Similar to the FB–

ZVS–PWM converter, the FB–ZCS–PWM converter also uses phase–shift control at


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constant switching frequency to achieve required converter operation.

Fig.2.29.Idealised waveforms of the FB ZCS PWM converter

The switches must have reverse-voltage blocking capability. The switch can be

implemented by an IGBT or a MOSFET in series with a reverse blocking diode, an

IGBT with reverse–voltage blocking capability, a MCT, or a GTO. An important

advantage of the circuit is that the rectifier diodes do not suffer from reverse recovery

problem since they commutate with zero–voltage switching. This feature makes the

converter attractive for applications with high output voltage e. g. power factor

correction circuits, where the rectifiers suffer from severe reverse–recovery problems

when conventional PWM, ZVS–QRC, or ZVS– PWM converter techniques are used.

The efficiency of the converter drops significantly at low line and heavy load since

the switches begin to lose zero current switching.

Fig.2.30.Switch S1 voltage and current during turn-on and turn-off


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Fig.2.31.Switch S3 voltage and current during turn-on and turn-off

Table 2.1 some dual characteristics of the FB ZVS PWM converter and FB ZCS

PWM converter

2.11 Zero-Voltage Zero-Current Switching PWM Converters

The operating frequency of IGBTs is normally limited to 20-30 kHz because of their

current tailing problem. To operate IGBTs at higher switching frequencies, it is

required to reduce the turn-off switching losses. ZVS with substantial external

capacitor or ZCS can be a solution. The ZCS, however, is deemed more effective

since the minority carriers are swept out before turning off.

The zero-voltage zero-current switching (ZVZCS) PWM converters are derived from

the full-bridge phase-shifted zero-voltage (FB-PS-ZVS) PWM converters. The PS-

ZVS PWM converter is often used in many applications because this topology permits

all switching devices to operate under zero-voltage switching by using circuit

parasitics such a transformer leakage inductance and devices junction

capacitance.


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However, because of phase-shifted PWM control, the converter has a disadvantage

that circulating current flows through a transformer and switching devices during

freewheeling intervals.

The circulating current is a sum of the reflected output current and transformer

primary magnetizing current. Due to circulating current, RMS current stresses of the

transformer and switching devices are still high compared with that of the

conventional hard-switching PWM full-bridge converter . To decrease the circulating

current to zero and thus to achieve zero-current switching for lagging leg, various

snubbers and/or clamps connected mostly at secondary side oftransformer are applied.

Fig.2.32.Principle of the ZVZCS Converter Operation

Hence, the converter achieves nearly zero-current switching for the lagging leg

(transistors T2, T3) due to minimised circulating current during interval of lagging leg

transition and achieves zero-voltage switching for leading leg (transistors T1 and T3).

An example of ZVZCS PWM converter is shown in Fig.2.27. ZVS of the leading leg

is achieved by the same manner as that of the ZVS full-bridge PWM converter, while

the ZCS of the lagging leg is achieved by resetting the primary current during

freewheeling period by using active clamp in the secondary side, which needs an

additional active switch. Oscillogram of the collector-emitter voltage uCE2 and

collector current iC2 in the lagging leg at turn–on and turn–off . The transistor is

turned-on at zero voltage and turned-off at zerocurrent. The circulating current does

not occur, only negligible magnetizing current flows during freewheeling interval

through primary winding of transformer. This combination of switching is very

effective for IGBT transistors, which have problems at turn-off due to tail current

effect. The converter is operating very well at nominal load, but it is not capable

operating over wide load range (from no-load conditions to short circuit) with zero-voltage or zero-current switching for all power switches.


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secondary winding of the auxiliary transformer TR2. Soft switching and reduction of

circulating currents for full load range are achieved in this converter. The converter is

especially suited for application where short circuit and no-load are normal states of

the converter operation, e.g. arc welding.


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CHAPTER 3

ZERO VOLTAGE SWITCHING PWM CONVERTERS

3.1INTRODUCTION

As one of the typical resonant- type ZVS techniques, the ZVS-QRC technique

eliminates the capacitive turn on loss which plagues zcs-QRCS and pwm converters.

the drain-to-source voltage of the power MOSFET in a ZVS-QRC is shaped to zero

prior to turn on, thus eliminating turn-on switching loss and the miller effect. In

addition the active switch in a ZVS-QRC technique for high frequency conversion

where MOSFETS are employed. however,the ZVS-QRC technique has several

limitations first,the power switch in a single-ended ZVS-QRC suffers from a high

voltage stress which is proportional to the load range. Using the buck ZVS-QRC as an

example, for a 10% to 100% load range, the peak voltage stress of the power switch

can be 11 times the input voltage. Therefore, a high voltage MOSFET accompanied

by on resistance and large input capacitance has to be used, resulting in a substantial

increase in conduction losses and the gate drive loss. Second, a wide switching

frequency range is required for ZVS-QRC to operate with a wide input voltage and

load range. The wide frequency range makes optimization of the power transformer,

input and output filters, control circuit, and gate drive circuit difficult. For example, to

decrease the conduction loss power MOSFET’s with a low on resistance are

preferred. However, MOSFET’S with low on resistance are accompained by large

input capacitances which can cause significant drive losses at high frequency

operation especially at high line and light load.

Another n limitation of the ZVS-QRC technique is severe parasitic ringing

between the resonant inductor and the diode junction capacitance. Due to the presence

of the large resonant inductor, this parasitic ringing is enhanced as compared to its

PWM counterpart. In a practical circuit, the severe parasitic ringing not only increases

switching loss and switching loss and switching noise, but may result in possible

instability in the closed-loop system.

In principle, the voltage stress of the power switch in a ZVS-QRC can be reduced

at the expense of a partial loss of ZVS at light load. This may not cause a thermal

problem, since the switch conduction loss is low at light load. In a real circuit,

however to operate a ZVS-QRC in an adequate frequency range, an external capacitor


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as part of the resonant capacitor usually needs to be placed in parallel with the power

switch. In this case, the partial loss of the ZVS at light load may not be allowed,

considering both high switching loss and high switching noise. In particular, the

switching frequency increases as load current decreases, thus the capacitive turn-on

loss can easily become intolerable at light load. With a wide load range, optimization

of ZVS-QRCs is very difficult to achieve. Employing an auxiliary switch across the

resonant inductor in a ZVS-QRC allows the new converter to operate with much

reduced circulating energy and with a constant frequency. It is also shown that the use

of a saturable inductor can further improve the performance of the proposed ZVS-

PWM converters.

3.2 A Family Of ZVS-PWM Converters

3.2.1 ZVS-PWM Switching Cell

The concept of ZVS quasi-resonant switch was introduced to perform a systematic

analysis of topologies and features of the ZVS_QRC’s. By incorporating the PWM

switching cell concept, a ZVS quasi-resonant switching cell can be derived. In this

figure, S is the active switch, D is the rectifying diode, Lf is the energy storage

inductor, Cr is the resonant capacitor, and Lr is the resonant inductor. To achieve ZVS,

the off-time of the power switch is fixed. The output voltage is regulated by varying

the on-time of the switch. By adding an auxiliary switch(S1) across the resonant

inductor, the ZVS-PWM switching cell shown in fig.3.2(b) is obtained. This auxiliary

switch makes the off-time of the power switch(S) controllable. It enables the

converter to regulate the output while operating at a fixed switching frequency.

In the ZVS quasi-resonant switching cell, the resonant inductor begins to oscillate

with the resonant capacitor after the power switch is turned off. The power switch is

turned on with ZVS after the resonance bringsr

C voltage zero. The off-time of the

power switch is determined by the resonant period of the resonant components. Thus

a ZVS-QRC operates with constant off-time control. Consequently, a ZVS-QRC

operating with a wide input voltage or load range has a wide frequency range.

In fig.3.1(b), S1 is turned on before the power switch is turned off. When the

power switch is turned off, the resonant inductor current freewheels through S1 for a

period of time. During this freewheel time , the energy stored in the resonant inductor

remains unchanged until S1 is turned off, when the resonant inductor begins to

oscillate with resonant capacitor. The power switch is turned on after the resonance


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brings the capacitor voltage to zero. By controlling the time interval of the

freewheeling stage, the off-time of the power switch can be varied, enabling the

converter to operate with a fixed frequency.

Fig.3.1(a) ZVS quasi-resonant switching cell

Fig.3.1(b) ZVS-PWM Switching cellTo introduce the principle of operation of the ZVS-PWM converters, the buck ZVS-

PWM converter is used as an example. The circuit schematic and key waveforms of

the buck ZVS-PWM converter are shown in fig.3.2. It can be seen that the new circuit

differs from a buck ZVS-QRC by an additional auxiliary switch placed in parallel

with the resonant inductor. The output filter inductor is considered as a current source

O I in the analysis. As shown in fig.3.3 five topological stages exist with in one

switching cycle.

(1)T0-T1: Before time To, the power switch S is conducting, and the rectifier diode D

is off. At time To, S is turned off. The freewheeling diode D is off, and the resonant

inductor current remains at Io value during this interval. The resonant capacitor is

charged linearly by Io until its voltage reaches the input voltage. The equivalent

circuit of this topological stage is shown in fig.3.3(a).This time interval is given by:

1r i

o

o

C V T

I ∆ = (3.1)


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(2)T1-T2:At time T1, D starts to conduct. The r L current still remains constant by

circulating through the auxiliary switch S1. Therefore, the energy stored in the

resonant inductor (which is used to achieve ZVS for S) stays unchanged.

(3)T2-T3: At time T2, S1 is turned off, and the resonance between r L and r C begins.

This interval lasts until T3, when resonance brigs Cr V to zero and the antiparallel

diode of S starts to conduct. This time period is approximately three quarters of the

resonant period,i.e.,

23

3

4r r

T L C ∆ = (3.2)

(4)T3-T4: S is turned on with ZVS during this time interval. The r L current increases

linearly while the diode D current decreases. At T4, diode D is turned off with ZCS.

(5)T4-T0: S1 is turned on with ZVS before S is turned off. This interval lasts until t0,

when S is turned off and the cycle is turned off.

Fig.3.2(a) Buck ZVS-PWM converter


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Fig.3.2(b)Buck ZVS-PWM converter waveforms


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Fig.3.3 Equivalent Circuits for five operation stages

From the above description, it can be seen that the operation of th ZVs siS-PWM buck

converter differs from that of the buck ZVS-QRC by possessing an extra freewheeling

stage (T1-T2) during which the resonant inductor current flows through S1 and

remains constant. Constant-Frequency operation is achieved by controlling this

freewheeling diode time interval(T1-T2). Furthermore, the resonant interval (T2-T3)

can be relatively short with respect to the switching period. In this way, the operation

of the proposed circuit is similar o that of the conventional PWM converter during

most portions of a switching cycle. Compared to a ZVS-QRC, the size of the resonantcomponents becomes much smaller, an circulating energy of the circuit is much


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reduced.

The design strategy for the proposed buck ZVS-PWM converter is quite different

from that of the buck ZVS-QRC. To limit the switch stress voltage, the circuit can be

designed to operate with ZVS only relatively heavy load (e.g., above 50 % load).

Thus the maximum voltage stress of the active switch is approximately three times the

input voltage at full load. At light load, ZVS is partially lost. This does not cause a

thermal problem since the conduction loss is generated.


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Chapter 4

Zero- current Switching PWM converters

4.1 IntroductionDue to continuous improvement of switching characteristics, lower conduction losses,

and lower cost, IGBT’s are gaining wide acceptance in switched-mode power

converters/inverters. since IGBT is a minority- carrier device, it exhibits a current tail

at turnoff, which causes considerably high turn-off switching losses. To operate

IGBTs at relatively high switching frequencies, either the ZVS or the ZCS technique

can be employed to reduce switching losses. Basically, the ZVS eliminates the

capacitive turn-on loss and reduces the turn-off switching loss by slowing down the

voltage rise and thereby reducing the overlap between the switch voltage and the

switch current. This technique can be effe, tive when applied to a fast IGBT with a

relatively small current tail.

For slow IGBTs, however, a large external resonant capacitor is required to

reduce the turn-off switching loss effectively. But this may not be tolerable from the

circuit point of view because of topology and design constraints. The ZCS technique

eliminates the voltage and current overlap by forcing the switch current to zero before

the switch voltage rises. Thus ZCS is deemed more effective than ZVS in reducing

IGBT switching losses, particularly for slow devices.

Compared to the ZVS converter topologies the ZCS converter topologies are

less mature. For high- frequency power conversion, the ZCS-QRC technique is most

frequently used. This technique offers ZCS power transistor and ZVS for the rectifier

diode. The diode junction capacitance and the transformer leakage inductance are

utilized to achieve soft-switching.

One of the major limitations of the ZCS-QRC technique is high circulating

energy caused by the resonant inductor which is in series with the power transistor.

As a result, the power switch suffers from a high current stress, and the rectifier from

a high voltage stress. The second limitation is severe parasitic ringing on the power

switch. Since the output capacitance of the power switch is not utilized, it oscillates

with the resonant inductor when the switch is turned off. This low frequency parasitic

ringing not only causes significant switching loss and noise, but also increases the

voltage stress of the power switch. The third limitation of the ZCS-QRC technique is


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capacitive turn-on loss and noise of the power switch. The capacitive turn-on loss

problem may not be severe for power converters using IGBTs or BJTs, since these

devices have relatively low output capacitance, and the operating frequency is also

relatively low. For high frequency power conversion in which power MOSFETs are

used as the power switch, however, this capacitive turn-on loss can be significant.

Another limitation is variable frequency operation, since the ZCS-QRCs operate with

constant on-time control.

This chapter presents a family of ZCS-PWM converters. Employing an

auxiliary switch in series with the resonant capacitor in a ZCS-QRC allows the new

converter to operate with constant frequency and much reduced circulating energy.

The ZCS-PWM converters can also be derived by simply applying circuit duality to

the ZVS-PWM converters. The ZCS-PWM technique is an extension of the ZCS-

QRC technique.

4.2 A Family of ZCS-PWM Converters

4.2.1. ZCS-PWM Switch

Figure 4.1(a) shows the basic configuration of the ZCS quasi-resonant

switching cell, where Lr and Cr are the resonant inductor and resonant capacitor,

respectively. In the ZCS quasi-resonant switching cell, the resonant inductor begins to

oscillate with the resonant capacitor when the power switch is turned on. The power

switch is turned off with ZCS after the resonance brings switch current to zero. To

achieve ZCS for the power switch, the on-time of the power switch, which is

determined by the resonant period of the resonant tank, is fixed. The output voltage is

regulated by varying the off-time of the switch. Thus a ZCS-QRC operates with

constant on-time control. Consequently, a ZCS-QRC operating with a wide input

voltage or load range has a wide frequency range.

By inserting an auxiliary switch (S1) in series with the resonant capacitor, the

ZCS-PWM switching cell is shown in fig.4.1(b) is obtained. In the ZCS-PWM

switching cell, S1 is is off when the power switch is turned on. The resonance

betweenr

L andr

C does not occur until S1 is turned on. When S1 is turned on,r

L

starts to resonate withr

C . After the resonance brings ther

L current to zero, S is

turned off with ZCS. Therefore, the function of S1 is to hold off the resonance for a

period of time. By controlling the hold-off time period, the on-time of the power

switch can be varied, enabling the ZCS-PWM converters to regulate the output while


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operating at a fixed switching frequency.

In principle, the power switch in fig.4.1 can be implemented with either a half-

wave or a full-wave switch. Since all the ZCS-PWM converters can be operated with

fixed frequency , different implementation of the power switch does not significantly

affect the basic characteristics of the converter. In practice, since the implementation

of the half-wave switch requires the use of a diode in series with the power transistor,

it will increase the conduction loss of the converter. To simplify the analysis, only the

full-wave version is discussed in the following the section.

Fig.4.1(a) ZCS quasi-resonant switching cell

Fig.4.1(b) ZCS-PWM switching cell

For each ZCS-PWM converter, the auxiliary switch can be implemented in two

ways as shown in 4.2. With different implementation, both the operation and the

performance of a ZCS-PWM converter are slightly different. In the following section,

the buck ZVS-PWM converter is used as an example to introduce the principle of

operation of the ZCS-PWM converters.


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Fig.4.2 Two ZCS-PWM switching cells with different implementation of the

auxiliary switch

4.2.2 Buck ZCS-PWM converter

4.2.2.1 Buck ZCS-PWM Converter

Figure 4.3 shows the circuit schematic and the waveforms of the ZCS-PWM

buck converter. This converter can be derived by replacing the PWM switching cell of

the buck converter with ZCS-PWM switching cell shown in fig.4.2(a). This converter

differs from the buck ZCS-QRC by the introduction of an auxiliary switch which in

series with the resonant capacitor. To simplify the analysis, the input filter inductor

and the output filter capacitor are assumed to be sufficiently large to be considered a


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negative peak value if: io

n

V I

Z >

At this moment, the gate-drive signals for S and S1 can be disabled at the same time,

so both S and S1 are turned off with ZCS. It can be seen that the worst operating

condition occurs at full load and low line. The resonant inductor current flows

through the anti-parallel diode S when it goes negative. This interval lasts until T3 ,

when the current through the anti-parallel diode S decays to zero and is turned off.

This operating stage is topologically identical to the previous one.

(e)T4-T5: At T4,r

C is still biased with certain voltage. During this time interval,r

C

is quickly discharged by currento

I .

(f)T5-T0: r C is discharged to zero voltage at T4. The L current freewheels through

D during this time interval. This operating stage is identical to the freewheeling stage

of the PWM buck converter. This interval lasts until To, when S is turned off, and the

switching cycle is repeated.


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Fig.4.3 Buck ZCS-PWM converter and its waveforms


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Fig.4.4 Equivalent Circuits for different operating modes of the buck ZCS-PWM

converter


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CHAPTER 5

SIMULATION RESULTS

Fig.5.1a.Proposed Circuit Diagram

The PWM controlled soft single switched boost converter circuit diagram as shown in

the fig.5.1a. It is used to less DC voltage to high DC voltage with minimum switching

loss and less voltage stress. 5.1b shows the input voltage and fig.5.1c shows the gate

pulse and Vds voltage across switch. fig 5.1d shows the output voltage and fig.5.1f

shows the comparison graph between input (vs) output voltage.

Fig.5.1b.Input DC Voltage


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Fig.5.1c.Switching Pulse (ii) Corresponding Voltage across the Switch

Fig.5.1d.Output Voltage

Fig.5.1e.Output current


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Fig.5.1f.comparison of input vs output voltage

Voltage Stress:

In fig 5.2a and 5.2c shows the convention and propsed boost circuit diagram.

Fig 5.2b and 5.2d shows the gate pulse and switch voltage for conventional and

proposed circuit diagram. From fig 5.2b conventional circuit has high voltage stress

and no ZVS switching. But fig 5.2d active the ZVS and low voltage stress across

switch thus the proposed circuit reduce the voltage stress.

Fig.5.2a.Conventional boost converter

Fig.5.2b. Gate pulse and Vds voltage


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Fig.5.3b.Input voltage with Disturbance

Fig.5.3c.Output voltage with disturbance

Closed loop system:

Closed loop system is used to control the output voltage. The output voltage is

compared with set value. Commparator gives the difference batween two signals it is

called as error. Error signal is given to PI controller. PI controller adjust Kp and Ki

gains based on the error signals. It is compared with one triangle carrier signal. It

generates a PWM pulse. It is given to MOSFET thus PWM pulse is generated. It is

control the output voltage. Fig.5.4a shows the closed loop system and fig.5.4b shows

the regulated output voltage with disturbance.


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Fig.5.4a.Closed Loop Circuit Diagram

Fig.5.4b.output voltage with Disturbance


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CONCLUSION

The PWM soft single switched boost converter is simulated in both open and closed

loop system. Open loop system explains about switching stress. Voltage stress should

be reduced using soft switching technique. This converter does not require any extraswitch to achieve soft switching, which considerably simplifies the control circuit.

PWM technique is used to control the output voltage. Thus PWM and soft switching

technique are improving the proposed converter performance. The output side voltage

regulation is achieved through closed loop system.


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FUTURE SCOPE

The proposed topology can be extended to other non isolated single switch DC-DC

converters such as buck, buck-boost, Cuk, Sepic, and Zeta. It can be extended to

isolated single switch DC-DC converters such as forward, Flyback, isolated Cuk,

isolated Sepic, isolated Zeta.


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REFERENCES

1) Mohammad Reza Amini, Student Member, IEEE, and Hosein Farzanehfard,

Member, IEEE “ Novel family of PWM soft- single switched DC-DC

converters with coupled inductors” IEEE Trans. Industrial electronics,Vol.56, no.6,june 2009.

2) E. S. da Silva, E. A. A. Coelho, L. C. Freitas, J. B. Vieira, and V. J. Farias,

“A soft-single-switched forward converter with low stresses and two derived

structures” IEEE Trans. Power Elect'ran., vol. 19, no. 2, pp. 388—395, Mar.

2004.

3) E. Adib and FF Farzanehfard, “Family of isolated zero-voltage transition

PWM converters” JET Power Electron., vol. I, no. I, pp. 144-153, Mar. 2008.

4) T. F. Wu, Y. S. Fai, J. C. Hung, and Y. M. Chen, “Boost converter with

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soft single switched dc-dc converters with coupled inductors

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