rf module design - [chapter 4] transceiver architecture

RF Transceiver Module DesignChapter 4

RF Transceiver Architectures李健榮助理教授

Department of Electronic EngineeringNational Taipei University of Technology

Outline

• General Considerations

• Frequency Conversion

• Receiver Architectures Heterodyne Receiver

Direct-Conversion Receiver (DCR)

Image-Reject and Low-IF Receiver

• Transmitter Architectures Direct-Conversion Transmitter (DCT)

Heterodyne and Sliding-IF Transmitter

Open-loop and Closed-loop PLL-based Transmitter

Envelope Tracking and Envelope Following Transmitter

Polar Transmitter

Department of Electronic Engineering, NTUT2/110

Front-end General Considerations

• TX: Adjacent channel leakage

• RX: Rejection of inband and out-of-band interference

BPF

Power Amplifier (PA)Transmitted Channel

Adjacent Channels

ω

BPF

Low Noise Amplifier (LNA) Bandpass Filter

Response

AdjacentChannel

Alternate AdjacentChannel

1ff


Interferer Suppression

• High linearity to accommodate interferes without experiencingcompression or significant intermodulation. Filtering theinterferer can relax RXlinearity requirements.

• BPF high selectivity is required for near channel rejection.

• Variable BPF is required for different carrier frequencies, andit is difficult.

900 900.4( ) MHzf

20 dB35 dB

BPF Response

Hypothetical filter to suppress an interference


Channel-Selection Filter

• All of the stages in the RXchain that precede channel-selection filtering must be sufficiently linear to avoidcompression or excessive intermodulation

• Since channel-selection filtering is extremely difficult at theinput carrier frequency, it must be deferred to some other pointalong the chain where the center frequency of the desiredchannel is substantially lower and hence the required filterQ’sare more reasonable.


Band-Select Filter

• A band-select filter selects entire RXband and reject out-of-band interferers, thereby suppressing components that may begenerated by users that do not belong to the standard ofinterest.

• Trade-off between selectivity and in-band loss (higher-orderfiltering sections and arise NF).

BPF

LNADesiredChannel

Receive Band

f

f

Band-selection filtering


TX-RX Feedthrough

• TX leakage in a CDMAtransceiver (full duplex). The RXmust meet difficult linearity requirements.

• A BPF following the LNAcan alleviate the leakage.

Du

ple

xer

−20 dBmLNA

PA

1 W (+30 dBm)

−50 dB

Du

ple

xer

LNA

PA

−50 dB

f

f

TX Leakage

f

BPF Response

BPF

10

dB

/div

.

20 MHz/div.

1f2f

TX Band RX Band

50 dB30 dB


Frequency Conversion (I)

• Recall Chapter 1 (double sideband amplitude modulation)

( ) ( )cos2m cs t A t f tπ=t( ) ( )BBs t A t=

ff

cf0 Hzcf−0 Hz

USBLSBUSBLSBLSBUSB

cos2 cf tπ“real signal”

Real signal

f0 Hz

Complex conjugate

USBLSB

1f1f−cos2 cf tπ

0 Hzcfcf−

USBLSBLSBUSB

IF cf f+c IFf f−IF cf f−c IFf f− −

Double sideband (DSB)

Double sideband (DSB)


Frequency Conversion (II)

• Recall Chapter 1 (linear modulation)

• Yes, a modulated signal sm(t) is a real signal.

( ) ( ) ( ) ( )1 12 2

2 2j t j tj f t j f tA t A t

e e e eφ φπ π− −= +

( ) ( ) ( )( )1cos 2ms t A t f t tπ φ= +

( ) ( ) 12Re j t j f tA t e eφ π= ⋅

f1f0 Hz1f−

“complex”“complex” “real”

Complex conjugate

( )I t

1cos tω1sin tω−

( )Q t

( )ms t

Real signal

Complex envelope


Frequency Conversion (III)

0 Hz2f2f−

0 Hz2f2f−

USBLSBLSBUSB

Real signal

f0 Hz

Complex conjugate

USBLSB

1f1f−1 2f f+2 1f f−1 2f f−2 1f f− −2cos2 f tπ

RFIF

( )I t

cos IF tωsin IF tω−

( )Q t

( )IFs t

Modulated signal (real signal)

f0 Hz

USBLSB

IFfIFf−

cos 2 cf tπ

RF 0 Hzcfcf−

USBLSBLSBUSB

IF cf f+c IFf f−IF cf f−c IFf f− −

Double sideband (DSB) mixing

upconversion

upconversion

IF

LO

LO By filtering, you can choose only USB or LSBtransmission, which is call the single-sideband(SSB) transmission.


Frequency Conversion (IV)

0 Hz2f2f−

0 Hz2f2f−

Real signal

f0 Hz

Complex conjugate

1f1f−1 2f f+1 2f f−2 1f f−2 1f f− −2cos2 f tπ

IFRF

downconversion

2 1f f<

2f2f−

0 Hz2f2f−

0 Hz2f2f−

Real signal

f0 Hz 1f1f− 1 2f f+2 1f f−1 2f f−2 1f f− −2cos2 f tπ

IFRF

downconversion

2 1f f>

2f2f−

High-side injection

Low-side injection

LO

LO


Receiver Architecture

• Basic Heterodyne Receiver

• Modern Heterodyne Receiver

Hetero-dyne

Different-freq. Mixing


Basic Heterodyne Receivers (I)

• Translating the desired channel to a much lower centerfrequency to permit a channel-selection filtering with areasonableQ.

ωinωinω− 0

ωLOωLOω− 0

Downconversion by mixingRF input

inω ω

Mixer

0 cos LOA tω

vLPF IF Output

in LOω ω− − in LOω ω− + 0 in LOω ω+in LOω ω−

Filtered-outFiltered-out

LO

( ) ( )1 1cos cos cos cos

2 2in LO in LO in LOt t tω ω ω ω ω ω⋅ = + + −

Low freq.High freq.

Two IF frequencies:


Basic Heterodyne Receivers (II)

• Use of LNA to reduce noise

• Variable IF:

• Constant IF:

Mixer

0 cos LOA tω

vLPF IF OutputRF input

LNA

IFj RFj LOf f f= − (Constant LO freq. and variable IF freq.)

IF RFj LOjf f f= − (Variable LO freq. and constant IF freq.)

Precise LO freq. and steps provided by a “frequency synthesizer”

Constant IF approach is more common to simplify the design ofIF path; e.g., it does notrequire a variable-frequency channel selection filter.

LO


Basic Heterodyne Receivers (III)

• Constant-LO downconversion mixing

• Constant-IF downconversion mixing

1RFff f

f1LOf

IFf0

1RFff f

fLOf

1IFf0 2RFff f

f

2IFf0

LOf

2RFff f

f

IFf0

2LOf


• While each wireless standard impose constrains upon theemissions by its own users, it may have no control over thesignals in other bands. The image power can therefore bemuch higher than that of the desired signal, requiring proper“image rejection.”

Image Problem in Heterodyne RX

cos LOtω

vLPF

Desired signal

Image

inω imω ωIFω ω

IFω IFω

LOωω

High-side injection

( ) ( )cosd d dA t t tω φ + ( ) ( )cosim im imA t t tω φ +

( ) ( ) ( ) ( ) ( ) ( ) ( )1 1cos cos

2 2IF d LO d LO d d LO d LO dx t A t A t t A t A t tω ω φ ω ω φ = + + − − +

( ) ( ) ( ) ( ) ( ) ( )1 1cos cos

2 2im LO im LO im im LO im LO imA t A t t A t A t tω ω φ ω ω φ + + + − − +


DownconvertedSpectrum (I)

1ω−2ω− 1ω+ 2ω+0ω

ω

ω

0 LOω+LOω−

0

1ω−2ω− 1ω+ 2ω+0ω

ω

ω

0 LOω+LOω−

0

Downconversion for 1LOω ω< Downconversion for 2 1LOω ω ω> >


DownconvertedSpectrum (II)

1ω−2ω− 1ω+ 2ω+0ω

ω

ω

0 LOω+LOω−

0

1ω−2ω− 1ω+ 2ω+0ω

ω

ω

0 LOω+LOω−

0

Downconversion for 2 1LOω ω ω> > Downconversion for 2LOω ω>

1 2

2LO

ω ωω +=


• The most common “image rejection “ approach is to precedethe mixer with an “image-rejection filter.”

• The filter exhibits a relatively small loss in the desired bandand a large attenuation in the image band, two requirementsthat can be simultaneously met if 2ωIF is sufficiently large.

• A filter with high image rejection typically appears betweenthe LNA and the mixer to lower the noise contribution to theRX NF (The NF increases while the filter precedes the LNA).

Image Rejection

Image Reject Filter

inω imωω

2 IFωcos LOtω

v

ImageRejectFilter

LNA


Image Rejection v.s. Channel Selection

Image Reject Filter

inω imωω

2 IFω

cos LOtω

v

ImageRejectFilter

LNA

v

ChannelSelectFilter

Desired channel

Interference

ImageChannel Select Filter

(high-Q needed)

IFωω

0

0ω

IFωinω imω

2 IFω

ω

High IF

Low IF

• If the IF is high, the image can besuppressed but complete channelselection is difficult, and vice versa.


Image Noise Increases Noise Figure

• Even in the absence of interferes, the thermal noise producedby the antenna and the LNAin the image band arrives at theinput of the mixer. The thermal noises in the desired channeland image band are downconverted to IF (unless the LNAhasa limited bandwidth to suppresses the noise in the image band).

LOω

LNA

inω

Thermal Noise

LOω inωω

2 in LOω ω−

ω


Dual IF Receiver (I)

• The concept of heterodyning is extended to multipledownconversions, each followed by filtering and amplification,to resolve the trade-off between “image rejection” and“channel selection.”

• This technique performs partial channel at progressively lowercenter frequencies, thereby relaxing theQ required of eachfilter.

1LOω

vBPF2

LNA

vBPF1 vBPF3

2LOω

vBPF4

Band Select Filter

ImageReject Filter

RF MixerMX1

ChannelSelect Filter

IF MixerMX2

ChannelSelect Filter IF Amp.

A C E GB D F H


Dual-IF Receiver (II)

1LOω

vBPF2

LNA

vBPF1 vBPF3

2LOω

vBPF4

Band Select Filter

ImageReject Filter

RF MixerMX1

ChannelSelect Filter

IF MixerMX2

ChannelSelect Filter IF Amp.

A C E GB D F H

Department of Electronic Engineering, NTUT

Desired Channel Image

fA

C

E

G

B

D

F

H

BPF1BPF2

Image

ImageBPF3

BPF4

f

f

f f

f

f

f

23/110

Mixing Spurs (I)

• In practice, the mixing is the multiplication of the RF input byall harmonics of the LO. Thus the RF mixer producescomponents at and IF mixer, wherem andn are integers.

• For the desired signal, only is of interest. But ifan interferer, , is downconverted to the same IF, it corruptsthe signal; this occurs if .

1in LOmω ω± 1 2in LO LOm nω ω ω± ±

1 2in LO LOω ω ω− −

intωint 1 2 1 2LO LO in LO LOm nω ω ω ω ω ω± ± = − −


Mixing Spurs (II)

• An example of a 2.4 GHz dual conversion RX:

1 1.98 GHzLOω =

LNA

vBPF

2 400 MHzLOω =

420 MHz2.4 GHz

20 MHz

2.7

6 G

Hz2.4 GHz

2.8

GH

z 4.38 GHzf

Received Spectrum

820 MHz780 MHz

2800 MHz2760 MHz

22 800 MHzLOω =12 3.96 GHzLOω =

4380 MHz

20 MHz420 MHz

2 400 MHzLOω =20 MHz


Modern Heterodyne Receivers

ω

ω0

0

1IFω− 1IFω+

2LOω+2LOω−

ω

ω

Desired ChannelInterferer

0

0

• Zero Second IF: Avoid secondary image(assume no interferers aredownconverted as an image to a zero center frequency).

• Interferer appears in the adjacent channel

2LOω ω

ω1IFω ω

02 1LO IFω ω=

2 1LO IFω ω=


Signal Becomes its Own Image

• For symmetrically-modulated signal:

• For asymmetrically-modulated signal:LOf

f

f0 f

LOf

( )S f ( )LOS f f−

vVCO

( )BBx t

t fcf

1IFω+1IFω− 0ω

ω0

same information on both side of the carrier

downconversion

Corruption occurs if the signal spectrum is asymmetric


Avoid Self-corruption of Asymmetric Signals (I)

• One can downconvert signal to an IF equal to half of the signalbandwidth to avoid self-corruption of a signal with asymmetricspectrum.

1IFω+1IFω− 0ω

ω0

BWω

2BWω+

2BWω−

1 2BW

IF

ωω ≥


Avoid Self-corruption of Asymmetric Signals (II)

• Zero second IF with quadrature downconversion.

( )IFx t

( ),BB Ix t

( ),BB Qx t

2cos LO tω

2sin LO tω

2 1LO IFω ω=

Quadrature baseband signal

Though xBB,I(t) and xBB,Q(t) exhibit identical spectra, they are separated in phaseand together can reconstruct the original information


Zero Second IF Heterodyne RX

• Zero second IF heterodyne RX with quadrature downconverison

( )IFx t

( ),BB Ix t

( ),BB Qx t

2cos LO tω

2sin LO tω

1LOω

vBPF

LNA

No image rejection filter LNA/mixer interface can be optimized (need not 50 Ohms) for gain, noise, and

linearity with little concern for the interface impedance values. The lack of image rejection filter requires careful attention to the interferers in

the image band, and dictates a narrow-band LNA design (suppress imagenoise).

No channel-selection filter is shown at the first IF, but some “mild” on-chipBPF is usually inserted to suppress out-of-band interferers.


Sliding-IF Heterodyne RX (I)

( )IFx t

( ),BB Ix t

( ),BB Qx t

2,ILO

1LOω

vBPF

LNA

vLO1 v2÷

2,QLO

t

2,ILO

2,QLO

1LO

90

RF Input

1st LO

1st IF

2nd IF

inff

f

f

f

1LOf

1in LOf f−

11 2

LOin LO

ff f− −

For an input band [f1, f2], the LOmust cover a range of [(2/3)f1, (2/3)f2].

11 2

LOLO in

ff f+ = 1

2

3LO inf f=

The 1st IF is not constant, because

1

1

3IF in LO inf f f f= − =


Sliding-IF Heterodyne RX (II)

• As fin varies fromf1 to f1, fIF1 goes fromf1 /3 to f2 /3 (slide IF).

RF Range

LO Range

1st IF Range

f1f 2f

f

f

1

2

3f 2

2

3f

1

1

3f 2

1

3f

( )2 1

2 1

1 13 3% *100%

1 1 12 3 3

IF

f fBW

f f

−=

+

( )( )

2 1

1 2

% *100%12

RF

f fBW

f f

−=+


Sliding-IF Heterodyne RX (III)

• Image band of the sliding-IF heterodyne RX

Image Band RF Band

LO Band1

1

3f 2

1

3f 1f 2f

f

f

1

2

3f 2

2

3f


Direct-Conversion Receivers

• Direct-conversion receiver (DCR) is also called the “zero-IF”,or the “homodyne” receiver.

• As mentioned previously, downconversion of an asymmetric-modulated signal to a zero IF leads to self-corruption unlessthe baseband signals are separated by their phases.

I

Q

cos LOtωsin LOtωvBPF

LNAvLPF

vLPF

inω

LO inω ω=


DCR Advantages

• The absence of an image greatly simplifies the design process.

• Channel selection is performed by low-pass filters, which canbe realized on-chip as active circuit topologies with relativelysharp cut-off characteristics.

• Mixing spurs are considerably reduced in number and hencesimpler to handle.

• The LNA/mixer interface can be optimized for gain, noise, andlinearity without requiring a 50-Ω impedance.


DCR Issues − LO Leakage

• A DCR emits a fraction of its LOpower fromits antenna, andthe LO emission is undesirable because it may desensitizeother receivers operating in the same band. Typical acceptablevalues range from−50 to−70 dBm(measured at the antenna).

• In heterodyne receivers, since the LOfrequency falls outsidethe band, it is suppressed by the front-end band-select filters inboth the emitting receiver and the victimreceiver.

PadLNA

LOSubstrate

LO

LNA

LO leakage Cancellation of LO leakage by symmetry


DCR Issues − DC Offset

• The leakage causes “LOself-mixing” at the mixer to produce adc component in the baseband (because multiplying a sinusoidby itself results in a dc term). The zero second IF architecturealso suffers fromthis issue.

• LO leakage yields a very large dc offset due to the high gain ofthe receiving chain, and this saturates the baseband circuits(degrades the dynamic range), prohibiting signal detection.

• Time-varying dc offset

RF LOV kV+

PadLNA

IF DCV V+


Effect of DC Offset in Baseband Chain

• Since and , thus . Amplifiedby another 40 dB, this offset reaches 1-V at the basebandoutput.

1 31.6vA = ( )632 2 VleakV µ= 10 mVdcV =

cos LOtω

LPF

sin LOtω

LNA

1 30 dBvA = 2 40 dBvA =

0 cos inV tω ( )cosbb in LOV tω ω−

1 0bb vV A V=cosleak LOV tω

1dc v leakV A V=


Leakage of Quadrature Phases of LO

• The dc offset measured in the basebandI and Q outputs areoften unequal.

LOLNA

( )cosleak LO leakV tω φ+

( ), cosdc I leak LO leak circuitV V Vα φ φ= +

( ), sindc Q leak LO leak circuitV V Vα φ φ= − +


Cancelling DC Offset

LNA

cos LOtω

LPF1C

1R

bv

1ABaseband SignalHPF Response

1f− 1f+0f

• Using a HPF (ac coupling) removes dc offset but also removesa fraction of the signal’s spectrumnear zero frequency, therebyintroducing distortion.

• As a rule of thumb, the corner frequency of the HPF must beless than 1/1000 of the symbol rate for negligible distortion.This may require very large capacitance and thus difficult tointegrate on chip (especially for lowsymbol rate). For the slowresponse to transient inputs (LOswitch, LNA gain change), accoupling is rarely used in today’s receivers.


DCR Issues − Even-order Distortion

• DCRs are additionally sensitive to even-order nonlinearity inthe RF path (IM2 falls around DC to corrupt the desired signaland mixer feedthrough), and so are the heterodynearchitectures having a second zero IF .

Beat Component

cos LOtω

1 2ω ω−Feedthrough

LNA

DesiredChannel

Interferers0

ω

0ω

1ω ω2ω


Mixer Feedthrough– Simple Mixer

inVoutV

LO

1R

( ) ( ) ( ) ( ) ( ) ( )1 1

2 2out in in inV t V t S t V t S t V t = ⋅ = − + ⋅

where is the RF input and is the Ideal LO toggling between 0 and1 with 50%duty cycle, and

( ) 1

2S t −

DC-free square wave

( ) 1

2inV t ⋅ is the RF feedthrough to the output

( )inV t ( )S t

represents a


Mixer Feedthrough– Differential Mixer

• If the output is sensed differentially, the RF feedthroughVout1(t)andVout2(t) are cancelled while the signal components add.

• This cancellation is sensitive to asymmetrics, e.g., if theswitches exhibits a mismatch between their on-resistance, thena net RF feedthrough arises in the differential output.

( ) ( ) ( )1out inV t V t S t= ⋅

LO

LO1R

1R

1outV

2outV

inV( )S t

( )1 S t−

( ) ( ) ( )2 1out inV t V t S t= ⋅ −


Even-order Distortion (I)

• The 2nd order nonlinear effect cause the beat amplitude whichgrows with the square of the amplitude of the input tones.

(log scale)

(log scale)inA

IIP2A

IIP2A1Aα

22Aα

( ) ( ) ( )21 2out in inV t V t V tα α= +

( ) ( ) ( )2 21 1 2 2 1 2 2 1 2cos cos cos cosA t t A t A tα ω ω α ω ω α ω ω= + + + + + +⋯

( ) 1 2cos cosinV t A t A tω ω= +


• Since the net feedthrough of thebeat depends on the mixer andLO asymmetries, the beatamplitude measured in thebaseband depends on the derivedimensions and the layout and istherefore difficult to formulate.

44/110

Even-order Distortion (II)

• Even-order distortion may manifest itself even in the absenceof interferers. Suppose in addition to frequency and phasemodulation, the received signal also exhibits amplitudemodulation.

• Both of the terms and are low-pass signals and,like the neat component, pass through the mixer with finiteattenuation, corrupting the downconverted signal.

( ) ( ) ( )0 cosin cx t A a t t tω φ= + +

( ) ( ) ( )21 2out in inV t V t V tα α= +

( ) ( ) ( ) ( )2 2 22 2 0 0

1 cos 2 22

2c

in

t tx t A A a t a t

ω φα α

+ + = + +

( )2 0A a tα ( )22 2a tα


DCR Issues − Flicker Noise (I)

• Linearity requirements limit the cascaded LNA/mixer gain, thedownconverted signal in a DCR is still relatively small andhence susceptible to noise in the BB circuits. Since the signalis centered around zero frequency, it can be substantiallycorrupted by the flicker noise.

• The mixers themselves may also generate flicker noise at theiroutput.

(log

sca

le)

fBWfCf

1000BWf

( )1 fS f

thS

,

1

2BW RF BWf f=1 fS

f

α= thc

Sf

α = th cS fα = ⋅

Assume noise components below fBW/1000 are unimportant

( )1 0.0015.9 ln

BW

BW

fc

n BW c th c th BW thfBW

fP df f f S f S f S

f f

α = + − = + +

∫

If no flicker noise 2n BW thP f S≈

Flicker noise penalty 1

2

n

n

P

P


DCR Issues − Flicker Noise (II)

• In a good design, the thermal noise at the end of the basebandchain arises mostly fromthe noise of the antenna, the LNA,and the mixer. Thus, a higher front-end gain directly raisesSth,thereby loweringfc and hence the flicker noise penalty.

• Flicker noise penalty:

An 802.11g RX with fc of 200 kHz:

10 MHzBWf =

1

2

1.04n

n

P

P=

A GSM RX with fc of 200 kHz:

1

2

16.4n

n

P

P=

(log

sca

le)

BWf200100

( )1 fS f

( ) kHzfthS

DownconvertedGSM Channel

Effect of flicker noise on a GSM channel

Flicker noise makes it difficult to employ DCR for a narrow channel bandwidth.In such cases, the “low-IF” architecture proves a more viable choice.


DCR Issues − I/Q Mismatch

• DCR require 90o shift of the RF signal and this generallyentails severe noise-power-gain trade-offs.

I

Q

LOV

vLPF

vLPF

90

RFV

I

Q

LOV

vLPF

vLPF

RFV90

Shift of RF signal or LO waveform by 90o


I/Q Mismatch (I)

• Errors in the 90o phase shift circuit and mismatches betweenthe quadrature mixers result in imbalance in the amplitudesand phases of the basebandI andQ outputs.

• The BB stages themselves may also contribute mismatches.

I

Q

LOV

vLPF

vLPF

RFV

90

Phase and Gain Error





I/Q Mismatch (II)

• I/Q mismatches tend to be larger in DCRs than in heterodynetopologies, because:

Propagation of a higher frequency experiences greater mismatches

LO quadrature phases suffer from greater mismatches at higher frequencies

cos LOtωsin LOtω

LNA

5 GHz

10 ps 18 @5 GHzT∆ = ⇒

( )IFx t

( ),BB Ix t

( ),BB Qx t

LNA

vLO1 4÷

5 GHz

4 GHz

1 GHz

10 ps 3.6 @1 GHzT∆ = ⇒

DCR Heterodyne RX


I/Q Mismatch – QPSK Example (I)

( ),BB Ix tvLPF

vLPF

( )inx t 90 ( )LOx t2

θ+

2

θ−

12

ε+

12

ε−

( ),BB Qx t

( ) cos sinin c cx t a t b tω ω= +

, 1a b = ±

( ), 2 1 cos2 2LO I cx t tε θω = + +

( ), 2 1 cos2 2LO Q cx t tε θω = − −

( ), 1 cos 1 sin2 2 2 2BB Ix t a bε θ ε θ = + − +

( ), 1 sin 1 cos2 2 2 2BB Qx t a bε θ ε θ = − − + −

The mismatch causes crosstalk between I and Q BB signals.


I/Q Mismatch – QPSK Example (II)

I

Q

t

tI

Q

Ideal

I

Q

t

t

I

Q

Ideal

Only amplitude mismatch : 0, 0ε θ≠ =

Only phase mismatch : 0, 0ε θ= ≠

( ), 12BB Ix t aε = +

( ), 1 cos2 2BB Qx t bε θ = −

( ), cos sin2 2BB Ix t a bθ θ= −

( ), sin cos2 2BB Qx t a bθ θ= − +


Correction of I/Q Mismatch

cos LOtωsin LOtω

LPF

LPF

LNAI

Q

IQ

PhaseMismatch

AmplitudeMismatch

t

LPF

LPF

ADC

ADC

Logic

φ

φ

cos LOtω

sin LOtω

LNA

• Calibration of quadrature phase and gain either at power up orcontinuously is usually needed for high performance system.

Test signal

Analog adaption

Digital adaption is more popularin nowadays systems


Mixing Spurs

• Unlike heterodyne systems, DCRs rarely encounter corruptionby mixing spurs. This is because, for an input frequencyf1 tofall in the baseband after experiencing mixing withnfLO, wemust havef1 ≈ nfLO.

• Since fLO is equal to the desired channel frequency,f1 lies farfrom the band of interest and is greatly suppressed by theselectivity of the antenna, the band-select filter, and the LNA.

• The issue of LOharmonics does manifest itself if the receiveris designed for a wide frequency band (greater then twooctaves). Examples include TVtuners, “software-definedradios,” and “cognitive radios.”


Image-Reject Receivers

• “Image-reject” architectures are another class of receivers thatsuppress the imagewithout filtering, thereby avoiding thetrade-off between image rejection and channel selection.

• Benefits froma 90o shifter (Hilbert transform, −j for f > 0, +j for f <0)

Re

Im

Re

Im

Re

Im

2

Aj+

2

A

2

A

2

Aj−

cω−

cω+ω

( )cos2

c cj t j tc

AA t e eω ωω −= +

Illustration of 90o phase shift for a cosine

( ) ( )9090cos 902

ccj tj t

c

AA t e e

ωωω − −− − = +

2 2c cj t j tA A

je jeω ω−= − +

sin cA tω=


Hilbert Transform(I)

• Hilbert transformation pair:

• The transform means a 90 degree phase shift in time domain, the impulse response of the Hilbert transformation .

( ) ( )( )

ˆx

x t dt

ττ

π τ∞

−∞=

−∫ ( ) ( )( )x t

x dt

τ τπ τ

∞

−∞= −

−∫

( ) 1h t tπ=

( ) 1h t

tπ=( )x t ( )x t

90( )x t ( )x t


Hilbert Transform(II)

• Simple relation between sine and cosine functions:

• It simply shows that if we want to make a transformbetweencosine and sine waveforms, a 90 degrees phase shift isrequired.

• FromEuler’s formula:

( ) ( )cos 90 sint tω θ ω θ+ − = + ( ) ( )sin 90 cost tω θ ω θ+ − = − +and

0 0

0cos2

j t j te et

ω ω

ω−+=

0 0

0sin2

j t j te et

j

ω ω

ω−−=

( ) ( )0 0

12

δ ω ω δ ω ω− + +

( ) ( )0 0

1

2 jδ ω ω δ ω ω− − +

F.T.

F.T.


Hilbert Transform(III)

• We like to find a transfer function, which is able transfer thecosine to since and since to cosine function.

( ) ( ) ( ) ( ) ( )0 0 0 0

1 12 2

H j jδ ω ω δ ω ω ω δ ω ω δ ω ω− + + ⋅ = − − − +

( ) ( ) ( ) ( ) ( )0 0 0 0

1 1

2 2H j

jδ ω ω δ ω ω ω δ ω ω δ ω ω − − + ⋅ = − − + +

( ) ( )sgnH j jω ω= − ⋅

( )1 , 0

sgn 0 , 0

1, 0

ωω ω

ω

>= =− <

( ) ( ) 0

0

1 1sgn

2 2 2j t j t j tj j

h t j e d e d e dt

ω ω ωω ω ω ωπ π π π

∞ ∞

−∞ −∞= − ⋅ ⋅ = − =∫ ∫ ∫

cosine

sine negative cosine

0 0ω ω= > ( ) ( ) ( )0

1 10 0

2 2H j jδ ω δ⋅ = −

0 0ω ω= − < ( ) ( ) ( )0

1 10 0

2 2H j jδ ω δ⋅ − =

0 0ω ω= = ( ) ( ) ( ) ( ) ( )1 10 0 0 0 0

2 2H jδ δ δ δ+ ⋅ = − − ( )0 0H =

( )H j jω = −

( )H j jω =

:

:

:

sine

phase

f90+

90−


90o Phase Shift (I)

• 90o phase shift for a modulated signal

Re

Im

( )X ω

ωcω+

cω−

( )X ω( )jX ω

( )jX ω−

( ) ( ) ( )cos cx t A t t tω φ= +

( ) ( ) ( ) ( )( ) ( )( ) ( ) ( )( ) ( )( )90 90cos 90

2 2c c c c

j t t j t t j t t j t t

c

A t A tA t t t e e je je

ω φ ω φ ω φ ω φω φ + − − + − + − + + − = + = − +

( ) ( )sin cA t t tω φ= +

( ) ( ) ( )90sgnX X jω ω ω= −


90o Phase Shift (II)

Re

Im

2

Aj+

2

A

2

Aj−

cω−

cω+ω

sin2

c cj t j t

c

e eA t A

j

ω ω

ω−−=

2

A−

sin cjA tω

Re

Im

2

A

2

Acω−

cω+ω

cos ctω

Re

Im

Acω−

cω+ω

cos sinc ct jA tω ω+

• Plot the spectrum of cos sinc cA t jA tω ω+

(SSB spectra)


sine is the Hilbert transform of the cosine

To get a SSB spectra: (1) real signal (2) its Hilbert transform (3) = (1)+j(2)

60/110

90o Phase Shift (III)

Re

Im

( )S ω

ωcω+

cω−

( )S ω( )jS ω

( )jS ω−

Re

Im

( )S ω

ωcω+

cω−( )S ω−

( )jS ω

( )jS ω−

Re

Im

( ) ( )ˆS jSω ω+

ωcω+

• A narrowband signal with a real spectrumis shifted by 90o

to produce . Plot the spectrumof which is calledthe “analytic signal,” or the “pre-envelope” of .

( )s t

( )s t ( ) ( )ˆs t js t+( )s t

( )S ω ( )ˆjS ω


90o Phase Shift (IV)

• Use RC-CRnetwork to implement the 90o phase shifter

1C

1R

1R

1C

1outV

2outVinV

HPFH

LPFH

11

2

1 1

1

R C

ω

11 1tan

2HPFH R Cπ ω−∠ = −

11 1tanLPFH R Cω−∠ = −

ω2

π

2

π−

02

π

( ) 1 1 1

1 1 1out

HPFin

V R C sH s

V R C s= =

+

( ) 2

1 1

1

1out

LPFin

VH s

V R C s= =

+

We can consider Vout2 asthe Hilbert transform ofVout1 at frequencies closeto (R1C1)−1


Amplitude response

Phase response

62/110

Quadrature Downconversion (I)

• Quadrature downconversion translate the spectrumto anonzero IF as a 90o phase shifter.

RFV

IFI

cos LOtωsin LOtω

IFQ

0cω− cω+ ω

0LOω− LOω+ ω

1

2+1

2+

1

2+ 1

2+

0IFω− IFω+ω

0cω− cω+ ω

2

j+

2

j−

LOω+0LOω− ω

0IFω−IFω+

ω

2

j−

2

j+

High-side injection


Quadrature Downconversion (II)

• Quadrature downconversion translate the spectrumto anonzero IF as a 90o phase shifter.

2

j+

2

j−

LOω+0LOω− ω

0

IFω−

IFω+ ω

2

j−

2

j+

1

2+ 1

2+

0IFω− IFω+ω

Low-side injection

RFV

IFI

cos LOtωsin LOtω

IFQ

0cω− cω+ ω

0cω− cω+ ω

0LOω− LOω+ ω

1

2+1

2+


Distinguish Desired Signal and Its Image

( )sig im IFI I+

cos LOtωsin LOtω

( )sig im IFQ Q+

Re

Im

sigI

IFω−sigQ

sigQ

sigI

IFω+ ω

Re

Im

imI

imQ

imQ

imI

IFω+ ω

Re

Im

cω+ ωimω+imω−

cω−

Signal Components

Image Components

IFω−


Signal: low-side injection

Image: high-side injection

LOω+

LOω−

65/110

Image Reject RX – Hartley Architecture (I)

• Negate image


C

A

LPF

LPF

B

cos LOtωsin LOtω

90

sigQ

imQ,90im

Q

,90sigQ

sigI

imI

IF Outputω

imω+ cω+cω− imω− 0

sigI

sigQ

imI

imQ

Re

Im

IFω−

IFω+ ω

Re

Im

IFω−IFω+

ω

Re

Im

IFω−

IFω+ ω

Re

Im

IFω−

IFω+ ω

66/110

Image Reject RX – Hartley Architecture (II)

C

A

LPF

LPF

B

cos LOtωsin LOtω

90

sigQ

imQ,90im

Q

,90sigQ

sigI

imI

IF Output


sigI imI

Re

Im

IFω−

IFω+ ω

Re

Im

IFω−

IFω+ ω

,90sigQ

,90imQ

Re

Im

IFω−

IFω+ ω

Re

Im

IFω−IFω+

ω

Re

Im

IFω−

IFω+ ω

67/110

• Negate image

Image Reject RX – Hartley Architecture (III)

• Realization of 90o phase shift in Hartley receiver

1cos tω1sin tω

RF Input

LPF

LPF

IF Output

1R

1R

1C

1C


Image Reject RX – Hartley Architecture (IV)

• Downconversion of Hartley receiver output to baseband

LPF

LPF

1cos LO tω1sin LO tω

90

2sin LO tω2cos LO tω

I

Q

RF Input


Image Reject RX – Weaver Architecture (I)

A

1cos tω1sin tω

RF InputLPF

LPF

2cos tω2sin tω

LPF

LPFB

C

D

E

F

−

+

IF Input


ωimω+ cω+cω− imω− 0

sigI imI

Re

Im

1IFω−

1IFω+ ω

Re

Im

1IFω−

1IFω+ ω

sigQ imQ

Re

Im

1IFω−

1IFω+ ω

Re

Im

1IFω−1IFω+

ω

70/110

• Negate image

Image Reject RX – Weaver Architecture (II)

A

1cos tω1sin tω

LPF

LPF

2cos tω2sin tω

LPF

LPFB

C

D

E

F

−

+

IF


sigI imI

Re

Im

2IFω−

2IFω+ ω

Re

Im

2IFω−

2IFω+ ω

sigQ imQ

Re

Im

2IFω−

2IFω+ ω

Re

Im

2IFω−

2IFω+ ω

Re

Im

IFω−

IFω+ ω

Low-side inj.

71/110

• Negate image

Image Reject RX – Weaver Architecture (III)

1inω ω−

ω2ω

2 12 inω ω ω− +0

FirstIF

ω

1 2inω ω ω− −0

SecondIF

ω

2 12 2inω ω ω− +1ω inω0

RF Input

SecondaryImage

DesiredChannel


• Problem of secondary image

72/110

Image Reject RX – Weaver Architecture (IV)


1cos tω

1sin tω

RF Input

LPF

LPF

2÷ ( ),BB Qx t

−

+−

+

( ),BB Ix t

• Double quadrature downconversion Weaver architecture toproduce BB outputs. The second downconverion produceszero IF to avoid secondary image.

73/110

Low-IF Receiver (I)

cf

GSM Adjacent Channel Spec.

9 dB

ff0 100 kHz

cff

LOff

200 kHz

• It is undesired to place image within the signal band becausethe overall NF would raise by approximately 3 dB.

• In “low-IF” RXs, the image indeed falls in the band but can besuppressed by image rejection techniques.

• For a GSMRX, signal would be corrupted by flicker noise in azero-IF architecture. The noise penalty can be lower by usinglow-IF architecture (attractive for narrow-channel standards).

Moderate IRR is ok.


Low-IF Receiver (II)

2sin tω

LPF

2cos tω

LPF

1R

1R

1C

1C

IF OutputRF Input

Quadrature Phases of Image and Signal

sin LOtω

LPF

cos LOtω IF OutputRF Input

LPF

ADC

ADC

90


• Adopt image cancellation technique with low-IF architecture

75/110

Low-IF Receiver (III)

• Low-IF receiver with double quadrature downconverter


sin ctωcos ctω

IF Output

RF Input

( ),IF Qx t

−

+ +

+

( ),IF Ix t90

76/110

Polar Receiver

• Using the oscillator injection locking technique to accomplishmagnitude and phase extraction of the complex envelope. Thistechnique was also published to performenvelope eliminationand restoration in the Kahn EER transmitter.

LPF

LPFILO1 ILO2

Magnitude

Phase


Transmitter Architectures

• Basic Direct Conversion Transmitter (DCT)

• Modern DCT

• Heterodyne Transmitters

• OOK Transceivers

• Open-loop Phase Modulation Techniques

• Closed-loop Phase Modulation Techniques

• Polar Transmitter


• Quadrature upconverter:Server as the modulator.

• Power amplifier:Amplify the signal.

• Matching network :Provide maximum power delivery to antenna and filter out-of-band componentsthat result from the PA nonlinearity.

• xBB,I(t) and xBB,Q(t) are generated by BB circuits and hence has a sufficientlylarge amplitude, the noise of the mixers is much less critical here than inreceivers.

• A predriver is typically interposed between the upconverter and the PA to serveas a buffer.

Direct-Conversion Transmitter (DCT)

cos ctω MatchingNetwork

PA

sin ctω−( ) ( ) ( )cos cx t A t t tω φ= +

( ) ( ) ( ) ( )cos cos sin sinc cA t t t A t t tφ ω φ ω= −

( ) ( ) ( ), cosBB Ix t A t tφ=

( ) ( ) ( ), sinBB Qx t A t tφ=


I/Q Mismatch

• I/Q mismatch in the DCT:

• Constellation:

( ) ( ) ( )1 2cos sinc c c c cx t A A t A tα ω θ α ω= + ∆ + ∆ +

( ) ( )1 2 1cos cos sin sinc c c c c c cA A t A A A tα θ ω α α θ ω = + ∆ ∆ + − + ∆ ∆

1 2, 1α α = ±

1 21 cos , 1 1 sinc c

c c

A A

A Aβ θ β θ

∆ ∆= + + ∆ = − + ∆

1 21 cos , 1 1 sinc c

c c

A A

A Aβ θ β θ

∆ ∆= + + ∆ = − − + ∆

1 21 cos , 1 1 sinc c

c c

A A

A Aβ θ β θ

∆ ∆= − + ∆ = + + ∆

1 21 cos , 1 1 sinc c

c c

A A

A Aβ θ β θ

∆ ∆= − + ∆ = − + + ∆


I/Q Calibration (I)

• Apply a single sinusoidal to both inputs of the upconverter toreveal phase mismatch.

It can be shown that the output contains two sidebands of equal amplitudes andcarries an average power equal to

We observe thatε is forced to zero as described above, then .

Thus, the calibration of phase mismatch proceeds to drive this quantity to zero.

cos ctωsin ctω0 cos inV tω 3outV

+

−

( ) ( ) ( )3 0 01 cos cos cos sinout in c in cV t V t V tε ω ω θ ω ω= + + ∆ −

( )0 cos 1 cos cosin cV tω ε θ ω= + ∆

( ) ( )2 23 0 1 1 sinoutV t V ε θ= + + ∆

2 23 1 sinout outV V θ− = ∆

( )0 cos 1 sin 1 sinin cV t tω ε θ ω− ⋅ + ∆ +


I/Q Calibration (II)

• Applying a sinusoidal to one BB input while the other is set tozero for gain mismatch calibration.

The gain mismatch can be adjusted so as to drive this difference to zero.

0 cos inV tω

cos ctωsin ctω 1outV

cos ctωsin ctω

0 cos inV tω

2outV

( ) ( ) ( )1 0 1 cos cosout in cV t V t tε ω ω θ= + ⋅ + ∆ ( )2

2 201 02out

VV t V ε= +

( )2 0 cos sinout in cV t V t tω ω= ⋅ ( )2

2 02 2out

VV t =

( ) ( )2 2 21 2 0out outV t V t V ε− =


Carrier Leakage (I)

• The analog BB circuitry producing the quadrature signalsexhibits dc offsets, and so does the baseband port of eachupconvertion mixer:

• The upconverter therefore contains a fraction of theunmodulated carrier, called “carrier leakage”:

( ) ( ) ( )1 2cos cos sin sinout OS c OS cV t A t V t A t V tφ ω φ ω= + − +

( ) ( ) ( ) 1 2cos cos sinout c OS c OS cV t A t t V t V tω φ ω ω= + + −

( )

2 21 2

2Relative Carrier Leakage OS OSV V

A t

+=


Carrier Leakage (II)

• RX BB outputs suffer from dc offsets.

• In the presence of carrier leakage, ifthe TX power is controlled byvarying BB signals, it is difficultforThe base station to measure theactual signal power.

0 2OSV V+ +

Q

I

0 1OSV V+ +0V− 0V+0V−

0V+

cos ctωsin ctω

PA

Receiver

Base Station

Bas

eban

dP

roce

sso

r

Carrier Leakage

cωω


Reduction of Carrier Leakage

Bas

eban

dP

roce

sso

r

DACQ

DACI

Register ADC

PowerDetector

cos ctωsin ctω

• The BB swing, A(t), must be chosen sufficiently larger toreduce carrier leakage. However, asA(t) increases, the inputport of mixers becomes more nonlinear. A compromise istherefore necessary.

( )

2 21 2

2Relative Carrier Leakage OS OSV V

A t

+=

• Use BB offset control to reducethe carrier leakage. Duringcarrier leakage cancellation, theBB processor produces a zerooutput so that the detectormeasures only the leakage. Thus,the loop can use the DACs todrive the leakage toward zero.


Transmitter Linearity (I)

• Upconversion mixers in TXsense no interferers, however,excessive nonlinearity in the mixer BB port can corrupt thesignal or raise the adjacent channel power.

• In most cases, as the BB signal swings increase, the PAoutputbegins to compress before the mixer nonlinearity manifestsitself.

• Power back-off is required for

variable envelope signal to avoid

spectrumregrowth at PAoutput.

1-dB Compression PointoutV

inV0V

t


Transmitter Linearity (II)

• In the TX chain, the signal may experience compression in anyof the stages. Since the largest voltage swing occurs at theoutput of the PA, this stage dominates the compression of theTX. In a good design, the preceding stages must remain wellbelow compression as the PAoutput approaches P1dB. Toensure this, we must maximize the PAgain and minimize theoutput swing of the predriver and the stage preceding it.

cos ctωsin ctω

,BB IV

,BB QV

PAPredriver

XV drV outVXV

drV

outV

BBV


Oscillator Pulling

• The PA output (very large swing) would couple to variousparts of the systemthrough the silicon substrate, packageparasitics, and traces on the printed-circuit board. Thus, it islikely that an appreciable fraction of PAoutput couples to theLO to pull the oscillator.

outφLO

ω∆

LOω ω

LOω ω

ω∆PA

LO

I

LOω

Q

Output Spectrum

injω ω


Avoid LO Pulling (I)

• Most of today’s DCTs avoid an oscillator frequency to the PAoutput frequency by using frequency division and mixing.

• Since the PAnonlinearity produces a finite amount of power atthe second harmonics of the carrier, the LOmay still be pulledby using the following architecture.

• Very high speed divider is needed, but even a substantial efforton divider design to enable this architecture is well justified.

I

2LO cω ω=

Q

LO 2÷PA

cω 2 cωω


Avoid LO Pulling (II)

• Use a frequency double is possible to avoid LOpulling, butthe doubler typically dose not provide quadrature phases,necessitating additional quadrature generation stages such asthe poly phase filter.

• Advantage: no harmonic can pull the LO.

• Disadvantage: the doubler and polyphase filter suffer fromahigh loss, requiring the use of power-hungry buffers.

I

2c

LO

ωω =

Q

LO 2X PolyphaseFilter

PA

cω ω


Avoid LO Pulling (III)

• Use of mixing to avoid LO pulling.

• It is difficult to remove the unwanted carrier by means offiltering because the two frequencies are only differ by only afactor of 3. Even the filter is applied, the unwanted sidebandwould corrupt other channels or bands.

1

2

ωLO

2÷

1ω

1

2

ω 13

2

ωω

PAQuadratureUpconverter

( ),BB Ix t

( ),BB Qx t

chI chQLO

1

2

ω 13

2

ωω

1

2

ω 13

2

ωω


Suppress Unwanted Sideband (I)

• Use the single-sideband (SSB) mixing technique to suppressthe unwanted sideband instead of filtering.

• The harmonics of the input frequencies also corrupt the outputof an SSB mixer.

2cos tω

2sin tω

1cosA tω

1sinA tω

outV1ω

2ω

outV

( )1 2 1 2 1 2cos cos sin sin cost t t t tω ω ω ω ω ω− = +

Symbol of a SSB mixer

2ω 1ω1 2ω ω−0 2 13ω ω− 1 2ω ω+ 1 23ω ω−ω


Suppress Unwanted Sideband (II)

• For use in a DCT, the SSB mixer must provide the quadraturephases of the carrier. This is accomplished by nothing that

2cos tω

2sin tω

1cos tω

1sin tω

2cos tω

2sin tω

+

−

+

+

( )1 2sin tω ω+

( )1 2cos tω ω+

( )1 2 1 2 1 2sin cos cos sin sint t t t tω ω ω ω ω ω− = +( )1 2 1 2 1 2cos cos sin sin cost t t t tω ω ω ω ω ω− = +

SSB mixer providing quadrature outputs


Suppress Unwanted Sideband (III)

• Carrier provided by SSB mixing for a DCT.

• While suppressing the carrier sideband atω1/2, thisarchitecture presents two drawbacks: (1) the spurs at 5ω1/2 andother harmonic-related frequencies prove troublesome, and (2)the LOmust provide quadrature phases, a difficult issue.

( ),BB Ix t

( ),BB Qx t

12

3

ω1

3

ωI/QI/QLO 2÷

PA

1ω

DCT using SSB mixing in LO path


Heterodyne Transmitter

• Another approach to avoiding injection pulling involvesperforming the signal upconversion in two steps so that theLO frequency remains far fromthe PAspectrum.

• Smaller I/Qmismatch

1sin tω

1cos tω

2cos tω

I

BPF

PA

Q

1ωω

2ωω

1 2ω ω+ω


Sliding-IF TX

• The carrier frequency is equal to 3ω1/2.

1ω

PA

BPF

LO2÷

I

Q

1

1

2ω

1

32

ω

RF Mixer


Carrier Leakage

• The dc offsets in BB yield a component atω1/2 at the output ofthe quadrature upconverter, and the dc offset at the input of theRF mixer produces another component atω1. The former canbe minimized, and the latter (lower sideband) atω1/2 must beremoved by filtering. The leakage atω1 is closer to the uppersideband than the lower sideband is, but it is also much smallerthan the lower sideband. Thus, the filter following the RFmixer must be designed to attenuate both to acceptably lowlevels.

1

2

ω

IF Output

1

2

ω 13

2

ω1ω

ω

RF Output

Carrier leakage in heterodyne TX


Mixing Spurs (I)

• Heterodyne TXdisplays various mixing spurs that must bemanaged properly. The spurs arise fromthe mechanismwith1st LO and 2nd LO.

1

2

ω+

IF Output

13

2

ω+ 15

2

ω+015

2

ω− 13

2

ω− 1

2

ω−ω

1

2

ω+ 13

2

ω+ 15

2

ω+0 17

2

ω+13

2

ω− 1

2

ω−ω

RF Output

1

2

ω+ 13

2

ω+17

2

ω− 015

2

ω− 13

2

ω− 1

2

ω−ω

LO mixed with 2LO, 5LO

IF mixed with 2nd LO


Mixing Spurs (II)

• Effect of harmonics of 2nd LO on TX output. Upon mixingwith +3ω1, the IF sideband at−3ω1/2 is translated to+3ω1/2,thereby corrupting the wanted sideband (if the modulation isasymmetric). Similarly, the IF sideband at−5ω1/2 is mixedwith +5ω1 and falls atop the desired signal.

13

2

ω+

ω13

2

ω− 015

2

ω−

13

2

ω+

ω0


Reduce Unwanted Components

• Use of BB quadrature SSB mixing and IF SSB mixing toreduce the unwanted component.

( ) ( ), cosBB Ix t A t θ=

( ) ( ), sinBB Qx t A t θ=

+

−

+

+

1ω LO2÷PA

RF SSB Mixer

I

Q

13

2

ω


OOK Transceivers

• On-off keying (OOK) modulation is a special case of ASKwhere the carrier amplitude is switched between zero andmaximum.

• Less bandwidth-efficient as unshaped binary pulses modulatedon one phase of the carrier occupy a wide spectrum.

LO

PA

LO

PA

LNAEnvelopeDetector

Direct LO switching PA switching

OOK RX

OOK TX


Open-loop Modulation

• Open-loop modulation based-on a frequency synthesizer (orphase-locked loop).

• Wideband (high data rate).

• Poor accuracy due to VCOfrequency drifting.

reff

DAC

VCO

PFD Loop Filter

Div-by-N

[ ]BBs n

( )BBs t

( )ms t


Closed-loop Modulation (I)

• Closed-loop modulation based-on a frequency synthesizer (or phase-locked loop).

• Narrowband (lowdata rate).

• Good frequency accuracy.

• No DACs required.

∆ −∑

VCO

PFD Loop Filter

[ ]BBs n

/ 1N N÷ +

( )ms t

reff

Modulator


Closed-loop Modulation (II)

• Use the compensated filtering to increase the data rate.

∆ −∑

VCO

PFD Loop Filter

[ ]BBs n

/ 1N N÷ +

( )ms t

reff

ModulatorCompensated

Filter


Closed-loop Modulation (III)

• Use the two-point ∆-Σmodulation to increase the data rate.

∆ −∑

Two-point VCO

PFD Loop Filter

[ ]BBs n / 1N N÷ +

( )ms t

reff

Modulator

DAC


Envelope Detector

Envelope Following/Tracking Transmitter

• Dynamically adjusting bias to improve efficiency.

( )BBA t′

( )ms t

Linear PA

Antenna

Matching

( )BBA t

I/Q Modulator

AmplitudeModulator/Regulator


( )I t

cos ctωsin ctω−

( )Q t

106/110

Polar Transmitter (I)

• Envelope Elimination and Restoration Scheme (Kahn EER TX,1952):


Envelope Detector ( )BBA t′

( )ms t

Switching-mode PA

Antenna

Matching

( )BBA t

I/Q Modulator

AmplitudeModulator/Regulator

( )I t

cos ctωsin ctω−

( )Q t

Limiter

107/110

Polar Transmitter (II)

• Polar Transmitter


( )BBA t

cos ctω

( )ms t

Switching-mode PA

Antenna

PhaseModulator

Matching

( )BBA t

( )BB tφ

Bas

eban

d

Pro

cess

orAmplitudeModulator

( ) 2Re c BBj f t te

π φ+

• Linear modulator to generate PM signal• Frequency synthesizer or PLL-based PM modulator

108/110

Polar Transmitter (III)

• Hybrid Quadrature and Polar Modulation TX (HQPM-TX):


Bas

eban

d

Pro

cess

or

( )BBA t′

( )ms t

Switching-mode PA

Antenna

Matching

( ),BB DSMA t

I/Q Modulator

AmplitudeModulator/

Class-S

( )I t

cos ctωsin ctω−

( )Q t

109/110

Summary

• In this chapter, many receiver and transmitter architectureswere introduced. For receiving or transmitting, there are twomain categories including heterodyne and direct conversionarchitectures.

• For these transceivers, the modulation and demodulation canbe classified as “I/Q” and “polar” schemes. I/Qmodulator isan universal modulator with high linearity and signal quality,and the polar modulator is adopted for improving powerefficiency. I/Q demodulator is the conventional scheme todemodulate signals, and the polar demodulator is proposed forlow-cost and low-power applications.
