research article a low-power ultrawideband low-noise ...a low-power ultrawideband low-noise...
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Hindawi Publishing CorporationActive and Passive Electronic ComponentsVolume 2013, Article ID 953498, 10 pageshttp://dx.doi.org/10.1155/2013/953498
Research ArticleA Low-Power Ultrawideband Low-Noise Amplifier in0.18 ๐m CMOS Technology
Jun-Da Chen
National Quemoy University, Department of Electronic Engineering, University Road, Jinning Township, Kinmen 892, Taiwan
Correspondence should be addressed to Jun-Da Chen; [email protected]
Received 11 July 2013; Revised 17 October 2013; Accepted 4 November 2013
Academic Editor: Rezaul Hasan
Copyright ยฉ 2013 Jun-Da Chen.This is an open access article distributed under the Creative Commons Attribution License, whichpermits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.
This paper presents an ultrawideband low-noise amplifier chip using TSMC 0.18๐m CMOS technology. We propose a UWB lownoise amplifier (LNA) for low-voltage and low-power application. The present UWB LNA leads to a better performance in termsof isolation, chip size, and power consumption for low supply voltage. This UWB LNA is designed based on a current-reusedtopology, and a simplified RLC circuit is used to achieve the input broadband matching. Output impedance introduces the LCmatching method to reduce power consumption. The measured results of the proposed LNA show an average power gain (S
21) of
9 dB with the 3 dB band from 3 to 5.6GHz. The input reflection coefficient (S11) less than โ9 dB is from 3 to 11 GHz. The output
reflection coefficient (S22) less than โ8 dB is from 3 to 7.5 GHz. The noise figure 4.6โ5.3 dB is from 3 to 5.6GHz. Input third-order-
intercept point (IIP3) of 2 dBm is at 5.3 GHz. The dc power consumption of this LNA is 9mW under the supply of a 1 V supply
voltage. The chip size of the CMOS UWB LNA is 1.03 ร 0.78mm2 in total.
1. Introduction
The ultrawideband (UWB) system has become one of themajor technologies for wireless communication systemsand local area networks. The IEEE 802.15.3a ultrawideband(UWB) system uses a specific frequency band (3.1 GHzโผ10.6GHz) to access data and employs the system of orthogo-nal frequency-divisionmultiplexing (OFDM)modulation [1โ3].The frequency band consists of four groups: A, B, C, andD,with thirteen channels. Each channel bandwidth is 528MHz,as shown in Figure 1.The system operates across a wide rangeof frequency 3.1โ5GHz or 3.1โ10.6GHz. The low frequencyband from 3.1 to 5GHz has been allocated for developingthe first generation of UWB systems [4]. The system hasseveral advantages such as low complexity, low cost, and ahigh data rate for the wireless system. In the front-end systemdesign, a low-noise amplifier (LNA) is the first block in thereceiver path of a communication system.The chief objectiveof the LNA is to reach the low-noise figure to improve theoverall system noise [5]. The LNA must minimize the noisefigure over the entire bandwidth, feature flat gain, goodlinearity, wideband input-output matching, and low power
consumption. In the radio frequency circuit design,GaAs andbipolar transistors have performed fairly well. Nevertheless,these processes lead to increased cost and greater complexity.The RF front-end circuits using CMOS technology canprovide a single-chip solution, which greatly reduces the cost[6]. With the rapid improvement of CMOS technology, itcan implement RF ICs with CMOS. Three major topologiesof the present wideband low-noise amplifiers are used. First,distributed amplifiers [7โ9] can provide good linearity andwidebandmatching but require high power consumption anda large chip area because of the multiple amplifying stages.Second, resistive shunt feedback amplifiers [4, 10] providegoodwidebandmatching andflat gain but require high powerconsumption. Last, Chebyshev filter amplifiers [11, 12] adopt awideband LC bandpass filter for input matching and a sourcefollower for outputmatching.This type of wideband amplifierdissipates a small amount of dc power. The Chebyshevfilter matching design requires three inductors. Therefore, avery large chip area is required. This paper proposes RLCbroadband matching for LNA design. The primary goal isto obtain wideband input-output matching and to reducethe chip area and power consumption. The LNA topology is
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2 Active and Passive Electronic Components
proposed in Section 2. The complete analysis of the designmethodology of the widebandmatching network is presentedin Section 3. In Section 4, implementation and measurementresults are presented. The conclusion is presented in the lastsection.
2. Low-Voltage WidebandCascode LNA Design
Figure 2 shows the basic cascode LNA. The current-reusedconfiguration can be considered as two common-sourceamplifiers (๐
1and๐
2).The current shared cascode amplifier
provides a low-power characteristic under the low voltagesupply. Because ๐
1and ๐
2share the same bias current,
the total power consumption is minimized [6, 12, 16]. For acascode amplifier, the overall noise figure can be obtainedin terms of the noise figure (NF) and gain of each stage asfollows:
๐นtotal = ๐น1 +๐น2โ 1
๐บ๐ด1
, (1)
where ๐นtotal is the total NF and ๐น1 and ๐น2 are the NF values ofthe first and second stage; respectively. ๐บ
๐ด1is the power gain
of this first stage, therefore, in the LNA circuits, the noise ofthe first stage is more critical. Consequently, the gain of thetransistorโs ๐
1must be high enough to suppress the noise.
The dominant noise sources for active MOSFET transistorsare flicker and thermal noises.The flicker noise is modeled asthe voltage source in series with the gate of value:
๐2๐(๐) =
๐พ
๐๐ฟ๐ถ๐๐ฅ๐, (2)
where the constant ๐พ is dependent on device characteristicsand can vary widely for different devices in the same process.The variables ๐, ๐ฟ, and ๐ถ
๐๐ฅrepresent the width, length,
and gate capacitance per unit area, respectively. The flickernoise is inversely proportional to the transistor area,๐๐ฟ. Inother words, a larger device reduces this noise. The noiseprocess of thermal noise is random. The MOSFET thermalnoise includes threemain noise sources, as shown in Figure 3:distributed gate resistance noise (๐2rg), drain current noise(๐2nd), and gate current noise (๐
2
nd).The thermal noise source mainly comes from the drain
current noise and gate-induced noise of the transistors.Multifinger layout technology is applied to reduce the gate-induced noise. According to the MOSFET noise analysis in[17, 18], we must express the noise figure in a way that explic-itly considers power consumption. In order to determine thewidth of๐
1and find out the best NF, we use the dependency
among noise factor (๐น), power dissipation (PD), and qualityfactor (๐
๐ฟ) to find out the optimal value [17]. Based on
these parameters, the related curves, NF (dB) versus ๐๐ฟ, are
plotted in Figure 4 by MATLAB simulation software. Thereis a compromise between noise performance and power gainin Figure 4. Therefore, the ๐
๐ฟcan be determined to be 4, as
PD equals to 10mW. The optimal width of MOSFET can beobtained by [17, 18]:
๐opt =3
2
1
๐๐๐ฟ๐ถ๐๐ฅ๐ ๐๐๐ฟ
โ1
3๐๐๐ฟ๐ถ๐๐ฅ๐ ๐
, (3)
where ๐๐is the operation frequency, ๐ฟ is the gate length, and
๐ถox is the gate capacitance per unit area. Equation (3) showsthat the optimal width for ๐
1is 240๐m, for ๐o = 3.4GHz,
๐ฟ = 0.18 ๐m, ๐ถ๐๐ฅ= 8.62 F/m2, ๐
๐= 50ฮฉ, and ๐
๐ฟ= 4.
Figure 5 shows the proposed UWB LNA schematic.The current sharing cascode amplifier provides low-powercharacteristics with low voltage supply. Note that ๐
2is
the common-gate stage of the cascode configuration, whicheliminates theMiller effect and provides better isolation fromthe output return signal [18, 19]. The passive components ๐ถ
1,
๐ถ2, ๐ถ3, ๐ 1, and ๐ฟ
1are adopted for the matching network
at the input to resonate over the entire frequency band. Theresistors ๐
2and ๐
๐are used to provide bias voltage for the
transistors ๐1and ๐
2. The capacitor ๐ถ
4provides signal
coupling between the two stages. The capacitor ๐ถ5bypasses
the ac current to the ground and avoids coupling to the firststage. This affects gain flatness; therefore, it is possible toprovide an ideal ac ground, but gain flatness is not affectedin the design. ๐ฟ
2is the inductor load of the first stage. The
output matching network is composed of ๐ฟ3, ๐ฟ4, ๐ถ6, and ๐ถ
7.
In the cascode stages, the first stageโs noise figure contributionis more than the second stageโs. The first stageโs transistorsize and bias point should be optimized for the low NF [17].The second stageโs transistor size and bias point should beoptimized for high linearity. The cascode LNA design is fullof trade-offs between optimal gain, low noise figure, inputand output matching, linearity, and power consumption.Moreover, to avoid oscillation, the parasitic couplings haveto be minimized [16]. The sizes of the devices of the LNA areshown in Table 1.
3. Proposed Wideband MatchingNetwork Analysis
3.1. Input Impedance Matching. As shown in Figure 2, in theconventional narrow band LNA design, the input impedanceof the input stage (๐
1) can be written as
๐in = ๐ (๐ฟ๐ + ๐ฟ๐) +1
๐ ๐ถgs1+ (
๐๐1
๐ถgs1)๐ฟ๐ , (4)
where ๐ถgs1 is the gate-source capacitance and ๐๐1 is thetransconductance of the input transistor๐
1. We choose ap-
propriate values of inductance (๐ฟ๐+๐ฟ๐ ) and capacitance๐ถgs1,
which resonate at a certain frequency. The real term can bemade equal to 50ฮฉ. Equation (3) determines the width of๐
1
and finds the best NF. For wideband design, it is difficult tolet the imaginary part of (4) remain zero for a wide range.To discuss UWB input impedance matching, we need toconsider the standard form of the second-order filter:
๐ (๐) =๐2๐2+ ๐1๐ + ๐๐
๐2 + ๐ (๐๐/๐in) + ๐
2
๐
=๐2๐2+ ๐1๐ + ๐๐
๐2 + ๐B + ๐2๐
, (5)
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Active and Passive Electronic Components 3
3432 3960 4488(MHz)
5016 5808 6336 6864 7392 7920 8448 8976 9504 10032
Group A Group B Group C Group D
Figure 1: The IEEE 802.15.3a spectrum.
Table 1: Device sizes of the proposed UWB LNA.
Device ๐ถ1(pF)๐ถ2
(pF)๐ฟ1
(nH)๐ถ3
(pF)๐ 1
(kฮฉ)๐ ๐
(kฮฉ)๐ถ4
(pF)๐ฟ2
(nH)๐ถ5
(pF)๐ 2
(kฮฉ)๐ฟ3
(nH)๐ฟ4
(nH)๐ถ6
(pF)๐ถ7
(pF)๐1-๐2
(๐m)Size 9.51 0.11 0.91 7.6 0.14 3.84 5.7 5.53 5.7 3.84 1.19 0.59 0.16 1.23 240/0.18
M2
M1
Ls
Lg
Vbias
Zin
Figure 2: Cascode low noise amplifier.
V2rg
gsi2ng i
2nd
Cgs rogmV
Rg
Figure 3: CMOS noise model.
where ๐2, ๐1, and ๐
๐are numerator coefficients determining
the type of second-order filter function. ๐๐is called the pole
frequency,๐in is termed the pole factor, and ๐ต is called band-width. According to ๐ต = ๐
๐/๐in, the bandwidth is inversely
proportional to the ๐in if the value of the pole frequency ๐๐is fixed.
0 1 2 3 4 5 6 7 80.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
NF
(dB)
PD = 5 mWPD = 10 mW
PD = 15 mWPD = 20 mW
QL
Figure 4: Noise Figure versus ๐๐ฟfor several power dissipations at
3.4GHz.
R2
L3 L4 C7
C6
C5
C4L2
M2
M1
C3
R1C2
C1 L1
RgVbias
RFin
RFout
VDD
Figure 5: Schematic of the proposed UWB LNA.
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4 Active and Passive Electronic Components
Rs = 50ฮฉ
Vs
Zin
C1
C2
C3
R1
L1Cgs1
Figure 6: The proposed wideband matchingapproximate first stageinput matching small-signal equivalent circuit.
Figure 6 shows the proposed wideband matching and asmall-signal equivalent circuit in the first stage. The inputimpedance of the RLC network can be written as
๐in =1
๐๐ถ1
+1
๐๐ถ2
// [๐๐ฟ1+ (๐ 1+
1
๐๐ถ3
) //1
๐๐ถgs1]
= ๐ in + ๐๐in.
(6)
Because ๐ถ3capacitance value is much larger than ๐ถgs(๐ถ3 โซ
๐ถgs), the (๐ 1 + (1/๐๐ถ3))//(1/๐๐ถgs1) will approximate (๐ 1 +(1/๐๐ถ
3)):
๐in =1
๐๐ถ1
+1
๐๐ถ2
// (๐๐ฟ1+ (๐ 1+
1
๐๐ถ3
)) = ๐ in + ๐๐in,
๐in โ(๐ + (๐
1/๐ฟ1)) ((๐ถ
1+ ๐ถ2) /๐ถ1๐ถ2)
๐2 + ๐ (๐ 1/๐ฟ1) + ((๐ถ
2+ ๐ถ3) /๐ฟ1(๐ถ2๐ถ3)),
(7)
where ๐ถgs1 is the gate-source capacitance of๐1. This equiv-alent circuit can roughly be an RLC second-order filterstructure. In (7), the quality factor of the filter circuit can begiven by
๐๐โ
1
2๐โ
๐ถ2+ ๐ถ3
๐ฟ1(๐ถ2๐ถ3), (8)
๐ โ1
๐ 1
โ๐ฟ1(๐ถ2+ ๐ถ3)
๐ถ2๐ถ3
, (9)
where ๐๐is the resonant frequency. In (8) and (9), to obtain
broader bandwidth, a low-quality factor needs to be added tothe passive devices. The narrowband LNA can be convertedinto a wideband amplifier by properly selecting passivedevices. According to (8), the capacitors ๐ถ
2, ๐ถ3and inductor
๐ฟ1will be optimized at the resonant frequency. We use the
CAD of Agilent ADS to analyze the circuit performanceof input impedance matching. According to (8) and (9), ๐
๐
is inversely proportional to ๐ฟ1while ๐ is proportional to
๐ฟ1. Therefore, bandwidth is inversely proportional to ๐ฟ
1.
Figure 7 is fixed as ๐ถ1(9.51 pF), ๐ถ
2(0.11 pF), ๐ถ
3(7.6 pF), and
๐ 1(140ฮฉ), showing that the resonant frequency is inversely
proportional to the ๐ฟ1. Bandwidth is inversely proportional
3 4 5 6 7 8 9 102 11Frequency (GHz)
โ20
โ25
โ15
โ10
โ5
0
L1 = 1.3nHL1 = 1.15nHL1 = 0.91 nH
S 11
(dB)
Figure 7: Fixed ๐ถ1, ๐ถ2, ๐ถ3and ๐
1and simulated ๐ฟ
1variation input
reflection coefficient.
to ๐ฟ1. According to (9), the input matching circuit uses ๐
1
and ๐ถ3to reduce the number of inductors and make the ๐
factor smaller. ๐ is inversely proportional to ๐ 1. Therefore,
bandwidth is proportional to ๐ 1. Figure 8 is fixed as ๐ถ
1
(9.51 pF), ๐ถ2(0.11 pF), ๐ถ
3(7.6 pF), and ๐ฟ
1(0.91 nH), showing
that the bandwidth is proportional to the ๐ 1. Figure 9 is fixed
as ๐ถ1(9.51 pF), ๐ถ
2(0.11 pF), ๐ถ
3(7.6 pF), and ๐ฟ
1(0.91 nH),
showing that the noise figure is inversely proportional to the๐ 1. ๐๐is inversely proportional to ๐ถ
3while ๐ is inversely
proportional to ๐ถ3. Therefore, the bandwidth is proportional
to the ๐ถ3. Figures 10 and 11 are fixed as ๐ถ
1(9.51 pF), ๐ถ
2
(0.11 pF), ๐ 1(140ฮฉ), and ๐ฟ
1(0.91 nH), showing that the
bandwidth and noise figure are proportional to the ๐ถ3. The
size of the transistors ๐1, ๐ 1, and ๐ถ
3must be carefully
selected. There is a trade-off in the design between widebandmatching and the noise figure. On the other hand, thedevice size must yield a sufficient noise performance andpower gain. As can be seen, the second-order filter of๐ฟ1(0.91 nH), ๐
1(140ฮฉ), and ๐ถ
3(7.6 pF) is employed as the
inputmatching network.The inputmatching network has lesscomplexity and better reflection coefficient from 3GHz to 10GHz.
3.2. Output Impedance Matching. Low-noise amplifiers relyon output impedance matching to achieve maximum powergain. A source follower technique has been widely used toprovide wideband output matching. Output impedance issimilar to 1/๐
๐๐ , in which ๐
๐๐ is a gate-source transconduc-
tance of the source follower [4, 10]. However, it consumesmore power. For these reasons, we introduce an outputimpedance matching method suitable for wideband LNAdesign. Figure 12 shows the approximate output impedancesmall-signal equivalent circuit. The output impedance of theRLC network can be written as
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Active and Passive Electronic Components 5
1 2 3 4 5 6 7 8 9 100 11Frequency (GHz)
R1 = 500 ฮฉR1 = 140 ฮฉR1 = 50 ฮฉ
โ20
โ15
โ10
โ5
0
S 11
(dB)
Figure 8: Fixed ๐ถ1, ๐ถ2, ๐ถ3, and ๐ฟ
1and simulated ๐
1variation input
reflection coefficient.
1 2 3 4 5 6 7 8 9 100 11
3456789
2
10
Frequency (GHz)
NF
(dB)
R1 = 500 ฮฉR1 = 140 ฮฉR1 = 50 ฮฉ
Figure 9: Fixed ๐ถ1, ๐ถ2, ๐ถ3, and ๐ฟ
1and simulated ๐
1variation noise
figure.
๐out = ((๐๐2//1
๐๐ถ๐2
//๐๐ฟ3) + ๐๐ฟ
4) //
1
๐๐ถ6
+1
๐๐ถ7
= ๐ out + ๐๐out,
(10)
๐out
โ(๐+(1/๐
๐2๐ถ๐2)) ((๐ถ
7+๐ถ6) /๐ถ6๐ถ7)
๐2+๐ (1/๐๐2๐ถ๐2)+((๐ถ
6๐ฟ4+๐ถ6๐ฟ3+๐ถ๐2๐ฟ3) /๐ถ๐2๐ถ6๐ฟ3๐ฟ4),
(11)
where ๐ถ๐2
is the parasitic capacitance of ๐2at the drain
node, and ๐๐2
is the resistor of ๐2at the drain node. This
equivalent circuit can approximately be an RLC second-order
1 2 3 4 5 6 7 8 90 10 11Frequency (GHz)
โ20
โ15
โ10
โ5
0
C3 = 7.6 pFC3 = 0.95 pFC3 = 0.15 pF
S 11
(dB)
Figure 10: Fixed๐ถ1,๐ถ2,๐ 1, and ๐ฟ
1and simulated๐ถ
3variation input
reflection coefficient.
1 2 3 4 5 6 7 8 90
3456789
2
10
10 11
NF
(dB)
Frequency (GHz)C3 = 7.6 pFC3 = 0.95 pFC3 = 0.15 pF
Figure 11: Fixed๐ถ1,๐ถ2,๐ 1, and ๐ฟ
1and simulated๐ถ
3variation noise
figure.
filter structure. In (11), the quality factor of the filter circuitcan be given by
๐๐โ
1
2๐โ๐ถ6๐ฟ4+ ๐ถ6๐ฟ3+ ๐ถ๐2๐ฟ3
๐ถ๐2๐ถ6๐ฟ3๐ฟ4
, (12)
๐ โ ๐๐2๐ถ๐2โ๐ถ6๐ฟ4+ ๐ถ6๐ฟ3+ ๐ถ๐2๐ฟ3
๐ถ๐2๐ถ6๐ฟ3๐ฟ4
, (13)
where ๐๐is the resonant frequency. In (12) and (13), in order
to obtain broader bandwidth, a low-quality factor needs tobe added to passive devices. ๐
๐is inversely proportional to
๐ถ๐2
while bandwidth is inversely proportional to ๐๐2๐ถ๐2. The
๐ถ๐2capacitance is proportional to the width of the transistor.
The ๐๐2
resistance is inversely proportional to the width ofthe transistor. UWB system uses a specific frequency band
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6 Active and Passive Electronic Components
Rs = 50 ฮฉ
outZ
C7L4
C6L3rd2 Cd2
Figure 12: The wideband output matching small-signal equivalentcircuit.
2 3 4 5 6 7 8 9 10 11Frequency (GHz)
โ30
โ20
โ25
โ15
โ10
โ5
0
S 22
(dB)
M2 = 160/0.18 ๐mM2 = 240/0.18 ๐mM2 = 320/0.18 ๐m
Figure 13: Fixed ๐ถ6, ๐ถ7, ๐ฟ3and ๐ฟ
4, simulated๐
2variation output
reflection coefficient.
(3.1โ5GHz or 3.1โ10.6GHz) to access data. Generally theresults of UWB return loss are less than โ10 dB. The powerconsumption is proportional to the width of the transistor.According to (13), the narrowband LNA can be convertedinto a wideband amplifier by properly selecting ๐
2. This
provides good wideband matching and flat gain. Figure 13is fixed as ๐ถ
6(0.16 pF), ๐ถ
7(1.23 pF), ๐ฟ
3(1.19 nH), and ๐ฟ
4
(0.59 nH), showing that the simulated ๐2variation output
reflection coefficient. As can be seen, the second-order filterof ๐2(240/0.18 ๐m) is employed as the output matching
network. According to (12), the capacitor ๐ถ6and inductor
๐ฟ3-๐ฟ4will be optimized at the resonant frequency. Figure 14
is fixed as ๐ถ6(0.16 pF), ๐ถ
7(1.23 pF), ๐
2(240/0.18๐m),
and ๐ฟ4(0.59 nH), showing that the resonant frequency is
inversely proportional to the ๐ฟ3. The output network has less
complexity and a better reflected coefficient from 3.1 GHz to10.6GHz.
3.3. Gain. At a high frequency, the current gain of commonsource configured transistor ๐
1is ๐๐1/๐ ๐ถgs1 [11]. Figure 15
shows the approximate first stage ๐1load of the simplified
2 3 4 5 6 7 8 9 10 11Frequency (GHz)
โ20
โ15
โ10
โ5
0
S 22
(dB)
L3 = 1.19nHL3 = 1.29nH
L3 = 1.09nH
Figure 14: Fixed ๐ถ6, ๐ถ7, ๐2, and ๐ฟ
4and simulated ๐ฟ
3variation
output reflection coefficient.
Id1
Vout1
C5
L2
Cd11/gm2
Figure 15: The first stage simplified small-signal equivalent circuit.
small-signal equivalent circuit [12]. The first stage load canbe approximated as
๐load1 โ1
๐๐ฟ2๐ถ5๐ถ๐1
โ ๐2๐ฟ2๐ถ5+ ๐๐๐2๐ฟ2+ 1
๐2 + ๐ (๐๐2/๐ถ5) + ((๐ถ
5+ ๐ถ๐1) / (๐ฟ2(๐ถ5๐ถ๐1)))
.
(14)
The transfer function of the input matching network is๐in(๐ ). The first stage voltage gain can be approximated as
๐ด๐1
โ โ๐๐1๐in
๐๐ถgs1๐ ๐ โ ๐Load1, (15)
where ๐ ๐ is the source resistance, in which voltage gain is
determined by the load inductor ๐ฟ2, the total capacitance at
the drain of ๐1, and bypass capacitor ๐ถ
5. The second stage
voltage gain can be estimated as
๐ด๐2
โ โ๐๐2
๐๐ถgs2๐ in2โ ๐out, (16)
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Active and Passive Electronic Components 7
1 2 3 4 5 6 7 8 9 10
10
12
110
2
4
6
8
0
Frequency (GHz)
ฮ
K
ฮ K
Figure 16: ๐พ (stability factor) and ฮ simulation result.
where ๐ in2 is the input resistance at the gate of๐2 and ๐outis the output load impedance. The LNA voltage gain can beapproximated as
๐ด๐โ๐๐1๐๐2๐in๐Load1๐out
๐2๐ถgs1๐ถgs2๐ ๐ ๐ in2. (17)
According to (17), the narrowband LNA can be convertedinto a wideband amplifier by properly selecting the devicesize. This provides good wideband matching, noise figureoptimization design, and flat gain.
3.4. Stability. Amplifier stability is important to preventnatural oscillation. The stability can be determined by the ๐-parameters, input and output matching network, and circuitterminations. The simpler tests show whether or not a devicecan be unconditionally stable [20]. One of these is the ๐พ-ฮtest, which shows if a device can be unconditionally stable ifRolletโs condition is defined as
๐พ =1 โ
๐112
โ๐22
2
+ |ฮ|2
2๐12๐21
> 1,
|ฮ| =๐11๐22 โ ๐12๐21
< 1.
(18)
The ๐พ-ฮ test cannot be used to compare the relative sta-bility of twoormore devices because it involves constraints ontwo separate parameters. A new criterion has been proposedwhich combines the ๐-parameters in a test involving only asingle parameter, ๐, defined as
๐ =1 โ
๐112
๐22 โ ฮ๐โ
11
+๐12๐21
> 1. (19)
The performance of LNA is simulated using the advancedesign system (ADS). To consider the stability of the LNA,the๐พ factor andฮ should be considered. Figure 16 shows thatthe ๐พ factor is always larger than 1 and the ฮ is smaller than1 all the time. Hence, this circuit is stable for all frequency
1 2 3 4 5 6 7 8 9 10 11
11
0Frequency (GHz)
3
5
7
9
1
๐load๐source
๐lo
ad๐
sour
ce
Figure 17: ๐ factor simulation result.
RFin RFout
Vbias
DC power supply(HP 4142B)
Network analyzer(HP 8510C)
S-parameters test set(HP 8517B)
VDD
Figure 18: Test setup used for the LNA ๐-parameter measurements.
bands. There is another way to diagnose whether the LNAis unconditionally stable. The ๐ factor at input and outputshould be larger than 1 in all frequency bands, as shown inFigure 17.
4. Measurement Results
On-wafer measurement is performed by an HP 8510C net-work analyzer and an HP 8517B is to test the ๐-parameter, asshown in Figure 18. A network analyzer is used to measurethe frequency response and input matching of the LNA.The input and output impedance matchings are both 50ฮฉ.To ensure that the LNA still provides a conversion gainwhen process deviation occurs, the other process corners,namely, typical-NMOS typical-PMOS (TT), fast-NMOS fast-PMOS (FF), and slow-NMOS slow-PMOS (SS), are also usedto simulate this LNA. The results are shown in Figure 19.Figure 19 shows the measurement forward gain (๐
21), from
3 to 5.6GHz; the measured gain is about 7โ10 dB. Themeasurement data is 6 dB lower than the pre-simulation data
-
8 Active and Passive Electronic Components
201816141210
86420
2.8 3.3 3.8 4.3 4.8 5.3 5.8Frequency (GHz)
S 21
(dB)
Pre-simulation TTPost-simulation FFPost-simulation TT
Post-simulation SSMeasurement
Figure 19: Simulation and measurement results of conversion gainversus frequency of the proposed LNA. RF stands for simulationresults of modifiedMOSFET RFmodel. FF, TT, and SS represent thesimulation results of fast-NMOS fast-PMOS, typical-NMOS typical-PMOS, and slow-NMOS slow-PMOS process corners.
โ10
0
โ20
โ30
โ40
MeasurementSimulation
3 4 5 6 7 8 9 102 11Frequency (GHz)
S 11
(dB)
Figure 20: Measured and simulated input reflection coefficient.
because of the process drift and parasitic effect in the layout.The result of conversion gain measurement is situated in thepost-simulation (SS) process corner.
Figure 20 shows the input return loss (S11), from 3 to
11 GHz, and the measured S11
is less than โ9 dB. Figure 21shows output return loss (S
22), from 3 to 7.5GHz, and the
measured S22is less thanโ8 dB. Figure 22 shows reverse isola-
tion (S12), from 3 to 11 GHz, and the measured S
12is less than
โ40 dB. The simulation measured input of 1 dB compressionpoint at 5.3 GHz shown in Figure 23 is about โ8 dB. Figure 24shows the measured fundamental output power and third-order intermodulation (IM3) for the RF input frequencyspacing of 1MHz, and the IIP
3is 2 dBm at 5.3 GHz. The
IM3 is measured, using two Agilnet E8247C continuous wave
โ10
โ15
โ20
โ25
โ30
โ5
0
MeasurementSimulation
3 4 5 6 7 8 9 102 11Frequency (GHz)
S 22
(dB)
Figure 21: Measured and simulated output reflection coefficient.
โ10
โ20
โ30
โ40
โ50
โ60
โ70
โ80
MeasurementSimulation
3 4 5 6 7 8 9 102 11
0
Frequency (GHz)
S 12
(dB)
Figure 22: Measured and simulated reverse isolation.
โ40 โ35 โ30 โ25 โ20 โ15 โ10 โ5 0 5 100
1
2
3
4
5
6
7
8
9
Gai
n (d
B)
Input power (dBm)
P1dB
Figure 23: Measured input 1 db compression point at 5.3 GHz.
-
Active and Passive Electronic Components 9
Table 2: Recently reported performance of UWB LNAs.
Reference Process CMOS(๐m) ๐11 (dB)Avg. gain
(db) Freq. (GHz) NF (dB) IIP3 (dBm)Die area(mm2)
Power(mW) Topology
[4] 0.18๐m CMOS
-
10 Active and Passive Electronic Components
reduces the chip area. References [10โ12] only show core LNApower consumption, excluding the output source follower.References [14, 15] only show simulation. It should show thelinearity of the experiment. The LNA plays an important rolein improving overall system linearity. Table 2 clearly showsthat the proposed LNA has a very small chip area and thelowest power consumption.
5. Conclusion
An RLC wideband matching UWB LNA has been presentedin the above results, which can operate at a supply of 1 Vvoltage in a 0.18 ๐m CMOS technology. In the proposedtopology, the narrowband LNA can be converted into awideband amplifier by the RLC matching method. The RLCinput matching circuit reduces the chip area. The outputmatching method reduces power consumption. The mainadvantages of the LNA topology are low power use, moderatenoise, linearity, power gain, and a small chip area.
Conflict of Interests
The author indicated no potential conflict of interests.
Acknowledgments
The author would like to thank the National Science council(NSC) for funds support NSC99-2221-E-507-005 and theNational Chip Implementation Center (CIC) for technicalsupport.
References
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[8] X. Guan and C. Nguyen, โLow-power-consumption and high-gain CMOS distributed amplifiers using cascade of inductively
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[11] A. Bevilacqua and A. M. Niknejad, โAn ultrawideband CMOSlow-noise amplifier for 3.1-10.6-GHz wireless receivers,โ IEEEJournal of Solid-State Circuits, vol. 39, no. 12, pp. 2259โ2268,2004.
[12] M.-F. Chou, W.-S. Wuen, C.-C. Wu, K.-A. Wen, and C.-Y. Chang, โA CMOS low-noise amplifier for ultra widebandwireless applications,โ IEICE Transactions on Fundamentals ofElectronics, Communications and Computer Sciences, vol. e88-A, no. 11, pp. 3110โ3117, 2005.
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