recent developments in saw devices

Recent developments in SAW devicesM.F. Lewis, M.A., D.Phil., F.lnst.P., C.L. West, M.A., D.Phil., J.M. Deacon,

B.Sc, Ph.D., and R.F. Humphryes, B.Sc.(Eng.), M.Sc, Ph.D., C.Eng., F.I.E.E.

Indexing terms: Ultrasonics, Surface-acoustic-wave devices (SAW)

Abstract: Surface-acoustic-wave (SAW) delay lines employing photolithographically fabricated transducerswere first described in 1965. Since then there has been a continuous stream of publications on the physics ofSAW propagation, and on the applications of these planar acoustic elements as high-frequency electrical com-ponents. By 1975 the principles of many potentially useful SAW devices had been described, and the firstdispersive delay lines were being employed in pulse compression radar. In the paper the advances made in morerecent years are reviewed, which have seen the development of numerous SAW devices for military and civilapplications, and the realisation of their mass-production capability in the TV IF filter. The paper alsodescribes the latest trends towards smaller and cheaper devices, lower insertion losses and more sophisticatedperformance, which will be the hallmarks of future generations of SAW devices.

1 Introduction

Historically, the ability of a semi-infinite isotropic solidsubstrate to support a mechanical disturbance localised atits surface was first demonstrated theoretically by LordRayleigh (circa 1885) in connection with his investigationsinto earthquakes. Such waves are known as Rayleighwaves, but their counterparts on anisotropic, and, in parti-cular, on piezoelectric, substrates are more commonlydescribed as surface acoustic waves (SAW). The particledisplacements involved usually resemble those obtaining inthe more familiar case of wave propagation on the surfaceof the sea: thus the mechanical disturbance is concentratedwithin a few wavelengths of the surface, and, for a forwardpropagating wave, the motion of a particle at the surface isretrograde elliptical. This description provides a conve-nient picture to bear in mind and, as discussed in the fol-lowing, is usually quite adequate to permit an appreciationof the principles of operation of SAW devices.

Although bulk acoustic wave (BAW) devices have longplayed an important role in signal processing in suchdiverse applications as delay lines, frequency filters and theubiquitous quartz-crystal oscillator, SAW devicesremained virtually unexploited until White and Voltmer[1] demonstrated the ability of the interdigital transducer(IDT) to excite SAW efficiently and selectively onpiezoelectric substrates such as single-crystal quartz. (Asimple SAW device employing IDTs is shown schemati-cally in Fig. 1 of Section 2.) Such devices offer three impor-tant attractions:

(a) SAW devices operate in the frequency range from~ 10 MHz to > 1 GHz, and so fill an awkward gapbetween the 'natural' regimes of lumped-component andmicrowave technologies. Further, SAW device bandwidthsrange from ~ 10 kHz to > 500 MHz, and so embracemany of the requirements of modern communications andradar industries.

(b) In a SAW device, the signal (in acoustic form) is con-centrated at the substrate surface, and is, therefore, readilyavailable to be tapped, guided, reflected, focused, absorbedetc. This leads to an extremely wide device-design flex-ibility, a point which will be a recurring theme throughoutthis review. The range of SAW devices demonstrated is

Paper 3163A, received 2nd March 1984Dr. Lewis and Dr. West are with the Royal Signals & Radar Establishment, St.Andrews Road, Great Malvern, Worcs. WR14 3PS. Dr. Deacon and Dr. Hump-hryes are with Signal Technology Ltd., Crompton Road, Groundwell IndustrialEstate, Swindon, Wilts., England

already diverse and there is still abundant scope forimprovements in performance (in the widest sense, e.g.including size, cost etc.), and for the use of SAW devices toperform ever more sophisticated functions.

(c) SAW device fabrication exploits the technique ofphotolithography developed by the semiconductorindustry. In mass production, SAW devices are, therefore,cheap, reproducible, rugged, planar components, and arequite suitable for integration with modern microelectroniccircuitry.

Once these points were appreciated (circa 1970) manygroups began an intensive worldwide investigation intoSAW devices, which has continued unabated to date; andwhich will doubtless continue for years to come. The firstSAW component to be exploited was the dispersive delayline which is used for the generation and matched filteringof the swept-frequency 'chirp' waveforms employed inpulse-compression radar. In the past decade, this com-ponent has established itself as a vital component at theheart of many modern radar systems. In the radar andcommunications industries, SAW bandpass filters havefound numerous applications; and the SAW TV IF filternow enjoys a market of many millions per annum, havinguniversally replaced its voluminous lumped-componentpredecessor. SAW delay lines have found various morespecialised applications, while narrowband delay lines andSAW resonators are beginning to find significant applica-tions as frequency-control elements.

Not surprisingly, the literature on SAW, on SAWdevices and on their applications is extensive. A specialIEE publication edited by Morgan [2] has reprinted 100of the key papers up to 1976, and several recent specialistbooks have a significant SAW content, Auld [3], Die-ulesaint and Royer [4], Matthews [5] and Oliner [6].

Annual developments are largely covered in the Pro-ceedings of the IEEE (USA) conferences on ultrasonics,1972 to date. In addition, over the years, a number ofexcellent reviews have appeared, especially those by White[7], Maines and Paige [8, 9] and Ash (Reference 6, Chap.4). The authors' intention in this present review is to avoidunnecessary repetition of the content of these earlierreviews, but rather to concentrate on more recent develop-ments and achievements, and, finally, to speculate on thedirections of future SAW research and development activ-ities. We have, therefore, assumed some degree of familiar-ity with the basic aspects of SAW devices and applicationsas outlined, for example, by Maines and Paige in an earlierIEE review [8].

186 IEE PROCEEDINGS, Vol. 131, Pt. A, No. 4, JUNE 1984

2 Review of SAW devices employing interdigitaltransducers (IDTs)

The simple SAW device shown schematically in Fig. 1employs unweighted IDTs, i.e. IDTs comprising metallic

earthedscreenbetweentransducers

out

substrate endsare angled andabsorber addedto eliminateunwanted SAW

following the frequency response of the device of Fig. 1would be dominated by the input IDT, because it hasmore periods (finger pairs) than the output IDT.

Let us now consider the acoustic and electrical proper-ties of the IDT. The problem is best tackled by consideringthe consequences of applying to the terminals of the inputIDT an RF voltage Vo from an electrical source of verylow impedance, Fig. 2a. In these circumstances, the voltageacross the IDT is frequency-independent and substantiallyequal to Vo. This voltage generates an (essentiallyelectrostatic) spatially and temporally periodic electric-fieldpattern close to the substrate surface, which in turn causesa periodic distortion of the substrate surface through the(inverse) piezoelectric effect. Thus, each period of the IDTacts as an elementary source of SAW, generating waveletsof amplitude a (ocFo), which propagate away from the

period P

piezoelectricsubstrate

._ bottom roughenedto scatter bulk waves

SAW SAW Ga(f)[] Ba(f)

Fig. 1 Schematic of SAW device employing interdigital transducers(IDTs)

In this device all finger pairs have the same overlap A which defines the aperture ofthe acoustic beam

fingers of constant overlap (A) which excite and receiveSAW over a uniform acoustic aperture A. In a typicaldevice, A is made many wavelengths wide (say ~ 50 l0) toensure the excitation of a substantially plane wave, and soavoid the complications of acoustic diffraction. As eachIDT is bidirectional, a SAW device of this type incurs aminimum loss of 6 dB when the IDTs are matched to thesource/load. In practice, such devices are often operateduntuned and incur a loss of 20-30 dB. As discussed in the

Table 1 : Basic properties of some SAW substrates

Ra(f)

Xa(f)

Fig. 2 (a) Schematic of IDT driven from voltage source Vo of internalimpedance Z o , (b) and (c) show shunt and series equivalent circuits of theIDT, each representation being appropriate in specific-devices

As discussed in the text, the limiting case Zo —• 0 has a special significance in rela-ting the impulse response to the IDT geometry

Substrate(cut and propagation)direction)

QuartzST-cut, X-propagationLiNbO3

K-cut, Z-propagation128°-rotated /-cut,X-propagation

LiTaO3

K-cut, Z-propagationX-cut, propagation112° from V towards Z

BGO(110)-cut.<001 >-propagation

AIP04

K-cut, X-propagation

Li2B407

X-cut, Z-propagation

FreesurfacevelocityV

m/s3157

34883977

32543288

1620

2750

3515

Piezoelectriccouplingcoefficient/r2

%0.14

55.6

0.740.6

0.85

0.3

1

Optimum numberof finger pairsper transducer'"opt

23

44

1011

10

16

9

Optimumfractionalbandwidth

%4

2527

109

10

6

11

1 st-ordertemperaturecoefficientof delay

*106/deg C0

9172

3518

130

0

0

2nd-ordertemperaturecoefficientof delay

x109/(deg C30

——

——

—

270

230

Comments

\ 2

Available up to25 cm long

Available up to25 cm long.3 in (7.62 cm) discs arereadily availableand cheap.Widely used forTV filters.

Little used.except byToshiba

Suitable forlong delays[35]

Just becomingavailablecommercially

Just becomingavailablecommercially.Most suitable fordelay lines.

IEE PROCEEDINGS, Vol. 131, Pt. A, No. 4, JUNE 1984 187

source in both directions at a velocity V which is typically3000 m/s (Table 1). At a certain synchronous frequency f0,the SAW wavelength Xo (= V/f0) is just equal to thespatial period P of the IDT; at this frequency the waveletsgenerated by each period of the IDT interfere construc-tively, leading to a strong net excitation of SAW. To agood approximation we may neglect regeneration withinthe individual transducers, if they satisfy the conditionNk2 < 1, where k2 is the piezoelectric coupling constant ofthe substrate concerned, Table 1. Thus at the synchronous(or centre) frequency f0 the emergent SAW amplitude isA' ~ Na, while at other frequencies the interference is notfully constructive and the net SAW excitation is corre-spondingly weaker. If this SAW generation is monitoredby a broadband output transducer, such as the single-finger-pair IDT of Fig. 1, the frequency response of thiselementary SAW bandpass filter takes the form

\H(f)\ oc(sinx/x) (1)

where

x = Nn(f-fo)/fo

This result is directly analogous to the response of anoptical diffraction grating comprising N slits, which latterresponse arises from the interference of N sources of lightwith equal incremental delays, e.g. Jenkins and White [10].As in the optical case, we have neglected the frequencyresponse of the individual SAW sources (periods of theIDT). This is usually quite justifiable, as the array factor ofeqn. 1 is invariably narrower than the element factor.Notice that the frequency response does not depend on thedetailed nature of the wave, e.g. on the particle motion.

The conventional treatment of the optical diffractiongrating leading to eqn. 1 is performed in the frequencydomain [10]. It turns out, however, that in the case ofSAW devices employing IDTs, greater physical insight isobtained by treating the problem in the time domain. Thisarises because when it is connected to a low-impedanceelectrical source/load the impulse response h(t) of an IDTclosely resembles its physical construction, Hartmann et al.[11]. Of course H(f) and h(t) are related through theFourier transform, so either can be determined uniquely ifthe other is known. To illustrate these points, imagine thatthe input IDT of Fig. 1 is impulsed by a very short electri-cal pulse approximating a ^-function in time. As discussedearlier, the electric field established instantaneously in eachgap of the IDT acts as a source of surface acoustic waves.These mechanical disturbances travel away from eachsource in a manner reminiscent of the ripple generated bydropping a stone in a pond. These wavelets in turn gener-ate a voltage as they pass through the output transducer ofFig. 1. Ignoring for the moment the finite extent of theoutput IDT, it is clear that the output waveform (i.e. theimpulse response h(t) of the input transducer) comprises anRF tone burst of duration N cycles at the fundamentalfrequency f0 = V/P with, in general, some harmoniccontent. As most SAW devices operate near the fundamen-tal frequency f0 with a fractional bandwidth A///o < 50%,we shall not usually be concerned with the harmonicresponses, details of which depend upon the internal struc-ture of each period of the IDT, Fig. 3. In the neighbour-hood of/0 the Fourier transform of h(t) is of the (sin x/x)form, reproducing eqn. 1 above. Of course SAW devicesare reciprocal, so this result could equally well have beenderived by reversing the roles of the input and outputIDTs.

In the preceding discussion we were concerned pri-marily with the response of the input IDT, and so ignored

the finite extent (and finite bandwidth) of the output IDT.In a real SAW device, the input and output IDTs are often

uu uu

Fig. 3 Some useful IDT geometriesa Normal IDT with finger and gap widths of A/4 (this suffers from reflections at thecentre frequency); b 'split-finger' or 'double-electrode' transducer with finger andgap widths of A/8 to eliminate the reflections in a (this pattern also displays a third-harmonic response [31]); c '2 into 1' geometry with finger and gap widths of A/6(this eliminates reflections and possesses a second harmonic response [32]) d three-phase transducer providing unidirectionality but requiring crossover technology[21, 23]; e 'bloomed' transducer employing A/4-offset strips to cancel reflectionsfrom the IDT fingers [33]; / single-phase unidirectional transducer (SPUDT)employing evaporated A/8-offset reflectors [22]; and g group-type version o f /requiring only single-stage fabrication [24]

of comparable physical dimensions and /i(t)-duration, andso the effects of both IDTs must be included in the overallresponse. In the simplest case (of at least one unweightedIDT, see Section 4.1), the overall frequency response is theproduct of the individual responses H(f) = Hl(f)H2(f),and the overall impulse response is the convolution of theindividual impulse responses, h(t) = h^t) * h2{t).

For future reference we shall extend the result of eqn. 1to the more general IDT structure illustrated schematicallyin Fig. 4. In this Figure, the central transducer 2 isweighted by apodisation, i.e. by varying the finger overlapA(x) with position x. Other weighting schemes aredescribed in Section 4. Further, the periodicity P isallowed to vary with x, as in the dispersive transducers ofSection 6. We shall consider the responses of IDT 2 whenemployed in conjunction with either the broadband IDT 1located to its left-hand side, or with IDT 3 located to itsright-hand side. As in the cases of the optical diffractiongrating, and the SAW device of Fig. 1, we shall ignore thefrequency response of the individual periods of the IDTs ofFig. 4, and for simplicity consider each acoustic source ofamplitude An to be localised at the centre of each livefinger of the IDT located at xn (n = 1, 2, 3, ..., N). Theimpulse responses (which may be obtained experimentally,e.g. by impulsing IDT 2) may, therefore, be approximated


mathematically by

hzi(t)~TtAmfi{t-xJV)

(2)

n

1

_L

- x —

)x,

Ir

rmi

x 2 x'3 x<

l l l rII11AW

•

L-X —

<3>rt]

11_

p.

iJL

time time

Fig. 4 (a) Schematic of generalised SAW transducer IDT 2 with vari-able overlap A(x) and (exaggerated) variation in period P(x), (b) schematicimpulse response of IDT 2 employed in conjunction with IDTs I or 3; notethat h2l(t) is the time reverse ofh23(t)

The relationship between IDT geometry and h(t) is discussed in more detail in Ref-erence 11. This Figure also illustrates the usefulness of SAW devices for the gener-ation of sophisticated waveforms with time duration up to ~ 100 /is or bandwidthsup to ~200 MHz. In the text we have lumped the acoustic sources in the gapseither side of a live finger into one equivalent source located at the centre of the livefinger

The frequency responses are given by the Fouriertransforms of the impulse responses

H(f) =

Thus

(-j2nft) dt (3)

(4)

and

H23(f) * {-j2nf(L - xn)/V}

(5)

where the separation of IDTs 1 and 3 is L = V/x. Notethat, apart from the constant group delay x, H2i and H23

are complex conjugates, a result of importance in Section 6on matched filters. It will also be noticed that, if transducer2 is symmetrical, the responses of eqns. 4 and 5 becomenondispersive, a common requirement in bandpass filters,see Section 4. Finally, we mention that the result of eqn. 4could have been derived equally easily in the frequencydomain by feeding IDT 1 with a signal of frequency / andsumming the voltages monitored by the live fingers of IDT2, assuming weak coupling to the output circuit.

The frequency responses of eqns. 1, 4 and 5 are knownas the acoustic responses of the transducers, because theyarise as a result of acoustic interference. They describe thefrequency response of a SAW device when the transducersare connected to a source/load of low impedance Zo rela-tive to the electrical impedance of the transducers Z(/).However, such a device would necessarily display a highinsertion loss owing to the impedance mismatch, and, inpractice, one invariably makes the impedance of eachtransducer comparable to Zo (usually 50 Q) to obtain anacceptable loss, say < 25 dB. The combination of Z o ,Z(/) and any tuning or matching elements introduces anelectrical frequency response to each IDT; and, to evaluateits effect on the overall response, it is obviously necessary

to know Z(/). In the literature, Z(/) is usually representedeither by the shunt equivalent circuit of Fig. 2b, or by theseries equivalent circuit of Fig. 2c, and is often dominatedby the static capacitance CT. The resistive elements Ga(f)or Ra(f) arise from the acoustic power generation, whereasthe reactive terms Ba(f) or Xa(f) are demanded by cau-sality, Nalamwar and Epstein [12]. In the case ofunweighted IDTs, e.g. Fig. 1, CT is given by CT = NCS

where Cs is the capacitance per finger pair. We can readilydeduce the form of Ga(f) from the earlier discussion, byequating the SAW power generated Pa oc (Na)2(sin x/x)2

to the electrical power dissipated Pe = VlGa, Fig. 2.Recalling that a oc Vo, we see that

GJJ) oc N2(sin x/xf (6)

The absolute value of Ga depends on the choice of sub-strate material and orientation. A convenient expressionfor its peak value is

Q tf\ _ gy Q N2k2 (7)

Values of k2 for the most commonly used SAW substratesare included in Table 1, and more extensive tables havebeen generated by Slobodnik [13]. The reactive part ofZ(f) is related to the real part through the Hilbert trans-form [12] and, for an unweighted IDT, is given by Smithet al. [14] as

Ba(f) = GJLf0)sin 2x — 2x

2x~2 (8)

The expressions for R^f) and Xa(f) are a little more com-plicated, and may be derived by equating the real and ima-ginary parts of the impedances of the equivalent circuits ofFigs. 2b and c. For example, Ra{f) is given by

= Ga(f)\Z(f)\: (9)

It frequently happens that Z is dominated by the IDT'stotal static capacitance C r = NCS and is, therefore, givenby Z ~ (jcoCT)~l. In these circumstances, Ra is indepen-dent of N and given by

RJJ) * (10)

This result is useful as it shows that IDTs with differentnumbers of finger pairs N can be matched to the samesource/load impedance (e.g. 50 Q) by tuning each with asingle inductor of appropriate value.

Furthermore, series-tuning is often especially conve-nient because an Ra(f0) of 50 ft holds for an acoustic aper-ture A ~ 50 Xo, which is suitable for other reasons,principally to limit acoustic diffraction.

As discussed above, the overall frequency response of aSAW device requires inclusion of the acoustic responseand the electrical response. An important result to emergefrom these considerations is that there is an optimum(maximum) fractional bandwidth that any givenpiezoelectric substrate can sustain without incurring anexcess insertion loss; i.e. over and above the minimumvalue of 6 dB arising from bidirectionality. This result isreadily derived by introducing the acoustic quality factor,Qa = N, from eqn. 1. The electrical quality factor of atuned IDT is, from eqn. 7,

a>CT _ n

Gn ~ ANk2 (11)

Notice that Qe has an inverse dependence of N, in contrastto Qa which is proportional to N. As the overall bandwidth


is determined by the higher Q, the optimum bandwidthholds for Qa = Qe, when N = Nopt = (n/4k2)i/2. Values ofNopt and the corresponding fractional bandwidth N~p] aregiven in Table 1. It should be emphasised that larger band-widths can always be produced (e.g. by using IDTs withsmall values of N, or the dispersive IDTs of Section 6), butone cannot simultaneously optimally match such IDTs(without reducing the bandwidth!), so an increased inser-tion loss must be accepted. This result is analogous to theBode-Fano theorem for RC circuits. A technique which issometimes useful is to leave the IDTs untuned and adjustthe aperture (A in Fig. 1) for minimum loss; it is straight-forward to show that this requires \Z\ to be made equalto the source/load impedance (assumed real).

Although the preceding discussion gives an excellentfirst-order approximation to the frequency response of asimple SAW device, in practice there are numeroussecond-order complications which need to be 'designedout' when devices are aimed at tight specifications. Theseproblem include: (a) the slowing and reflection of SAW bythe IDT fingers, (b) acoustic diffraction and (c) the excita-tion of unwanted bulk acoustic waves. These effects can allbe reduced to acceptable levels, for example by modifi-cations to the IDT patterns and by the correct choice ofpiezoelectric substrate and its back face treatment, Fig. 1.It is also amusing to note that most of these problematiceffects have subsequently been turned to advantage inother devices, e.g. reflections in SAW resonators [19] andbulk waves in SSBW devices, Section 7.

Undoubtedly one of the most serious problems encoun-tered in SAW devices arises from the triple-transit (andhigher-order) signals which derive from multiple reflectionsof SAW between the input and output IDTs. There arevarious causes of such reflections; for example, even asmall perturbation of the SAW velocity by the IDT fingerscan cause strong reflections at the centre frequencybecause the periodicity (A/2 separation) satisfies the Braggcondition for normal-incidence reflection. The physicalmechanism of the reflection may be through mass loading,topography or piezoelectric shorting. This particularproblem is largely overcome by the use of split-fingertransducers, Fig. 3b, or the other antireflecting designs ofFigs. 3c and e. However, even when such reflections areovercome, one is left with a reflection arising from the verynature of the IDT—in a lossless symmetrical 3-port junc-tion (with one electrical and 2 identical acoustic ports) it isimpossible to match all 3 ports simultaneously [15]. ThisSAW problem has been described by various workers[14]; including Rosenberg [16], whose recent extensiveanalysis derives many other basic properties of IDTs fromfundamental considerations. Assuming an otherwise loss-less device, a plot of the minimum possible level of reflec-tion against insertion loss is provided in Fig. 5a, while theresulting phase and amplitude ripple are shown in Fig. 5b.This shows that the problem is especially severe in devicesexhibiting the lowest losses of ~6 dB. It is primarily forthis reason that, to date, many SAW devices have beendesigned with an insertion loss of order 20 dB. It shouldalso be noted that, for various reasons, the level of triple-transit signal obtained in practice is higher than theminimum possible level shown in Fig. 5b by up to 10 dB.Various schemes have been devised to reduce the effect ofthis triple-transit signal [17], but the most recent trend istowards devices which either employ unidirectional trans-ducers (UDT), or which utilise both acoustic ports intransmission and reception. Such devices are often morecomplex than the simple structure of Fig. 1, but have theadded attraction of a very low insertion loss, typically

insertion loss of filter or delay line.dB10 20 30

5 10 15insertion loss of one transducer,dB

20-100

10 20 30 40 50level of spurious delayed signal,dB

60

Fig. 5 (a) Plot of minimum possible levels of reflected waves againstinsertion loss, for devices employing symmetrical SA W transducers, and (b)plot of phase and amplitude ripple arising from any time-delayed spurioussignals in SA W devices

2 dB. Some of these arrangements are shown schematicallyin Fig. 6. In Fig. 6a, the input and outputs IDTs areeffectively made unidirectional by adding reflectors in theform of additional tuned-and-shorted IDTs [18]. Thepenalty is the additional complexity, and some reductionin bandwidth. In the 2-port resonator, Fig. 6b, the reflec-tors are replaced by a large array of weakly reflectingstrips or grooves as originally described by Ash [19]. Suchdistributed reflector banks introduce minimal loss, so theresulting devices can exhibit extremely high Q-values andare suitable for sustaining stable.oscillators; their attrac-tions and limitations are discussed in more detail inSection 7. The weakness of the individual reflectors pre-cludes their use in wideband devices.

The multistrip coupler (MSC) unidirectional transducer


out

1in

• •out

out

UDT UDT

out

out

in

^H|i--^

out

in

Iout

Fig. 6 Some approaches to low-loss SAW devicesa Use of external reflecting transducers [18]; b two-port SAW resonator [19];c MSC unidirectional transducer [20]; d other unidirectional transducers, e.g. thoseof Figs. 3d,/and g; e interdigitated transducers (IIDTs) [25-27];/'Ring' filter [28]and g reflecting-MSC filter [29, 30]

(UDT) of Fig. 6c is suitable for use on high-/c2 materials(Table 1) but occupies additional substrate area [20].More economical in real estate are the multiphase [21]and single-phase unidirectional transducers (SPUDT) [22]of Fig. 6d. 3-phase IDTs are, however, more difficult tofabricate and match. Most such devices have employedbandwidths < 10%, but, in conjunction with work onwideband convolvers (Section 8), West [23] has demon-strated bandwidths up to ~40%. The properties of theSPUDT have not yet been fully assessed. Their principalattractions are that they occupy no additional substratearea and that they are readily matched with a single induc-tor. As originally described, their fabrication is 2-stage[22], but a recent variant overcomes this problem [24].The interdigitated interdigital transducer HDT [25] of Fig.6e offers low loss with simple fabrication, but a full designprocedure has not yet been developed. Nevertheless, tworecent papers have described encouraging results at fre-quencies approaching 1 GHz, Wadaka et al. [26] andHikita et al. [27]. The 'ring' filter arrangement of Fig. 6/introduced by Sandy and Parker [28] and the reflectingMSC version of Feldmann and Henaff [29], Fig. 6g, havethe disadvantage of occupying more substrate area, but areotherwise capable of yielding an attractive performance,Pollock et al. [30]. A fuller discussion of some of thesetechniques is given in Section 4.

By now, it will be quite apparent to the reader thatthere is currently a worldwide effort underway to devisesimple and practicable low-loss transducer structures.Results to date are encouraging, and we have little hesita-tion in predicting that in a few years' time a new gener-ation of SAW components will be available which offer aninsertion loss of only a few decibels. This will permit theiremployment in a wider range of applications; for example,as front-end filters in receivers operating up to at leastlGHz.

3 Delay lines and multistrip couplers

Two of the earliest SAW devices to receive extensive inves-tigation are delay lines and multistrip couplers (MSC), anda detailed account of their design, performance and appli-cations was provided in an earlier IEE review [8]. In thisSection, we provide a summary of more recent develop-ments in these relatively well established topics.

3.1 SAW delay linesIt is apparent from Table 1 that most materials have aSAW velocity of order 3000 m/s, and so are capable ofproviding delays of about 3 /xs per cm of path. High-quality single-crystal-quartz and lithium-niobate substratesare currently available commercially in lengths up to~25 cm, so that delays up to ~75 us can be achievedwithout folding the acoustic path; these substrates providetemperature stability or wideband operation, respectively,Table 1. Simple 2-port delay lines employing these sub-strates have found limited u.se, e.g. as memories [34], infusing and in radar return simulation. In such devices, thetrade-off between insertion loss and spurious triple-transitsignal may be inferred from Fig. 5. In this context, itshould be noted that the propagation loss encountered inlong delay lines causes additional triple-transit suppress-ion, and for all practical purposes it is adequate to includethis loss in the device loss (upper abscissa) of Fig. 5a. Arecent addition to the list of available SAW delay line sub-strates is lithium tetraborate (Table 1) which has a higherpiezoelectric coupling than quartz, but much better tem-perature stability than lithium niobate [35]. The principal


problem precluding the use of this material in most SAWdevices, e.g. bandpass niters, concerns the excitation of aspurious bulk wave, but this causes no difficulty in longdelay lines. This material is relatively easy to grow andshould soon become available in 25 cm lengths, making itattractive for applications requiring substantial delays.

Some applications require delays of greater than 75 ^sand, to meet this demand, a number of techniques havebeen devised to fold the acoustic path, as discussed by Ash(Reference 6 Chap. 4). While the performance achieved isoften impressive, e.g. demonstrating delays up to 1 ms, wedo not include details here, as most such devices have been'one-off' prototypes, and a production capability has yet tobe established. Instead we include References to the bestpublished results, to which the interested reader may refer:

(a) spiral waveguides, Adkins and Hughes [36](b) MSC reflecting trackchangers, Browning and Mar-

shall [37](c) helical delay lines, Fortunko and Shaw [38](d) disc delay lines, Mason et al. [39].

A great attraction of SAW devices is their planar structurewhich allows the signal to be tapped at any point in itspath [8], and Fig. 7 shows a simple such tapped delay line

DELAY L.INCftp. TURN t

CSIGfSlAt. TECHNOLOGY t-Tl

Fig. 7 Photograph of SAW tapped delay line for radar return simulation

for radar return simulation. This concept has recently beenextended to provide a device with 120 MHz bandwidthand 1 ̂ s incremental delays up to 42 /is, Warne et al. [40].Further, the signal/noise ratio was sufficiently great toallow these modules to be cascaded, to provide amaximum delay of 400 /is. A photograph of the arrange-ment is provided in Fig. 8.

An obvious extension of the SAW tapped delay lineconcept is the programmable tapped delay line, in whichthe amplitude and/or phase of each output tap is prog-rammed electronically and the outputs summed. Such anarrangement is potentially useful as a programmablematched filter (PMF), e.g. for biphase modulated wave-forms [41], and has also been investigated recently as aprogrammable bandpass filter, e.g. for use in rejecting nar-rowband interferences [42]. Although, at first sight thePMF appears to be an attractive concept exploiting theplanar construction of SAW devices, in practice the hybrid[41] and monolithic versions [43, 44] all face a variety ofdifficulties in respect of fabrication complexity, time andbandwidth limitations, and/or power consumption. Inaddition, the PMF faces competition from digital tech-niques at low bandwidths and from the SAW convolver athigh bandwidths. The most recent work in this area is thatof Lattanza et al. [45] who described a hybrid deviceemploying custom LSI-chips and acting as a 256-tap PMFfor a 64 Mbit/s MSK waveform. The performance of thisdevice was close to theoretical over the full military tem-perature range, but it will be noted that the parametersreported are also available with the monolithic SAW con-volver of Section 8.

The hybrid programmable SAW bandpass filterdescribed by Panasik [42] employs a basic SAW delayunit containing 16 taps (IDT fingers) controlled by dual-gate FETs. Each tap weight has a dynamic range of> 50 dB and the use of inphase and quadrature channelsallows complete control of the amplitude and phase ofeach tap. The assembly can produce up to 15 nulls, eachwith a depth of 30-40 dB, across its operating bandwidthof 100 MHz.

This device is still at a relatively early stage of develop-ment and its full capability has yet to be demonstrated.

One difficulty faced by such programmable tappeddelay lines concerns electrical breakthrough of the signaland/or switching waveforms; this problem is not shared bythe SAW convolver of Section 8, which can eliminate spu-rious signals by frequency filtering.

Another application which fully exploits the planarformat of SAW device technology is the acoustic beamformer for use in phased-array radar. A 2-dimensionalSAW transducer array of the form described in the follow-ing text can be used for generating or receiving the signalsfrom an antenna array. The principle of operation is illus-trated in Fig. 9 which shows how the different delays of thereturn echo received at each antenna element are compen-

EMwavefronts antenna

elements SAW substrate

) " !

e\

\

inininiff\

0Oi

9>0

i"i"iffiff

iD029=0

1 7Ml

1 /lWo03

6<0

Fig. 8 Photograph of Plessey SAW-tapped-delay-line unit providing1 fis outputs up to 40 us, and 9 such units cascaded to provide delays up to360 us

Fig. 9 Schematic of 2-dimensional SAW IDT array useful in the gener-ation and reception of signals from a linear antenna arrayTo appreciate the principle of operation, imagine that the antennas are illuminatedby an EM wave incident at an angle 6 as shown. The peak output will emerge fromthat output for which the differential EM delays are compensated by acousticdelays, e.g. output O, (0 > 0) in this figure.


sated by acoustic delays at IF. A prototype beamformer ofthis kind has been described by Warne and Morgan [46],who employed 32 antenna elements and achieved aminimum beam width of ~3°. Although the sidelobe sup-pression fell short of the original aim, it is believed that itcan be improved to better than 25 dB in all circumstances,by correcting for an amplitude ripple present in the proto-type.

3.2 Multistrip couplersIn its simplest form the multistrip coupler (MSC) com-prises an array of parallel metallic strips coupling SAWfrom one acoustic path to another, as shown schematicallyin Fig. 10 and described in detail by Marshall et al. [47]. A

Fig. 10 Schematic of MSC used to couple SAW from one acoustic trackto anotherIn this Figure, the wave coupled to the lower track has a 90° phase advance overthe wave remaining in the upper track [47, 52].

basic property of this arrangement is its ability to achievecomplete energy transfer from one track to the other overa bandwidth of ~ 100% on any piezoelectric substrate.However, the number of strips required, NT, variesinversely with k2 (Table 1) and is prohibitively large onmost substrates. Even on YZ-LiNbO3, NT < 100, and theMSC often occupies a significant fraction of the substratearea, a factor of economic importance in some applica-tions. In the earlier IEE review by Maines and Paige [8], itwas shown that the MSC can form the basis of numeroususeful and ingenious components including the UDT ofFig. 6c, SAW reflectors, reflecting track changers, acousticbeam compressors and the 'magic T".

Its operation may be viewed as follows: each neigh-bouring pair of fingers may be regarded as an elementaryIDT which develops a voltage difference as a result of theSAW incident in the upper track. This voltage difference istransferred to the lower track where it, in turn, excitessurface acoustic waves. As discussed in Section 2 the band-width of the individual finger-pair IDT is ~ 100%, and alittle thought shows that the SAW excitation in the lowertrack is constructive in the forward direction and destruc-tive in the reverse direction at all frequencies (except closeto the stopband [8] when the strip spacing is k/2, or amultiple thereof, when there is also strong coupling in thebackward direction). This arises because in the forwarddirection all paths suffer the same delay, so the impulseresponse of the MSC approximates to a <5-function in time.Thus the operation of the MSC is inherently very broad-band, providing the narrow stopband is avoided. It is alsostraightforward to see that the SAW beam excited in thelower track of Fig. 10 is of uniform amplitude across theaperture, even if the incident beam in the upper track has aspatial variation due to the apodisation of the input IDT.

This result is important as it ensures that the overall fre-quency response of the device of Fig. 10 is the product ofthe responses of the individual IDTs, Section 4. In general,this is not true for two apodised IDTs in the absence of theMSC [48]. Another useful property of the device of Fig. 10is that any unwanted bulk waves generated by the inputIDT tend to remain in the upper half of the device ratherthan causing a spurious output signal. For these tworeasons, the MSC has been used in many high-qualitySAW bandpass filters over the past decade. By contrast,few of the other components described in Reference 8 havefound widespread use, but the MSC-UDT of Fig. 6c maysoon see a revival as result of current interest in low-lossSAW devices.

In recent years, several variants of the original MSChave been described which have one feature in common:they deliberately modify the simple MSC structure of Fig.10 to alter its impulse response from the ^-function formdescribed above, i.e. they introduce a frequency-filteringaction into the MSC. In the offset MSC [49] shown sche-matically in Fig. I6d, this is achieved by stepping one half ofan MSC pattern in the direction of SAW propagation. Asdiscussed in Section 5, this causes different incoming fre-quency components to traverse different paths, and can beused to implement a SAW filter bank. In the fanned MSCdescribed by Solie [50] some strips of, say, the lower trackof Fig. 10 are stepped in the direction of SAW propagationby various integer multiples of Ao, the SAW wavelength atsome chosen frequency f0. Clearly this stepping processdoes not affect the response at / 0 , but it does extend theimpulse response of the MSC, and so introduces abandpass response around f0. The idea of the fanned MSCis, then, to add an additional frequency-selective elementto the basic SAW bandpass filter of Fig. 10, giving greaterdesign flexibility; A somewhat similar process is employedin the reflecting MSC, RMSC [29], and its variants [30],except that the stepping is chosen to ensure constructiveinterference in the reverse direction and destructive inter-ference in the forward direction at the frequency of inter-est. The use of two such RMSCs leads naturally to thelow-loss configuration shown earlier in Fig. 6g.

The only other new MSC component of importance isthe 2-period MSC of Maerfeld and Farnell [51], which canbe used to perform beam compression with any ratio ofinput and output apertures in a single step; this com-ponent retains the wideband property of the original MSCand is of importance in some SAW convolvers, asdescribed in Section 8.

4 Bandpass filters

It is seen from Section 2 that the design of a SAWbandpass filter essentially reduces to the problem of deriv-ing a suitable impulse function h(t), related to the filter fre-quency response H(f) by the Fourier transform. Ingeneral, h(t) is infinite in extent, so it must be truncated inorder that the filter be physically realistic. Procedures forthe derivation of finite impulse response functions whoseFourier transforms give optimal approximations to adesired 'ideal' frequency response have been the subject ofconsiderable research effort over the past decade (Rabinerand Gold [53], Chap. 3). As a result, several very usefulcomputer-aided-design procedures are now generally avail-able, e.g. Parks and McClellan [54], McCallig and Leon[55]. Although developed for use in relatively low-frequency digital signal processing systems, such algo-rithms are directly applicable to SAW bandpass filter


design as the interdigital transducer is itself a transversalfilter, at least to first order.

The basic transversal filter consists of a delay line whichis tapped at uniform time intervals. The output from eachtap is suitably weighted and summed on a common bus(Tancrell [56]). Such a structure is readily interpreted inthe interdigital transducer (IDT). Transducer fingers arepositioned at uniform intervals and tied together electri-cally by busbars.

4.1 Weighting methodsWeighting of the taps is most commonly accomplished byvarying the lengths of the transducer fingers, a techniqueknown as 'apodisation' (Fig. 4). Generally, it is more con-venient to regard gaps between fingers rather than fingersthemselves as sources of surface acoustic waves (Lewis etal. [57]), in which case the tap weight at a particular gapis a linear function of the overlap between the adjacentfingers comprising that gap.

The simplest structure for an apodised transducer isthat in which the finger pitch is a half wavelength, as forthe uniform transducer of Fig. 3a. However, apart from theproblem of reflections described in Section 1, such a struc-ture suffers the disadvantage that its amplitude-frequencyresponse can only be symmetrical about the centre fre-quency. In many applications, for example the TV IF filter[69], an asymmetrical response is required. Hence, in mostSAW bandpass filters, photolithographic considerationspermitting, each transducer consists of more than two elec-trodes per period. Common choices are four or three elec-trodes per period (Figs. 3b and c). It is evident thatapodisation provides a very convenient method for weigh-ting IDTs. It adds no extra processing complexity and, inprinciple, provides a very wide weighting dynamic range.However, there are some drawbacks to the use of apodisedtransducers in bandpass filters.

The first, most obvious drawback was pointed out byTancrell and Holland [48], as long ago as 1971. This is therestriction that, in order that the filter frequency responsebe simply the product of the individual responses of theinput and output IDTs, only one IDT can be apodised.From the point of view of filter synthesis, it is highly desir-able that the overall filter response be amenable to sub-division in this manner, so that the design of each IDT cantake place virtually independently of the other. Later itwas shown by Deacon et al. [58] that both transducersmay be apodised if used in conjunction with a multistripcoupler (MSC). Since then, this configuration has beenused successfully in a wide variety of bandpass filters (seeFig. 11, for example).

The incorporation of the MSC brings the extra advan-tage that bulk wave responses are suppressed (Tancrell andEngan [59]) and extension to the use of the fanned multi-strip coupler (FMSC) described in Section 3 is possible(Solie [50]) giving greater design flexibility. However, thedisadvantages are, first, that it is confined to those sub-strates having sufficiently high piezoelectric coupling factork2 to warrant its use, and, secondly, it roughly doubles thesurface area (and, therefore, chip cost) of the device. Forthose cases where a low-/c2 material (most commonlyquartz) must be used, or where chip cost is a particularlysignificant factor in determining the selling price of thefilter, the restriction to a single apodised transducer, atmost, will still apply.

A second disadvantage in the use of apodised IDTs isthe presence of diffraction. It has been shown (Szabo andSlobodnik [60]) that, for those materials and orientationsfor which the slowness surface closely approximates a

parabola (all common materials other than YZ lithiumniobate), the rate of diffraction scales inversely as thesquare of the aperture. The universal curves for suchmaterials are shown in Fig. 12. In apodised transducers,

-8034 42-36 38 40

b frequency, MHz

Fig. 11 (a) General SAW filter layout comprising two apodised IDTswith an intervening multistrip coupler {MSC) and {b) response of a filter on128°-rotated lithium niobateTransducers contain 390 and 408 electrodes, MSC has 100 strips

- 1

-2

3 mT3in-erto

5.2

"6?

8

9

10

40

35

30

o>o,25

•o

0,-20o

10

5

•

\ phase \

"Namplitude

\

\ \ :

\ \

0.01 0.1 1.0normal ised d is tance CX/Ap2

Fig. 12 Universal diffraction curves showing amplitude and phase varia-tion against separation of two transducers with identical aperturesThe normalised quantities X and Ap are the separation and apertures measured inwavelengths, respectively. The parameter C is a measure of the material's acousticanisotropy and is given by C = 11 + dtp/d91, where 9 defines the propagationvector and <j> the beam steering angle [60].


where some finger overlaps will inevitably be very small (ofthe order of a wavelength) diffraction is likely to have asignificant effect on filter performance. Consequently, therehas been much published literature on the subject of dif-fraction correction in SAW bandpass filters, most notablyby Savage and Matthaei [61], Mader et al. [62] and Peachet al. [63]. Although such correction procedures have beenshown to be quite successful, it is always preferable,wherever possible, to avoid diffraction effects by main-taining a large acoustic aperture.

Other forms of weighting have been proposed to over-come these two problems of apodised transducers. Of thesethe most widely implemented has been the 'withdrawalweighting' (WW) method of Hartmann [64]. Here thetransducer weighting function is approximated by taking aperfectly uniform IDT and selectively removing electrodesor sources (Fig. 13). This is evidently a very coarse weigh-ting method, giving best approximation to the originalfunction for transducers having many fingers, i.e. narrow-band filters. However, when used in such cases, in conjunc-tion with diffraction-corrected apodised transducers,excellent results can be obtained (Fig. 14).

Other methods for uniform transducer synthesis havebeen proposed, e.g. the 'phase interference' method ofAtzeni et al. [65], the 'series weighting' method of Engan[66] and its use in combination with withdrawal weighting

by Sandy [67], the use of a pair of dispersive transducers(see Section 6) in a nondispersive configuration and the

Fig. 13 Three schematic weighted SAW transducers with approx-imately the same frequency responsesa Apodized IDT, b finger-withdrawal weighted IDT, c source-withdrawal weightedIDT

Fig. 14 High-selectivity SAW filter on ST-cut quartz

Horizontal scale: 0.6 MHz/div, vertical scale: 10 dB/div. The centre frequency is 200MHz with a reproducibility of ±60 kHz. Courtesy GEC-MEDL, Lincoln.

'phase weighting' method of Hikita et al. [68]. These allawait exploitation on the same scale as withdrawal weigh-ting. An alternative to weighting the transducers is toprovide frequency shaping external to the transducers. Onesuch method is the FMSC described in Section 3.2. Whenused in either transmission or reflection mode, this givesthe benefit of adding frequency selectivity at the cost of aminimal increase in loss. An alternative external frequencyselective element is the reflective dot array (RDA).Although developed primarily as an alternative to theRAC technology for SAW dispersive delay lines, the RDAhas been shown to give very impressive results when usedfor bandpass filters. Solie [76] has demonstrated abandpass filter on ST quartz having nearly 90 dB out-of-band rejection. This rejection has never been achieved witha pair of IDTs alone on ST quartz.

4.2 TV IF filterThe application of SAW bandpass filters is almost exclu-sively at IF where the high insertion loss (typically20-30 dB) does not impose a great handicap in terms ofsignal-to-noise degradation. By far the largest single appli-cation (defined in numbers of devices sold per year) is fordomestic TV receivers. The mass production of such filtersstarted in the mid 1970s and has now reached a pointwhere the majority of TV sets made throughout the worldcontain SAW rather than conventional LC filters in the IFsection.

In such a price-sensitive market the most importantparameter for the designer is cost and, therefore, substratearea. The progress made in this respect over the lastdecade may be judged by recalling that in the mid 1970san advanced TV IF filter design produced 36 chips on a2 in (5.08 cm) circular wafer of lithium niobate (de Vries etal. [69]). Nowadays, between 200 and 500 chips areaccommodated on a 3 in (7.62 cm) wafer, dependent on theparticular broadcasting system for which the filter isintended (Fig. 15).

A key factor in the reduction of chip area was theintroduction of low bulk wave cuts of lithium niobate(Shibayama et al. [70]) and lithium tantalate (Kodama[71]). Currently, the most commonly used substratematerial is the 128° rotated cut of lithium niobate. This notonly gives reduced bulk levels compared with YZ lithium


niobate but also has a lower temperature coefficient ofdelay and a higher piezoelectric coupling factor (see Table1). Its only significant drawback relative to YZ lithium

Fig. 15 3 in (7.62 cm) diameter wafer of 128°-rotated lithium niobatecontaining 260 TV IF filters

[Signal Technology Ltd., 1984]

niobate is the increased diffraction rate, a property whichmakes the design of such small aperture filters particularlytroublesome. However, the rewards of reducing chip areain such large-scale applications, by even the smallestamount, can justify the enormous computing effortinvolved in correcting for diffraction effects.

An alternative route to reducing material costs has beenthe development of zinc-oxide thin-film technology. Workon the growth of thin ZnO films on glass for SAW applica-tion has a long history [77], but only in the last five yearshas its use in the TV IF filter become an economic propo-sition [78]. It remains to be seen whether, in the long term,ZnO can compete in cost and performance with the single-crystal materials.

4.3 Low lossIt was said earlier that application of SAW filters is con-fined chiefly to the IF stage, and that this is purely becauseof their high insertion loss. High insertion loss is not aninherent property of SAW devices, but comes about chieflyfor two reasons. First, IDTs are bidirectional and, hence,in the simplest filter configuration consisting of a singleinput and a single output transducer the minimum pos-sible loss is 6 dB (Fig. 5a). Secondly, as shown earlier, if thetransducers are matched to the external loads so as toachieve this minimum loss, then the triple transit echo willbe at a level of only —12 dB, with respect to the directSAW, giving rise to unacceptably high amplitude andphase ripple in the frequency domain. The natural remedyis to overcome the triple-transit problem the easy way, bymismatching the transducers to increase the insertion lossand, consequently, reduce the triple-transit signal. This is

usually achieved very simply by failing to use any externalmatching component at all.

To reduce insertion loss to values acceptable for incor-poration as front-end filters, a solution to the bidirectionalloss must be found, together with a means of adequatelysuppressing triple and other multiple transit echoes. Theseproblems are solved simultaneously if the IDTs can bemade unidirectional, i.e. reduced to a 2-port device havingone electrical port and only one acoustic port, the otheracoustic port being totally decoupled. Early implementa-tions, e.g. with MSC (Marshall [20]), were followed by theinvention of the three-phase unidirectional transducer(UDT), resulting in the demonstration of filters having2 dB insertion loss only (Rosenfeld et al. [72]). However,because of the need for two metallisation stages and theprovision of low-capacitance air-gap crossovers betweenthe metal layers, processing complexity is considerablyincreased. In addition, a larger number of matching com-ponents is normally required for each transducer toprovide the 3-phase drive necessary for unidirectionaloperation.

The first of these objections was overcome by Yama-nouchi et al. [73] who demonstrated 1 dB loss filters usinggroup-type unidirectional transducers (GUDT). The quad-rature drive required for these transducers still generallyrequires more than one tuning component per transducer,however (Malocha and Hunsinger, [74]). The most recentdevelopment in unidirectional transducers attempts tominimise external tuning components. The single-phaseunidirectional transducer (SPUDT) of Hartmann et al.[22] relies on SAW reflection internal to the transducer toachieve unidirectionality. Part of the processing complex-ity is restored, however, as the technique involves a secondmetallisation of gold requiring careful alignment with thefirst aluminium layer. No crossovers, as in the 3-phaseUDT, are required, however, and matching can beachieved with a single inductor.

A different approach of low loss is illustrated by theinterdigitated interdigital transducer (IIDT) configurationproposed by Lewis [25]. Here the transducers are of thesimple bidirectional type, but more than two are involved.It has long been recognised that half the minimum 6 dBinsertion loss of a single bidirectional pair can be reco-vered by the addition of a third transducer, and, if such aconfiguration is made symmetrical, then perfect matchingof the central IDT to its electrical load results in the com-plete suppression of multiple-transit echos (Lewis [17, 75]).In the IIDT device, this idea is extended to include four ormore IDTs. As more transducers are added, the minimumtheoretical insertion loss reduces according to

IL = 10 log10{(N + 1)/JV} dB

where N + 1 is the number of input transducers and N thenumber of output transducers. (An odd total number oftransducers is normally considered, although an evennumber would be equally valid.) Single IIDT filters with2 dB losses have been demonstrated [25] and a cascadedpair with less than 4 dB loss and good stopband per-formance has also been shown by Hikita et al. [27]. Theadvantages of this technique are that it requires a singleprocessing stage only and a minimal matching componentcount. A disadvantage might be thought to be the increasein substrate area due to the extra IDTs, but the likelyapplication of such filters at the front end of systems oper-ating at UHF implies that the size of each 'subtransducer'is likely to be very small anyway; so total chip area wouldbe acceptable from the point of view of material cost andpackage requirements.


Table 2: Review of experimental data for SAW filter banks

Centre frequency

MHz200

211200333

414

147200350253125325325

3 dB bandwidth ofsingle channel

MHz14

353

3

2410.80.6-1.0

1020

Number ofchannels

9

810

9

9

516

1002

101616

Channelseparation

MHz20

653

12

36.7181

1010

Insertion loss

dB-21

- 1 0-35-20

-26

-25-32- 5 5 *

-2.9- 4 0-19.5J-22.3J

Ultimaterejection

dB30

43—

47

61

2530

-40-40—

Reference

79

79, 828081, 90

81, 90

8385, 8687, 88899198)98]

Comments

Individual filters: electricalpower splittersOffset MSCIndividual filters: constant KIndividual filters: series/parallel connectedIndividual filters: series/parallel connectedFan-shaped MSCReflective grooveMultipassband* 4 stagesMultipassband—low lossProgrammable oscillator

Butterworth filters

% individual filter

5 Filter banks

5.1 GeneralIn the preceding Section we have seen that SAW bandpassfilters have many attractive properties, including anextremely flexible design capability which can lead tophysically small components of high performance and lowcost. It is, therefore, natural to attempt to apply the tech-nology to the more demanding task of providing a filterbank, contiguous or otherwise, whose function is toseparate signals of different frequencies into differentoutput channels. Ideally, such a device would provide aseries of rectangular passbands in the frequency domain,but, in reality, one is limited by fundamental consider-ations (i.e. the uncertainty principle, as in Section 4) and bythe practical constraints of the particular application, e.g.the need to handle pulsed waveforms whose spectra maycover more than one channel, leading to spurious outputsin the time domain. The applications of such devices arevery diverse, and, under the generic heading 'filter banks',we have included a variety of components generally knownas multiplexers/demultiplexers, channelised receivers andspectrum analysers, each of which is amenable to an SAWimplementation [79, 92]. These applications provide agreat challenge and opportunity to the SAW designerbecause they cover such a wide variety of specification par-ameters (Table 2); for example, multiplexers [79, 82] andfrequency synthesisers [90, 91] require excellent isolationbetween channels, while spectrum analysers are 'wide open'receivers in which adjacent channels usually overlap at the3 dB points. In specific applications, other desirable fea-tures may include small physical dimensions, high effi-ciency, good temperature stability and minimumprocessing delay. A great advantage of the SAW tech-nology, in general, is the high degree of reproducibility ofperformance, and this aspect is of particular significance incomplex components like filter banks, because competingtechnologies often require the trimming of many com-ponents. Another attraction of the SAW device concernsthe degradation of performance with a change of tem-perature; this amounts to a simple and predictable fre-quency scaling, Table 1.

The most straightforward way to realise a SAW filterbank is to use a series of individually designed bandpassfilters in parallel. The interconnection can be performed (i)by using power splitters [79, 81], (ii) by feeding the IDTsin parallel, Fig. 16a, in series, Fig. 166, or in a series/parallel configuration for impedance matching purposes,or (iii) by implementing a constant-/c ladder network [80].

If the frequency selectivity is built into the output IDTs,the input transducer can become a single wideband large-aperture IDT, Fig. 16c. However, this latter technique, as

mmi

iii i

I i

mi

rED1Ii

Fig. 16 Different design for SA W filter banksa Individual bandpass filters fed in parallel from a split electrical input [79, 80, 81];b individual bandpass filters fed in series from a single electrical input [80]; c single-input transducer feeding many output transducers; d offset multistrip-coupler-basedfilter bank [79, 82]; e fanned reflecting multistrip-coupler-based filter bank [83] and

/reflective groove array filter bank [85].

with the use of electrical power splitters, introduces a losswhich increases with the number of channels. This chan-nelisation loss can be reduced significantly by using afrequency-selective offset multistrip coupler [79, 82]. Thisapproach employs a broadband input IDT in conjunctionwith a series of offset MSCs that direct the different fre-quency components towards a series of narrowbandoutput IDTs, Fig. 16d. The operation of the MSC multi-plexer is based on the fact that the signal transferred by anormal 3 dB MSC (i.e. one with NT/2 periods, Section 3.2)from one track to another has a n/2 phase advance over


the signal remaining in the original track [52]. Conversely,when two equal signals in adjacent tracks have a relativephase shift of n/2, the signal emerges from the MSC in onetrack only (Reference 20, Fig. la). In Fig. 16d the equiva-lent effect is built into the MSC itself by incorporating anoffset (or jog) of (N + %)XQ. The two output channels ofsuch a jogged MSC have interleaved multiple passbandresponses [82] similar to Fig. 18a and b shown later. Theseparation of adjacent bands in one channel is A/= 1/T, Tbeing the temporal offset, i.e. T = (N + £)A0/F. By exten-sion of this principle, as in Fig. 16d, Solie developed an8-channel multiplexer with a centre frequency of 232 MHzand an insertion loss of only 10 dB. This may be comparedwith the expected loss for the straightforward design, Fig.16c, which would be about 20 dB.

Another frequency-selective MSC arrangement whichcan be used in filter bank applications was first describedby Feldmann and Henaff [83, 84]. In this device, theperiodicities of the upper and lower tracks are chosen toensure constructive interference in the reverse direction,and destructive interference in the forward direction. If theoutput track is also fanned as in Fig. 16e, the different fre-quency components emerge at different points across theaperture, whence they are received by an array of narrow-band narrow-aperture output IDTs. The prototype devicedemonstrated by Feldmann and Henaff had five channels,each of bandwidth ~2 MHz, at a centre frequency of150 MHz.

A further technique capable of providing a modestnumber of channels has been described by Melngailis [85,86]. This is illustrated in Fig. 16/and employs a number ofgroove arrays to Bragg reflect different frequency com-ponents towards a series of output IDTs of appropriatelystaggered frequencies. To a good approximation, thesearrays are transparent to frequencies other than thosesatisfying the 90° Bragg reflection conditions, and thenumber of channels is limited primarily by the substratedimensions: Melngailis's device provided 16 output chan-nels with a resolution of 6.7 MHz. We believe that the lossof 32 dB in this device could be reduced significantly bymatching the output transducers.

5.2 Spectrum analysisBy direct detection of the outputs of the various channelsany of the previous approaches, we can form the basis of aspectrum analyser with parallel readout capability. Alter-natively, the channelised receiver may be used in conjunc-tion with an IFM (instantaneous frequency meter) toprovide more precise frequency information, but its suc-cessful operation relies on a priori knowledge of the likelychannel densities, as the IFM cannot cope with simulta-neous signals. Two approaches to spectrum analysis whichovercome this limitation are the compressive receiver [92]of Section 6, which provides an output serially in time, andthe acousto-optic spectrum analyser described below,which provides its outputs in parallel.

SAW (as distinct from bulk acoustic waves) form thebasis of the integrated acousto-optic spectrum analyser(IAOSA) [94], which is potentially a small, rugged and effi-cient component. In this device, light confined to within afew micrometres of the substrate surface, by a planar wave-guide, is Bragg deflected by the periodic refractive indexchanges induced by surface acoustic waves excited by theinput IDT, the angle of deflection being proportional tothe input frequency, Fig. 17. The angular distribution ofthe deflected light is transformed to a spatial distributionat the focal plane of a collection lens, where it is detectedby a linear array of photodiodes. These diodes provide

198

outputs from the various frequency channels in parallel. Ifrequired these outputs may be integrated for up to ~ 1 ms,

lensi

acoustic absorber

• i \ diffracted light —

(ens 2

spatialfilter 1

gt==^Q SAW transducer

photodetectorarray

spatialfilter 2

Fig. 17 Acousto-optic spectrum analyser

Refractive index changes induced by the SAW Bragg diffract the incident laser lightwhich is then focused onto a photodetector array. The angular distribution of dif-fracted components is proportional to the SAW frequency.

to facilitate serial readout and/or the detection of weaksignals. For input (and SAW) frequencies of ~1 GHz,there is a close spatial match between the SAW and opticalfields, leading to a strong acousto-optic interaction. Unfor-tunately, at higher SAW centre frequencies (which arenecessary to sustain bandwidths > 500 MHz), the degree ofoverlap is reduced and with it the interaction efficiency.For this and other reasons, such as increasing fabricationdifficulties and acoustic propagation losses, the maximumbandwidth sustainable by the IAOSA is likely to be limitedto <1 GHz. It is also fair to say that the IAOSA facesstrong competition from bulk acousto-optic devices whosecapability is developing rapidly; for example, a recentdevice with 1 GHz bandwidth has shown a deflection effi-ciency of 44%/W through improved acoustic transduction[95]. Ironically, this has highlighted the problem of acous-tic intermodulation effects; and, in this respect, it isamusing to realise that the IAOSA has an inherent advan-tage because it is amenable to the introduction of con-trolled acoustic dispersion: indeed this may well occurnaturally through the presence of the optical waveguide[100].

5.3 Frequency synthesisThe discussion up to this point has been concerned withfilter banks, in which all outputs are available simulta-neously and continuously. By contrast, for frequency syn-thesis it is usual to employ only one output at a time inorder to select a single tone from the spectrum of a combgenerator. It is important to recognise that, when usingthis technique, the speed of frequency hopping is limitedonly by the switching circuitry. Budreau et al. [90] usedthe 18-channel filter bank of Slobonik [81] with two combgenerators and three local oscillators to produce throughmixing a frequency-hopped output, covering the rangefrom 1369 MHz to 1601 MHz intervals, and with a spu-rious suppression of 47 dB. Obviously such schemes canbecome quite complex and an approach which reduces thenumber of SAW filters by making each programmable isdescribed in the following.

The principles of switchable filters using programmablemultipassband transducers have been outlined by Hays etal. [82, 87]. A full implementation of this approachreduced the number of filters necessary to select one out of2" tones from 2" to n/2, which can be very significant forlarge n. The operation of a single programmable filter withtwo programmable transducers to select one tone fromfour is illustrated schematically in Fig. 18. A set of suchfilters connected in series, and with successively decreasingchannel bandwidths, can be used to extend the selectivity

IEE PROCEEDINGS, Vol. 131, Pt. A, No. 4, JUNE 1984

rapidly. The device described by Hays employing thesegeneral principles demonstrated 100 addressable channels

transducer Astate 1

transducer Astate 2

transducer Bstate 1

transducer Bstate 2

b*c

comb offrequencies

selected tonee*f

Fig. 18 Switchable multiple passband transducers used to select tonesfrom comb generatorsTransducer A has two states a and b which are interleaved to give selection betweenadjacent tones/. Transducer B also has two states c and d, but this time the pass-bands select from adjacent pairs of tones. The series combination of these two trans-ducers, say, transducer A in state 1 and transducer B in state 2, can select a singletone from four e and g.

in the range 300-400 MHz. The additive losses in suchdevices can be reduced by the use of 3-phase transducers,as demonstrated by Potter and Shoquist [89].

5.4 Banks of matched filtersSo far all the devices in this Section have been concernedwith banks of bandpass filters. There are, however, poten-tial applications for banks of matched filters for the gener-ation and reception of waveforms with time-bandwidthproducts greater than unity. For example, a set of disper-sive filters of the type described in Section 6 could beemployed to add waveform agility to a pulse compressionradar.

In addition, banks of SAW matched filters can be usedto realise communications links with low error rate in thepresence of noise and interference from cochannel users. Insuch a modem system, the encoding and decoding involvesthe assignment of a particular codeword (signature) toeach message. These codewords are selected from anappropriate alphabet of signal waveforms, chosen suchthat their transmission makes the best use of a given noisychannel. Where a multiple word alphabet is employed, thisencoding and decoding can be most successfully accom-plished using matched SAW filter banks [97]. In theextreme case, total versatility can be built into the systemusing (SAW) convolvers as the decoding elements (Section8).

6 Dispersive delay lines

6.1 Pulse compression systemsSAW dispersive delay lines have a delay that is a noncon-stant function of the instantaneous frequency of the inputsignal. Historically, dispersive delay lines were one of thefirst applications [101] to which SAW techniques wereapplied. Indeed, to date, the most successful application ofSAW devices in radar is in pulse-compression systems.

A pulse-compression radar [102-106] is a practicalimplementation of a matched filter system. It involves the


transmission of a long coded pulse and the subsequentprocessing of the received echo through the recognition ofthe transmitted code to obtain a narrow pulse. Thus,increased detection is achieved by a long-pulse system,while maintaining the range resolution capability of anarrow-pulse system.

Various configurations exist for such a system [105].Each represents an alternative description of the matchedfilter conditions defined in eqn. 5. All use a coded signalwhich, after suitable modulation, frequency up-conversionand amplification, is applied to an antenna. On reception,the target echo undergoes down-conversion and amplifica-tion, before being applied to a matched filter. A typicalconfiguration is shown in Fig. 19.

1H U transmitter

amplifier

LO m

TR

i

mixer

DET weighting filter

c

mismatch section matchedfilter

Fig. 19 Schematic of pulse-compression radar set employing conjugate(SAW) filters for waveform generation and matched filtering

Pulse compression using SAW dispersive delay lines hasbecome the accepted means of optimising the range,resolution and signal-to-noise performance of pulsedradar. In Fig. 19 the coded signal is represented by thefrequency response of an unweighted SAW dispersivedelay line. This signal is obtained by exciting the dispersivedelay line by a unit impulse. On reception, a conjugateSAW device is used to perform the matched filtering. Thecompressed pulse at the output of the matched filter isaccompanied by responses that occur in time, and are adirect result of the matched filtering. Because time is rangein radar, these additional responses are termed range side-lobes [102].

When many targets with differing radar cross-sectionalareas are interrogated by the pulse-compression radar,such range sidelobes represent a source of unwanted signaland must be suppressed as much as possible. This isachieved by amplitude weighting the output of thematched filter. The required weighting function to achievethe desired range sidelobe performance is normally incor-porated into the SAW device design. However, this effec-tively mismatches the filter and degrades thesignal-to-noise performance. For most systems, a compro-mise must be reached between spurious free dynamic rangeand signal-to-noise degradation. Recent work on nonlinearFM has shown that this problem can be reduced consider-ably.

6.2 Waveform design considerationsThe ability and freedom to design various characteristicsinto the radar signal is an important factor in the develop-ment of pulse compression. To date, linear FM waveforms

199

have received a considerable amount of attention; how-ever, nonlinear FM waveforms are now being activelyinvestigated. The advantage and disadvantages of thesewaveform types are described in the following:

6.2.1 Linear FM: This is the simplest waveform type cur-rently in use. The ideal matched-filter response for a linearFM system is a SINC function in time, with —13.5 dBclose-in sidelobes. Weighting filters are used to reducethese range sidelobes from the —30 dB to —50 dB level.The consequence of this is to degrade the filter correlationgain and to broaden the compressed pulse width. Thiseffect is well documented [102] and there exists a completerange of possible weighting functions [107].

To accommodate the pulse-width broadening due torange sidelobe reduction, the required system bandwidthmust be increased over the ideal system bandwidth [102,106]. The latter is directly related to the reciprocal of theresolution defined by the — 4 dB width of the compressedpulse. For a — 40 dB Taylor weighting function [108] thepulse-width broadening at the — 4 dB level is 43%, with asignal-to-noise degradation of 1.15 dB.

Although the compressed pulse shape and signal-to-noise ratio are fairly insensitive to Doppler shifts, linearFM does suffer from excessive range-Doppler cross-coupling. This effect manifests itself as a range error unlessa priori information is available, on either the range or theexpected Doppler. The Doppler frequency shift of theradar echo further degrades the matched-filter responsefrom the ideal. This introduces an additional pulse-widthbroadening factor and an additional signal-to-noise ratiodegradation.

If the output of the pulse expander is gated and limitedin time, then the pulse compressor is no longer its matchedfilter. Far out sidelobes are produced that are higher thanexpected. Such Fresnel sidelobes are generated around thecompressed pulse within the interval ± T/2, where T is theexpanded pulse width. These are suppressed by additionalweighting in the compressor to about 20 log10 TB + 3 dB,where TB is the time bandwidth product. In such cases,the effects of frequency responses outside the useful bandof interest must be considered [108]. Where TB is small, alarge amount of asymmetry is introduced into the Fresnelsidelobes by the Doppler frequency shift.

For time bandwidth products below 100, significant dis-tortions occur in the SINC function. This problem is over-come by computing a spectrum for the combined encodingand receiving filters that gives the desired sidelobe level.The receiver spectrum is obtained by dividing this desiredspectrum by the spectrum of the encoded signal, a processwhich is referred to as reciprocal ripple [109]. The impulseresponse of the receiving filter is then obtained from theinverse Fourier transform. This response maintains a sig-nificant amplitude outside the time duration of the trans-mitted signal. For a time bandwidth product below 20 andsidelobe suppression better than 30 dB this extension canbe as much as 60% [108].

6.2.2 Nonlinear FM: A radar signal with a linear fre-quency time law has a quadratic phase variation. Nonlin-ear radar signals have a phase variation that shows minordepartures from quadratic. The instantaneous frequency,therefore, also exhibits minor departures from linear [10].The type of nonlinearity depends on the required per-formance. An improvement of about 1 dB in signal-to-noise is available from a nonlinear waveform compared toa linear waveform. It must be noted that a 1 dB gain in

signal-to-noise is equivalent to a 25% increase in transmit-ted power.

Because the frequency-time law is nonlinear, there is anincreased sensitivity to Doppler; i.e. the time shift foroptimum correlation is greater for nonlinear waveformsthan for linear waveforms for the same Doppler shift. Thedispersive slope changes over the frequency band andincreases with the weighting introduced. Thus, sidelobesuppression is directly related to Doppler degradation.

The design procedure to optimise the nonlinear wave-form for Doppler performance is not straightforward asthere is no exact analytical method. Approximations exist[108, 110] and successful designs have been produced.SAW technology has allowed the realisation of thesedesigns because of the ability to implement complicatedpatterns into the transducer, Fig. 4.

For Doppler frequency shifts less than 1% of band-width nonlinear FM can be used. The nonlinear FM wave-form [102] requires no time or frequency weighting forrange sidelobe suppression. This is because the FM modu-lation of the waveform is designed to provide the desiredpower spectrum. Thus, a true matched filter between theexpander and the compressor results. Signal-to-noise lossassociated with weighting is eliminated.

When symmetrical FM modulation is used with timeweighting to reduce the sidelobes, the resulting nonlinearFM waveform will have an ideal ambiguity function [105].This waveform has a frequency time function that invertshalf way through the expanded pulse. By using one half ofa symmetrical waveform, a nonsymmetric waveformresults, but this retains some of the range-Doppler cross-coupling of the linear FM waveform. The exactness of theambiguity function implies a rapid deterioration of per-formance with increasing Doppler and demands a knowl-edge of the range and Doppler variation. The resultingmatched filter response is the inverse of the Fourier trans-form of the squared magnitude of the transmitted spec-trum. Also it is proportional to the autocorrelationfunction of the transmitted waveform.

A SAW expander and compressor has been built [111]with 45 fis duration and a bandwidth of 830 kHz. It oper-ated at 60 MHz and was fabricated on lithium niobate.The desired range sidelobe specification was 30 dB degrad-ing to 26 dB with 10 kHz Doppler. This was achievedusing a nonlinear frequency-time law with a chirp rateclose to a reciprocal Hamming function. The mismatchloss was 0.15 dB. Nonlinear compression from 12.4 fxs to0.2 /is at 60 MHz has been achieved with 45 dB sidelobesuppression [108]. Ripple compensation was used in thedesign.

6.3 Dispersive delay-line realisation

6.3.1 Interdigital electrode structures: The design ofSAW dispersive delay lines using interdigital electrodestructures has been well documented [103, 112, 113]. Thereader is referred to these texts for such aspects of designas electrode geometry, apodisation and matching. Alsocovered are the physical effects of diffraction, electrodeinteraction and spurious bulk wave generation. The weigh-ting functions required for electrode apodisation aredescribed by Harris [107].

The inline transducers shown in Fig. 20 have been usedto produce large bandwidth dispersive delay lines. Such adevice with a bandwidth of 500 MHz centred on 1.3 GHz[104] exhibited an insertion loss <40 dB. The materialwas YZ lithium niobate and electron-beam lithographywas used to achieve the submicron pattern. The response


of a similar device operating at a fundamental frequencyof 750 MHz is illustrated in Fig. 21. The lower centre fre-quency allowed conventional photolithography, but

V//////////////A '//////////////A

WZ////////////A V7777Z//////////Fig. 20 Schematic SAW inline dispersive delay line

-20

-30

-40

or5 0

-S-60J

| - 7 0 -E

-80

-90

-100450 500 550 600 650 700 750 800 850 900 950 1000 1050 1100

frequency, MHz

Fig. 21 Measured insertion loss of dispersive SAW delay line with>500 MHz bandwidth

proved difficult to design because of the large fractionalbandwidth. Both devices used linear FM. The problemwith inline transducers lies in the Fresnel ripple in the fre-quency domain, caused by the abrupt truncation of thetransducer and the convolution of the two transducerwaveforms. This can be overcome by using the inverse-transform technique described in Section 6.2.1. However,the resulting truncation of the time domain functiondegrades the time sidelobes. Performance is furtherdegraded by interelectrode reflections.

These problems are overcome by the slanted transducer[105, 116] of Fig. 22. A flat power response over a widefrequency range is achievable with minimum insertionloss. Unwanted interference effects are removed, becausethe 'acoustic image' of the input transducer only overlaps

the receiving transducer in the region to which it ismatched.

However, inline transducers are used for time delaysexceeding 20 us. This is to reduce the area of substrateoccupied by the device. An example with 37 /is delay and4 MHz bandwidth on ST-quartz is illustrated in Fig. 23.

Fig. 22 Schematic of dispersive SAW delay line using slanted IDTs


Fig. 23 Typical SAW dispersive delay line with large time dispersionfabricated on ST-cut quartz

The frequency-time law is nonlinear, and conventionalphotolithography was used. To avoid positional errorsassociated with photolithography on long delay time, dis-persive transducers composed of groups of periodic ringershave been employed [117].

The slanted transducer has been the subject of consider-able interest because of its obvious attributes. Very widebandwidth devices are realisable. Fractional bandwidths of63% have been recorded at 320 MHz [108] and1600 MHz [119]. The latter was a linear down-chirp withan insertion loss of 45 dB on 128°-rotated lithium niobate.Because the slanted transducer configuration allows pro-pagation paths for each frequency to be distributed spa-tially, it is possible to phase compensate with metal stripesin the acoustic path [120]. Reduction of time sidelobes byas much as 5 dB [120] have been observed for linear FM.

Simulated sidelobe levels as low as — 50 dB have beenreported without distortion at moderate compressionratios [108]. Realised time sidelobes after truncation of— 41 dB have been achieved using inline transducers andthe inverse Fourier transform technique. Slanted trans-ducers eliminate the inherent limiting factors associatedwith inline transducers, and with the incorporation of thephase compensation plate should realise very low sidelobelevels close to theoretical predictions.

Projection printing is simpler and, therefore, cheaperthan electron-beam lithography. However, its use islimited to frequencies below 1 GHz. Third-harmonic oper-ation extends this frequency range beyond 1 GHz. Suchoperation is achieved normally through a wide [122] elec-trode pattern (mark to space 4:1) or by double electrodes[123]. Both inline [122] and slanted transducers [123]have been used for third-harmonic operation. Bandwidthsof 500 MHz [124] have been achieved at 750 MHz, withan insertion loss of 38 dB.

High-frequency operation introduces a severe matchingproblem. This has been reduced by the adoption of micro-strip matching circuits [119]. A similar problem occurswith long delay devices. Here transmission-line matchingtechniques [125] are used. The current trend in radar istowards modules containing SAW devices, matching net-

201

works, amplifiers and logic circuitry. An example of anexpander and compressor module is illustrated in Fig. 24.

Fig. 24 Expander and compressor subsystem modules incorporatingSAW dispersive delay lines

A system block diagram is shown in Fig. 25 and typicalresults are shown in Fig. 26.

6.3.2 Reflective array compressor (RAC): This techniquehas been widely investigated in recent years and the readeris referred to Matthews [112] for a review of the relevant

design details. The basic structure is shown in Fig. 27. Asignal emanating from one transducer is transferred to theother transducer by two corner-turning manoeuvres. Thereflection is spatially frequency dependent and is achievedby deposited metal stripes or grooves.

As the propagation path is folded, an RAC is normallyused for long time delays and, thus, large time-bandwidthproducts. Exceptionally good phase and amplituderesponses can be obtained primarily due to the lack ofsensitivity of these devices to fabrication defects andsecond-order effects.

Groove RACs require ion-beam etching. A groove-depth-weighted linear RAC with a time-bandwidthproduct of 16200 has been reported [126]. The centre fre-quency was 400 MHz and bandwidth 180 MHz. The inser-tion loss was 42 dB and the material was YZ lithiumniobate. Time delays of 125 /is have been produced [127]with groove-depth-weighted RACs on bismuth germaniumoxide. Amplitude weighting by uniform etching using ion-beam or chemical means is achievable through the use of achevron structure [128]. Such constant groove depthdevices have been produced at a 1 GHz, with a bandwidthof 500 MHz and a dispersion of 1 ^s [129].

It is possible to adjust the phase response of an RAC bymeans of a metal-film pattern deposited in the regionbetween the reflectors [130]. This enables the device to betrimmed to obtain the desired performance. Sidelobe levels

trigger

COHOO

70MHz c

1

>

timing

impulsegen

timing

i

impulsegen

SAWexpander

limitingamplifier

SAWexpander

limitingamplifier

O O

L* .̂

oo

UP

1

\

5

matchingpad

matchingpad

SAWcompressor

SAWcompressor

amplifier

amplifier

ooOO

O^OO

i

Fig. 25 Expander and compressor subsystem block schematic


approaching 40 dB are readily achieveable [112] using thistechnique.

The design of the groove-depth profiles and the phase

compensating metal film for linear FM RAC devices isbased on the linear relationship between frequency andposition. Clearly, for nonlinear RACs this is not the case.By relating the groove depth and position to the impulseresponse of the device [131], it is possible to design nonlin-ear RACs. A nonlinear RAC expander-compressor pairoperating at 60 MHz with a 45 fis expanded pulse andsidelobe level of 31 dB in the absence of Doppler has beenreported [111].

The typical RAC substrate is lithium niobate. For ade-quate temperature stability such devices must be operatedin an oven. Alternative wide-bandwidth temperature-stableRACs are of considerable interest to system designersbecause of the size, weight and power consumption oflithium niobate RACs. Two cuts of quartz look promising,namely ST-cut and RAC-cut [132]. Devices have beenmade on both cuts [133] and exhibit chirp slope changesof 400 p.p.m. for ST-cut and 50 p.p.m. for RAC-cut over100°C.

The majority of RAC devices employ grooves. Metalstripes were first used in the chevron pattern [128] toachieve amplitude weighting. An alternative approach is tophase weight the stripes by using different widths [134].RAC devices using metallised fingers with time-bandwidthproducts greater than 400 have been manufactured [135]on YZ lithium niobate. Similar devices have been reportedon bismuth germanium oxide [136]. The advantage of thisdevice is the one-step photolithographic process for bothreflectors and transducers.

The slanted transducer produces a flat power response.When used as the input transducer to an RAC, this pro-perty enables weighting for pulse compression to be putinto the transducer. However, as the transducer is slantedthen the RAC must also be slanted [116]. The slantedRAC so produced eliminates unwanted interstripe reflec-tions. Such an SRAC with 100 MHz bandwidth and atime-bandwidth product of 1000 has been produced [116]at 150 MHz. Metal stripes were used as the reflectors.

Recently [137], a 180° reflection device has been manu-factured. The incident and reflected beams are separatedby a multistrip coupler. A device with a bandwidth of50 MHz and a time-bandwidth product of 1000 has beendemonstrated on YZ lithium niobate using grooved reflec-tors. This technique offers the promise of a lower insertionloss.

Fig. 26 Typical SAW expander and compressor subsystem measure-ments

a Waveform of expanded pulse; b compressed pulse; c expanded version of bdemonstrating the sidelobe structure

reflective arrays

phase plate

Fig. 27 Schematic construction of a SAW reflective array compressor(RAC) employing groove or metal strip reflector banksSignals travel from input to output via two Bragg reflections. A fine correction tothe performance can be made by means of the phase plate, and typically results inan RMS phase error of only a few degrees.

6.3.3 Dot array devices: This is a technique for producinglow-cost, high-performance reflective-array devices andwas first investigated by Solie [138]. The array of reflectingetched grooves is replaced by an array of reflecting metaldots. The principal advantage is single-step fabrication, asthe reflecting dot array is a part of the same mask and thesame metallisation. In addition, weighting can be incorpo-rated into the mask without the use of apodisation. Bothround and rectangular dots have been used and theoreticalmodels exist [138, 139].

The metal dot introduced by Solie [138] relies for itseffectiveness on mass loading, and is thus dependent onmetal thickness. Both aluminium and gold dots have beeninvestigated [140]. The number of dots in a row deter-mines the reflectivity. Such reflective dot arrays have beenapplied to bandpass filters [141], linear FM filters [142]and nonlinear FM filters [140] on both lithium niobateand ST quartz [140, 143]. Techniques for reducing reflec-tions due to unwanted Bragg interactions exist [139]. Alinear FM dispersive filter with a time-bandwidth productof 2000 at a centre frequency of 350 MHz on lithiumniobate has been reported [143].


The reflectivity of a dot can arise from the surfacetopography, the impedance change due to mass loading orto electrical shorting of the surface piezoelectric fields. Therelative importance of these factors is dependent on themechanical properties of the dot and the piezoelectricproperties of the substrate. Thick metal dots require strin-gent processing control as both reflectivity and wave velo-city in the array are dependent on metal thickness. Thisrequirement is removed using thin metal dots whose reflec-tivity is derived from electrical shorting [144, 145]. Recentwork at Oxford University [145] has led to the estab-lishment of a design procedure for the dot reflective-arraycompressor with thin aluminium dots, for which the reflec-tivity is independent of metal thickness. Initial results ona device with a time-bandwidth product of 1000, centredon 160 MHz and having 25 /xs dispersion, were in goodagreement with theory.* Phase error is approximately±6° RMS with chirp slope correct to 1 part in 600. A 180°reflecting metal dot array has been reported by the samegroup [146].

6.3.4 Compressive or microscan receiver: The SAW dis-persive filter and the principles of pulse compression canbe used to produce a powerful signal analysis tool. Thecompressive receiver [147] represents an instrumentcapable of performing the Fourier transform in real time atvery high data rates. Fourier transformation of signalsoccupying a definite frequency channel may be regarded asmatched filtering or correlation against reference wave-forms (correlation processor [105]). This technique is rep-resented by the chirp transform [147] and can beimplemented in two distinct ways as shown in Fig. 28.

convolvingfilter

Cout

dispersive (chirp)generator


convolvingfilter

C

convolvingfilter

C-•-out


Fig. 28 Two configurations commonly employed to perform the chirptransform operation

SAW dispersive devices are ideally suited for the convolverand chirp generation functions. Standard mixers can beused for the required mixing functions. Thus, a real-timespectrum analyser is easily realisable using available com-ponents.

For optimum processing, the convolving filter in theMCM case (see Fig. 28) must have twice the time-bandwidth product of the signal to be analysed. Con-versely, for the CMC case (see Fig. 28), the multiplicationfunction is twice as large. Normally, the MC of either theMCM or CMC configurations is used, because, in mostcases, only the modulus of the Fourier transform isrequired. The duty cycle is 50%. However, theoreticallythe full CMC processor which includes input band limitingis capable of 100% duty cycle.

* E.G.S. Paige—private communication

204

A compressive receiver having 1 GHz of bandwidth isillustrated in Fig. 29. The overall bandwidth is divided into

Fig. 29 Photograph of SAW compressive receiver in operation

four channels each analysing 250 MHz. The SAW devicesreferred to in Fig. 21 are used in this instrument. Theresolution over 1 GHz of bandwidth is 4 MHz with adynamic range greater than 60 dB. The disadvantage ofthis technique is the output data rate which requiresspecialised high-speed data processing. A priori informa-tion on the incoming data greatly assists this problem.However, the advantages include large dynamic range, lowcost and high MTBF due to the simplicity of the systemimplementation. In addition, the data is available for pro-cessing in microseconds.

7 Oscillators and resonators

Most oscillators can be modelled as an amplifier with afrequency filter as the feedback element, providing positivefeedback at the frequency of interest. In the case of SAWoscillators (SAWO) the construction is literally of the formshown in Fig. 30A, the feedback element usually being a2-port delay line or resonator (Fig. 6b) on ST-cut quartzfor temperature stability, Table 1. This arrangement wasoriginally devised by Maines et al. [148] to aid the mea-surement of the temperature coefficient of SAW delaytimes on various substrates (Table 1), but has subsequentlybeen developed into a useful single-mode oscillator struc-ture [33]. For oscillation at frequency f0 the following con-ditions should be satisfied:

(a) The open-loop gain G should exceed unity(b) The total-loop phase shift, acoustic plus electrical,

should equal 2nn where n is an integer, i.e.

2nf0 x + = Inn

phaseshifter

(12)

out

Fig. 30A Schematic of SAW oscillator employing a delay line as thefeedback elementThe phase shifter <j> is optional but may be used for frequency adjustment, tem-perature compensation or frequency modulation.


This equation is valid for a SAW oscillator controlled by adelay line of delay T. For a 2-port resonator the effective

f i - f 2out tocounter

Fig. 30B Schematic of a two-SAWO scheme configured for sensorapplications

This provides temperature compression and a digital readout from the counter. Thefrequencies of the two halves are offset initially to provide a monotonic output andto avoid frequency-locking phenomena.

delay is xeff = Q/nf0, where Q is the loaded quality factor,and which may be as high as 104 [149]. In either case, theacoustic phase shift and its first derivative with frequencytypically dominate 0amp and its frequency derivative byseveral orders of magnitude, resulting in good frequencystability as discussed in the following.

For single-mode operation it is sufficient to ensure thatconditions (a) and (b) above are only satisfied simulta-neously at or near / 0 . This is readily achieved by thedesign of the SAW feedback element which, in the case ofthe 2-port delay line, is merely an example of the design ofSAW bandpass filters as described in Section 4. Fig. 31shows the electrical response of such a 2-port delay lineemploying the bloomed IDTs of Fig. 3e. The sidelobe sup-pression of order 20 dB is more than adequate to ensuresingle-mode oscillation at the centre frequency of958 MHz, and the phase characteristic is ideal to imple-ment a linear FM capability [33] by the inclusion of anelectrically controlled phase-shift element in series with theamplifier, (Fig. 30A). The linear variation of frequency with(j)amp is evident from eqn. 12. The passband distortion dis-cernible in Fig. 31A arises from triple-transit effects (Fig. 5)and can be virtually eliminated by removing the IDT

Fig. 31A Measured response of 958 MHz SAWO delay-line filter

The ST-cut quartz chip together with the 5 mm long gold wire bonds, which shunttune the 'bloomed' transducers, are housed in a T08 package. The upper trace(10 dB/div) shows a minimum loss of 12.9 dB, and the lower trace (180°/div) showsan almost linear phase slope corresponding to Q ̂ 1000. The horizontal scale is 1MHz/div.

Fig. 31 B Response of untuned 75 MHz 2-port SAW resonator on STquartz, illustrating the nonlinear phase response (lower trace, 45°/div)

If the transducers are tuned, the loss (upper trace, 10 dB/div) is reduced fromc^20 dB to 4 dB, and the Q from ~ 12000 to 2000. NB The horizontal scale is4 kHz/div; compare with Fig. 31 A.

tuning and accepting a higher loss of ~ 17 dB. However, inmany applications, especially above 500 MHz, low loss ismore important than the fidelity of the passband charac-teristics, whose only significant effect is on the FM linear-ity. This is one reason for the interest in 2-port SAWresonators, which have a highly nonlinear phase character-istic, but which can often be operated with a loss of ~6dB. In the near future, the SPUDT [22, 24] employingmass-loaded reflections on quartz may offer the best ofboth worlds, with low loss and an agreeably linear phaseresponse.

The attractions of SAWO include most of those outlinedearlier for SAW devices in general. For example, the SAWchips are small and planar, and are readily integrated witha microelectronic amplifier. A typical application is as arugged miniature UHF FM transmitter, or as the receiverlocal oscillator. Another important application is as theprimary source for microwave generation, e.g. for use in aradar transmitter or receiver. The principal attraction ofthe SAWO over a conventional quartz-crystal oscillator,for such applications, concerns the high fundamental fre-quency of operation. As a consequence, the microwavesource requires a smaller multiplier chain, and enjoys theadditional advantage of a greater separation between theoutput frequency and the closest spurious sidebands whichare inevitably generated in the multiplication process. Suchan SAWO at a fundamental frequency of 1030 MHz hasoperated satisfactorily at the mast head of an RSRE radarsince 1974, where its advantages of smallness, low powerconsumption and reliability have proved especially benefi-cial. In radar applications another important aspect of per-formance is the short-term stability, and, in this respect,SAWO, provide an excellent and largely predictable per-formance. Nowadays, the short-term stability of oscillatorsis usually described in terms of the output spectrum, whosegross features are illustrated in Fig. 32. The output ofSAWO (and other oscillators) is, essentially, highly filteredwhite noise, with the AM component stripped off by thelimiting action of the saturated sustaining amplifier. Thecentral region of Fig. 32 derives from Johnson noise andshows the classical decline in sideband power at 20 dB/decade as a result of the action of the feedback loop [33].For greater offsets, the feedback is negligible (e.g. for>2 MHz in Fig. 31 A) and the noise plateau simply arisesfrom white Johnson noise in the amplifier. The 30 dB/


decade region in Fig. 1 arises from the ubiquitous pheno-menon of 1//noise. This is presumed due to relatively slow

fc - 8 0

oo

o

~ -100a

--120-

o*550/

~ -160

.a I/f noise (30dB/decade)

Johnson noise(20 dB/decade)

c plateau

0.1 10 100offset. kHz

1000

Fig. 32 Schematic output spectrum of SAW oscillatorThe absolute offsets and sideband levels indicated are typical of an SAWO of mod-erate stability at a centre frequency of a few hundred megahertz.

fluctuations in the device parameters (e.g. the amplifierand/or SAW device phase shift), but the precise physicalmechanisms occurring are not well understood in SAWOor any other device [150, 151].

In the case of SAWO, it has been shown that the noiselevels in regions b and c of Fig. 32 are minimised by theuse of low-loss high-phase-slope SAW components, andlow-noise high-power amplifiers. The 1//noise is best mini-mised experimentally, e.g. by comparing makes of ampli-fier. In optimum circumstances, the spectral purity of SAWoscillators can be extremely impressive and comparable toany competition [151, 152]. The principal development inSAWO, in recent years, has come about from the use ofthe SAW resonator as the feedback element. It has notproved possible to make a direct SAW analogue of thefamiliar bulk wave resonator, as no one has yet devised ahighly efficient broadband reflector of SAW equivalent tothe free faces of a quartz crystal. To overcome this defi-ciency, Ash [19] introduced the distributed reflector shownschematically in Fig. 3b and comprising a periodic array ofweak reflectors spaced 1/2 apart. Such an arrangementprovides essentially lossless reflection albeit over a narrowfrequency band, and two such reflector banks form ahigh-Q cavity. Fortunately, that is all that is necessary (ordesirable) to sustain a stable oscillator. Many of the earlyinvestigations into SAW resonators employed LiNbO3substrates, and it remained for Marshall [153] to demon-strate the practicability and design details of resonatorsusing metal-strip reflectors on quartz substrates. Suchdevices are fabricated by conventional single-stage photo-lithography and have a respectable temperature coefficient(Table 1). SAW resonator-controlled oscillators haverecently been incorporated into production equipment,Elliott et al. [149], Ebata et al. [154] and two companies(SAWTEK and R.F. Monolithics, USA) have each sold> 1 million such devices to date.

It might be imagined, from this discussion, that the res-onator has replaced the delay line as the principal SAWOcontrolling element. This is far from true, however, thedelay line offering advantages in various other respects,such as power handling, FM capability and tuning range.For example, the device of Fig. 31A can evidently be tuned

over ~1 MHz, which is more than adequate to take upmanufacturing tolerances and its own variation with tem-perature, allowing the device to be electronically trimmedand temperature compensated, and still provide a wideFM capability. There are so many factors involved intypical oscillator specifications that it is not possible togive simple guidelines on the choice between resonatorsand delay lines.

Numerous other aspects of SAWO performance havebeen studied in the past, and in the following we review themost widely investigated topics of recent years. Theseinclude attempts to optimise the temperature stability andlong-term stability (aging), to optimise the vibration sensi-tivity, and the use of SAWO as sensors. In addition, weinclude brief mention of a closely related family of oscil-lators employing surface-skimming bulk waves rather thanSAW.

As long ago as 1970, the ST-cut of quartz with propaga-tion along the X-axis was identified as an orientation witha rather favourable combination of SAW properties,Schultz et al. [155], and it has remained popular eversince. In particular, the first-order temperature coefficientof delay vanishes for this cut, making it of importance inmany applications, especially in oscillators. However, it isapparent from Table 1 that the change in delay arisingfrom the second-order coefficient amounts to ~ 100 p.p.m.over the temperature range from — 40°C to +70°C. Thistranslates directly into the fractional change in frequencyof an SAW oscillator and is about an order of magnitudeworse than that of the familiar AT-cut used in convention-al bulk-wave quartz-crystal oscillators. To remedy thissituation, numerous attempts have been made to identifyother temperature-compensated materials; and while thesehave often been successful in so far as the first-order tem-perature coefficient vanishes, it happens that the second-order coefficients are significantly worse than those ofquartz, e.g. A1PO4 and lithium borate, Table 1. Otherworkers have sought superior orientations of quartz itselfby extending the earlier investigations on singly rotatedcuts (of which the ST-cut is an example) to the moregeneral case of doubly rotated cuts with propagation in anarbitrary direction thereon. The most complete search ofNewton [156] has revealed several cuts which offer animprovement over the ST-cut by a factor of about 2. Sucha modest improvement has not proved sufficiently attrac-tive, however, for any of these new cuts to be adopted in alarge-scale production. In the case of delay-line-controlledoscillators, an alternative approach to improved tem-perature stability is simply to incorporate in the oscillationloop an electronically variable phase shifter, Fig. 30A,driven for example by a temperature-dependent correctionvoltage, which is itself derived from a thermistor circuit.This approach is frequently used on conventional bulk-wave quartz-crystal oscillators. The solution is, however,not usually feasible with a resonator-controlled SAW oscil-lator whose response may well be narrower than its overallchange with temperature.

Two other novel approaches to improved temperaturestability of SAW devices on quartz have been described,both of which exploit the planar nature of the substrate.The first involves the use of two, or more, delay lines onthe same substrate, but with different propagation direc-tions and temperature coefficients of delay. When these areconnected in parallel electrically, the combination canhave markedly superior temperature stability to either ofthe individual delay lines [157]. This approach has alsobeen extended to SAW resonators [158]. The secondapproach exploits the anisotropy of the crystalline sub-


strate, one consequence of which is a temperature-dependent beam-steering angle; i.e. the angle between thegroup velocity and wave vector. This phenomenon can beexploited to provide temperature compensation in anSAW device by deploying on the substrate auxiliaryoutput transducers connected in parallel electrically withthe main output transducer. The scheme operates becausethe contribution to the overall response from these aux-iliary transducers varies with temperature due to theirvariable degree of illumination by the input transducer,Ballato et al. [159]. The feasibility of each of these schemeshas been demonstrated experimentally, but a full designprocedure has not yet been reported for either.

An extensive review of the various techniques devisedfor the temperature compensation of SAW oscillators (andmany other SAW devices and subsystems) has beenpublished by Lewis [160]. Even so, it is a simple fact thatthese temperature-compensation schemes have not beenwidely used, because they detract from one of the principalattractions of SAW devices; namely, their simplicity.Instead, devices or systems demanding great temperaturestability commonly employ ovens with their obvious pen-alties of size, cost and power consumption. It will, there-fore, be quite apparent to the reader that the discovery ofmore temperature-stable materials would be of greatbenefit to the SAW activity, and nowhere more so than inoscillator applications.

A common requirement in oscillator specifications is fora certain frequency stability from all causes, i.e. includingvariations in temperature, supply voltage and aging overthe device's lifetime (typically, 5 to 25 years). As discussedin the preceding Section, various techniques exist for thetemperature compensation of SAW oscillators and, ifnecessary, they can be ovened. Similarly, the supplyvoltage can be stabilised. However, it is not possible tocorrect for aging unless another frequency standard isemployed; e.g. if the SAWO output is divided down to~5 MHz and locked to a conventional quartz-crystaloscillator. This latter procedure obviously detracts greatlyfrom the simplicity of the SAWO, although it often pro-vides a better solution to high-frequency generation thanthe use of multiplier chains. So it is that aging is often oneof the limiting factors in the overall stability of SAWOs.Early investigations showed that laboratory-fabricateddevices could age by as much as 100 p.p.m. in their firstyear of operation, probably as a result of surface contami-nation and/or the relief of mounting strains. Over the pastdecade, various investigators have demonstrated reducedaging as a result of improved processing, e.g. by copyingbulk quartz-crystal fabrication procedures, and it appearsthat aging levels of ~1 p.p.m./year can now be achievedunder normal operating conditions [152, 161]. Theseadvances are of great importance and suggest that the lim-iting factor in the overall stability of SAWOs is now thetemperature dependence, emphasising the desirability ofmore temperature-stable substrated materials.

When an oscillator is subjected to mechanical vibrationthe frequency becomes modulated at the vibration rate,leading to spurious sidebands in the output spectrum. Inmany applications, such as Doppler radar, it is vital tominimise this effect. At first sight, it appears that SAWOhave a great advantage over conventional quartz-crystaloscillators, in this respect, because the SAW element'scharacteristics (e.g. Fig. 31) are virtually unaffected by thesubstrate thickness and mounting arrangement, which cantherefore be chosen to minimise the vibration sensitivity.However, the preliminary results published to date onSAWO show comparable behaviour in this respect, with

sensitivities of order 10"8 to 10~10/g [162, 163]. Thisprobably reflects the vast effort expended on the designand mounting of conventional quartz crystals over the past40 years. We speculate that considerable improvements arepossible in this aspect of SAWO behaviour, but, of course,the mounting procedure adopted must not degrade theaging characteristics, e.g. through outgassing.

It is evident from the foregoing discussion that SAWOspossess various attractive features, e.g. their planar con-struction and high-frequency operation, but are currentlysomewhat inferior to conventional quartz-crystal oscil-lators in other respects like aging and temperature stabil-ity. In an attempt to combine the more attractive featuresof each technology, several groups have investigated theuse of surface-skimming bulk waves (SSBW) [164], other-wise known as SBAW [165]. Such devices retain theplanar construction of SAW, but the orientation of thepiezoelectric substrate is chosen such that the IDTs act as(acoustic) endfire array antennas launching and receivingbulk acoustic waves propagating close to the surface. Todate, the principal advantage demonstrated for SSBWoscillators is their higher fundamental frequency of>3 GHz [165]; other aspects like aging and the vibrationsensitivity have not yet been studied extensively.

We conclude this Section by discussing the use ofSAWOs as sensors. A well known procedure for measuringany physical parameter is to convert it to a frequency,which can itself be measured cheaply and with great preci-sion. The simplicity of construction of SAWO make themespecially suitable for this purpose, while the dual-channelscheme illustrated schematically in Fig. 30B provides ahigh degree of common mode rejection of the frequencyshift due to temperature changes. A further attraction ofsuch schemes is the essentially digital format of the output,which can be fed directly into a computer, e.g. for processcontrol. Such arrangements have been demonstrated forthe measurement of pressure [166, 167], acceleration [168]and voltage [169], and closely related devices have beendescribed for temperature measurement [170], gas detec-tion [171], rotation measurement [172] and for use asgraphic sensors [173] and hydrophones [174]. It isevident, from this brief discussion, that SAWO are poten-tially capable of meeting many modern sensor require-ments and form a significant part of the current worldwideactivity in sensor research and development.

8 Convolvers

In Section 6 of this review we described the use of SAWdevices as matched filters for signal processing. All thedevices described in that Section were designed for a fixedwaveform. The convolver is a nonlinear SAW device whichcan operate as a programmable matched filter, with com-plete waveform adaptability within its own time and band-width limitations. In this respect, it differs from theprogrammable tapped delay lines of Section 3.1, whichhave only been demonstrated for biphase-modulated andMSK waveforms. The physical basis of the convolver oper-ation lies in the generation of an electrical polarisation Pthrough the bilinear interaction of two counterpropagatingSAWs launched from transducers at opposite ends of adelay line. Owing to this parametric mixing under thecentral electrode, an output voltage is developed at twicethe input frequency, and which is spatially uniform (k = 0)as a result of /c-conservation in the interaction. The high-frequency signal on this electrode is filtered to form theconvolver output. Consider two (input) signals F(t — z/V)and G(t + z/V), propagating in opposite directions on the


substrate. When the two signals overlap, the voltage on theoutput electrode at any instant t is given by the spatialintegral

fi/2C(t) = B F(t - z/V)G(t + z/V) dz (13)

J-L/2

where L is the length of the electrode as shown in Fig. 33.

inputIDT 1

parametric electrode

i: inputIDT 2

output

Fig. 33 Simple schematic of principles involved in SAW monolithic con-volver

Two counter-propagating SAWs are launched from IDTs at each end of the device.These waves interact nonlinearly under a parametric electrode, producing a time-compressed output signal which corresponds to the convolution of the two acousticbeams provided they are completely contained within the output electrode.

If F and G are totally contained within the electrodeduring the interaction, the limits of the integral can beextended to ± oo without affecting the value of C(t). Usingthe transformation T = t — z/V, we obtain

C(i) = - VB \ F(x)G(2t - T) dx (14)

which is an expression for the convolution of the wave-forms F and G. The factor of two in the argument of Gdemonstrates that in the acoustic convolver the output iscompressed in time by a factor of two. Physically, thisarises because the counterpropagating waves have a rela-tive velocity of 2V. This compression introduces problemsdue to time segmentation effects [203] and due to theincreased bandwidth of the output signal. However, thesedisadvantages are offset by the significant advantage ofbeing able to filter the output signal from the input signals,an advantage not shared with the tapped-delay-line prog-rammable matched filter of Section 3.1. Further, theoutput is free of clocking spikes, as the modulation isapplied in a separate compartment of the overall system.In the convolver, we have seen that the fixed IDT patternof the conventional matched filter has been replaced by apropagating SAW whose waveform is the time reverse ofthe anticipated signal. It is worth mentioning, at this stage,the different definitions of convolution efficiency thatappear in the literature. Systems engineers are not con-

cerned with internal material efficiencies, but rather withthe external efficiency obtained by a 'black box' measure-ment in a 50 Q system. Conventionally, this efficiency isdefined by the relationship

(15)P1P2

where PY and P2 are the two input power levels and Pout isthe output power level. The efficiency defined in this wayhas units of inverse power (dBrcT1). Almost all recentpapers use an alternative definition, namely the outputpower when 0 dBm is applied to each input port. In thislatter case, the units of efficiency are dBm. Typically, theefficiency of modern convolvers is in excess of — 70 dBm(see Table 3). As typical input powers to the devices maybe of order 20 dBm this convolution efficiency results inoutput powers in excess of —30 dBm, allowing a usefulsignal/thermal-noise ratio of > 60 dB.

The convolution of two acoustic signals was firstdemonstrated by Quate and Thompson [175] in 1970.There followed several years of academic research into thedevice, but interest waned until systems were devised forwhich this device is uniquely suited, performing the equiv-alent of 1011 ten-bit complex multiplications/second in itsconvolution role. Recently [178-202] much work has beencarried out on the detailed structure of the device through-out the world, and relatively high-efficiency devices havebeen produced. Although, at the beginning of the renewedeffort, a wide variety of structures and materials wereemployed, the last three years have seen a convergence onmonolithic devices on YZ-LiNbO3, as originally demon-strated by Defranauld and Maerfeld [204]. Most of thesedevices employ acoustic beam compression to enhance theacoustic energy density and, hence, convolution efficiency.

The basic components of a monolithic SAW convolverare shown in Fig. 33 and comprise the two input trans-ducers and one output parametric electrode. This para-metric electrode is a simple metallised plane over theinteraction region. Such a metallic film on a piezoelectricsubstrate reduces the SAW velocity and also acts as a Av/vwaveguide, Coldren and Schmidt [176], resulting in acous-tic beam confinement during propagation. Batani andAdler [177] demonstrated, experimentally, that this con-finement resulted in improved external convolution effi-ciency for a device with fixed input powers. However, thefraction of the acoustic energy outside the waveguidebecomes more significant as the waveguide width W is

Table 3: Review of experimental data for acoustic-wave-based convolvers

Integrationtime

//S7.5

1212102010.810202222

2.5161722

CO

C

O

32

3 dB inputbandwidth

MHz1020

24095

10092

100100100100

76120

759021

10020

TBproduct

75240

2880950

200010001000200022002200

190182012751980

3361600

640

Inputfrequency

MHz1350

100500300300300290290

30300800300300300300300124

Convolutionefficiency

dBm-1

—- 5 6

-75-81-70- 6 1- 6 7- 7 0- 6 6- 6 9- 7 8- 6 7- 6 3- 7 0-53-65- 6 0

Suppression ofself-convolution

dB—

———

> 4 0—

> 3 0> 3 3—

> 4 3> 3 4> 3 5> 5 5> 2 6>40>40> 4 0

Reference Comments

175194194*1979,200199184184198*1981,197189•1982,•1982,23, 19 •23178

1 st acoustic convolver

> Si over lithium niobate.

p. 729Si over lithium niobate

p. 181

p. 149p. 124

> Variable bandwidth

* Proc. IEEE Ultrasonics Symp.


reduced, and results in an optimum width of W ~ 1.5A0,where Ao is the wavelength at the centre frequency [178].

The narrow acoustic beam needed to feed the para-metric electrode introduces its own problems, which haveonly recently been overcome. Conventional transducerswith low apertures have an inconvenient electrical imped-ance; namely, a series radiation resistance much greaterthan 50 Q (Section 2) and a low capacitance which rendersthe transducer susceptible to strays. Various alternativeapproaches have been investigated; namely, beam com-pression, chirp transducers and focused interdigital trans-ducers (Fig. 34). The most popular approach uses a

JFig. 34 Several methods which have been used by various workersaround the world to couple acoustic waves efficiently to a narrow waveguidea Parabolic horn compressor [188, 190], b multistrip coupler compressor [184,185], c prism coupler [209], d focused IDTs [205] and e chirped IDTs [189]

larger-aperture conventional transducer, with a convenientelectrical impedance and some beam-compression tech-nique. Three forms of beam compressors have beendemonstrated: horn compressors [190, 188] developedfrom planar optics [186], multistrip-coupler compressors[184, 185], and prism couplers. Alternatively, the para-metric electrode can be fed directly from either chirp trans-ducers with a narrow aperture, and with a convenientelectrical impedance [189], or from a focused interdigitaltransducer (FIT) [205] which relies on matching theacoustic beam profile at the focus of a curved IDT to theacoustic waveguide fundamental mode. An additionaladvantage of the chirp transducer approach is its ability tocompensate for any dispersion in the waveguide.

The discussion in this Section, so far, has consideredeffects which are of primary importance to the efficiency ofthe device. We now review some of the many secondaryeffects which have been studied over the past few years,

and which have transformed the convolver from a labor-atory curiosity to a viable component. Waveguide multi-moding, dispersion and attenuation can adversely affectperformance. When multimoding is present the variousmodes have different velocities, which results in inter-ference effects between these modes [179]. These effects areeliminated by reducing the waveguide width W to thepoint where only the fundamental symmetrical mode canpropagate freely. The group velocity dispersion of this fun-damental mode can also be minimised by the choice ofwaveguide width [180], although the dispersion inherentin a Av/v waveguide is modified by mass loading and isespecially important at higher frequencies [178]. Theeffects of dispersion and attenuation on SAW convolverperformance have been discussed by Adler and Cafarella[182]. Fortunately, the implementation loss introduced bythe observed dispersion is of modest proportion. Attenu-ation due to propagation loss only affects the overall effi-ciency of the device and does not degrade the fidelity of theconvolved signal. Another problem inherent to the outputplate concerns the spatial uniformity of the signal alongthe plate. An accurate technique for measuring the plateuniformity, with 25 ns resolution, has been described bySelviah et al. [181]. In an ideal output plate, the amplitudeand phase of the contribution from any point of the plateis independent of the position of that point, so ensuringspatial uniformity of the summation. Several effects mayadversely affect this uniformity; plate resistance and elec-tromagnetic long-line effects predominating.

For a typical film thickness of 1500 A, a parametricelectrode 3A0 wide at 300 MHz and 16 jus long has atypical end-to-end resistance of 400 Q. The electrode must,therefore, be connected at several points so that the resist-ance does not significantly reduce the output and the leveldoes not depend on location. Also, owing to the highdielectric constant of lithium niobate (e ~ 46), the velocityof electromagnetic waves along the parametric electrode isrelatively low, and it is common to have a structure whichis about one half of the electromagnetic wavelength at theoutput frequency. This leads to standing-wave problemsand significant plate nonuniformity. It has been shown byGoll [210] that inductive terminations to the ends of theplate lead to acceptable efficiency and plate uniformity.The effect of other terminations on the output has beenstudied by Adler [183].

Further design features of the convolver are necessaryto eliminate unwanted spurious signals; many of thesesignals and their solutions were originally described byMotz et al. [191]. The most important spurious signal inthese devices is the self-convolution signal. This is due toone input wave convolving with itself after reflection fromvarious features of the device, e.g. waveguide ends, com-pressors, transducers or even fabrication defects such aspinholes [190]. These reflections are minimised by conven-tional SAW procedures, e.g. the use of split finger IDTs(Fig. 3b). As these reflected waves can never be totallyeliminated, Fig. 5, to obtain larger suppression of spurioussignals, it is necessary to use the dual-track cancellationscheme described by Motz et al. [191]. This scheme essen-tially consists of two almost identical convolvers, with theinput transducers connected in parallel. One of the fourtransducers has its electrode polarity reversed so that thetwo convolution signals are in antiphase and are extractedas a difference signal. Because the features of the twotracks are nominally identical, selfconvolution signalssuffer common-mode rejection. In practice, however, minorfabrication defects destroy the ideal balance, but, typically,result in an acceptable selfconvolution suppression of


40 dB. The two-track scheme also provides a more repro- technological solution to the problem, there are moreducible output port impedance [187]. elaborate forms of convolver. In the acousto-optic convol-

multiple tapsangled waveguide ends

dual convolver schemeabsorber

waveguide terminationinput 2

input 1

split aperture transducer

sand-blastedrear surface

horn compressor earthedand with stepped ends

Fig. 35 Schematic of monolithic SAW convolver

The structure shown here represents the device designed at RSRE, Great Malvern,but the principles involved have been addressed by researchers all over the world.In particular, the various devices differ in methods of beam compression (see Fig.34). The inset is a photograph of a real device and demonstrates the small size of theconvolver. In this photograph the planar coils at either end of the device are used toseries tune the transducers.

An alternative two-track scheme has been described byLewis and West [190], in which the phase shift betweenthe tracks is introduced using a metallic phase plate. Thiscomponent is frequency-sensitive, only producing the 180°phase shift at the centre frequency, but it has sufficientbandwidth to be of value in a 100 MHz bandwidth devicenear 300 MHz. Operated as a delay line, this arrangementoffers the diagnostic advantage of giving a direct reading ofamplitude and phase imbalances between the two tracks,allowing defective devices to be rejected before fullassembly [187].

Several groups [178-202] have developed widebanddevices with comparable performance. In Fig. 35 we showthe features of our own particular arrangement which hasbeen incorporated into a commercially available module(Fig. 36). The performance of such a convolver with abiphase m-sequence code is shown in Fig. 37.

In this Section, we have concentrated on monolithicSAW convolvers. While these devices provide the simplest

Fig. 36 Dual-channel convolver module containing two 2-track convol-ver chips, amplifiers, filters and transducer tuning

This unit is commercially available from Signal Technology Ltd. and has an overallsize 114 x 26 x 33 mm. A convolver chip, for use in such a module, is shown in theforeground.

ver, SAWs propagating in two spatially separated channelsare illuminated by an optical beam and the doubly dif-fracted light is focused onto a fast detector. The maximumoutput of such a device is limited by the optical power andsaturation effects in the detector. The self-convolutioneffects present in the monolithic device are. not present asthe two counterpropagating SAWs cannot interact. In thesilicon-lithium-niobate gap-coupled acoustoelectric con-volver [194, 202], the mixing is accomplished through thenonlinear interaction of the SAWs with the mobile chargecarriers in the silicon. The integration is performed with aconducting plate on the back of the silicon. A furtherdevice which can perform the convolution integral is thememory correlator which uses a Schottky-diode array on a

aperiodic 255 chipm-sequencetheoretical ACF

starting <11111111>

Fig. 37 Two-channel convolver response to gated biphase modulated

255 chip m-sequence operating at 32 Mchip/s on a 300 MHz carrier

The lower trace shows the theoretical autocorrelation function for this code.


silicon strip for the storage of a reference waveform. Byelectrically impulsing the silicon plate when the referenceSAW is beneath it, a replica of the reference is stored as acharge pattern in the diode array. A second signal interactswith this stored-charge pattern producing the desired non-linear output on the silicon plate.

To date, only the monolithic convolver is availablecommercially, Fig. 33; the more sophisticated devices havenot yet been exploited, which is at least partly due to theirfabrication complexity.

The performance of the monolithic device (TB-productand efficiency) are quite acceptable in many systems andthe speed of this analogue device overshadows competingtechnologies; it should, therefore, find a place in thesystems designer's catalogue of available components.

As we have seen, the first generation of monolithic SAWconvolvers is only just becoming available. There is,undoubtedly, scope for further improvement in variousaspects of performance; for example, the transverse hori-zontal convolver of Monks et al. is a novel structure whichpotentially offers improved efficiency and lower spuriouslevels [201]. A further useful development for some appli-cations would be an extension of the integration time,some approaches to which are described in Reference 178.Another aspect of the convolver performance concerns thenecessity to provide a time-reversed reference waveformwhich can be an embarrassment in some applications, e.g.when processing the chirp waveforms of Section 6.However, this criticism does not apply to digitally encodedwaveforms, e.g. PSK. Indeed it is perfectly possible tovisualise the development of a SAW convolver modulewith digital input and output to exploit the enormousspeed of this analogue component in digital processors.

9 Conclusion and discussion

In this review, we have described, topic by topic, our viewof the most significant advances in the field of SAW tech-nology since the earlier review by Maines and Paige [8].Here we shall take a sideways look at the activity as awhole to review the overall trends. First, we have seen dra-matic improvements in the performance of most SAWcomponents; for example, (i) delay times have beenextended to 1 ms [38], (ii) bandpass filters routinely offeran out-of-band rejection of 60 dB, (iii) oscillators havebeen extended in frequency to 3 GHz, and have demon-strated short-term stabilities that could not be measuredten years ago [151, 152], (iv) dispersive delay lines haveprovided TB-products of 20000 with an RMS phase errorof only a few degrees [126], while (v) monolithic convol-vers with 100 MHz bandwidth and 16 [is integration timehave shown a dynamic range of 60 dB; as programmablematched filters their performance greatly exceeds that ofany forseeable digital competition. Secondly, and largely asa result of these activities, we have seen the acceptance ofSAW components in a variety of applications, both civiland military. The prime examples are, of course, the dis-persive delay line in pulse-compression radar and the TVIF filter, each of which now dominates its own field, butnumerous other applications have been reported. Forexample, by 1977, Williamson had listed 37 USA Govern-ment systems employing SAW devices [206].

A third trend, which is increasingly discernible, istowards simpler and cheaper components to meet therequirements of larger markets. The foremost example isthe TV IF filter which currently occupies one third of theoriginal substrate area, enabling a packaged device to besold for 50 pence (sterling). Even more dramatic simplifica-tions in convolver design have occurred recently (Table 3),

and there is currently an extensive research and develop-ment activity aimed at simpler and cheaper low-lossbandpass filters, e.g. the SPUDT [22], and dispersers, e.g.through printed dot array reflectors [142]. These trendswill doubtless continue for many years and constitute thenormal improvement phase of a successful commercialproduct.

Concerning future research and development topics, itseems to us that there are still many areas of SAW whichare ripe for exploitation, and we mention some of thesebelow:

(i) It is apparent from Section 5 that SAW devices aretechnically capable of providing high-performance filterbanks, but that smaller and simpler structures are reallyneeded; it should not be beyond the capability of SAWengineers to devise such.

(ii) As discussed in Section 7, we may well see furtherdevelopments in SAW sensors in the immediate future.

(iii) The development of viable monolithic convolvershas forced a detailed understanding of the properties ofSAW waveguides. This, in turn, may lead to the realisationof some of the early prophecies of waveguide-based SAWcomponents and subsystems [207].

(iv) To date, only a few SAW devices have exploited thepresence of the second substrate dimension; namely, RACs[130], array processors [46] and temperature-compensation schemes [157]. It would be surprising ifother applications do not arise. An extension of the prin-ciples of SSBW devices to the third dimension also offerssome interesting device possibilities [208].

(v) Several advantages would follow the integration ofSAW and semiconductor devices on a single piezoelectricsubstrate such as GaAs. These include:

(a) the addition of selectivity to microelectronics, e.g. theincorporation of front-end and IF filters on integratedreceiver chips

(b) the addition of greater programmability to SAWdevices, e.g. to tapped delay lines, bandpass filters andmemory correlators [211]

(c) improved ruggedness and reliability, e.g. in missile-borne transmitters and submarine cable repeaters.

In summary, we see a healthy future for SAW, with abun-dant scope for the development of new components and acontinuing activity to simplify and reduce the cost of exist-ing devices to access larger markets.

10 References

1 WHITE, R.M., and VOLTMER, F.W.: 'Direct piezoelectric coup-ling to surface elastic waves', Appl. Phys. Lett., 1965, 7, pp. 314-316

2 MORGAN, D.P. (Ed.): 'Key papers on surface acoustic wave passiveinterdigital devices' (IEE Reprint Series 2, 1976)

3 AULD, B.A.: 'Acoustic fields and waves in solids—Vols. I and II'(Wiley, 1973)

4 DIEULESAINT, E, and ROYER, D.: 'Elastic waves in solids'(Wiley, 1980)

5 MATTHEWS, H. (Ed.): 'Surface wave filters' (Wiley, 1977)6 OLINER, A.A. (Ed.): 'Topics in applied physics—Vol. 24, Acoustic

surface waves' (Springer-Verlag, 1978)7 WHITE, R.M.: 'Surface elastic waves', Proc. IEEE, 1970, 58, pp.

1238-12768 MAINES, J.D., and PAIGE, E.G.S.: 'Surface-acoustic-wave com-

ponents, devices and applications', Proc. IEE, 1973, 120, (IOR), pp.1078-1110

9 MAINES, J.D., and PAIGE, E.G.S.: 'Surface acoustic-wave devicesfor signal processing applications', Proc. IEEE, 1976, 64, pp.639-652

10 JENKINS, F.A., and WHITE, H.E.: 'Fundamentals of optics'(McGraw-Hill, 1957)

11 HARTMANN, C.S., BELL, D.T., and ROSENFELD, R.C.:'Impulse model design of acoustic surface-wave filters', IEEE Trans.,1973, SU-20, pp. 80-93


12 NALAMWAR, A.L., and EPSTEIN, M.: 'Immitance character-ization of acoustic surface-wave transducers', Proc. IEEE, 1972, 60,pp. 336-7

13 SLOBODNIK, A.J, CON WAY, E.D, and DELMONICO, R.T.:'Microwave acoustics handbook—Vol 1A: Surface wave velocities'(Air Force Cambridge Research Labs, USA, 1973)

14 SMITH, W.R., GERARD, H.M., COLLINS, J.H., REEDER, T.M,and SHAW, H.J.: 'Analysis of interdigital surface wave transducersby use of an equivalent circuit model', IEEE Trans., 1969, MTT-17,pp. 856-864

15 KERNS, D.M., and BEATTY, R.W.: 'Basic theory of waveguidejunctions and introductory microwave network analysis' (PergamonPress, 1967)

16 ROSENBERG, R.L.: 'Wave-scattering properties of interdigitalSAW transducers', IEEE Trans., 1981, SU-28, pp. 26-41

17 LEWIS, M.F.: 'Triple-transit suppression in surface-acoustic-wavedevices', Electron. Lett., 1972, 8, (23), pp. 553-554

18 SMITH, W.R., GERARD, H.M., COLLINS, J.H., REEDER, T.M,and SHAW, H.J.: 'Design of surface wave delay lines with interdigi-tal transducers', IEEE Trans., 1969, MTT-17, pp. 865-873

19 ASH, E.A.: 'Surface wave grating reflectors and resonators', IEEEsymposium on microwave theory and techniques, Newport Beach,llth-14th May, 1970

20 MARSHALL, F.G., PAIGE, E.G.S., and YOUNG, A.S.: 'New uni-directional transducer and broadband reflector of acoustic surfacewaves', Electron. Lett., 1971, 7, (21), pp. 638-640

21 ROSENFELD, R.C, HARTMANN, C.S., and BROWN, R.B.:'Low-loss unidirectional SAW filters'. Proceedings 28th annual fre-quency control symposium, 1974, pp. 299-303

22 HARTMANN, C.S, WRIGHT, P.V, KANSY, R.J, and GARBER,E.M.: 'An analysis of SAW IDT's with internal reflections and theapplication of the design to single-phase unidirectional transducers'.Proc. IEEE Ultrasonics Symposium., 1982, pp. 40-45

23 WEST, C.L.: 'SAW convolver employing unidirectional transducersfor improved efficiency'. Ibid., 1982, pp. 119-123

24 LEWIS, M.F.: 'Group-type unidirectional SAW devices employingintratransducer reflector banks', Electron. Lett., 1983, 19, (25), pp.1085-1087

25 LEWIS, M.F.: 'SAW filters employing interdigitated interdigitaltransducers'. Proc. IEEE Ultrasonics Symp., 1982, pp. 12-17

26 WADAKA, S., MISU, K., and KATO, T.: 'A low loss 0.9 GHz BandSAW filter'. Ibid., 1983, pp. 59-61

27 HIKITA, M., TABUCHI, T., KOJIMA, H., NAKAGOSHI, A., andKINOSHITA, Y.: 'Low loss SAW filter for antenna duplexer'. Ibid.,1983, pp. 77-82

28 SANDY, F., and PARKER, T.E.: 'Surface acoustic wave ring filter'.Proceedings of the 30th annual control symposium, 1976, pp.334-339

29 FELDMANN, M, and HENAFF, J.: 'Design of multistrip arrays'.Proc. IEEE Ultrasonics Symp., 1977, pp. 686-690

30 POLLOCK, W, SCHOFIELD, J., MILSOM, R.F., MURRAY,R.J., and FLINN, I.: 'Low-loss SAW filter using single-phase IDT'sand no external tuning'. Ibid., 1983, pp. 87-92

31 BRISTOL, T.W., JONES, W.R, SNOW, P.B., and SMITH, W.R.:'Applications of double electrodes in acoustic surface wave devicedesign'. Ibid., 1972, pp. 343-345

32 KERBEL, S.J.: 'Design of harmonic surface acoustic wave (SAW)oscillators without external filtering and new data on the tem-perature coefficient of quartz'. Ibid., 1974, pp. 276-281

33 LEWIS, M.F.: 'The surface acoustic wave oscillator—a natural andtimely development of the quartz crystal oscillator'. Proceedings ofthe 28th annual frequency control symposium, 1974, pp. 304-314

34 ZIMMERMAN, R.L., SCHWEITZER, B.P., and BENDER, R.C:'High data rate, high bit density acoustic digital memory'. Proc.IEEE Ultrasonics Symp., 1972, pp. 359-364

35 SHORROCKS, N.M, WHATMORE, R.W., AINGER, F.W., andYOUNG, I.M.: 'Lithium tetraborate—a new temperature compen-sated piezoelectric substrate material for surface acoustic wavedevices'. Ibid., 1981, pp. 337-340

36 ADKINS, L.R., and HUGHES, A.J.: 'Long delay lines employingsurface acoustic wave guidance', J. Appl. Phys., 1971, 42, pp. 1819—1822

37 BROWNING, T.I., and MARSHALL, F.G.: 'Compact 130 fis SAWdelay line using improved MSC reflecting track changers'. Proc.IEEE Ultrasonics Symp., 1974, pp. 189-192

38 FORTUNKO, CM., and SHAW, H.J.: 'One millisecond surfaceacoustic wave delay line'. Ibid., 1975, pp. 537-538

39 MASON, I.M., PAPADOFRANGAKIS, E., and CHAMBERS, J.:'Acoustic-surface-wave disk delay lines', Proc. IEEE, 1976, 64, pp.610-612

40 WARNE, D.H., PURCELL, J.J, and MORGAN, D.P.: 'Cascadabletapped delay line module'. Proc. IEEE Ultrasonics Symp., 1981, pp.53-57

212

41 WRIGLEY, C.Y., HAGON, P.J., and SEYMOUR, R.N.: 'Prog-rammable SAW matched filters for phase coded waveforms'. Ibid.,1972, pp. 226-228

42 PANASIK, CM.: 'SAW programmable transversal filter for adapt-ive interference suppression'. Ibid., 1982, pp. 100-103

43 HAGON, P.J., MICHELETTI, F.B. and SEYMOUR, R.N.: 'Inte-grated programmable analogue matched filters for spread spectrumapplications'. Ibid., 1973, pp. 333-335

44 GREEN, J.B., KINO, G.S., WALKER, J.T., and SHOTT, J.D.: 'TheSAW/FET: a new programmable SAW transversal filter'. Ibid., 1982,pp. 436-441

45 LATTANZA, J., HERRING, F.G., KRENCIK, P.M., and CLERI-HEW, A.F.: '240 MHz wideband programmable SAW matchedfilter'. Ibid., 1983, pp. 143-150

46 WARNE, D.H, and MORGAN, D.P.: 'SAW beam former'. Ibid.,1982, pp. 202-206

47 MARSHALL, F.G., NEWTON, C O , and PAIGE, E.G.S.: 'Theoryand design of SAW multistrip coupler', IEEE Trans., 1973, MTT-21,pp. 206-215

48 TANCRELL, R.H, and HOLLAND, M.G.: 'Acoustic surface wavefilters', Proc. IEEE, 1971, 59, pp. 393-409

49 SOLIE, L.P.: 'A surface acoustic wave multiplexer using offset multi-strip couplers'. Proc. IEEE Ultrasonics Symp., 1974, pp. 153-156

50 SOLIE, L.P.: 'A SAW bandpass filter technique using a fannedmultistrip coupler', Appl. Phys. Lett., 1977, 30, pp. 374-376

51 MAERFELD, C, and FARNELL, G.W.: 'Nonsymmetrical multi-strip coupler as a surface-wave beam compressor of large band-width', Electron. Lett., 1973, 9, (18), pp. 432^34

52 MARSHALL, F.G, NEWTON, C O , and PAIGE, E.G.S.: 'SAWmultistrip components and their applications', IEEE Trans., 1973,MTT-21, pp. 216-225

53 RABINER, L.R., and GOLD, B.: 'Theory and application of digitalsignal processing' (Prentice-Hall, 1975)

54 PARKS, T.W, and McCLELLAN, J.H.: 'A program for the designof linear phase finite impulse response digital filters', IEEE Trans.,1972, AU-20, pp. 195-199

55 McCALLIG, M.T, and LEON, B.J.: 'Constrained ripple design ofF.I.R. digital filters', ibid., 1978, CAS-25, pp. 893-902

56 TANCRELL, R.H.: 'Analytical design of surface wave bandpassfilters'. Proc. IEEE Ultrasonics Symp., 1972, pp. 215-217

57 LEWIS, B, JORDAN, P.M., MILSOM, R.F, and MORGAN, D.P.:'Charge and field superposition methods for analysis, of generalisedSAW interdigital transducers'. Ibid., 1978, pp. 709-714

58 DEACON, J.M, HEIGH WAY, J, and JENKINS, J.A.: 'Multistripcoupler in acoustic-surface-wave filters', Electron. Lett., 1973, 9, (10),pp. 235-236

59 TANCRELL, R.H, and ENGAN, H.: 'Design condiderations forSAW filters'. Proc. IEEE Ultrasonics Symp., 1973, pp. 419-422

60 SZABO, T.L, and SLOBODNIK, A.J.: 'The effect of diffraction onthe design of acoustic surface wave devices', IEEE Trans., 1972,SU-20, pp. 240-251

61 SAVAGE, E.B, and MATTHAEI, G.L.: 'A study of some methods• for compensation for diffraction in SAW IDT filters', ibid., 1981,SU-28, pp. 439-448

62 MADER, W, STOCKER, H, and VEITH, R.: 'Compensation ofdiffraction effects on group delay time and stop-band rejection'.Proc. IEEE Ultrasonics Symp., 1980, pp. 391-395

63 PEACH, R.C, DOGGETT, N.H., McCLEMONT, F.S, KAT-SELLIS, A, and DYER, A.J.: 'The diffraction analysis and correc-tion of narrow band SAW transversal filters'. Ibid., 1981, pp. 58-62

64 HARTMANN, C.S.: 'Weighting interdigital surface wave trans-ducers by selective withdrawal of electrodes'. Ibid., 1973, pp. 423-426

65 ATZENI, C, MANES, G, and MASOTTI, L.: 'Synthesis ofamplitude—modulated SAW filters with constant—length fingers'.Ibid., 1973, pp. 414-418

66 ENGAN, H.: 'Series—weighting of surface acoustic wave trans-ducers'. Ibid., 1974, pp. 422-424

67 SANDY, F.: 'Combining series sections weighting with withdrawalweighting in surface acoustic wave transducers', IEEE Trans., 1979,SU-26, pp. 308-312

68 HIKITA, M, KINOSHITA, Y, KOJIMA, H, and TABUCHI, T.:'Phase weighting for low loss SAW filters'. Proc. IEEE UltrasonicsSymp., 1980, pp. 308-312

69 De VRIES, A.J, SREENIVASAN, T, SUBRAMANIAN, S, andWOJCIK, T.J.: 'Detailed description of a commercial surface-waveTV IF filter'. Ibid., 1974, pp. 147-152

70 SHIBAYAMA, K, YAMANOUCHI, K, SATO, H, andMEGURO, T.: 'Optimum cut for rotated Y-cut LiNbO3 crystalused as the substrate of acoustic-surface-wave filters', Proc. IEEE,1976, 64, pp. 595-7

71 KODAMA, T.: 'Optimisation techniques for SAW filter design'.Proc. IEEE Ultrasonics Symp. 1979, pp. 522-526

72 ROSENFELD, R.C, BROWN, R.B., and HARTMANN, C.S.: 'Uni-


directional acoustic surface wave filters with 2 dB insertion loss'.Ibid., 1974, pp. 425-428

73 YAMANOUCHI, K., NYFFELER, F.M., and SHIBAYAMA, K.:'Low insertion loss acoustic surface wave filter using group type uni-directional interdigital transducer'. Ibid., 1975, pp. 317-321

74 MALOCHA, D.C. and HUNSINGER, B.J.: Tuning of group typeunidirectional transducers', IEEE Trans., 1979, SU-26, pp. 243-245

75 LEWIS, M.F.: 'A different approach to the wave scattering proper-ties of interdigital transducers', ibid., 1983, SU-30, pp. 55-57

76 SOLIE, L.P.: 'The development of high performance RDA devices'.Proc. IEEE Ultrasonics Symp., 1979, pp. 682-686

77 SANDBANK, C.P., and BUTLER, M.B.N.: 'Acoustic surface waveson isopaustic glass', Electron. Lett., 1971, 7, (17), pp. 499-501

78 SHIOSAKI, T.: 'High-speed fabrication of high-quality sputteredZnO thin films for bulk and surface wave applications'. Proc. IEEEUltrasonics Symp., 1978, pp. 100-110

79 VAN de VAART, H., and SOLIE, L.P.: 'SAW multiplexing tech-nique', Proc. IEEE, 1976, 64, pp. 688-691

80 WEBB, D.C, and BANKS, C : 'Properties of a constant-K ladderSAW contiguous filter bank', IEEE Trans., 1976, SU-23, pp.386-393

81 SLOBODNIK, A.J., ROBERTS, G.A, SILVA, J.H., KEARNS,W.J, SETHARES, J.C., and SZABO, T.L.: 'Switchable SAW filterbanks at UHF', ibid., 1979, SU-26, pp. 120-126

82 HAYS, R.M.: 'Switchable multichannel surface wave devices'.ECOM contract DAAB 07-73-0094, 1974

83 FELDMAN, M., and HENAFF, J.: 'The reflective ASW multistriparray: a frequency sensitive device for filter and channel bank'. Proc.European Microwave Conf., 1976, pp. 239-241

84 HENAFF, J., and FELDMAN, M.: 'Design and capabilities of SAWfilters: synthesis and technologies'. Proc. ISC AS, 1979, pp. 617-620

85 MELNGAILIS, J., and FLYNN, G.T.: '16-channel surface-acoustic-wave grating-filter bank for real-time spectral analysis', Electron.Lett., 1974, 10, (7), pp. 107-109

86 DOLAT, V., and MELNGAILIS, J.: '16-channel filter bank'. Proc.IEEE Ultrasonics Symp., 1973, pp. 756-759

87 HAYS, R.M, ROSENFELD, R.C., and HARTMANN, C.S.: 'Select-able bandpass filters—multichannel surface wave devices'. Ibid.,1973, pp. 456-459.

88 CLAIBORNE, L.T.: 'Surface wave bandpass filters: componentsand subsystems'. Optical and Acoustical Microelectronics Symp.,1974, pp.461-469

89 POTTER, B.R., and SHOQUIST, T.L.: 'Multi-passband low lossSAW filters'. Proc. IEEE Ultrasonics Symp., 1977, pp. 736-739

90 BUDREAU, A.J, SLOBODNIK, A.J., and CARR, P.H.: 'Review ofSAW-based direct frequency synthesisers', IEEE Trans., 1982,MTT-30, pp. 686-693

91 HARTEMANN, P.: 'Programmable acoustic-surface-wave oscil-lator', Electron. Lett., 1975, 11, (5), pp. 119-120

92 MOULE, G.L.: 'SAW compressive receivers for RADAR intercept',IEE Proc. F, Commun., Radar & Signal Process., 1982, 129, (3), pp.180-186

93 ALLEN, D.E.: 'Channelised receiver—A viable solution for EW andESM systems', ibid., 1982, 129, (3), pp. 172-179

94 TSAI, C.S.: 'Guided wave acoustic optic Bragg modulators for wideband integrated optic communications and signal processing', IEEETrans., 1979, CAS-26, pp. 1072-1098

95 CHANG, I.C.: 'Acoustic-optic devices and applications', ibid., 1976,SU-23, pp. 2-22

96 OVERBURY, A.P., PITT, C.W., and SKINNER, J.D.: 'A gratingintegrated optic spectrum analyser'. IEE Colloquia on SAW devicesand systems, Dec. 1982, Digest 1982/86, pp. 4.1-4.6

97 GRANT, P.M., COLLINS, J.H, DARBY, B.J., and MORGAN,D.P.: 'Potential applications of acoustic matched filters to air trafficcontrol systems', IEEE Trans., 1973, MTT-21, pp. 288-300

98 SLOBODNIK, A.J, FENSTER MACHER, T.E., KEARNS, W.J.,ROBERTS, G.A., SILVA, J.H, and NOONAN, J.P.: 'SAW Butter-worth contiguous filters at UHF', ibid., 1979, SU-26, pp. 246-253

99 CHANG, C, and LEE, S.: 'Efficient wideband acoustooptic Braggcells'. Proc. IEEE Ultrasonics Symp., 1983, pp. 427-430

100 SCHMIDT, R.V.: 'Acoustic surface wave velocity perturbations inLiNbO3 by diffusion of metals', Appl. Phys. Lett., 1975, 27, pp. 8-10

101 TANCRELL, R.H, SCHULZ, M.B, BARRETT, H.H., DAVIS, L.,Jun, and HOLLAND, M.G.: 'Dispersive delay lines using ultrasonicsurface waves', Proc. IEEE, 1969, 57, pp. 1211-1213

102 COOK, C.E, and BERNFELD, M.: 'Radar signals' (AcademicPress, 1967)

103 MAINES, J.D, and JOHNSTON, J.N.: 'SAW devices andapplications—Part 2: Pulse compression systems', Ultrasonics, 1973,11, pp. 211-217

104 SKOLNIK, M.I.: 'Introduction to radar systems' (McGraw-Hill,1962)

105 SKOLNIK, M.I.: 'Radar handbook' (McGraw-Hill, 1970)

106 'Handbook of acoustic signal processing—Vol. II: Pulse expansion/compression IF subsystems for radar' (Andersen Labs Inc.)

107 HARRIS, F.J.: 'On the use of windows for harmonic analysis withthe discrete Fourier transform', Proc. IEEE, 1978, 66, pp. 51-83

108 BUTLER, M.B.N.: 'Radar applications of s.a.w. dispersive filters',IEE Proc. F, Commun., Radar & Signal Process., 1980, 127, (2), pp.118-124

109 JUDD, G.W.: 'Techniques for realising low time sidelobe levels insmall compression ratio chirp waveforms'. Proc. IEEE UltrasonicsSymp., 1973, pp. 478-481

110 NEWTON, CO. : 'Nonlinear chirp radar signal waveforms for SAWpulse compression filters', Wave Electron., 1974, 1, pp. 387-401

111 SANDY, F.: 'A RAC using a non-linear chirp'. Proc. IEEE Ultra-sonics Symp., 1975, pp. 385-389

112 GERRARD, H.M.: 'Surface wave interdigital elctrode chirp filters',in MATTHEWS, H. (Ed.): 'Surface wave filters, design, constructionand use' (J. Wiley, 1977), Chap. 8

113 ASH, E.A.: 'Fundamentals of signal processing devices', in OLINER,A.A. (Ed.): 'Topics in applied physics—Vol. 24: Acoustic surfacewaves' (Springer Verlag, 1978)

114 WEGLEIN, R.D., WAUK, M.T., and NUDD, G.R.: '500 MHzbandwidth SAW pulse compression filter'. Proc. IEEE UltrasonicsSymp., 1973, pp. 482-485

115 POTTER, B.R., and HARTMANN, C.S.: 'SAW slanted correlatorsfor linear fm pulse compressors'. Ibid., 1977, pp. 607-610

116 POTTER, B.R., and HARTMANN, C.S.: 'SAW slanted device tech-nology', IEEE Trans., 1979, SU-26, pp. 411^18

117 STOCKER, H.R., VEITH, R, WILLIBALD, E., and RIHA, G.:'SAW pulse compression filters with long chirp time'. Proc. IEEEUltrasonics Symp., 1981, pp. 78-82

118 MACDONALD, D.B., and DANIELS, W.D.: 'Wide bandwidthSAC delay lines'. Ibid., 1978, pp. 749-755

119 POTTER, B.R, DANIELS, W.D., and BROWN, S.: 'Design andanalysis tools for the slanted array correlator'. Ibid., 1983, pp.200-204

120 COOPER, J.B., and POTTER, B.R.: 'Phase compensation of linearfm slanted transducers by use of metallised stripes'. Ibid., 1982, pp.96-99

121 ENGAN, H.: 'Excitation of elastic surface waves by spatial harmo-nics of interdigital transducers', IEEE Trans., 1969, ED 16, pp.1014-1017

122 MELLON, D.W, and BELL, D.T.: 'Development of SAW pulsecompressor'. Proc. IEEE Ultrasonics Symp., 1973, pp. 486-489

123 MININ, J , and KOGAN, E.: 'Third harmonic chirp delay line'.Proc. IEEE Ultrasonics Symp., 1982, pp. 109-112

124 STOKES, R.B., LAU, K.F., YEN, K.H., KAGIWADA, R.S., andKONG, A.M.: 'Wideband third harmonic chirp filters'. Proc. IEEEUltrasonics Symp., 1981, pp. 28-32

125 MATTHAEI, G.L, YOUNG, L, and JONES, E.M.T.: 'Microwavefilters impedance—matching networks and coupling structures'(McGraw-Hill, 1964)

126 OTTO, O.W., and GERARD, H.M.: 'On Rayleigh wave reflectionfrom grooves at oblique incidence and an empirical model for bulkwave scattering in RAC devices'. Proc. IEEE Ultrasonics Symp.,1977, pp. 596-601

127 DOLAT, V.S, and WILLIAMSON, R.C: 'BGO RAC with 125 /isof dispersion'. Ibid., 1975, pp. 390-394

128 MARSHALL, F.G, PAIGE, E.G.S, and YOUNG, A.S.: 'Amplitudeweighting of SAW reflecting array structures'. Ibid., 1974, pp.202-204

129 GERARD, H.M., and JUDD, G.W.: '500 MHz bandwidth RACfilter with constant groove depth'. Ibid., 1978, pp. 734-737

130 WILLIAMSON, R.C: 'Large time bandwidth product devicesachieved through the use of SAW reflective gratings'. Proc. Int. Sp.Sem. on Comp. Prof, and Sys. App. of SAW Devices, IEE, 1973, pp.181-190

131 JUDD, G.W, and OTTO, O.W.: 'A design synthesis and analysisprocedure for linear and non linear fm RAC devices'. Proc. IEEEUltrasonics Symp., 1977, pp. 602-606

132 OATES, D.E.: 'A new cut of quartz for temperature stable SAWdispersive delay lines', IEEE Trans., 1979, SU-26, pp. 428-430

133 BOROSON, D.M., and OATES, D.E.: 'Experimental and theoreti-cal analysis of temperature dependence of wideband SAW RACdevices on quartz'. Proc. IEEE Ultrasonics Symp., 1981, pp. 38-43

134 GODFREY, J.T., NOTHNICK, C.E., SCHELLENBERG, J ,MOORE, R.A., and GRAULING, C.H.: 'Phase-weighted metallisedreflective arrays'. Ibid., 1976, pp. 406-410

135 GODFREY, J.T., GRAULING, C.H., NOTHNICK, C.E., andREAMS, R.: 'Low-time sidelobe metallic RAC structures'. Ibid.,1978, pp. 739-743

136 KITANO, T., NISHIKAWA, K, and ASAKAWA, K.: 'Amplitudeweighted metal strip RAC. Ibid., 1977, pp. 585-589

137 CHAPMAN, R.E., CHAPMAN, R.K., MORGAN, D.P., and


PAIGE, E.G.S.: 'Inline reflective array devices'. Ibid., 1978, pp.728-733

138 SOLIE, L.P.: 'Reflective dot arrays'. Ibid., 1977, pp. 579-584139 BLOCH, P.D., PAIGE, E.G.S., and STOVE, A.: 'Selective reflection

of SAW by periodic dot arrays'. Ibid., 1979, 687-690140 THOSS, J.L., PENUNURI, D , THOSTENSON, M.: 'Implementa-

tions of reflective array matched filters for radar applications'. Ibid.,1981, pp. 63-68

141 SOLIE, L.P.: 'SAW reflective dot arrays', Appl. Phys. Lett., 1976, 28,pp. 420-422

142 VAN DE VAART, H., and SOLIE, L.P.: 'A SAW pulse compressionfilter using reflective dot arrays', ibid., 1977, 31, pp. 1-3

143 SOLIE, L.P.: 'The development of high performance RDA devices'.IEEE Ultrasonics Symp., 1979, pp. 682-686

144 HUANG, F., and PAIGE, E.G.S.: 'Reflection of SAW by thin metaldots'. Ibid., 1982, pp. 72-82

145 HUANG, F., and PAIGE, E.G.S.: 'Influence of size and shape onreflection of surface acoustic waves by thin aluminium dots onlithium niobate', Electron. Lett., 1982,18, (5), pp. 232-233

146 WOODS, R.C.: 'Dispersive delay lines using 180° reflecting metaldot arrays'. Proc. IEEE Ultrasonics. Symp., 1982, pp. 88-91

147 ROBERTS, J.B.G., MOULE, G.L., and PARRY, G.: 'Design andapplication of real-time spectrum-analyser systems', IEE Proc. F,Commun., Radar & Signal Process., 1980, 127, (2), pp. 76-91

148 MAINES, J.D., PAIGE, E.G.S., SAUNDERS, A.F., and YOUNG,A.S.: 'Simple technique for the accurate determination of delay-timevariations in acoustic-surface-wave structures', Electron. Lett., 1969,5, (26), pp. 678-680

149 ELLIOTT, S., MIERZWINSKI, M., and PLANTING, P.: 'The pro-duction of surface acoustic wave resonators'. Proc. IEEE UltrasonicsSymp., 1981, pp. 89-93

150 PARKER, T.E.: 7 / / phase noise in quartz delay lines and reson-ators'. Ibid., 1979, pp. 878-881

151 PENAVAIRE, L., SEGUIGNES, D., LARDAT, D., BONNIER,J.T., CHEVALIER, J.Y., and BESSON, Y: 'A 120 MHz SAW reson-ator stabilized oscillator with high spectral purity'. Ibid., 1980, pp.256-259

152 PARKER, T.E.: 'Precision surface acoustic wave (SAW) oscillators'.Ibid., 1982, pp. 268-274

153 MARSHALL, F.G.: 'SAW resonators constructed of Al on ST-quartz for use in high stability feedback oscillators'. Ibid., 1978, pp.290-292

154 EBATA, Y, SATO, K., and MORISHITA, S.: 'A LiTaO3 SAW res-onator and its application to video cassette recorder'. Ibid., 1981, pp.111-116

155 SCHULZ, M.B, MATSINGER, B.J., and HOLLAND, M.G.: 'Tem-perature dependence of surface acoustic wave velocity on a-quartz',J. Appl. Phys., 1970, 41, pp. 2755-2765

156 NEWTON, C O . : 'A study of the propagation characteristics of thecomplete set of SAW paths on quartz with zero temperature coeffi-cient of delay'. IEEE Ultrasonics Symp., 1979, pp. 632-636

157 BROWNING, T.I., and LEWIS, M.F.: 'A novel technique forimproving the temperature stability of SAW/SSBW devices'. Ibid.,1978, pp. 474-477

158 HOHKAWA, K., and YOSHIKAWA, S.: 'Temperature stable oscil-lators employing parallel connected SAW resonators'. Ibid., 1979,pp. 623-626

159 BALLATO, A, LUKASZEK, T., WILLIAMS, D.F., and CHO,F.Y.: 'Power flow angle and pressure dependence of SAW propaga-tion characteristics in quartz'. Ibid., 1981, pp. 346-349

160 LEWIS, M.F.: 'Temperature compensation techniques for SAWdevices'. Ibid., 1979, pp. 612-622

161 GRISE, W.R., and MEEKER, T.R.: 'Packaging and reliability ofSAW filters'. Ibid., 1983, pp. 117-124

162 TANSKI, W.J.: 'High performance SAW resonator filters for satel-lite use'. Ibid., 1980, pp. 148-152

163 PARKER, T.E., and CALLERAME, J.: 'Sensitivity of SAW delaylines and resonators to vibration'. Ibid., 1981, pp. 129-134

164 LEWIS, M.F.: 'Surface skimming bulk waves, SSBW'. Ibid., 1977,pp. 744-752

165 LAU, K.F., YEN, K.H., KAGIWADA, R.S., and KONG, A.M.:'High frequency temperature stable SBAW oscillators'. Ibid., 1980,pp. 240-244

166 REEDER, T.M., CULLEN, D.E, and GILDEN, M.: 'SAW oscil-lator pressure sensors'. Ibid., 1975, pp. 264-268

167 WEIRAUCH, D.F., SCHWARTZ, R.J., and BENNETT, R.C.:'SAW resonator frit-bonded pressure transducer'. Ibid., 1979, pp.874-877

168 MEUNIER, P.L, and HARTEMANN, P.: 'Cantilever SAW acceler-ometers'. Ibid., 1982, pp. 299-302

169 JOSHI, S.G.: 'A temperature compensated high voltage probe usingsurface acoustic waves'. Ibid., 1982, pp. 317-320

170 HAUDEN, D , JAILLET, G., and COQUEREL, R.: 'Temperature

sensor using SAW delay line'. Ibid., 1981, pp. 148-151171 BRYANT, A., LEE, D.L, and VETELINO, J.F.: 'A surface acoustic

wave gas detector'. Ibid., 1981, pp. 171-174172 TIERSTEN, H.F., STEVENS, D.S., and DAS, P.K.: 'Circulating

flexural wave rotation rate sensor'. Ibid., 1981, pp. 163-166173 ISHII, A., and HASHIMOTO, S.: 'Graphic sensor using a Lamb

wave'. Ibid., 1981, pp. 167-170174 STAPLES, E.J., WISE, J , SCHOENWALD, J.S., and LIM, T.C.:

'Surface acoustic wave underwater sound sensors'. Ibid., 1979, pp.870-873

175 QUATE, C.F., and THOMPSON, R.B.: 'Convolution and correla-tion in real time with non-linear acoustics', Appl. Phys. Lett., 1970,16, pp. 494-496

176 COLDREN, L.A., and SCHMIDT, R.V.: 'Acoustic surface waveAV/V waveguides on anisotropic substrates', ibid., 1973, 22, pp.482-483

177 BATANI, N.K., and ADLER, E.L.: 'Acoustic convolver usingribbon waveguide beamwidth compressors'. Proc. IEEE UltrasonicsSymp., 1974, pp. 114-116

178 LEWIS, M.F., and WEST, C.L.: 'The design and performance ofSAW convolvers with extended integration times'. Proc. IEEE Ultra-sonics Symp., 1983, pp. 189-194

179 COLVIN, R.D., and CHARLSON, E.J.: 'Multimoding in SAW con-volver waveguides', IEEE Trans., 1980, SU-27, pp. 385-388

180 DARBY, B.J., GUNTON, D.J., and LEWIS, M.F.: 'The design andperformance of a small efficient SAW convolver'. Proc. IEEE Ultra-sonics Symp., 1980, pp. 53-58

181 SELVIAH, DR., WARNE, D.H., and MORGAN, D.P.: 'Spatialuniformity measurement of SAW convolvers', Electron. Lett., 1980,18, (9), pp. 837-839

182 ADLER, E.L., and CAFARELLA, J.H.: 'The effect of acoustic dis-persion on SAW convolver performance'. Proc. IEEE UltrasonicsSymp., 1980, pp. 1-4

183 ADLER, E.L.: 'Electromagnetic long-line effects in surface waveconvolvers'. Proc. IEEE Ultrasonics Symp., 1980, pp. 82-87

184 GAUTIER, H., and MAERFELD, C : 'Wideband elastic convol-vers'. Proc. IEEE Ultrasonics Symp., 1980, pp. 30-36

185 AMERI, S., and GUNTON, D.J.: 'Efficient three-wavelength-outputSAW multistrip beam compressor for convolver applications', Elec-tron. Lett., 1981, 17, (22), pp. 836-838

186 BURNS, W.K., MILTON, A.F., and LEE, A.B.: 'Optical waveguideparabolic coupling horns', Appl. Phys. Lett., 1977, 30, pp. 28-30

187 HODGE, A.M., and LEWIS, M.F.: 'Detailed investigations intoSAW convolvers'. Proc. IEEE Ultrasonics Symp., 1982, pp. 113-118

188 YAO, I.: 'High performance elastic convolver with parabolic horns'.Ibid., 1980, pp. 37-42

189 MORGAN, DP., WARNE, D.H., and SELVIAH, D.R.: 'Narrowaperture chirp transducers for SAW convolvers'. Ibid., 1981, pp.186-191

190 LEWIS, M.F, and WEST, C.L.: 'High performance SAW convol-vers'. Ibid., 1981, pp. 175-180

191 MOTZ, M.D., CHAMBERS, J., and MASON, I.M.: 'Suppression ofspurious signals in a degenerate SAW convolver'. Ibid., 1973, pp.152-154

192 GANGULY, A.K., and DAVIS, K.L.: 'Nonlinear interactions indegenerate SAW elastic convolvers', J. Appl. Phys., 1980, 51, pp.920-926

193 WEST, C.L.: 'High-performance SAW convolver using 3-phase uni-directional transducers', Electron. Lett., 1982,18, (10), pp. 401-403

194 BECKER, R.A., RIEBLE, S.A., and RALSTON, R.W.: 'Comparisonof acousto-electric and acousto-optic signal processing devices',Proc. Soc. Photo-Opt. Instrum. Eng., 1979, 209, pp. 126-133

195 GREEN, J.B., and KHURI-YAKUB, B.T.: 'A 100 nm beamwidthZnO on Si convolver'. Proc. IEEE Ultrasonics Symp., 1979, pp.911-914

196 COMER, A.E., and MULLER, R.S.: 'A new ZnO-on-Si convolverstructure', IEEE Electron Device Lett., 1982, EDL-3, pp. 118-120

197 COLVIN, R.D., CARR, P.H., ROBERTS, G.A, and CHARLSON,E.J.: 'UHF elastic convolver with improved efficiency', IEEE Trans.,1982, SU-29, pp. 115-117

198 GOLL, J.H., and MALOCHA, D.C.: 'An application of SAW con-volvers to high bandwidth spread spectrum communications', ibid.,1981, MMT-29, pp. 473-483

199 ENGAN, H., INGEBRIGTSEN, K.A, and RONNEKLEIV, A.:'Design of S.A.W. convolver for processing M.S.K. modulated wave-forms', Electron. Lett., 1980,16, (24), pp. 908-909

200 CAFARELLA, J.H.: 'Surface acoustic wave devices for spread spec-trum communication', Jpn. J. Appl. Phys., 1980, 19, Suppl. 19, pp.667-674

201 MONKS, T.J., PAIGE, E.G.S., and WOODS, R.C.: 'SAW andSSBW convolvers utilizing bilinear field in various directions'. Proc.IEEE Ultrasonics Symp., 1983, pp. 402-405

202 YAO, I., and REIBLE, S.A.: 'Wide bandwidth acousto-electric con-


volvers'. Ibid., 1979, pp. 701-705203 MORGAN, D.P., COLLINS, J.H., and SUTHERLAND, J.G.:

'Asynchronous operation of a surface wave convolver'. Ibid., 1972,pp. 296-299

204 DEFRANOULD, P., and MAERFELD, G: 'A SAW planarpiezoelectric convolver', Proc. IEEE, 1976, 64, pp. 748-751

205 GREEN, J.B., and KINO, G.S.: 'SAW convolver using FIT', IEEETrans., 1983, SU-30, pp. 43-50

206 WILLIAMSON, R.C.: 'Case studies of successful SAW devices'.Proc. IEEE Ultrasonics Symp., 1977, pp. 460-468

207 STERN, E.: 'Microsound components, circuits and applications',IEEE Trans., 1969, MTT-17, pp. 835-844

208 LEWIS, M.F.: 'High-frequency acoustic plate mode device employ-ing interdigital transducers', Electron. Lett., 1981, 17, (21), pp.819-822

209 DAVIS, K.L., and WELLER, J.F.: 'Elastic convolver using planarprism couplers'. Proc. IEEE Ultrasonics Symp., 1980, pp. 74-76

210 GOLL, J.H., and BENNETT, R.C.: 'Reactive output tuning of highBT product SAW convolvers'. Ibid., 1978, pp. 44-47

211 WAGERS, R.S., and MELLOCH, M.R.: 'GaAs strip-coupledmemory correlators'. Ibid., 1983, pp. 377-386


recent developments in saw devices

Documents