Propagation Modeling at 60 GHz for Indoor Wireless LAN Applications

Nektarios Moraitis, Philip Constantinou

National Technical University of Athens Mobile RadioCommunications Laboratory

9 Heroon Polytechniou 157 73 Zografou, Athens, Greece Tel: +3010 7723849/3974, Fax: +3010 772 3851

email: {morai, fkonst}@mobile.ece.ntua.gr

ABSTRACT This paper reports narrowband and wideband results derived by propagation modeling at 60 GHz for indoor WLAN applications. A multi-ray model is proposed and verified through a simulation process. The propagation in the site-specific environment can be described using 4-5 rays without reducing the accuracy of the results. RMS delay spread varies with distance from 0.57 ns up to 2.32 ns. Coherence bandwidth for 0.9 correlation was found 65 MHz, 116 MHz and 87 MHz for isotropic, omni-omni and horn-horn antenna configurations respectively.

I. INTRODUCTION

Multimedia and computer communications are playing a major role in today’s society, creating new challenges for the development of telecommunications systems. The pressure for wireless systems to cope with increasing data rates is enormous and Wireless Broadband Systems (WBSs), which are systems with rates greater than 2 Mbps, are emerging rapidly [1]. We can mention enough system cases that may modify the WBSs’ perspectives, but two main approaches have come to light. The Wireless Local Area Networks (WLANs), which are focused on computer communications and Mobile Broadband Systems (MBSs) [2] focusing on cellular systems providing full mobility to B-ISDN users.

For the development of the mentioned broadband systems, wide frequency segments have been allocated in the millimeter region of the spectrum and especially around 60 GHz. In millimeter wave frequencies, the propagation modeling, apart from the known empirical models, can be realized based on geometrical optics using ray-tracing theory. In the 60 GHz region the diffraction phenomenon can be neglected, and the sum of the direct ray and the reflected rays is enough to describe the behavior of the propagation channel with great accuracy [1,3]. The modeling in extremely high frequencies poses the problem of the accurate description of the propagation scenarios at the wavelength scale (5 mm at 60 GHz). Hence the main target is to describe the main obstructions and the surfaces that affect the signal propagation. The description is not only in terms of the geometric characteristics of the propagation environment but also in terms of the surface electromagnetic parameters

(relative dielectric constant, losses etc) in order to extract the surface reflection coefficients.

In this paper we present a multi-ray model in order to describe the signal propagation at 60 GHz in an indoor environment. The model’s validity is verified through a simulation process from the narrowband and wideband point of view of the channel respectively. The propagation mechanisms are explained analytically whereas channel parameters such as RMS delay spread and coherence bandwidth are calculated.

This paper is organized as follows. Section II deals with the propagation modeling describing analytically the geometry of the environment under consideration, the proposed multi-ray model and the simulation procedure. In Section III, LoS propagation results are presented from the narrowband point of view, explaining the mechanisms and the behavior of the signal propagation. Section IV, deals with results from the wideband point of view in order to evaluate the wideband channel parameters. Finally, Section V gives some initial thoughts for future work and Section VI is devoted to discussion and conclusions derived by the entire simulation procedure.

II. PROPAGATION MODELING A. Description of the Environment Geometry

The simulation environment is a long corridor with dimensions 44 x 2.20 x 2.75 m3 as shown in Figure 1. In Figure 1 the Left Wall surface is made of brick and plasterboard with wooden doors every 3 m but in order to simplify the simulation procedure we assume the surface as a uniform wall made of brick and plasterboard with its dielectric characteristics given in Figure 1. Furthermore, in Figure 1 all the material characteristics are provided as well as the propagation geometry and the terminal positions. The beginning and the end of the corridor are open areas (very large spaces) and are not taken into account. B. Multi-Ray Model

The multi-ray model is a general case of the two-ray model [4,5] for more than two reflected components. The reflected components may exhibit single or double reflection from a plane surface. Third or fourth order reflections, especially at 60 GHz, are negligible contributors to the average power and are not taken into

Tx - Rx Horizon tal Separation (corridor 0-44 m )

Ceiling: fu rred ceiling m ade of alum inum

Floor: concrete covered w ith marb le

Left Wall : ligh t w all m ade of b rick and p lasterboard

Rx

Tx

2.75 m

2.2 mTx

Rx

ht

hr

x

y

Single Reflection

Direct Com ponen t

Doub le Reflection

Right Wall : external w all m ade of brick and concrete

x : Tx d istance form the r igh t vertical su rface (0.8 m)

y : Rx d istance form the r igh t vertical su rface (1.4 m )

h t : Tx height (2 m )

hr : Rx heigh t (1.5 m)

M aterial Dielectric Characteristics

Left W all : er = 4.44, σ = 0.001Right W all : er = 5, σ = 0.001Floor : er = 3, σ = 0.269Ceiling : er = 1, σ = 2e6

Figure 1: Simulation environment, propagation geometry and material dielectric characteristics.

account. The reflection geometry can be described in the horizontal as well as in the vertical plane as shown in Figure 1. Hence, if we know the geometry of the environment where the signal propagates (length, width, height) and the surface reflection coefficients, one may calculate the propagation losses. The propagation losses are calculated by the summation of N single reflected and M double reflected rays given by:

( )

⋅+++= ∑ ∑

= =

∆∆N

i

M

j

j

j

jbjaj

i

ip

ji ed

RRe

dR

ddL

1 10

1log2068 φφ

(1)

where d is the horizontal separation between Tx and Rx, 0d is the path length of the direct component and

id , jd are the path lengths each of the i single reflected and j double reflected rays. Moreover, iR is the reflection coefficient of i single reflected ray whilst

jaR , jbR are the reflection coefficients of the j double reflected rays on a and b reflecting surfaces

respectively. Finally ii l∆=∆λπφ 2 and jj l∆=∆

λπφ 2

are the phase differentials between the direct and the reflected rays with il∆ and jl∆ the differential path lengths between the direct and the i single and j double reflected rays, and λ is the wavelength (5 mm). B. Simulation Procedure

For the simulation of the signal propagation at 60 GHz the multi-ray model given by equation (1) will be used for the specific environment shown in Figure 1. The received signal (in dBm), is given by:

( ) ( )dLGGPdP prttr −++= (2)

where ( )dLp is given by equation (1). In equation (1) we use 4 single reflected ( 4=N ), plus 4 double reflected ( 4=M ) rays and the direct component. Additionally: • The diffraction is not taken into account since at 60

GHz the phenomenon is almost negligible and the diffracted power does not contribute to the total received power.

• The non-uniformities of the surface materials in indoor environments are such that the produced scattering has not a substantial contribution to the received power. The most significant contribution is from the 9 rays previously reported.

• Further reflected rays are not taken into account since their contribution to the total received power is insignificant. It will be shown that 9 rays in total can describe with great accuracy the signal propagation in the specific environment.

• Up to second order reflections are taken into account since third or fourth order reflections, especially at 60 GHz, are negligible contributors to the average power.

• Atmospheric propagation losses are not taken into account since in indoor environments the attenuation is very small (0.00116 dB/m).

• The antenna radiation patterns are taken into account in vertical and horizontal plane respectively for both transmitter and receiver, determining the incidence angle of each ray ( )itiG θ , ( )iriG θ .

• The rays that are within the 3 dB beam-width of the antenna’s main lobe are assumed to have a constant gain. Rays outside the main lobe are neglected. In other words we do not take into account the side lobes of the antenna radiation pattern since at 60 GHz the rays that emerging from the antenna’s side lobes do not contribute to the total received power.

The most significant side lobes are at least 15 dB below the main lobe.

During the entire simulation procedure vertical polarization is assumed. Hence, for the rays reflected from vertical walls we use the perpendicular reflection coefficient ( ⊥sR ), whereas for the rays from floor and ceiling surfaces we use the parallel reflection coefficient ( ||sR ). Both reflection coefficients are given in [4] equations (3.24) and (3.25). In the reflection coefficient equations the complex dielectric constant [4] is given by:

σλεε 60jr −= (3)

where rε is the relative dielectric constant of the reflecting surface, σ is the conductivity of the surface in Siemens/m and λ is the wavelength. The values of rε and σ are given in Figure 1 [6,7].

We assume three different transmission systems with different antenna characteristics and transmitted power. This is done so as to examine how the antenna radiation patterns affect the signal propagation in the indoor environment. The systems are: 1. System 1: Isotropic antennas on both transmitter and

receiver and 20 dBm output power. 2. System 2: tP = 20 dBm, tG = rG = 8.5 dBi omni-

directional antennas with vθ = 8o. 3. System 3: tP = 10 dBm, tG = rG = 20.8 dBi horn

antennas with vθ = 15o and hθ = 28o. Finally, the simulation is conducted with MatLab script, using 9 rays in total. The initial transmitter position is at the beginning of the corridor and the receiver is moving away from Tx with 1 m initial separation. We assume that the receiver is moving at almost constant speed of 0.2 m/sec and we collect a signal sample as a function of distance every 0.00125 m ( λ /4). The total number of samples for the entire corridor (44 m) is 34401.

III. LINE-OF-SIGHT PROPAGATION RESULTS

In Figure 2 are depicted the received power as a function of distance as well as the received signal strength of the various reflected rays in respect to the LoS component as a function of distance. The specific plots will give us information concerning the percentage of the single and double reflected ray contribution to the total received signal. The provided results concern the third transmission system (horn-horn antennas).

It is observed in Figure 2a, that the received power decays with distance at a rate of nd/1 . This rate depends on the number of the reflections that contribute to the total signal, as well as the their amplitude with respect to the LoS component. Using fitting in a minimum mean square error sense we found n=1.75 suggesting wave guiding effects along the corridor, as it is expected. The electric field of the reflected rays with respect to the LoS component is calculated as a function of distance along the corridor. The curves in some cases exhibit discontinuities since the propagation paths fade due to the environment geometry. From Figure 2a,b we observe that at the first 8 m, only the direct component

contributes to the total signal, whilst the ceiling single reflection starts to contribute (100% or equally with the direct ray) after 8 m until the end of the corridor. The ceiling reflection contributes significantly because the transmitter is closer to the ceiling. Moreover, the material of the ceiling (aluminum) is a perfect reflecting surface ( 1−≅R ). The ground single reflected ray contributes after 26 m with 70%, reaching 80% at the end of the corridor. The single reflections from the left and right walls start to contribute after 6 m reaching 60% at the end of the corridor. Their contribution is almost equivalent and the tiny differences are due to the different material characteristics.

(a)

(b)

(c)

Figure 2: (a) Received power as a function of distance, (b) Normalized received field of the single reflected rays as a function of distance, (c) Normalized received field of the double reflected rays as a function of distance.

The double reflected rays contribute to the total received power but in a smaller amount in comparison with the single reflections. From Figure 2c we observe that the double reflections from the right-left (i.e. first

and second reflection) and left-right wall start to contribute after 15 m and 20 m respectively, since the horizontal plane beamwidth ( hθ = 28o) includes these components after the specific distances. Their contribution though, reaches only 10% and 18%. The double reflection from ground-ceiling is zero due to the narrow beamwidth of the antenna at the vertical plane ( vθ = 15o) and the fact that the transmitter is closer to the ceiling. The same effect occurs for the ceiling-ground double reflection, which contributes only at the last 6 m of the corridor, though with a significant percentage (72%).

It is worthwhile to mention that in order to simulate the propagation in the specific environment we may reduce the number of rays, using 4-5 rays instead, without affecting the accuracy of the results. Hence we may use rays that contribute with 50% or more to the total received power. Apart from the direct component one can use the 4 single reflections (Figure 2b) and only one double reflection (ceiling-ground in Figure 2c). It is clear from the results that third or greater order reflections would not have significant effect to the signal propagation.

IV. WIDEBAND RESULTS

The wideband multipath channel is often modeled as a time varying linear filter with complex impulse response (CIR) [4]:

( ) ( ) ( )( ) ( )( )∑=

−=N

iiii ttjtath

1

,exp,, ττδτφττ (4)

For the indoor propagation environment where the time varying factors of the impulse response typically are human movement, it is appropriate to treat the channel as quasi-stationary. Assuming that the phase variations in the CIR have a uniform distribution we may consider only the amplitude and the delay components.

The most significant parameter derived from the procedure is the received power as a function of the time delay known also as the power delay profile (PDP). The power delay profile can be expressed as:

( ) ( ) ( )∑=

−=N

iir dPP

1

ττδτ (5)

where ( )dPr is the received signal of the ith ray and N is the total number of the rays used in the simulation procedure. The time resolution is assumed 1 ns. During the data binning process we assigned each echo to the nearest value of delay equal to a multiple of 1 ns. The average received power in every bin is normalized to the maximum received power.

A significant parameter to be evaluated is the RMS Delay Spread ( RMSτ ), which characterizes the frequency selective behavior of the channel and is given by:

( )

( )

( )

( )

22

⋅−

⋅=

∑∑

∑∑

kk

kkk

kk

kkk

RMS P

P

P

P

τ

ττ

τ

τττ (6)

where k is the number of the bin, kτ is the time delay of the k -bin and ( )kP τ is the average power of the k -bin. The Fourier transform of the normalized PDP gives the normalized frequency correlation function:

( ) ( )

∆−⋅=∆ ∑

−

= Nfn

jPfRN

nnH

πτ

2exp

1

0

(7)

where Ν is the number of the bins of which the normalized PDP is consisted. Noise-padding techniques are applied, prior to DFT, in order to refine the achieved frequency resolution. Based on the real value of

( )fRH ∆ , we calculate the coherence bandwidth for 90% 75% and 50% correlation.

Figure 3 depicts a normalized power delay profile derived by simulation procedure using equations (1), (2) and (5). During the process the horn-horn antenna system is used, whilst the receiver is 30 m away from the base station.

Figure 3: Normalized PDP concerning the horn-horn

transmission system with 30 m Tx-Rx separation.

From Figures 2 and 3, it is evident that the first two arrived components are the direct and ceiling reflection rays with equal power contribution as mentioned in Section III. The reflections from the left and right walls arrive together at 4 ns but they appear as one component due to the binning procedure. The strong peak that arrives at 6 ns is the single reflection from the ground, which is more delayed since it starts to contribute after 27 m as shown in Figure 2b. This is attributed to the narrow beamwidth at the vertical plane (15o) hence the radiation pattern includes the ground ray after the specific distance, therefore exhibiting greater delay. Finally, in Figure 3 the RMS delay spread is 1.58 ns and the maximum delay spread is 7 ns.

Figure 4 depicts the frequency correlation functions with 30 m separation between Tx and Rx comparing the three alternative transmission configurations reported in Section II. It is clear that with isotropic antennas the correlation drops faster implying larger delays since all the multipath components contribute to the total received signal. This has to do with the antenna radiation patterns; therefore isotropic antennas include all the reflected rays in their diagrams. On the other hand the horn-horn configuration, which has very narrow antenna beamwidths, does not include all the rays suppressing some of the multipath components that arrive to the receiver. This reduces the RMS delay spread, increasing

the channel quality. This is testified on Table 1 where the channel parameters are indicated for the three antenna configurations.

Figure 4: Normalized frequency correlation functions for the

three transmission configurations. The separation is 30 m between Tx and Rx.

From Table 1, is clear that the best results are provided by the omni-omni configuration due to its very narrow vertical pattern (only 8o) reducing significantly the reflections from the ceiling and the ground. Increasing the antenna directivity, RMS delay spread is reduced from 2.31 ns to 1.18 ns and 1.58 ns as shown in Table 1.

Table 1. Channel parameters for the three transmission configurations.

τRMS (ns)

B0.5 (MHz)

B0.75 (MHz)

B0.9 (MHz)

Isotropic 2.13 162 108 65 Omni-Omni 1.18 310 197 116 Horn-Horn 1.58 222 147 87

Finally Figure 5 depicts the values of RMS delay

spread as a function of distance for the alternative antenna configurations. Apart from the isotropic antenna configuration, which presents almost constant curve from 10 m to 37 m, RMS delay spread increases with distance as shown in omni-omni and horn-horn cases.

Figure 5: RMS delay spread as a function of Tx-Rx separation

for the three transmission configurations. Using isotropic antennas, all the reflected rays contribute significantly to the received signal along the entire corridor, therefore τRMS is larger compared to the other cases. When the antennas are more directive the initial τRMS value is very small (mainly direct component), but

as their patterns start to include other reflected rays, τRMS starts to increase reaching its maximum at the end of the corridor.

V. FUTURE WORK

The main target concerning future work is to compare the simulation results with measurements in the site-specific environment. Narrowband (CW) as well as wideband measurements will take place in order to verify and compare the presented results.

VI. CONCLUSIONS

Narrowband and wideband parameters were evaluated through a simulation process at 60 GHz for indoor WLAN applications. The signal can propagate with 4-5 rays (horn-horn antennas) to contribute to the total received signal. Hence less than 9 rays may be enough to describe, with great accuracy, the indoor signal propagation. The signal decays with distance at a rate of nd/1 where n=1.75 for the specific geometry suggesting wave guiding effects. The antenna radiation patterns affect significantly the signal propagation, since different patterns may alter the results changing the space and the time contribution of the reflected rays. Furthermore, the material characteristics affect in great extent the propagation either reflecting perfectly the signal (aluminum) or absorb significant amount of energy depending always on the incidence angle. Increasing the antenna directivity, RMS delay spread values can be reduced suppressing the multipath components. It is also found that the RMS delay spread varies significantly, increasing as a function of distance when directive antennas are used.

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