on design and analysis of broadband 2-segment dielectric...

Turk J Elec Eng & Comp Sci

() : 1 – 10

c⃝ TUBITAK

doi:10.3906/elk-1206-33

Turkish Journal of Electrical Engineering & Computer Sciences

http :// journa l s . tub i tak .gov . t r/e lektr ik/

Research Article

On design and analysis of broadband 2-segment dielectric resonator array

antenna for 5–6 GHz applications

Mohammed Fadzil AIN1∗, Ubaid ULLAH1, Zainal Arifin AHMAD2

1School of Electrical and Electronic Engineering, Universiti Sains Malaysia, Nibong Tebal,Pulau Pinang, Malaysia

2School of Material and Mineral Resources Engineering, Universiti Sains Malaysia, Nibong Tebal,Pulau Pinang, Malaysia

Received: 09.06.2012 • Accepted: 08.01.2013 • Published Online: ..2014 • Printed: ..2014

Abstract:A 2-segment dielectric resonator with an 8-element array energized with a corporate feed network is designed

and evaluated. It is well known that microstrip antennas are characteristically narrow bands due to 2 nonradiating

edges out of 4. In this work, 2-segment dielectric materials with diverse permittivity are used as a resonator that can

resonate in an omnidirectional pattern. A modified microstrip wrap-around parallel feed line with λ/4 transformer

etched on a single side of a copper-grounded substrate (εs = 3.38) is used to excite the dielectric resonator antenna.

The 2-segment dielectric resonators are loaded over the feed line by optimizing their position with respect to the open

ends of the feeding circuit. With this arrangement, approximately 17% (5.05–5.9 GHz) impedance bandwidth is achieved

with 13.8 dBi directivity and a reasonably directional radiation pattern. For comparative purposes, a microstrip patch

antenna array excited with same feeding network is also designed and evaluated. Simulation is performed using computer

simulation technology and close agreement between the simulation and the measured results is observed.

Key words: Dielectric resonator antenna array, 2-segment dielectric resonator, 8-element array, corporate feed line

1. Introduction

Planar antennas designed with different approaches and techniques have been reported in the literature [1–3].

Among these planar symmetry antennas, dielectric resonator antennas (DRAs) are consistently proving to be

promising antennas in the world of wireless communication. In the past few decades, numerous studies have

been carried out, reporting on different aspects of DRAs [4–11]. A number of feeding techniques can be easily

employed to excite DRAs with different geometrical shapes [12]. Rectangular dielectric resonators (DRs) have

an advantage over DRs with other shapes, i.e. spherical or cylindrical, having more degrees of freedoms in terms

of flexibility in dimension with 2 aspect ratios of width/height and width/depth [13]. Antenna engineers are

more prone to use rectangular DRs to achieve their anticipated profile and impedance bandwidth characteristics

rather easily. Many researchers have studied rectangular DRs and have reported about them in the literature

[14,15]. A multisegment rectangular DRA array was reported in [16], where a complicated microstrip branch

line network with 2 layers of substrate was used to feed the DRA. The electromagnetic energy was coupled to

DRs by using slot coupling. An additional stub was also used for efficient coupling of electromagnetic energy

to the resonators.

∗Correspondence: [email protected]

1

AIN et al./Turk J Elec Eng & Comp Sci

In this paper, a different approach is used to feed a 2-segment rectangular DR. To avoid the complications

involved in fabrication of complex circuits, a feed line with rather simple geometry is used to excite the DRA. A

modified copper feed line is printed on a single face of a copper-grounded substrate, which can be etched easily.

A rectangular DR is chosen as a combination of 2 materials with different permittivity. Materials with high

permittivity have high quality factors and hence respond in a narrow band with strong coupling capabilities. On

the other hand, materials with low permittivity (≥20) have low quality factors, wider impedance bandwidth,

and low coupling to the feed line. A study on single-element multisegment DRA antennas was published in 2000

[17]. The authors’ idea was to use multisegment rectangular DR with different permittivities stacked one above

the other. The lower segment with a high dielectric constant will act as in impedance transformer between the

feed line and the top segment, having a low dielectric constant and acting as a principle resonator. The same

idea was plotted out in the form of a modified microstrip wrap-around parallel feed array antenna using 2-

segment DRs. The resonant frequency of each DR was predicted using the modified wave guide model [17]. The

complete details of the modified dielectric waveguide model for predicting the resonant frequency of 2-segment

DRs can be found in [18]. For the top segment, a microwave laminate from the Rogers Corporations (Rogers RT

6010) (εr = 10.2) is used, which is easy to cut into different shapes and sizes. For the lower segment, internally

fabricated CaCu3Ti4O12 is used as an impedance transformer. This 2-segment rectangular DR is loaded over

the parallel corporate feed line etched on Rogers RO4003 copper-grounded substrate with permittivity of εs

= 3.38. A quarter-wave impedance transformer is used to transform line impedance and to split the power

equally among the 8 feeding arms of the array antenna. Due to the absence of a metallic patch, the chances of

surface waves are reduced, which could cause serious mutual coupling between the 2 immediate DR elements

and hence diminish the performance of the array antenna. For further control over mutual coupling due to space

waves between immediate resonators, separations between adjacent feeding arms of the parallel feed network

are attuned to λ/3 length at center frequency fc = 5.5 GHz. For comparative purposes, a microstrip patch

antenna (MPA) with an 8-element array energized with the same feeding network is also designed, and all the

results of the MPA are compared with the proposed array antenna.

2. Antenna configuration

An illustration of an elementary corporate feed network for a 4-element array is shown in Figure 1, which

was used to feed microstrip patch wrap-around antennas [19] mostly mounted on missiles for fixed-beam

communication with a radar system. As can be seen from Figure 1, 6 quarter-wave transformers are employed

to feed a 4-element patch antenna, which means more discontinuities and hence more power losses. The same

feed line with modified geometry is used to excite a 2-segment DR with 8-element array antenna efficiently.

To account for the effects of impedance mismatching and power losses due to discontinuities and bends in the

feed line, a slightly different approach has been used to design the feed line structure, as shown in Figure 2. In

our proposed feeding network architecture, only 2 quarter-wave impedance transformers are used to transform

impedances efficiently for a 4-element array, and the same network is extended to 8 feeding arms. This reduces

the number of discontinuities in the circuit and comparatively more power can transfer to the feeding arms. A

quarter-wave (λ/4) impedance transformer is used to split the power by transforming line impedances through

each junction. In general, the transmission line will transform the impedance of an antenna, making it very

difficult to deliver power, unless the antenna is matched to the transmission line properly. If the antenna is not

matched, the input impedance will vary widely with the length of the transmission line. Furthermore, if the

input impedance is not well matched to the source impedance, not much power will be delivered to the antenna.

2


Thus, in order to have a well-matched feeding network, a λ/4 transformer is used, as an interesting phenomenon

happens when the length of transmission line is λ/4. A well-known equation for impedance transformation in

transmission lines is as follows:

Z0 =√ZinZA (1)

Eq. (1) suggests that if a transmission line with impedance Z0 is employed with length L = λ/4, the input

impedance can be matched to the load impedance efficiently. Figures 3 and 4 respectively show the simulation

and prototype of a dielectric loaded parallel corporate feed array network with 8 feeding arms. Figure 5a

shows the geometry of the 8-element array MPA, while Figure 5b shows the dimension of the single patch.

By manipulating a quarter-wave impedance transformer line, impedances are matched efficiently in the entire

network.

100Ω

50Ω 70Ω 100Ω 70Ω 50Ω

100Ω 100Ω

50Ω 70Ω 100Ω70Ω 50Ω

100Ω

50Ω

50Ω70Ω100Ω70Ω

50Ω

50Ω

λ/4 λ/4

λ/4 λ/4

I1

Z1

Z1||Z2 I2

Z2 Z3 Z4

I3 I4

Za Zb Zc Zd

50Ω Input

Z3||Z4

100Ω70Ω

50Ω

50Ω

λ/4

Za Zb

50ΩInput

70Ω 50Ω

50Ω

100Ω

100Ω100Ω 100Ω 100Ω

Zc Zd

λ/4

100Ω

S=λ/3S=λ/3

Figure 1. Corporate parallel feed network. Figure 2. Modified corporate parallel feed network.

100 100 100 100

100

100 100

50

70

50

70 70

50

50 input

50 50 50

70 70 70

DR DR DR DR DR DR DR

Substrate =

Figure 3. Simulation profile of 8-element corporate feed antenna.

3


Figure 4. Prototype 8-element corporate feed array antenna.

(a) Simulation profile of eight element array MPA.

(b) Dimension of single patch.

50 input

100 100

70

70 70

50 50 50

50 50

100 100 100 100

Patch

100

15.51 mm

18 mm Metallic

Patch

Figure 5. a) Simulation profile of 8-element array MPA. b) Dimensions of a single patch.

Though ample amounts of losses occur in each junction of the feeding network, due to the absence

of metallic losses sufficient power is transmitted to each arm of the feeding network to excite the loaded DR.

Optimized and calculated dimensions for each pair of feeding arms and single-element 2-segment DRs are shown

in Figures 6 and 7, respectively. To optimize the separation (s) between 2 immediate elements for averting

phase and amplitude errors, a parameter sweep is performed with a step size of 0.05 in Computer Simulation

Technology (CST 2010). The 2-segment DR is excited by loading it over this optimized corporate feed parallel

network.

4


20 mm

100 Ω

100 Ωλ/4

S = λ/3

100 Ω

50 Ω70 Ω

1

1

z

y

x L

W

Ground

CCTO Microstrip

Line

Rogers RT6010

H

Figure 6. Optimized dimensions of each pair of feeding

arms.

Figure 7. Dimensions of single element 2-segment dielec-

tric resonator. W = 3.86 mm, L = 18.4 mm, H= 11.7 mm,

εL = 27.05 εr = 10.2, εs = 3.38, height ts = 0.813 mm,

tL = 2.6 mm, tr = 9.1 mm.

To theoretically predict the frequency of operation of the 2-segment DR, Eqs. (2) and (3) are used

to calculate the permittivity and height of the lower segment DR. The lower segment acts as an impedance

transformer between the feed line and the upper segment, which is our principle resonator.

εL=η0√ϵr

z0εL =

ηo√εr

zo(2)

tL =c

4fo√εL

(3)

The conventional dielectric waveguide model (DWM) [13] in Eq. (3) is then modified in such a way as

to compensate for the effect of the additional bottom segment and substrate on the main resonator.

kx tan (kxtr) =√

(εr − 1)k2o − k2x (4)

Effective permittivity (εeff )εeffεeff replaces εrϵr of the DR and effective height (Heff ) replaces height (tr)

of the DR in the DWM equation.Heff

εeff =Heff

tr/εr + tL/εL + ts/εs

(5)

Heff = tr + tL + tsHeff=tr+tL+ts (6)

Here, εr, ϵrεL , εL and εs are the dielectric constants of the top segment, bottom segment, and substrate,

respectively. tr, tL, tr,tL and ts represent the thicknesses of the top segment, bottom segment, and substrate,

respectively.

Eqs. (5) and (6) are substituted into Eq. (4) and consequently the modified dielectric wave guide model

becomes:

kx tan (kxHeff ) =√

(εeff − 1) k20 − k2x

5


where:

kx =√εeffk2o − k2y − k2z ,

k0 =2π

λ=

2πf0c

, ky =π

W, kz =

π

L.

C is the speed of light c = 3×108 , and W and L are the width and length of the 2-segment rectangular DR,

respectively. K is the wave number.

All the dimensions for the single resonator are measured in millimeters and they are stacked one above

another in such a way that no air gap remains between the 2 segments and there is strong coupling between

the DR and the feed line.

These calculated dimensions of the 2 segments are used for all elements used in the proposed array

antenna. Finally, different aspects of the designed 2-segment DR with 8-element array antenna are studied in

comparison with the 8-element MPA. All the results produced are discussed in the following section.

3. Results and discussion

Figure 8 shows simulated and measured return losses of the 8-element array antenna in comparison with the

MPA for the optimum positions of the 2-segment DRA over the feed line. For simulation, the minimum value

of return loss is –32.7 dB, while for measurement, it increases to –28.2 dB. The magnitude of 10 dB return loss

is from 5.0 GHz to approximately 6.0 GHz in both simulation and measurement, which shows 17% impedance

bandwidth of the antenna. Meanwhile, the graph of return loss of the 8-element microstrip patch array antenna

shows weak coupling and a narrower bandwidth up to 3.6%. Due to the enormous amount of metallic losses in

millimeter wave frequency, our proposed antenna array fed with the same feeding techniques surpasses the MPA.

With the 8-element 2-segment array antenna, close agreement is observed between simulation and measurement.

0

-10

-20

-30

-40

4.70 5.104.90 5.30 5.705.50 5.90 6.30

-10

6.10

Variable

Frequency (GHz)

Simulation

Measurment

MPA

Ret

urn

Lo

ss (

dB

)

Figure 8. Simulated and measured return loss for 8-element array antenna.

As the antenna is fed by direct microstrip line array, each element in the array is in phase with the others

and a broadside beam is expected to be produced. To give a clear picture of the radiation pattern for the

proposed antenna, a perspective view of a 3D radiation pattern with transparent antenna structure is shown

in Figure 9. A broadside radiation pattern with maximum directivity value of 13.87 dBi can be seen. The

Cartesian plot of the radiation pattern of the proposed 8-element array antenna in comparison with the MPA

is also measured in both the E-plane and H-plane. The results are shown in Figures 10 and 11 for the E-plane

and H-plane, respectively. Magnitude of the measured main lobe is 13.2 dB with broadside radiation at center

6


frequency and 3 dB angular width is approximately 16.5 , which suggests that the proposed antenna has a

relatively narrow beam width and so beam scanning of the antenna should be reasonably accurate. As can be

seen from the plot of the MPA patch array, the maximum value of the main lobe is 10 dB with a narrower beam

width, which is expected from metallic patches operating in the millimeter wave frequency range. A number of

side lobes also appear in the radiation pattern with an almost equal amount of current or equivalent voltage in

each element of the array.

Figure 9. Perspective view of 3D radiation pattern of array antenna.

Phi/ Degree

Dir

ecti

vity

(dB

i)

1801701601501401301201101009080706050403020101

15

10

5

0

–5

–10

–15

–20

–25

Variable

MPA

SimulationMeasured

Phi/Degree

Data

1801701601501401301201101009080706050403020101

10

5

0

-5

-10

-15

-20

-25

Variable

MPA

SimulationMeasurment

Figure 10. E-plane radiation pattern of 8-element array. Figure 11. H-plane radiation pattern of 8-element array.

Radiation patterns in both the E-plane and H-plane are almost the same, but not exactly, which is

because of the nonideal environment for measurements. The E-plane of the antenna shows discrimination in the

measured pattern, the beam width is slightly broad compared to simulation, and side lobes appear marginally

at different angles and with different amplitudes, which shows a minor phase shift in the measured radiation

pattern. Similarly, in the H-plane of the antenna most of the simulated and measured radiation is broadside

with the main lobe direction at 90 . A few side lobes on both sides of the main lobe are clearly visible with side

lobe level of approximately –1.2 dB. The dips present at random points in the radiation pattern confirm the

presence of phase and amplitude variations, which was expected due to radiation from each junction and bends

7


in the feed line. Furthermore, maximum directivity is achieved when the separation (s) between 2 immediate

elements is optimized to λ/3. It was observed that by further reducing the value of s , directivity decreased,

which shows mutual coupling between elements due to space wave flinches, which reduce directivity of the

antenna. In addition, for comparison, the radiation pattern of the MPA is plotted, which shows maximum

directivity of approximately 10 dBi and a number of side lobes on either sides of the main lobe. Both the E-

plane and H-plane of the MPA are closely similar, but with lower directivity and gain compared to the proposed

antenna.

Figure 12 shows the plot of simulated and measured gain in comparison with the MPA. The gain of the

antenna is measured by using the absolute-gain method compared to a standard antenna. The plot clearly

shows the dominance of the proposed antenna in both simulation and measurement compared to the MPA.

The maximum value of gain in simulation is 12.72 dB, while for measurement it is 12.5 dB. For the MPA, the

maximum gain in the antenna operating region is 9.7 dB, which means that our antenna has an advantage of

almost 3 dB gain over the MPA.

Frequency (GHz)

Ga

in (

dB

)

6.56.05.55.04.5

13

12

11

10

9

8

7

Variable

Measurment

MPASimulation

Figure 12. Gain of the array antenna.

4. Comparison with MPA

The Table shows a comparison between the 8-element arrays of the MPA and 2-segment DR array antenna. It

can be clearly seen that all the parameters of the proposed DRA surpass the MPA. Among these comparisons, a

noteworthy point is that by increasing the array factor, the number of elements increases and with that the size

of the dual-segment array is reduced compared to the MPA, which means that if the numbers of elements are

further increased for satellite or radar communication, a compact antenna can be designed at higher frequencies.

Table. Eight-element MPA array in comparison with proposed array.

Parameters MPA array Proposed array Difference (%)Dimensions of substrate (mm) L × W = 60 × 200 L × W = 65 × 155 19.25%Impedance bandwidth 3.6% 17% 13.4%Directivity 10.2 dBi 13.87 dBi 28%Gain (dB) 9.7 dB 12.8 dB 24.2%

8


5. Conclusion

In this paper, a 2-segment rectangular DR with 8-element array antenna excited with a microstrip corporate

parallel feed was addressed. A slight modification in the feed structure was made to have control over losses

in the feed line due to bends and junctions. An 8-element microstrip patch array antenna fed with the same

feeding network was also designed and we compared those results with our proposed antenna. It was found

that our proposed DRA outperformed the MPA array in almost all aspects of the antenna. Furthermore, the

simplicity in the modeling and fabrication of the feeding network makes this antenna superior to previously

reported multisegment DRAs. As this antenna covers a useful range of the frequency spectrum, it can be easily

employed for wireless communication systems operating in the range of 5–6 GHz. It was also found that if we

increased the number of elements in the array, the size of the antenna was reduced, and so this antenna has a

better chance to be used for radar or satellite communication with increased numbers of radiating elements in

the array antenna.

Acknowledgment

The authors gratefully acknowledge financial support from a USM short-term grant under project no. 304/PBA-

HAN/6039035 and a USM Research University (RU) grant under project no. 1001/PELECT/854004.

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