microwave wireless communication system - diva portal

Technical report, IDE0628, January 2006

Microwave Wireless Communication System

Master’s Thesis in Electrical Engineering

Carl Dagne, Johan Bengtsson, Ingemar Lindgren

School of Information Science, Computer and Electrical Engineering Halmstad University

Microwave Wireless Communication System

Master’s thesis in Electrical Engineering

School of Information Science, Computer and Electrical Engineering Halmstad University

Box 823, S-301 18 Halmstad, Sweden

January 2006

Description of cover page picture: A microwave antenna for the 2.4 - 2.5 GHz band.

Preface This project was performed during the autumn 2005 with Emil Nilsson and Arne Sikö as supervisors. We would like to sincerely thank both supervisors for their help with both theoretical and practical matters. Their knowledge and interest in the area has been a big aid to us in completing our work. We would also like to thank our opponent Ola Johnsson at FMTS, for his thoughts and comments. Carl Dagne, Johan Bengtsson & Ingemar Lindgren Halmstad University, January 2006

i

Abstract The purpose of the project was to develop the hardware to a microwave wireless system working at the frequency 2.45 GHz. The functionality of the system should also be easy to understand since the system is to be used in an educational purpose. Much time has been spent impedance matching components, a task that proved to be harder than we expected. Other work that has been is layout of all parts, filter construction and the writing of an easy to understand thesis. After the parts had been completed, they were tested in a network analyzer and/or spectrum analyzer. Successful full system test has been done up to 400 meters, the length the system is to be used for.

iii

Contents PREFACE......................................................................................................................................................................I

ABSTRACT................................................................................................................................................................III

CONTENTS................................................................................................................................................................IV

1 INTRODUCTION .............................................................................................................................................. 1 1.1 BACKGROUND.............................................................................................................................................. 1 1.2 ASSIGNMENT ............................................................................................................................................... 1 1.3 METHOD ...................................................................................................................................................... 1

1.3.1 Prestudy.................................................................................................................................................. 1 1.3.2 Design and Simulation ........................................................................................................................... 1 1.3.3 Measurements and Tests ........................................................................................................................ 1

1.4 READING INSTRUCTIONS.............................................................................................................................. 2 2 THEORETICAL BACKGROUND .................................................................................................................. 3

2.1 MICROSTRIPS ............................................................................................................................................... 3 2.2 IMPEDANCE MATCHING ............................................................................................................................... 4

2.2.1 The Smith Chart ..................................................................................................................................... 4 2.2.2 Scattering Parameters............................................................................................................................ 5 2.2.3 Impedance Matching with Lumped Elements (L-networks) ................................................................... 6 2.2.4 Impedance Matching with Microstrips................................................................................................... 9

2.3 FILTERS...................................................................................................................................................... 11 2.3.1 Bandpass IF Filter ............................................................................................................................... 11 2.3.2 Microstrip Filters ................................................................................................................................. 12 2.3.3 Lowpass Microstrip Filter.................................................................................................................... 15 2.3.4 Bandpass Microstrip Filter .................................................................................................................. 17

2.4 MODULATION TECHNIQUE......................................................................................................................... 18 2.4.1 Phase Modulation ................................................................................................................................ 19 2.4.2 Modulator............................................................................................................................................. 22 2.4.3 De-Modulator....................................................................................................................................... 22 2.4.4 Phase Locked Loop .............................................................................................................................. 23

2.5 OSCILLATORS ............................................................................................................................................ 26 2.6 LOW-NOISE AMPLIFIERS (LNAS) .............................................................................................................. 27

2.6.1 Impedance Matching of RF LNA.......................................................................................................... 29 2.6.2 The IF LNA........................................................................................................................................... 34

2.7 MIXERS...................................................................................................................................................... 34 2.7.1 Diodes .................................................................................................................................................. 34 2.7.2 Single-Ended Mixers ............................................................................................................................ 35 2.7.3 Balanced Mixers................................................................................................................................... 36 2.7.4 Double Balanced Mixers ...................................................................................................................... 36 2.7.5 Impedance Matching of Downconverter .............................................................................................. 36 2.7.6 Impedance Matching of Upconverter ................................................................................................... 41

2.8 ANTENNAS................................................................................................................................................. 42 2.9 CIRCULATORS............................................................................................................................................ 43 2.10 POWER AMPLIFIER..................................................................................................................................... 44

3 RESULTS.......................................................................................................................................................... 45 3.1 MIXERS...................................................................................................................................................... 45

3.1.1 Downconverter ..................................................................................................................................... 45 3.1.2 Upconverter.......................................................................................................................................... 46

iv

3.2 FILTERS...................................................................................................................................................... 46 3.2.1 Lowpass IF Filter ................................................................................................................................. 46 3.2.2 Lowpass Microstrip Filter.................................................................................................................... 47 3.2.3 Bandpass Microstrip Filter .................................................................................................................. 49

3.3 LOW-NOISE AMPLIFIER ............................................................................................................................. 51 3.3.1 Microwave Low-Noise Amplifier.......................................................................................................... 51 3.3.2 IF Low-Noise Amplifier........................................................................................................................ 52

3.4 POWER AMPLIFIER..................................................................................................................................... 53 3.5 OSCILLATOR .............................................................................................................................................. 54 3.6 MODULATOR ............................................................................................................................................. 54 3.7 DEMODULATOR ......................................................................................................................................... 55 3.8 SYSTEM TESTS ........................................................................................................................................... 55

3.8.1 Receiver................................................................................................................................................ 55 3.8.2 Transmitter ........................................................................................................................................... 55 3.8.3 Transmitter Receiver Test .................................................................................................................... 56

4 CONCLUSIONS............................................................................................................................................... 58

5 REFERENCES ................................................................................................................................................. 59

6 PERMISSION................................................................................................................................................... 62

v

Introduction

1 Introduction

1.1 Background Today wireless technology is used in many applications well integrated into our everyday life. One of the challenges for designing a microwave system is that the functionality of the electronic components changes when dealing with the upper frequencies of the UHF band. This means for example the filter design will differ much from conventional design methods, and transmission lines have to be short to make sure they do not work as antennas and thereby disturb the system.

1.2 Assignment The main purpose of this thesis is to construct a microwave link between Halmstad University and the Agellis laboratory. The system has to be fast enough to support a video link, which means a bandwidth of at least 4 MHz. The microwave link constructed will send signals at 2.45 GHz. This frequency was used since it belongs to the 2.4 GHz to 2.5 GHz band, which is one of the ISM frequency bands and is free to use. This thesis focuses on the RF part while the software interface used to run the complete system will be designed by another thesis group. The complete system will also be used in laboratory exercises in a microwave communication course at Halmstad University. The difference between this project and other similar available products is that this project will not try to make a compact system solution, but a system where all components are visible and the functionality is easy to understand.

1.3 Method The main goal of this thesis can be divided into the following three sub-goals: prestudy, design and simulation and measurements and test.

1.3.1 Prestudy An exhaustive study on existing material about microwave systems and the parts included in such a system was the first thing done. The studied material consisted of books, articles and web sites.

1.3.2 Design and Simulation After the prestudy, circuits were designed and simulated using PSpice and Orcad, produced by Cadence, and ADS 2004A, by Agilent.

1.3.3 Measurements and Tests Laboratory measurements and testing of the designed circuits were done continuously during the project using spectrum analyzer, signal generator and network analyzer to control the simulated results. Finally, a test of the whole system was carried out.

1

Microwave Wireless Communication Link

2

1.4 Reading Instructions This thesis is intended to be read by persons studying, or with degree in, Electrical or Computer Engineering. Basic knowledge of radio communication systems and filters is needed.

Theoretical Background

2 Theoretical Background A wireless communication system consists of several parts. Below is a block scheme of a transceiver. This chapter will explain the functionality of the parts in such a system and some basic theory needed for understanding these parts.

Figure 2-1 Components of a transceiver system.

2.1 Microstrips Microstrips are one type of transmission lines which are devices “used to transfer energy from one point to another efficiently” [4]. There are a number of different transmission lines, but the most important type for this project is microstrips. The microstrip consists of a ground plane and a strip conductor, separated by a dielectric substrate, as seen in figure 2-2.

Figure 2-2 Microstrip transmission line [3].

3


4

Microstrips are used at microwave frequencies where they are more efficient than conventional wiring. A major advantage when using microstrips is that it is easy to connect surface mounted components. As with all other wires conducting a current, the microstrip also has an electric field, which travels from the strip conductor to the ground plane. But since the electric field is not strictly confined to the area under the strip conductor, the field will also travel from the edge of the strip conductor to a place on the ground plane that is not situated directly under the conductor. This phenomenon is called fringing effect which will make the conductor seem electrically longer than it really is. This means that if a microstrip at quarter wavelength is wanted, the conductor has to be slightly shorter because of the fringing effects. Another thing that affects the length is the dielectric substrate that has a relative dielectric constant. But since the strip conductor has a dielectric substrate on one side and air on the other, an effective dielectric constant has to be calculated to compensate for the difference. A higher effective dielectric constant will then make the microstrip shorter because the speed of light is slower in a dens material, which in turn will make the wavelength shorter since λ=c/f. This project makes use of coaxial cables that have a conductor in the middle and a ground plane around the conductor with a dielectric substrate in-between. Like the microstrip, the electric field for the coaxial cable will travel from conductor to ground. When a coaxial cable and a microstrip transfer signals from one to the other, the electric field has to change form and it is therefore important to add an extra microstrip to make this change possible. This extra microstrip should also have the characteristic impedance which in this case is 50Ω, a number that can be altered by changing the width of the strip conductor. The relation is that the smaller the impedance, the larger the width [4], [2].

2.2 Impedance Matching The reason for doing impedance matching is to deliver maximum power to a load. The only requirement for a matching network to be found is that the load impedance, ZL, has a real part. When dealing with impedance matching, there are two different categories usually mentioned. The first one is matching of transmission lines, where it comes down to terminating the line with Z0, which is the characteristic impedance of the line. The other is matching a source or a load by deriving its complex conjugate [3].

2.2.1 The Smith Chart The impedance matching can be simplified by using a Smith chart which is an easy-to-use tool that can give an approximate solution when deriving an impedance matching network. It is represented by circles and lines as can be seen in figure 2-3.


Figure 2-3 The Smith chart

The real axis passes horizontally through the middle of the chart, while the bent lines from the right hand side of the chart to the outer circle stand for complex values. The complex lines above the real line are positive while the lines below are negative. The centre of the chart represents the real value 1. It is also possible to find admittance values by turning the impedance chart 180 degrees. When using the Smith chart, the matching is made by moving from the complex conjugate of the impedance to the centre of the chart by adding reactive elements that do not consume active power.

2.2.2 Scattering Parameters A type of parameter that often is used when talking about impedance matching of a system or device is the scattering parameter, also called S parameters. An S parameter can be calculated using the following equation.

Sij=Vi-/Vj

+

The above equation says that Sij can be determined by sending an incident wave of voltage Vj

+ into port j and measure the reflected voltage amplitude and phase Vi

-, coming out of port i. If considering a simple two-port device, S11 will give a value of how well the input is matched and

5


6

S22 will tell us how well the output port is matched. For a perfect match S11 and S22 should be zero, meaning that nothing of the signal will be reflected.

2.2.3 Impedance Matching with Lumped Elements (L-networks) The simplest way of impedance matching is by using the L section. This kind of network uses two reactive elements to match load impedance. There are two possible configurations as shown in figure 2-4 below.

Figure 2-4 Different configurations for L-section matching networks.

When deciding which network to use, the Smith chart is used. First, the normalized load impedance is derived, zL=ZL/Z0. If this number lies inside the 1±jx circle on the Smith chart, the network in figure 1a should be used and if it is outside the circle, the network in figure 1b should be used. The reactive parts in the figure above may be either inductors or capacitors. If the frequency is below ca 1 GHz, lumped elements can be used in the implementation. When dealing with frequencies above 1 GHz, the lumped elements should be converted into microstrips. This because of parasitic capacitances and inductances that becomes larger with increasing frequency [3]. Another way to see which network configuration to use when calculating the values of the components in the network is to find out which network works for which region. The allowed and the forbidden regions for the different kind of network configurations are shown in figure 2-5, where an impedance should be connected to the right of the networks. The determination of values for the reactive components can easily be approximated using the Smith chart. There are, however, some basic rules to be considered when dealing with the Smith chart in this way. A shunt component always moves on the admittance chart while a series component moves on the impedance chart. By themselves, an inductor always moves on the positive side of the smith chart while capacitors move on the negative side [2].


Figure 2-5 Allowed and forbidden regions for different network configurations [17].

Example Design an L section matching network to match the impedance Z=100-j50Ω, if Z0=50Ω at the frequency of 500 MHz. Solution After normalizing the impedance to z=2-j1Ω, it is plotted on the Smith chart of figure 2-6.

7


8

Figure 2-6 Impedance an admittance chart with Z and its complex conjugate plotted.

To achieve a match, the complex conjugate, Z*, of the impedance has to be known. It is easily found by changing the sign on the complex part of the impedance. As seen in figure 3, there are two possible network solutions to the problem. When using the network with the shunt inductor, the way to the centre of the chart is found by following the black lines b and c. b stretches from 0.2 to 0.5 on the admittance chart, i.e. it is 0.3 long, while c can be read to be 1.2. The values of the capacitor and the inductor can now be found.


05.532

0 ==fb

ZLa π

nH

305.52

1

0

==cfZ

Ca πpF

When calculating the values of the two reactances with the other matching network where there is a shunt capacitor, the formulas will differ slightly.

39.42 0

==fZxCb π

pF

1.192

0 ==f

yZLb π

nH

where x is the length of the line from Z* to the circle with the real value 1, but on the opposite side of the real axis. This gives x the number of 0.69. y is the same length as c but is measured on the opposite side of the real axis of the chart. The results of these calculations are shown in figure 2-7.

Figure 2-7 The two resulting matching networks.

The first two values should be applied to a and the second two values to b.

2.2.4 Impedance Matching with Microstrips When dealing with frequencies above ca 1 GHz, the normal performance of capacitors and inductors no longer applies. These lumped elements have to be substituted by microstrips to make a matching network. The most basic form of matching network is using single-stub matching. The aim of using single-stub matching is to make the length of the stub and the transmission line in such a way that a match is found for the admittance, Y. There are two ways of doing this when using a single-stub match. These are shown in figure 2-8.

9


10

Figure 2-8 Single-stub matching network with a) open circuit stub and b) short-circuit stub.

The stub will act differently depending on the length of the line. The open circuit stub will be capacitive when its length is from 0 to λ/4 and inductive when from λ/4 to λ/2. The short-circuit stub will act in the opposite way compared to the open circuit stub. In microstrips, short-circuit stubs are difficult to realize which means that the open circuit stub often is used [18]. Example Find the matching network for the admittance load, YL=0.3+j0.3, using microstrips.

Figure 2-9 Smith chart for the matching network.

Theoretical Background Starting at the load, the length of the microstrip is found by moving toward from point A to point B while keeping a constant distance from the centre of the chart. This distance is found to be 0.376 λ. The length of the open stub can then be found to be 0.152 λ by moving from point B to the centre of the chart and measuring the length of the stub on the outermost scale of the Smith chart.

2.3 Filters Several filters are needed in a microwave system like the one in this project where their main purpose is to shape the signal spectrum. As can be seen in figure 2-1, various filters are needed. These filters are bandpass IF filter and bandpass RF filter, but a RF lowpass filter will also be needed to filter out the harmonics from the local oscillator.

2.3.1 Bandpass IF Filter The IF bandpass filter made with the Cauer technique, has a centre frequency of 70MHz and a bandwidth of 10MHz. The lower stopband extends from zero to 60 MHz and the other stopband reaches from 80 MHz to infinity.

62

61

62

61

10*80*2

10*60*2

10*75*2

10*65*2

πω

πω

πω

πω

=

=

=

=

S

S

C

C

If multiplying the two passband frequencies and the two stopband frequencies and comparing these two products, it is easily seen that they differ from each other. One frequency then has to be changed to make them equal. The first stopband frequency is the chosen to be changed and will be given the new frequency 60.9375 MHz. The next step is to use frequency transformations to convert to lowpass filter.

612 10*10*2πωω =−=Ω CCC rad/s

612 10*0625.19*2πωω =−=Ω SSS rad/s

The order of the filter can then be found to be N=4 using MatLab. For the network configuration, a π-network was chosen and the normalized values were found to be C1’=0.9307, C2’=0.2191, C3’=1.631, L2’=1.063, L4’=0.8307, R1’=1, R2’=1, where R1’ and R2’ are the normalized values to the 50Ω impedances.

11


12

Figure 2-10 The normalized lowpass filter.

Frequency transformation was made to get the network of the bandpass filter. The values were calculated to be C1=296pF, C2=69.74pF, C3=6.14pF, C4=519pF, C5=7.86pF, L1=17.6nH, L2=74.5nH, L3=846nH, L4=10nH, L5=661nH, R1=50Ω, R2=50Ω.

Figure 2-11 The bandpass filter.

Since most of the values of the lumped elements are not available as standard components, the values had to be changed. The new values were found using ADS. C1=330pF, C2=82pF, C3=8.2pF, C4=470pF, C5=8.2pF, L1=15nH, L2=68nH, L3=680nH, L4=10nH, L5=680nH, R1=50Ω, R2=50Ω. Changing the values of these elements will of course change the characteristics of the filter. These changes will however be acceptable according to ADS. However, it is very important to test the filter in a network analyzer to confirm that it works. This is not certain since the components have 5% tolerance which means that the characteristic of the filter still can change.

2.3.2 Microstrip Filters When dealing with frequencies above ca 1 GHz, it is better to use filters constructed with microstrip transmission lines instead of lumped elements. There are lumped elements that can be used for high frequencies but since these have standard values another advantage using microstrips is that any length and width of the transmission line can be made. There are a number of different configurations to use when designing microstrip filters. When it comes to bandpass filters, two frequently used types are coupled line and capacitively coupled line filters [11].

Theoretical Background Figure 2-12 shows these two types, where both types can have different lengths, width and spacing between the lines.

Figure 2-12 a) Capacitively coupled line filter and b) coupled line filter.

Open stub filters are another type of filter that are used in microwave systems, but are used to create lowpass filters.

Figure 2-13 A third order open stub filter with 50 ohm transmission lines.

Designing open stub filters is done with the use of Richard´s transformation and Kuroda´s identities. Richard´s transformation is used to convert from lumped elements to transmission line sections.

13


14

Figure 2-14 Richard´s transformation for an inductor and a capacitor [3].

The length λ in 2-15 is for the centre wavelength ω0. Kuroda´s identities are used because it is difficult to implement series stubs in microstrip filters. This is done by separating filter elements by using transmission line sections [3].


Figure 2-15 The four Kuroda´s identities used to convert from open-circuited stubs to short-circuited stubs and vice

versa, where n2=1+Z2/Z1 [3].

2.3.3 Lowpass Microstrip Filter The reason to make a lowpass filter is mainly to eliminate the harmonics from the local oscillator (LO). Since the LO is oscillating at 2.38 GHz, the harmonics will be at N*2.38GHz, where N is an integer greater or equal to 2. Using MatLab, the lowpass was calculated to be a fifth order filter with the passband frequency fc=2.5 GHz. The normalized values for the filters can also be found using MatLab. g1=1.7058

15


16

g2=1.23 g3=2.5408 g4=1.23 g5=1.7058 A T-network was chosen for this filter. Richard´s transformation is used to transform the T-network from lumped elements to open- and short-circuited stubs, figure 2-16 a. Unit elements are added to make the use of Kuroda´s identities easy.

Figure 2-16 Transformation from short-circuited stubs to open stubs using Kuroda´s identities [3].

Theoretical Background Denormalization with the characteristic impedance 50Ω and the frequency 2.5 GHz gives the impedances of the final filter. The width of the transmission lines were then calculated using TX-line. After these calculations were done, the filter was implemented in ADS and the parameters were tuned to achieve best possible result.

2.3.4 Bandpass Microstrip Filter Coupled line filter technique was used and the centre frequency should be f0=2.45 GHz, with a bandwidth from 2.4 GHz to 2.5 GHz.

Where

17


18

k=N-1, Z0e1 means the first even mode characteristic impedance and Z0o

1 means the first odd mode characteristic impedance. Using the characteristic impedance Z0=50Ω gives the following impedances on the coupled lines.

Z0e1=62.09 Z0o

1=37.91 Z0e

2=49.93 Z0o2=44.38

Z0e3=49.93 Z0o

3=44.38 Z0e

4=62.09 Z0o4=37.91

The length, width and spacing between the lines can then be poorly estimated with TX-line and optimazation and/or tuning with ADS or some other computerized tool is a must to make this a working filter. The final filter can be found in figure 2-17.

Figure 2-17 The complete bandpass filter.

The filter is constructed of 4 regions, where the first and the fourth are the same, just as the second and the third are the same. The first and the last element are 50Ω impedances, the first and fourth region consists of 2 lines with width W1=27.5 mils, length L1=848 mils and the spacing S1=27 mils. The second and the third region are constructed with W2=39 mils, L2=838 mils and S2=75 mils. Mils is one thousandth of an inch and is a very common measure.

2.4 Modulation Technique The function of the modulator and de-modulator is of course to modulate and demodulate the signal. There are three major ways to modulate a signal. The first way is amplitude modulation (AM). AM work in such a way that the information signal causes the carrier signal to increase and decrease in amplitude, where high amplitude corresponds to a 1 and low amplitude corresponds to a 0. This form of modulation is very sensitive to noise and is mostly used for sending data at a low bit rate. The next way to modulate a signal is by frequency modulation (FM). The information signal makes the carrier signal to increase or decrease in frequency and two different frequencies correspond to 1 and 0. The last way of modulating a signal is by phase modulation.

Theoretical Background 2.4.1 Phase Modulation Phase modulation, or phase shift keying (PSK), is today one of the most used way of modulating data. There are also different ways of doing PSK, two examples are binary phase shift keying (BPSK) and quadrature phase shift keying (QPSK). BPSK uses different phase to represent binary 1 and 0.

The type of modulation used in this project is QPSK. QPSK uses four different phases to represent 00, 01, 10 and 11 and uses therefore the bandwidth more efficiently since every phase shift represents two bits instead of one bit, as in BPSK.

Figure 2-18 Phase diagram [1]

The modulation of the signal is done by dividing it into 2 signals, I and Q, multiplying the I with a sinusoidal signal at 70 MHz and multiplying the Q with a cosinusoidal signal at 70 MHz in a mixer. The results from the mixers are then added to each other to get the quadrature phase signal. In our case, the binary bits 0 and 1 will be represented by the logical levels -1 and +1. The theory of how to get the quadrature phase signal can be proven by the use of Euler’s relations.

jeet

tjtj

2)sin(

ωω

ω−−

= 2

)cos(tjtj eet

ωω

ω−+

=

If I and Q equals 1, then

19


20

)4/sin(2

)22(21

)2

1()2

1(

)21

21()

21

21(

22)cos()sin(

)4/()4/(

4/4/

πω

ωω

πωπω

πωπω

ωω

ωω

ωωωω

+=−

=

=−=

=−

−+

=

=−−+=

=+

+−

=+

+−+

−−

−

−

−−

tjee

eeeej

jje

jje

je

je

eejeett

tjtj

jtjjtj

tjtj

tjtj

tjtjtjtj

The result if doing similar calculations with the different kinds of input signal will be as shown in table 2-1.

Binary input I (after mixer) Q (after mixer) Sum 00 -sin(ωt) -cos(ωt) sin(ωt-135°) 01 -sin(ωt) cos(ωt) sin(ωt+135°) 10 sin(ωt) -cos(ωt) sin(ωt-45°) 11 sin(ωt) cos(ωt) sin(ωt+45°)

Table 2-1 Corresponding phase shift in the output signal for different binary input signals [1].

When the phase quadrature signal received on the other side of the system it has to be demodulated.

Figure 2-19 A simplified demodulator.

To make the demodulator demodulate the received signal correctly, it has to know when a signal starts. It is therefore important to use a synchronizer circuit which in this case will contain a

Theoretical Background phase locked loop (PLL), discussed later on, and an element that divides the frequency from the PLL by four. The basic function of the PLL is to multiply the IF signal frequency by four and by doing so eliminating the differences in phase shift as can be seen in the equations below [1]. One problem that could arise is that the local oscillator possibly could have a constant phase error because of the different path lengths between the signal going through the synchronizer and the one going through the divider.

)4sin()4*4/34*sin()()4sin()4*4/34*sin()(

)4sin()4*4/4*sin()()4sin()4*4/4*sin()(

πωπωπωπωπωπωπωπω

+=−=+=+=+=−=+=+=

tAtAtstAtAts

tAtAtstAtAts

After the signal has gone through the PLL, it is divided by four so that the original frequency is regained, without modulated phase shift. This means that the local oscillator in the demodulator is running at the correct frequency. The demodulation of the received signal is then be done by multiplying it in a mixer circuit. This can be mathematically proved by using Euler’s relations.

2)2cos(

2)cos(

2)cos(

2)2cos(

42)2cos(

4

22)sin()sin(

)2()()()2(

)()(

ϕωϕ

ϕϕω

ϕω

ϕωω

ϕϕ

ϕωωϕωϕωωϕω

ϕωϕωωω

+−=

=+−

+=

=−+

−−

+=

=−

+−−=

=−

∗−

=+∗

−

+−−+−−+

+−+−

t

t

eet

eeeejee

jeett

jj

tjttjttjtj

tjtjtjtj

The only problem with this method is that it cannot distinguish between the phase shifts with a + or a – sign. Therefore, to decode all four quadrants, the input signal has to be multiplied by both a sinusoidal and a cosinusoidal waveform. The higher frequency also has to be filtered out, which is done by using a lowpass filter. If multiplying the received signal with a cosinusoidal waveform at the same frequency, this will after a few steps give the following result [1].

2)sin(

2)2sin()sin()cos( ϕϕωϕωω +

+=+∗

ttt

21


22

2.4.2 Modulator The modulator, MAX2452, an integrated circuit (IC) was chosen from Maxim that filled the requirements needed for this project.

Figure 2-20 A block diagram of the modulator integrated circuit [7].

I and Q represent the two bits sent at every phase shift. These signals will have a frequency of 600 kHz. The I signal is then mixed with a sinusoidal signal and the Q with a cosinusoidal signal, both at 70 MHz, and added to each other to get the final quadrature phase signal. The local oscillator is run by a TANK-circuit that controls the operating frequency, which in our case will be oscillating at 140 MHz [1].

2.4.3 De-Modulator The demodulator used is the MAX 2451 includes many of the parts that can be found in the modulator.


Figure 2-21 Block diagram of the demodulator integrated circuit [26].

The input signal to the demodulator is the IF signal that is oscillating at 70.6 MHz. It is divided into two signals where one of these signals is mixed with a sinusoidal signal at 70 MHz and the other is mixed with a cosinusoidal signal at 70 MHz. The sinusoidal and the cosinusoidal signal are provided by the local oscillator. The difference between the modulator and the demodulator is that the frequency of the local oscillator in the demodulator is not decided by a TANK-circuit. Instead, a phase locked loop (PLL) is used. By using a PLL the demodulator will be synchronized with the modulator sending the data [1].

2.4.4 Phase Locked Loop Phase locked loops are analogue circuits that are commonly used in many analogue and digital systems. PLLs can be used for clock recovery in communication systems, frequency synthesizers in TVs and wireless communication systems to select different channels and much more. This is done by adjusting the PLL oscillating frequency to get it to match the desired frequency. PLLs are non-linear systems, but when in lock, their behaviour can be estimated with linear equations. A PLL usually consists of a phase detector, a loop filter, a voltage controlled oscillator (VCO) and a frequency divider as shown in the figure below.

23


24

Figure 2-22 Basic structure of a PLL [22].

The phase detector compares the input signal to the signal received from the voltage controlled oscillator (VCO). The bigger the phase difference, the bigger the output voltage from the phase detector. The loop filter makes the result from the phase detector smoother and passes it on to the VCO. The VCO then changes its oscillating frequency depending on the voltage at the input. If the PLL is stable and the system reaches a point where the two inputs to the phase detector are in phase, the signals will also have the same frequency since angular frequency is the derivative of the phase with respect to the time. When changing the number N in the frequency divider, the synthesizer output signal will change accordingly. The phase detector can be built in different ways. One way is to use a double balanced mixer to multiply the input signals, but a much better way is to use a phase frequency detector (PFD). It contains two D flip-flops and an AND gate as shown in figure 2-23. If fr or fv goes high, the Up port or the Dwn port will go high respectively. When they both have the logic value 1, the AND gate will cause the flip-flops to reset. This means that the two output ports only are 1 the amount of time it takes the AND gate to reset the flip-flops. The waveforms for the ports to the PFD can be found in figure 2-24. When fr and fv are the same, the output ports Up and Dwn will also be the same. This means that when calculating Up-Dwn, the result should be zero for the two input signals to be equal. If for example fr oscillates at a higher frequency than fv, FF1 will be set to one more often than FF2. The two flip-flops will however be reset to zero an equal number of times. The result of this will be that the Up port will have a higher average value than the Dwn port. Making the calculation Up-Dwn will then give a positive value which means that the VCO in the PLL will be fed with a higher voltage, making it oscillate faster [22].


Figure 2-23 A PLL with two flip-flops and an and-gate [22].

Figure 2-24 Waveform of the PLL shown in the figure above [22].

The PLL used in this project has the disadvantage that it has no memory to remember for example the number in the frequency divider. The PLL therefore has to be reprogrammed every time the device is turned on. The reprogramming of the PLL will be done using a program called Codeloader 2 via a parallel cable.

25


26

2.5 Oscillators Oscillators are complicated nonlinear circuits to design. A simplified and linearized version of an oscillator can look like the one in figure 2-25. The two port oscillator consists of an amplifier and a linear filter H(s), as a feedback component.

Figure 2-25 Basic structure of an oscillator.

The input to the oscillator can be thermal noise or a step response that is removed once the oscillator reaches steady state but is necessary to start the device. Since the networks is supposed to oscillate, it has to have a pair of complex conjugate poles in right-half complex plane of the unit circle. The oscillator loop gain should be exactly one to prevent the circuit from attenuating or increase uncontrollably. This project will be using a voltage controlled oscillator (VCO) MAX2753 to convert the signals to IF and RF. A VCO can be achieved by using a varactor in the filter and by doing this, the oscillation frequency is changed. The output signal produced by the MAX2753 will be made to oscillate at 2.38 GHz. This means that there will be harmonics at n*2.38 GHz, where n is an integer.

Figure 2-26 Harmonics for the MAX2753 [27].

Theoretical Background The harmonics of the oscillator can be seen in figure 2-26. The first harmonic is suppressed 30dBc (dBm compared to carrier) and the second harmonic 27dBc. The inductor and the varactor of the tank circuit controlling the frequency are integrated on the circuit which simplifies making of the external circuitry.

2.6 Low-Noise Amplifiers (LNAs) The smallest signal that can be received by a system defines the receiver sensitivity and is set by the noise. The largest signal that can be handled by the receiver without affecting the quality of the data, gives an upper power limit. The Low-noise amplifier (LNA) plays a very important role in the receiver design. Its main purpose is to amplify extremely low signals without decreasing the signal to noise ratio (SNR). When designing a LNA there are many parameters to consider and it is impossible to design a LNA without trade offs. Some of the parameters that give a description of how well a LNA performs are gain, noise figure, stability, linearity, low power consumption and input and output match. Parameters that interfere with each other are for example low noise figure and good input match, stability and gain, IP3 and current consumption. The first and most important step in a LNA design is the selection of transistor. There are three design parameters that first should be taken into consideration which are noise, gain, IP3 and decide what VCE and IC will give the best performance, information that can be found in the datasheet for the device. LNA linearity is also an important parameter and a measure of it is the 3rd order intercept point (IP3) which indicates how well the LNA performs in presence of strong nearby signals and how well it deals with harmonics. For bipolar junction transistors, the output-IP3 can be estimated using the following formula:

)5**log(*103 CCE IVOIP = [dBm] [25] where VCE is in volts and IC is in mA. The Input-IP3 can also be estimated taking the OIP3 subtracted by the gain. After having chosen a suitable transistor, the next step is to choose DC bias circuitry. It should give stable thermal performance, be cost effective and simple solution that occupies smallest possible area. One of the simplest forms of DC biasing that fulfils the major requirements is shown in figure 2-27.

27


28

Figure 2-27 Simple form of DC biasing [25].

This DC bias is good since a stable biasing point is wanted. If IC decreases, VC will increase which will result in a higher Ib. Higher Ib will in turn increase IC. If the temperature increases, IC will increase. This will lower VC causing IB to decrease which finally will cause IC to decrease. For Rb to have little influence on the source matching, which is important for the noise figure, the feedback network should be decoupled with an inductor. Another possibility bias feedback can be realized with an emitter resistor and a capacitor, shown in grey in figure 2-27. Ce should present a short at the operating frequency to make its influence on gain and noise performance as small as possible. Stability is the next thing to consider. The LNA should be unconditionally stable, which means that the device will not start to oscillate no matter what load is presented on the input or output port. Stability can be controlled using the s-parameters and by using them in the following equations.

21122211 SSSSS −=∆

||2||||||1

2112

2222

211

SSSSSK ∆+−−

=

If the intermediate value ∆S is less than 1 and the factor K (called Rollett Stability Factor K) is larger than 1, the circuit will be unconditionally stable. There are some methods to make the LNA stable. One of them is to use R-L-C feedback between collector and base in order to lower the gain at lower frequencies and thereby improving the stability. Another method is to use a matching filter, usually put at the output of the transistor to make the gain decrease for specific

Theoretical Background narrow bandwidth at high frequency. Yet another method is to use an emitter feedback inductor in order to make the LNA more stable at higher frequencies. The next thing to do is to match the noise and input return loss (IRL) of the source. A usual approach is to match the input impedance of the transistor with Γopt which gives the best noise match. Normally, this means that the input return loss of the LNA will be sacrificed since the optimal IRL only can be achieved with a matching network that terminates the complex conjugate of S11. How this is done is shown in the calculation part of this chapter.

2.6.1 Impedance Matching of RF LNA The low-noise amplifier used in this project will be the MAX2644 from Maxim. When calculating the matching network, there are several parameters that have to be known. These can be found in [20]. The scattering parameters are found to be S11=0.5619/-35.54° S21=5.6624/150.06° S12=0.0236/68.36° S22 = 0.4043/5.00° As can be seen in the scattering parameters, S21 and S12 differ a lot. This means that the device is active and is amplifying the signal travelling from port 1 to port 2. This also means that any matching circuitry placed at port two will affect the input reflection coefficient very little, compared to what a matching circuit at port 1 would affect the output reflection coefficient. In most passive devices, the two parameters, S21 and S12, would be the same. If looking at the suggested surrounding circuitry for the MAX2644 in figure 2-28 from the data sheet of the low-noise amplifier, it can be seen that there are already a shunt inductor and a series capacitor on the input, where the capacitor is to act as a DC block and the shunt inductor is a matching circuitry. This capacitor does not have to be taken into consideration when calculating the matching network since its value is very big (33pF) i.e. its reactance will be very small.

Figure 2-28 The surrounding circuitry of the MAX2644 low-noise amplifier [20].

29


30

The first thing that will be checked is if the amplifier is unconditionally stable by calculating the intermediate value |∆S| and the stability coefficient K.

°−=−=∆ 1558.6/3021.021122211 SSSSS

2901.2||2

||||||1

2112

2222

211 =

∆+−−=

SSSSSK

Since the intermediate value is less than 1 and the stability coefficient is larger than 1, the amplifier is stable. It is now possible to calculate the maximum available gain.

42.17)1log(10||||log10 2

12

21 =−−+= KKSSMAG dB

For a perfect match, S11 should be 0, which means that nothing of the incident wave is returned to port 1. This is naturally done by a matching network. But, since the matching network should consider the noise figure the calculations will differ from conventional matching. Also, it is important to take the reflection coefficients into consideration since the matching network added to the input will make a difference to the reflection coefficient on the output and vice versa. The next step is to calculate two intermediate quantities C2 and B2 so the load reflection coefficient can be found.

2

221122

21||2

CKSSB

L−±

=Γ

Where

°−=∆−= 678.10/25934.0*11222 SSSC

and 75646.0||||||1 22

112

222 =∆−−+= SSSB This gives

°=Γ 678.10/39682.L The next step is to find the source reflection coefficient by doing similar calculations.

1

221121

21||2

CKSSB

S−±

=Γ

Theoretical Background Where

°−=∆−= 048.44/46623.0*22111 SSSC

and

061.1||||||1 2222

2111 =∆−−+= SSSB

This gives

°=Γ 048.44/54733.0S Γin and Γout can now be calculated.

°−=Γ−Γ

+=Γ 576.43/547.01 22

211211

L

Lin S

SSS

°−=Γ−Γ

+=Γ 9474.9/4019.01 11

211222

S

Sout S

SSS

Since it is important to think about the noise, a noise figure circle is drawn to calculate the best match for the noise. According to the datasheet for the device the minimum noise figure of the transistor is Fmin=1.589 dB, the reflection coefficient for optimum noise figure is Γopt=0.408/70.63°, the equivalent noise resistance of the transistor is RN=21.94Ω and the noise figure is 2 dB at maximum gain. The first thing to do is to calculate the radius and the centre of the 2dB noise figure circle.

1172.0|63.70408.01|50/94.21*4

442.1585.1|1|/4

22

0

min =°∠+−

=Γ+−

= optN ZR

FFN

°∠=+

Γ= 63.704558.0

1NC opt

F

2988.01

)||1( 2

=+

Γ−+=

N

NNR opt

F

31


32

Figure 2-29 Smith chart with noise figure circle and constant gain circle.

The next step would be to draw a gain circle around S11

* and then match the point where the two circles intersects. However, this is not necessary since it can be seen that the complex conjugate to Γin lies within the noise figure circle. It is therefore only necessary to match this complex conjugate. More about how to match with lowest possible noise can be found in [2]. Since this device will be working in the microwave region, it is better not to use lumped elements. Instead, these elements will be substituted by microstrips. These will be derived from the Smith chart.


Figure 2-30 The deriving of microstrip lengths using the Smith chart.

The match will be done as explained in the impedance matching chapter. Working backwards, the first thing that will be calculated is the length of the line on the input. The length will be found by moving from 1 toward the load to 2. This length is read to be .269λ and with a λ that according to TX-line will equal 84.4504mm at 2.45 GHz for the laminate chosen for this project, this means that the length of the line will be 22.72mm. The length of the open stub is then found to by moving from 3 to 0 which gives 0.146* λ=12.33mm. The length of the line and the stub for the output is found if doing the same way as with the input. The line will be .328* λ=27.70mm and the open stub .117* λ=9.88mm. ADS was used to tune these parameters and the final lengths (in mm) can be found in figure 2-31.

33


34

Figure 2-31 The final matching network.

2.6.2 The IF LNA A low-noise amplifier for the intermediate frequencies will also be needed in the system. The MAX2611 was chosen to meet the requirements. The s-parameters of the LNA are already very good, thus, no external matching are needed. According to the datasheet the LNA will amplify the input signal 19 dB with a noise figure of 3.6 dB.

2.7 Mixers The purpose of using mixers is to convert a signal to a different frequency. To do this, a non-linear element is used that multiplies the two input signals. This element is most commonly a diode, but can also be a transistor. Mixers are three port circuits that have to be impedance matched at all ports to achieve good sensitivity and low noise. This can be complicated since several frequencies and their harmonics are involved. An important parameter for a mixer is its conversion loss, defined as

LC=10log (available RF input power/IF output power) dB [2]. Normally, the conversion loss is between 4 and 7 dB. A factor that affects the conversion loss of the mixer is the power level of the local oscillator signal. For minimum conversion loss, most LO powers should be more than 0 dBm and less than 10 dBm [2].

2.7.1 Diodes A diode can basically be seen as a non-linear resistor. Its DC V-I characteristic can be expressed as

1)−= VS (eII(V) α

where α=q/nkT, and q is the charge of an electron, k is Boltzmann’s constant, T is the temperature, n is the ideality factor and IS is the saturation current. If the diode voltage is set to be


v+= 0VV where V0 is the DC bias voltage and v is an AC signal voltage. This will change the V-I characteristic for the diode and can by using Taylor series be expressed as

...21)(

00

2

22

0 +++=VV dV

IdvdVdIvIVI

where I0=I (V0) is the DC current. The first derivative can be seen as

jdS

VS

V RGIIeI

dVdI 1)( 0

0

0

==+== αα α

which defines the junction resistance of the diode, Rj, and the dynamic conductance of the diode, Gd=1/Rj. The second derivative is then expressed as

'0

222

2

)(0

00

ddSV

SV

d

V

GGIIeIdVdG

dVId

==+=== ααα α

The V-I characteristic of the diode can now be rewritten as

...2

)( '2

00 +++=+= dd GvvGIiIVI

and is thus the three-term approximation for the diode current [2].

2.7.2 Single-Ended Mixers The single-ended mixer is the simplest type of mixers and is often used as a part in more complex mixers. If a RF signal is mixed in a downconverter with a signal from a local oscillator and the signals can be described as

)cos()(

)cos()(

tvtv

tvtv

LOLOLO

RFRFRF

ω

ω

=

=

From the three-term approximation for the diode current it is possible to see that the diode current will consist of a DC current, the RF and local oscillator frequencies. The v2 term will then give rise to the following output current.

35


36

)))cos((2))cos((2

)2cos()2cos((4

))(cos)cos()cos(2)(cos(2

))cos()cos((2

2222'

2222'

2'

tvvtvv

tvtvvvG

tvvvtvG

tvtvG

i

LORFLORFLORFLORF

LOLORFRFLORFd

LOLOLORFLORFRFRFd

LOLORFRFd

ωωωω

ωω

ωωωω

ωω

++−+

+++=

++=

+=

The DC terms can be ignored since it only changes the voltage level and the terms ωRF, ωLO and ωRF+ ωLO will be filtered out. This means that the term of importance, ωRF-ωLO, are left [2].

2.7.3 Balanced Mixers A balanced mixer is made of two or more identical single-ended mixers together with a 3 dB hybrid junction (90° or 180°) to give better input SWR or RF/LO isolation. A hybrid junction is a four-port device that when sending a incident wave into port 1 will couple to port 2 and 3 but not to port 4. The angle 90° or 180° tells how much the output ports will differ in phase. If the local oscillator port is in anti-phase (balanced) the mixer will reject all products where the RF signal has a harmonic with even number. If the RF port is in anti-phase the mixer will reject all products where the LO signal has a harmonic with an even number. Since the port that is in anti-phase also is cancelled at the IF port, the LO is often chosen to use this port because the LO should be driven at a higher level than the RF signal [2].

2.7.4 Double Balanced Mixers In a double-balanced mixer, both the LO and the RF are balanced. Another thing that makes it different from the single-balanced mixer is the fact that it normally uses four diodes in a ring or star configuration. The advantages of using a double-balanced mixer is that all ports are isolated to each other and that it has increased linearity which gives it improved suppression of harmonics. All even order products are suppressed. By using more diodes the IP3 is also improved. On the negative side, the LO has to be driven at a higher level and the mixer requires two hybrid junctions.

2.7.5 Impedance Matching of Downconverter MAX2680 is a low noise downconverter from Maxim that operates at low voltage. In this project the RF input signal has been selected to operate at 2.45 GHz and is mixed with a LO signal operating at 2.38 GHz using a double balanced mixer. This gives a down converted IF output signal operating at 70 MHz [23]. In the datasheet [23] it is possible to find the impedance loads for different frequencies which will be matched with a transmission line Z0 = 50 ohm. These impedance loads are found to be


ZRF-port = 33+j73 (for RFinput port at 2.45 GHz) ZIF-port = 803+j785 (for IFoutput port at 70 MHz)

In this case it is not necessary to match the LOinput port because it has a VSWR better than 1.5:1. This means that the impedance load lies somewhere inside a circle with a radius of 0.5 from the centre of the Smith chart (see figure 2-32). The LO-port is considered to be close enough to the centre of the chart, where it is known to be perfectly matched. It is more important to have a matched RF-port for the weak incoming RF-signal and a matched outgoing IF-port so the signal will be strong enough to be detected by the demodulator [23].

Figure 2-32 A VSWR better than 1.5:1 [24].

To match the RFinput port and IFoutput port there are rules to follow. They can be found in the impedance matching chapter. In the fig 2-33 you can see zin and zout which are the normalized complex conjugate to the impedance loads ZRF and ZIF respectively.

37


38

Figure 2-33 Impedance matching with Smith chart.

Figure 2-33 shows that the zin lies outside the 1±jx circle and will have the configured network like figure 2-4b. It also shows that zout lies inside the 1±jx circle and will have the network like figure 2-4a. To calculate the values of the capacitors and inductors for the matching network the distance from the impedance loads to the centre has to be analyzed. Starting at zin and following line a and moving in the negative direction indicates that it is a capacitor. Since line a is moving on the impedance chart this indicates that it is a serial capacitor. Following line b to the centre by moving on the admittance chart means that this is a parallel component. Because of the movement in positive direction indicates that it is an inductor. The same rule applies for zout which give us a parallel inductor and a serial capacitor. By measuring the lines, a, b, c and d, it is possible to calculate the inductors and the capacitors values.


667.010*2450*2*)48.046.1(*50

121

600

1 =+

==ππfaZ

C pF

33.410*2450*2*75.0

502 6

0

01 ===

ππfbZ

L µH

947.010*70*2*12.0

502 6

0

02 ===

ππfcZ

L µH

579.710*70*2*6*50

121

600

2 ===ππfdZ

C pF

The calculated values are an approximation and had to be simulated in ADS to get more precise values, C1= 0.669pF, L1= 4.39µH, L2= 0.96µH, C2= 7.73pF. The final matching network configuration is to be seen in the Fig 2-34.

Figure 2-34 Matching network for the downconverter.

Since ZRF has an incoming signal at 2.45 GHz it should be matched with microstrip component. It is not necessary to do this because there are lumped component that works at higher frequencies. But a good approach to get a precise matching is using microstrips. Methods for impedance matching with microstrips can be found in the impedance matching chapter. The elements that will be substituted by microstrips are C1 and L1. These will be derived from the Smith chart in figure 2-35.

39


40

Figure 2-35 Deriving the lengths of the microstrips using the Smith chart.

To calculate the length d you have to move from 1 towards the load to 2. This length is shown to be 0.403λ. The laminate chosen for this project has a wavelength that equals λ= 84.4504mm at 2.45 GHz. This means that the length of the line d =33.95mm. The length ℓ of the open stub is then found to by moving from 3 to 4 which gives 0.17* λ=14.36mm. Since a dc-block has to be placed between the microstrip and the RFinput port, a capacitor at 68 pF was added. This moved our impedance load on the Smith chart so instead of recalculate the values, a simulation was made in ADS. The new lengths were found to be


d = 34.065 mm ℓ =14.4725 mm

Figure 2-36 The microstrip matching network for the RFinput port.

2.7.6 Impedance Matching of Upconverter MAX 2671 is a low noise upconverter using a double balanced mixer operating at low voltage. The IF input signal has been selected to operate at 70 MHz and is mixed with a LO signal operating at 2.38 GHz. This gives an upconverted RF-output signal at 2.45 GHz [28]. In Maxim’s datasheet [28] you find recommended values for inductors and capacitors that will be used as a matching network at the RF-port. If you analyze these values the load will not be correctly matched to a 50Ω source. In this approach the recommended components will be used to get the correct bias current to the RF port and a complementary matching network to these components will be done. The impedance loads were found to be

ZRF-port= 31-j95 (by RFoutput port at 2.45 GHz) ZIF-port= 200-j300 (by IFinput port at 70 MHz) The same approach to calculate the impedance matching network were done like the downconverter, so the value will only be displayed. The final matching network was simulated by ADS and had the configuration like Fig 2-37.

41


42

Figure 2-37 Impedance matching network for the upconverter in ADS.

2.8 Antennas There are many different antennas to choose from when designing a wireless system. The challenge is to find an antenna that fit a particular system. The size of the antenna is related to the wavelength. Since these antennas are made in solid materials, the wavelength will be shorter. When designing antennas it is a big advantage if it is possible to make the antenna of such a length where it is resonant, which means that is has no reactance. Another aspect to take into consideration when choosing an antenna is the polarization it uses. A vertically polarized antenna cannot receive horizontally polarized radiation, just as a horizontally polarized antenna is not able to receive vertically polarized radiation. This can also be an advantage since less noise is received. Radio links can for example double its bit-rate by sending both horizontally and vertically polarized on the same frequency. One of the most basic antennas is the dipole antenna. The dipole antenna is fed at its centre and has normally the total length of half a wavelength. The dipole antenna can be good to use since it is easy to match its radiation resistance to a transceiver. A loop antenna can be a good choice when making a hand-held transmitter system since it can be printed on a small circuit board. The biggest disadvantage when using a loop antenna is that it is very inefficient. The patch antenna is an antenna used extensively for the 2.4GHz band and higher. Like the loop antenna, the patch antenna is also printed on a circuit board. The form of the antenna is normally circular or rectangular, although other shapes are also used. The radiation is generally maximum perpendicular to the board. The reason that the length L of the patch has to be a little bit shorter than a forth of a wavelength is because of the so called fringing effect. Fringing effect occurs because a fraction of the field generated by the antenna lies outside the physical patch dimensions W*L.


Figure 2-38 Rectangular microstrip antenna.

When designing the antenna, there are a number of things to consider. The length L of the patch, decides the resonance frequency, a critical parameter since the bandwidth of the patch is very small. The width of the antenna does not affect the resonance frequency much but is a major factor when calculating the input resistance and the bandwidth. When using a large patch, this will increase the power output and therefore decrease the resonance resistance but increase the bandwidth and the radiation efficiency [13], [5]. The antennas used are two patch antennas made by Svenska Antennspeciallisten AB with a gain of 9 dBi. Using the Friis power transmission equation it is possible to calculate how strong the signal will be at the receiver. The formula reads

)4( 2

2

RGG

PP rttr π

λ=

where Pr is the received power, Pt is the transmitted power, Gt is the transmitter gain, Gr is the gain for the receiver and R is the lengths between the antennas.

2.9 Circulators RF circulators are passive devices used to control the propagation of an RF signal. A three-port circulator can look as the device in figure 2-39. The interaction between magnets and ferrite materials give rise to a circular motion in the magnetic field which can be strong and will cause a signal on a port to follow the magnetic field around to another port and not being able to travel in the opposite direction.

43


44

Figure 2-39 Overview of a three-port circulator.

If a signal is sent from port 1 and the magnetic field travels in the clockwise direction as in the figure above, the signal will first come to port 2. If this port is matched, the signal will enter this port but if it is mismatched, the signal will be reflected at port 2 and continue to port 3 and enter it if this port is matched [4]. A circulator will not be constructed in this project, but will instead be bought.

2.10 Power Amplifier A Power Amplifier is placed before the filter at the end of a transmitter as near the antenna as possible to avoid attenuation of the signal being transmitted. Its purpose is to give a high output power of the modulated signal so it can be detected and decoded by the receiver. For this project a component produced by Minicircuit has been chosen. This component (ERA-3SM) is a broadband amplifier designed to deliver an output power of minimum 9dBm in the frequency range DC to 3GHz. The typical gain for a signal at 2 GHz is 18,7dB with a minimum of 16dB. For matching purpose, the scattering parameters were measured with bias circuitry only. A matching network was then calculated from these parameters.

Results

3 Results All the finished circuits were tested in a network analyzer, spectrum analyzer and/or oscilloscope to verify their functionality. The result of these measurements is presented in this chapter.

3.1 Mixers

3.1.1 Downconverter The downconverter was tested in the network analyzer and was matched at the IF side by tuning a trimmer capacitor and changing the inductor value to 0.27uH. The RF side also had to be adjusted by using a small amount of copper-tape. After being fairly matched a test using the spectrum analyzer was done to see how the downconverter worked. Driving the LOinput port with a power of -5 dBm at a frequency of 2.38 GHz, the IFoutput power was measured. The power at the RFinput port was changed from -24dBm to -50 dBm. 2.45 GHz 70 MHzRF: -24 dBm → IF: -21.3 dBm RF: -30 dBm → IF: -27.3 dBm RF: -40 dBm → IF: -37 dBm RF: -50 dBm → IF: -47 dBm As we can see the RFoutput signal is amplified but according to its datasheet [23] an amplification at around 7dB should occur. One of the reasons for this could be that the downconverter was fairly matched, but not properly. Since the VCO, which should drive the LO port, hade an output power of -11 dBm and could not be improved, another test was done with this power. 2.45 GHz 70 MHz RF: -24 dBm → IF: -24 dBm RF: -30 dBm → IF: -30 dBm RF: -40 dBm → IF: -40 dBm RF: -50 dBm → IF: -50 dBm It could be seen from the IFoutput port was that the amplification we had disappeared but no attenuation of the outgoing signal in proportion to the incoming signal was seen. The 2.38 GHz signal was also leaking through the IFoutput port. This signal was suppressed -20 dbm and had a power of -31dBm. The values were considered to be good enough for the receiver part.

45


46

3.1.2 Upconverter The test using the network analyzer of the upconverter showed that adjustments of the matching had to be done at both sides. The inductor and the capacitor was changed to 0.22 uH and 17.5 pF at the IF side while the RF side was adjusted with copper tape. After being fairly matched a test on the spectrum analyzer was done to see how the upconverter worked. By driving the LOinput port with a power of -11 dBm at a frequency of 2.38 GHz, the RFoutput power was measured. The power at the IFinput port was -24dBm. The result showed three signals. Frequency Amplitude2.31 GHz(fLO-fIF) -40 dBm 2.38 GHz(LO) -1 dBm 2.45 GHz(RF) -40 dBm The first signal with a frequency of 2.31GHz is a mixed signal between the LO and IF (fLO-fIF). It could be seen of the two other signals is an amplified LO signal and a suppressed RF signal. The wanted result was a -22 dBm suppressed LO signal and a 9 dB amplified RF signal according to the datasheet [28]. The problem was solved by removing the inductor and changing the capacitor to 220 pF. The test resulted in the following signals. Frequency Amplitude2.24 GHz(fLO-2*fIF) -42 dBm 2.31 GHz(fLO-fIF) -17 dBm 2.38 GHz(LO) -36 dBm 2.45 GHz(RF) -16 dBm 2.52 GHz(fLO+2*fIF) -42 dBm We got the expected result for the LO but the RF signal was not amplified as much as stated in the datasheet. Apart from this, two new signals were found at 2.24 GHz and 2.52 GHz. These new signals were rather weak and will not be any problem but the 2.31 GHz signal with an amplitude of -17 dBm has to be filtered.

3.2 Filters

3.2.1 Lowpass IF Filter The bandpass filter for the IF region was tested in the network analyzer. When looking at the result it was obvious that this could not be used since the filter response was extremely bad. Instead a new filter was designed with the help of ADS only. This time, a simple lowpass filter configuration was used. The filter response can be seen in figure 3-2.

Figure 3-1 The finished IF lowpass filter.

Results

IF lowpass filter

-90

-80

-70

-60

-50

-40

-30

-20

-10

0

10

0 1 2 3 4 5 6

Frequency (GHz)

Am

plitu

de (d

B)

S11 (dB)S21 (dB)

Figure 3-2 IF filter response.

At 70 MHz S11 is -20.7 dB and S21 is -1 dB.

3.2.2 Lowpass Microstrip Filter The theoretical result of the filter given by ADS can be seen in figure 3-3.

47


48

Figure 3-3 Theoretical lowpass filter response.

This can be compared to the filter response from the network analyzer seen in figure 3-4. The constructed filter is close to the theoretical filter response but there is a peak at approximately 5.5 GHz that need to be suppressed in order to make the filter work better. To eliminate this peak, the filter was improved by connecting it to the network analyzer and putting small patches of copper tape on it. The improved filter response can be seen in figure 3-5.

2.38 GHz lowpass filter

-50-45-40-35-30-25-20-15-10

-505

0 2 4 6 8 10 12 14

Frequency (GHz)

Ampl

itude

(dB)

S11 (dB)S21 (dB)

Figure 3-4 The characteristics of the lowpass filter.

Results

Tuned 2.38 GHz lowpass filter

-50-45-40-35-30-25-20-15-10

-505

0 2 4 6 8 10 12 14

Frequency (GHz)

Ampl

itude

(dB)

S11 (dB)S21 (dB)

Figure 3-5 Improved lowpass filter response using copper tape.

It is obvious that the final filter is much better than before the adjusting using copper tape. Since this filter is going to be used to filter out the harmonics of the local oscillator (LO), it is important that S21 is low for these frequencies. There is a small peak at 12.5 GHz but since this is very far away from the central frequency of the local oscillator.

Figure 3-6 The RF lowpass filter with tuning.

3.2.3 Bandpass Microstrip Filter This filter was also tested with the network analyzer and gave the graph in figure 3-7. Its theoretical response can also be seen in figure 3-9.

49


50

2.45 GHz bandpass filter

-90

-80

-70

-60

-50

-40

-30

-20

-10

0

10

0 5 10 15

Frequency (GHz)

Am

plitu

de (d

B)

S11 (dB)S21 (dB)

Figure 3-7 Characteristics for the bandpass filter.

It is clear that these two graphs differ because of irregularities when the filter was fabricated, but it is still a filter that will work for the purpose it was made for. The first peak on S21 is at 2.45 GHz, with the attenuation 1.9 dB. At this point, the return loss is 27.4 dB.

Figure 3-8 RF bandpass filter.

Results

Figure 3-9 Characteristics for the theoretical bandpass filter.

3.3 Low-Noise Amplifier

3.3.1 Microwave Low-Noise Amplifier The first time the RF LNA was tested, it did not amplity the signal at all. To make it work, the connections to the ground plane had to be shortened to be as small as possible, the shorter they where, the higher the gain. After making them as small as possible, a gain at about 15 dB was achieved, but not the 16 dB as promised in the datasheet for the device. The dip in S11 that was supposed to be at 2.45 GHz was at 2.8 GHz. The LNA impedance matching network was therefore tuned to get a better value. Figure 3-11 shows the scattering parameters for the tuned LNA.

Figure 3-10 RF Low noise amplifier.

51


52

LNA scattering parameters

-40

-30

-20

-10

0

10

20

0 0,5 1 1,5 2 2,5 3 3,5

Frequency (GHz)

Ampl

itude

(dB) S11 (dB)

S21 (dB)S22 (dB)S12 (dB)

Figure 3-11 Scattering parameters for the RF LNA.

At 2.45 GHz S21 is 13.91 dB, S11 -14.73 dB, S22 -13.58 dB and S12 -29.25 dB.

3.3.2 IF Low-Noise Amplifier

The IF LNA was not matched in any way, but simply put together with its bias circuitry. Its scattering parameters can be seen in figure 3-13. The gain of the LNA is approximately 19.5 dB and the reflected signal will be attenuated 13 dB.

Figure 3-12 The IF low-noise amplifier.

Results

IF LNA scattering parameters

-50

-40

-30

-20

-10

0

10

20

30

0 1 2 3 4 5 6

Frequency (GHz)

Ampl

itude

(dB

)

S11 (dB)S22 (dB)S12 (dB)S21 (dB)

Figure 3-13 S11 and S21 for the IF LNA.

3.4 Power Amplifier The result from the power amplifier can be seen in figure 3-15. The gain of the device can be read as ca 14 dB and the reflected signal was attenuated a bit more than 15 dB.

Figure 3-14 The power amplifier.

53


54

Power amplifier

-40

-30

-20

-10

0

10

20

0 0,5 1 1,5 2 2,5 3 3,5

Frequency (GHz)

Am

plitu

de (d

B)

S11 (dB)S21 (dB)S22 (dB)S12 (dB)

Figure 3-15 Scattering parameters for the power amplifier.

3.5 Oscillator The oscillator was fabricated as stated in the datasheet for the device. The control voltage was chosen to be controlled with a potentiometer. The oscillator is also frequency dependant which means that it has to be reset to 2.38 GHz every time it is location with different temperature. Looking at the oscillator frequency spectrum using the spectrum analyzer it could be seen that the frequency 2.38 GHz had an output power at -11 dBm. The first harmonic was suppressed 13.5 dBc and the second 23 dBc. Both these figures are less than stated in the datasheet but since they are far away from the carrier frequency, it will not be a problem to suppress them. Another two harmonics were found, the first at 11.9 GHz and the second at 14.28 GHz, both of them suppressed 50 dBm. No signal was found at 9.52 GHz.

Figure 3-16 Oscillator.

3.6 Modulator The modulator was tested with an oscilloscope and was found to be working rather poorly. When the prescaler output from the circuit was tested, a signal that should have been a square wave at 17.5 MHz was found to be a triangular wave that varied from 17 MHz to 18 MHz. The IF signal

Results was even worse and varied from 65 MHz to 75 MHz. After some improvements the output signal varied 3 MHz which is not good enough.

3.7 Demodulator The demodulation of the signal is done with three circuits, phase locked loop, frequency divider and the demodulator itself. The frequency divider divides the signal by two, from 280 MHz to 140 MHz and is working fine. But because the PLL is not working, we have not been able to test the demodulator.

3.8 System tests

3.8.1 Receiver After each part had been confirmed to work the receiver side was put together to control its functionality. During these tests, the demodulator was not used since there was no signal to demodulate. An input signal was created using a signal generator and sent to the first part of the receiver system, the LNA, using a cable. This means that there was no actual wireless transmission, but a simulated weak input signal via a cable. During these simulations we found that the local oscillator leakage through the downconverter to the IF side was significant. Two other signals at 4.83 GHz and 4.9 GHz could also be detected even if they where weak. To be sure to eliminate these signals another lowpass filter was added to the IF side. The first test with antennas was done indoors in a 20 meter long corridor. The transmitted power was set to 0 dBm and could be seen at the spectrum analyzer connected to the receiver. The only problem was that the received power varied some, something that probably could be explained by the interference of signals travelling different ways. To further test the system, a signal was sent between two buildings of approximately 35 meters. The received signal power was could be measured to -40.6 dBm and -49.91 dBm for a transmitted power of 0 dBm and -10 dBm respectively. A transmission was also done using antennas only, which gave a result of -52.59 dBm and -63.08 dBm for the same transmitted signal powers. These values can be controlled using the Friis equation which will give the result -52.03 dBm and -62.03 dBm. The next step was to check if the receiver could detect a signal sent from 400 meters away, the length it is supposed to be working at. Sending a signal of 0 dBm gave a received signal power of -59 dBm with a noise floor at -85 dBm. To get a better signal, another LNA was added to the RF side of the receiver, making the final value -48 dBm. It should be noted that the transmitted signal power is adjusted by the signal generator and that the antennas will cause an amplifying effect of the signal of 9 dB. When calculating the received power using the Friis equation, a higher value will be found. This depends on that the antennas are hard to align manually which means that the antenna amplification will not be 9 dB but a value slightly less than that.

3.8.2 Transmitter Since we knew that an output power of 0 dBm was sufficient for the receiver to detect, the transmitter side of the system only had to be tested by itself to make sure it fulfilled this criterion. Using only the power amplifier after the upconverter and feeding the upconverter with -27 dBm

55


56

at 70 MHz from a signal generator, gave a signal power of -1.5 dBm. Adding an extra LNA before the PA gave an output power of 10 dBm.

3.8.3 Transmitter Receiver Test Again, the 400 meter stretch was used to transmit our signals and two LNAs were used at the receiver side. This time, a -27 dBm, 70 MHz signal was used as input to the upconverter. The power at the receiver was measured to -36 dBm when using the PA after the upconverter and -25 dBm when adding the LNA in between these two circuits. A test was also done to see how the signal looked for the RF port of the downconverter. At the receiver side, the two LNAs and a bandpass filter was used.

Figure 3-17 RF input to the downconverter.

As can be seen in figure 3-17 the 2.31 GHz signal, with second largest amplitude, from the upconverter is not suppressed enough, as well as the LO signal, between the 2.31 GHz signal and the 2.45 GHz signal, which is the one with the highest amplitude. Since it is important to eliminate these signals, especially the 2.31 GHz signal, a new bandpass filter with smaller bandwidth was needed at the transmitter. The two signals to the far left are signals transmitted from elsewhere and not from our transmitter. Since we want to suppress these signals further, a new bandpass filter wither smaller bandwidth will be added to the receiver too. The same test with the new bandpass filters can be seen in figure 3-18.

Figure 3-18 RF input to downconverter using the new BP filters.

Results The 2.31 GHz signal, number 3, has been suppressed so it is no longer detectable, the local oscillator signal, number 2, is still detectable but is further suppressed. The 2.45 GHz signal is at number 1. When using the whole system and sending the downconverted 70 MHz signal into the spectrum analyzer the signal was seen to be quite noisy. This noise only occurs when doing transmissions outdoors. This means that it most likely comes from some other wireless application and can be eliminated by fixing the components to a common ground plane and using shielding boxes.

57


58

4 Conclusions The purpose of this thesis was to construct two transceivers that should communicate using QPSK that can be used in educational purpose. Because of too little time and money, the finished system only consists of one transmitter and one receiver, where the transmitter is missing a working modulator. If we ignore the modulation of the signal, the system works very well for the length it is designed for, 400 meters. One thing that has been difficult during this work is the impedance matching of components. The scattering parameters mentioned in datasheets that has been used to calculate the matching networks have never given a satisfying result when controlling them with the network analyzer. The result has actually been so bad that we have had to rematch the components with copper tape instead of finely adjust the matching that first was made. We are not sure why this is, but it probably has something to do with parasitic capacitances and inductances. Another thing is that the matching network solutions suggested by manufacturers of the components do not work either. This might have something to do thing the laminate and the type of components used. The matching was further complicated since the network analyzer used had some malfunctions and there were times we did not know if we could rely on the values given by it. For future work, the modulator needs to be stabilized and the PLL probably needs to be redesigned to make it work. During this time, we feel that we have learnt a lot about how microwave system works, but at the same time we have realized how much there is to learn about this area. The best thing about this work has been that we have gained own experiences by doing a practical work.

References

5 References [1] P Karlsson, F Svensson, Microwave Wireless Communication Link, Halmstad

University, Halmstad, Sweden 2005. [2] David M. Pozar, Microwave Engineering (second edition), John Wiley & Sons

Inc, 1998. ISBN: 0-471-17096-8 [3] Noyan Kinayman, Modern Microwave Circuits, Artech House Inc, 2005. ISBN: 1-58053-726-X [4] Thomas Laverghetta, Microwaves and Wireless Simplified (Second Edition), Artech House Inc, 2005. [Online] Available: Halmstad University’s E-brary. ISBN: 1-58053-944-0 [5] Alan Bensky, Short-range Wireless Communication: Fundamentals of RF System

Design and Application, L L H Technology Publishing, 2000. [6] Maxim Dallas semiconductor (2005-05-14) RF and wireless ASICs. [Online] Available: http://www.maxim-ic.com/products/asics/wireless/ [7] Maxim Dallas semiconductor (2005-05-15) MAX 2452 Quadrature Modulator. [Online] Available: http:// pdfserv.maxim-ic.com/en/ds/MAX2452.pdf [8] David A. Johns, Ken Martin Analog Integrated Circuit Design, John Wiley &

Sons Inc, 1997. ISBN: 0-471-14448-7 [9] RdE Radiowave Ltd, (2005-05-16). [Online] Available: http://www.radiowave.co.uk/RFhardware.html#FILTERS [10] Microwave Filters, (2005-05-16). [Online] Available: http://home.sandiego.edu/~ekim/e194rfs01/filterek.pdf [11] Richard Brown, RF/Microwave Hybrids: Basics, Materials and Processes [Online] Available: Halmstad University’s E-brary. ISBN: 1-20-207233-3 [12] University of San Diego, Microstrip Transmission Lines [Online] Available: http://home.sandiego.edu/~ekim/e194rfs01/mstrip.pdf [13] R. Garg, P. Bhartia, I. Bahl, A. Ittipiboon, Microstrip Antenna Design Handbook,

Artech House Inc, 2000. 59


60

ISBN: 0-89006-513-6 [14] Mona Mostafa, RF CMOS Power Amplifiers: Theory, Design and Implementation,

2001. ISBN: 0-7923-7628-5 [15] Maxim Dallas semiconductor (2005-05-23) MAX 2244 Power Amplifier. [Online] Available: http://www.maxim-ic.com/quick_view2.cfm/qv_pk/3239/ln/ [16] Microwave 101 (2003-05-23) Circulators. [Online] Available: http://www.microwaves101.com/encyclopedia/circulators.cfm [17] http://contact.tm.agilent.com/Agilent/tmo/an-95-1/classes/imatch.html [18] Arto Lehto, Radio Engineering for Wireless Communication and Sensor

Applications, Artech House Inc, 2003. ISBN: 1-58053-669-7 [19] Voinigescu, S.P.; Maliepaard, M.C.; Showell, J.L.; Babcock, G.E.; Marchesan, D.;

Schroter, M.; Schvan, P.; Harame, D.L., A Scalable High-Frequency Noise Model for Bipolar Transistors with Application of an optimal Transistor Sizing for Low-Noise Amplifier Design. Solid-State Circuits, IEEE Journal of Volume 32, Issue 9, Sept. 1997 Page(s):1430 – 1439.

[20] http://pdfserv.maxim-ic.com/en/ds/MAX2644.pdf [21] Trung-Kien Nguyen; Nam-Jin Oh; Choong-Yul Cha; Yong-Hun Oh; Gook-Ju

Ihm; Sang-Gug Lee, CMOS Low Noise Amplifier Design Optimization Technique. Microwave Theory and Techniques, IEEE Transactions on Volume 53, Issue 2, Feb 2005 Page(s):538 – 547.

[22] Bianchi Giovanni, Phase-Locked Loop Synthesizer Simulation, McGraw-Hill

Companies, 2005. [23] http://pdfserv.maxim-ic.com/en/ds/MAX2680-MAX2682.pdf [24] http://www.mathworks.com/products/demos/rftoolbox/imped_match/imped_match.html#6 [25] www.semiconductors.philips.com/acrobat/other/discretes/CDMA_LNA_design.pdf [26] Maxim Dallas semiconductor (2005-05-15) MAX 2451 Quadrature Demodulator. [Online] Available: http:// pdfserv.maxim-ic.com/en/ds/MAX2451.pdf [27] Maxim Dallas semiconductor (2005-05-15) MAX 2753 Quadrature Modulator.

http://www.maxim-ic.com/quick_view2.cfm/qv_pk/3239/ln/

http://www.microwaves101.com/encyclopedia/circulators.cfm

http://pdfserv.maxim-ic.com/en/ds/MAX2680-MAX2682.pdf

http://www.semiconductors.philips.com/acrobat/other/discretes/CDMA_LNA_design.pdf

References [Online] Available: http:// pdfserv.maxim-ic.com/en/ds/MAX2753.pdf [28] http://pdfserv.maxim-ic.com/en/ds/MAX2660-MAX2673.pdf

61

http://pdfserv.maxim-ic.com/en/ds/MAX2660-MAX2673.pdf


62

6 Permission 2.14, 2.15, 2.16 Reproduced by permission from Author (Noyan, Kinayman), Modern

Microwave Circuits, Norwood, MA: Artech House, Inc., 2005. © 2005 by Artech House, Inc.

2.20, 2.21, 2.26, 2.28 Copyright Maxim Integrated Products (http://www.maxim-ic.com).

Used by permission.

https://webmail.hh.se/horde-3.0.6/services/go.php?url=http%3A%2F%2Fwww.maxim-ic.com

microwave wireless communication system - diva portal

Documents