microwave photonics || photonic oscillators for thz signal generation

4

Photonic Oscillators for THz SignalGeneration

Andreas St€ohr and Dieter J€ager

4.1 Introduction

The photonic oscillator concept is a rather new technique for providing low-phase noise

continuous-wave signals in the THz regime. Compared to other electrical and optical

generation techniques, photonic oscillators exhibit a number of unique features such as

ultra-wideband tuneability, compactness and ability to operate over a broad temperature range

making it an interesting device for several THz applications. According to a recent study

initiated by the European Space Agency (ESA), photonic oscillators utilizing advanced

photodetectors are considered as one of the most promising candidates for the generation

of THz signals [1]. This chapter reviews the state-of-the-art in photonic oscillators for THz

generation and compares this new technique with other existing electrical and optical

approaches. Since the development of terahertz photonic oscillators is strongly related to the

invention of ultra high-frequency photodetectors we will also cover recent achievements in

high-frequency photodetectors in this chapter. Explicitly, we will discuss the high-frequency

performance of distributed travelling-wave photodetectors which exhibit a great potential for

efficient local oscillator (LO) generation at THz frequencies. This fact is experimentally proven

by demonstrating compact photonic oscillators employing advanced travelling-wave photo-

detectorswhich are indeed capable of providing sufficient LOpower at THz frequencies (e.g. to

pump a superconductor–insulator–superconductor (SIS) mixer at around 500GHz). A key and

unique feature of a photonic oscillator compared to other THz sources is its ability to allow for

tuning the LO frequency over a wide frequency range. As an example, compact photonic

oscillator modules exhibiting an amazingly large tuning range of almost 1 THz will be

presented.

Microwave Photonics: Devices and Applications Edited by Stavros Iezekiel

© 2009 John Wiley & Sons, Ltd. ISBN: 978-0-470-84854-8

4.2 THz Sources

Today, the term terahertz is commonly applied to submillimetre-wave (submm-wave) energy

that fills the wavelength range of 1000 mm–100 mm (300GHz–3 THz). Below 300GHz, we

enter the millimetre-wave (mm-wave) regime covering the wavelength region of 10–1mm

(30–300GHz). Beyond 3 THz and up to 30 THz (100–10mm)we enter the far-infrared (far-IR),

but the exact border between submillimetre and far-infrared is still rather blurred since the

frequency range 3–10 THz is more or less unclaimed territory. Above 30 THz there is the

infrared region covering thewavelength span 10–1mm(frequency 30–300 THz). In Figure 4.1,

the commonly assigned terms for the various frequency bands of the electromagnetic spectrum

between 30GHz and 3 PHz are indicated.

So far, the THz frequency regime has been of interest for some more or less niche

applications existing in that high-frequency range. Today, we notice a significantly increased

interest in THz technologies which is in part stimulated by some recent space and earth

exploration experiments such as the Atacama large millimetre array (ALMA) telescope

project [2]. The ALMA telescope will be the largest ground telescope for astronomers in the

next decade, comprising some 64 submm-wave high quality antennas. ALMA will make it

possible to scan and map the emission lines of a variety of lightweight molecules within the

frequency region from 30 to 950GHz for studying the formation of stars and planets. However,

besides astronomy there is also much interest in THz technology for imaging applications (T-

ray imaging), for security applications and even for some communication systems. An

excellent overview of the various applications in the THz frequency band can be found in

the review paper by Siegel [3].

Despite the fact that there is quite some interest in THz frequencies, the commercial use (e.g.

the development of compact THz sources) is only just beginning to emerge. Looking at the

various existing concepts for high-frequency LO power sources summarized in Figure 4.2, it is

noticeable that all-electronic solutions such as Gunn diodes, IMPATT diodes, resonant

tunnelling diodes (RTD) or frequency doublers and triplers are widely employed in the

mm-wave and ‘lower’ submm-wave regions. On the other hand, all-optical sources, namely

different types of lasers such as p-doped Ge lasers or quantum cascade lasers (QCLs) are

3PHz300THz30THz3THz300GHz30GHz frequency

1mm1cm 1 µm

1eV100meV10meV1meV energy

100nm

10eV

10 µm100 µm wavelength

THz(submm)

UVIRfar-IRmm visible

1.31.55

Figure 4.1 The electromagnetic spectrum between 30GHz and 3000THz. The term THz is commonly

applied to the submm-wave frequency range which extends from 300GHz up to about 3 THz

86 Microwave Photonics: Devices and Applications

employed in the far-IR. Among optical sources, QCLs are very promising devices for the lower

far-IR region, being currently limited in terms of their lower cut-off frequency at around a few

THz. For lower frequencies in the THz frequency range, there is almost no power source

available as can be seen from Figure 4.2, not to mention the complexity of the few sources

existing at all. Obviously, at this point the question arises as to whether optoelectronics could

provide a compact source technology not only for the THz frequency range but also for lower

frequencies in the mm-wave range. Recently, some promising high-frequency optoelectronic

or photonic solutions have been investigated and the achievements in this field of research

indicate that the photonic oscillators (PO) concept indeed has great potential for the generation

of continuous-wave THz signals with reasonable power levels. As compared to purely

electrical and optical solutions the PO concept exhibits some unique advantages such as

ultra-wide tuneability, compactness and the possibility of room-temperature operation. These

advantages have increased the interest in the PO concept but so far the implementation of

optoelectronics for THz signal generation has been delayed; this is mainly because of the

limited power level of POs and because of some technological uncertainties of this emerging

technique. However, in the course of ongoing research and developments, the power level

provided by POs is increasing while on the other hand the required power levels – for example

the power level to pump a THz mixer – are decreasing. This fact will foster the future

development and improvement of THz POs which do have the potential to become a baseline

technology for a wide variety of high-frequency applications as we will show later in this

chapter.

At first,we shall discuss the effect of optical heterodyning or photomixing in a photodetector,

which is the fundamental physical mechanism employed in a PO for generating a high-

frequency sinusoidal signal. Then, the basic constitution of a photonic oscillator for high-

frequency sinusoidal or quasi-sinusoidal signal generation is explained. Since the performance

of a PO in terms of maximum operating frequency and maximum available power is to a large

extent determined by the high-frequency photodetector being used, we will look at the various

Figure 4.2 Overview of available all-electrical and all-optical power sources versus operating

frequency; power slopes with frequency are also indicated (original data has been prepared by J.

Hesler [1])

Photonic Oscillators for THz Signal Generation 87

existing types of high-frequency photodetectors and compare them in terms of their high-

frequency performance. Next, the high-frequency performance of travelling-wave photode-

tectors (TWPDs) is investigated using an analytical model describing the vertical and

longitudinal carrier transport in a TWPD. By using this model, the major physical effects

determining the efficiency of TWPDs at high frequencies are studied. In Section 4.5, various

fabricated photonic oscillator modules will be presented, yielding either quasi-optical free

space radiation of the generated THz signals or guided transmission within a rectangular

metallic waveguide. Ultra-wideband (0.02–0.7 THz) bow-tie antenna integrated and narrow

band 0.46 THz slot antenna integrated photonic oscillators are presented in detail, as well as

rectangular waveguide coupled TWPDs that enable ultra-wideband continuous-wave signal

generation from 0.06 THz up to 1 THz.

4.3 Optical Heterodyning or Photomixing in a Photodetector

The fundamental physical principle that is used in photonic THz oscillators is to down-convert

optical infrared signals to the THz regime by employing a high-frequency photodetector. This

principle, which is also called optical heterodyning or photomixing, is illustrated in Figure 4.3.

Two phase-locked infrared waves with angular frequenciesv1 andv2 are superimposed and

injected into a high-frequency photodetector that down-converts the infrared input signal to the

THz frequency regime by generating an electrical signal at the frequencyv2-v1. To explain this

inmore detail, let us consider the relation between the generated electrical output signal and the

two superimposed optical input waves from a more physical point of view. For simplicity, we

assume that the two optical input waves are linearly polarized monochromatic plane waves in

the infrared which propagate in the þ z direction. Let

E1 ¼ E1ejðv1t� k1zþw1Þe1; ð4:1Þ

and

E2 ¼ E2ejðv2t� k2zþw2Þe2; ð4:2Þ

Photodiode

Beamsplitter

+z direction

ω2−ω1

ω2~ω1, ω2,ω1>>ω2−ω1

ω2

ω1

Figure 4.3 Principle for optical heterodyne generation employing a high-frequency photodetector. The

angular frequencies of the two optical input wavesv1 andv2 are in the IR region (e.g. 200THz), whereas

the difference frequencyv2 -v1 of the generated output signal is much lower, typically in the THz ormm-

wave region


be the complex electrical field vectors of the two opticalwaves,with field amplitudes E1 and E2,

angular frequencies v1 and v2 and wave numbers k1 and k2. The phase of each optical input

wave is considered byw1 andw2 and e1 and e2 are the unit vectors determining the orientation of

the electrical field vector of the linearly polarized optical input waves. The intensities of the

constituent waves are given by the magnitude of their Poynting vectors and are therefore

given by

I1 ¼ 1

2

ffiffiffiffiffiffiffiffiffi«r«0m0

rjE1j2; ð4:3Þ

and

I2 ¼ 1

2

ffiffiffiffiffiffiffiffiffi«r«0m0

rjE2j2: ð4:4Þ

If the two incident optical waves are perfect plane waves and have precisely the same

polarization (e1¼ e2), the resulting electrical field E0 of the optical interference signal is the

sum of the two constituent input fields and hence we can write E0¼E1 þ E2. Taking the

squared absolute value of the optical interference signal we obtain

jE0j2 ¼ jE1 þE2j2 ¼ jE1j2 þ jE2j2 þE1E2* þE1

*E2

¼ jE1j2 þ jE2j2 þ 2jE1jjE2jcosððv2 �v1Þ � t�ðw2 �w1ÞÞ: ð4:5Þ

FromEquation (4.5) and by usingEquations (4.3) and (4.4), it follows that the intensity of the

interference signal I0 is given by

I0 ¼ I1 þ I2 þ 2ffiffiffiffiffiffiffiffiI1I2

pcosððv2 �v1Þ � t�ðw2 �w1ÞÞ: ð4:6Þ

By launching this optical interference signal into a photodetector, a photocurrent i is

generated which can be expressed as

i ¼ h0 � q=hf1 �P1 þh0 � q=hf2 �P2 þ 2hfc� q=h �

ffiffiffiffiffiffiffiffiffiffiP1P2

f1f2

scosððv2 �v1Þ � t�ðw2 �w1ÞÞ; ð4:7Þ

where q is the electron charge and P1 and P2 denote the optical power levels of the two

constituent optical input waves. The photodetector’s DC and high-frequency quantum

efficiencies are represented by h0 and hf c. It is of course important to consider that the

detector’s quantum efficiency is not independent of the frequency. Several intrinsic and

extrinsic effects such as transit time limitations or microwave losses will eventually limit

the high-frequency performance of the detector and thus the detector’s DC responsivity h0 is

typically much larger than its high-frequency responsivity hf c. In our case, we can further

simplify the photocurrent equation (Equation (4.7)) by considering the fact that the two optical

input waves are close in frequency (f1� f2) whereas the difference frequency fc is by far smaller

( fc¼ f2� f1� f1, f2). As an example, if we employ a photodetector operating in the infrared at

1.55mmwavelength,wemight use optical inputwavelengths atl1¼ 1.55mm( f1¼ 193.4 THz)

and l2¼ 1.542 mm ( f2¼ 194.4 THz). In this case, the difference frequency would be exactly

fc¼ 1THz which is about 200 times smaller than the optical frequencies. A small detuning of


one input laser wavelength by only 0.8 nm (at 1.55mm) thus results in a remarkable change of

the beat frequency by about 100GHz.

If we further assume for simplicity that the power levels of the two optical input waves are

equal (Popt�P1�P2), Equation (4.7) becomes

i ¼ 2s0Popt þ 2sf cPoptcosð2pfctþDwÞ; ð4:8Þwhere

fc ¼ f2 � f1; ð4:9Þdenotes the difference frequency or beat frequency of the two constituent optical input waves

and Dw¼w2�w1. Here s0¼h0�q/hf and sf c ¼ hf c� q=hf are the photodetector’s DC and high-

frequency responsivities given in A/W.

Equation (4.8) is the fundamental equation describing optical heterodyning in a photode-

tector. The first term is the DC photocurrent generated by the constituent optical input waves

and the second term is the desired high-frequency signal oscillating at the difference frequency

fc (down-converter). From a physical point of view, it is important to note that by optical

heterodyning no signal is generated which oscillates at the sum of the two optical frequencies.

This is in contrast to nonlinear effects such as three-wave mixing in a second-order nonlinear

optical medium where not only the difference frequency but also a wave at a higher frequency

f1 þ f2 (up-converter) is generated.

According to the theoretical discussion above, a photonic oscillator for sinusoidal LO

generation thus consists of an optical heterodyne source generating the dual wavelength or dual

mode optical signal and a high-frequency photodetector. Schematically, this concept is

illustrated in Figure 4.4. Besides compact size, light weight and room temperature operation,

a large tuning range is another important advantage of the PO concept. Slightly shifting the

wavelength of one laser by only 0.8 nm results in a remarkable tuning of the beat frequency by

100GHz as discussed above. Themaximum tuning range of the PO ismainly determined by the

high-frequency responsivity of the employed photodetector and the locking range of the optical

heterodyne source. As wewill see later in this chapter, the tuning range can be quite large and a

tuning range of about 1 THz has already been achieved experimentally.

Instead of using a dual-mode optical input signal to the photodetector (PD) one can also use a

multimode optical input signal for generating a quasi-sinusoidal oscillator signal. This concept

is shown schematically in Figure 4.5. Here the optical spectrum consists not only of twomodes

but of a larger number of modes, with a constant difference frequency fc between all

phasephase lockedlockedopticaloptical heterodyneheterodyne

sourcesource

highhigh --frequencyfrequencyphotodiodephotodiode

λ

P

λ

λ λ

0λ∆

20

0 λλ∆⋅≈ cf

PhasePhase locked lockedopticaloptical heterodyneheterodyne

sourcesource

HighHigh--frequencyfrequencyphotodiodephotodiode

λ

Popt

0λ

1λ 2λ

λ∆

20 λλ∆⋅≈ cfc

Figure 4.4 Typical set-up of a photonic local oscillator consisting of a stabilized optical source

generating a phase locked optical heterodyne signal and a high-frequency photodetector generating the

electrical beat signal in the THz frequency regime


neighbouring modes. In this case, the photodetector generates a quasi-sinusoidal output signal

at the difference frequency fc. The multimode optical source employed in this approach could

be a mode-locked laser diode (MLLD) for example.

In order to generate an oscillator signal with high spectral purity and high stability the locking

of the opticalmodes is crucial.Different injection techniques for achieving opticalmode locking

have been proposed and investigated. The selection of two phase locked modes from a

multimode optical source, such as a Fabry–Perot laser, a mode-locked laser diode [4] or an

optical comb generator [5, 6] is a comparably straightforward approach. Other approaches are

based upon optical injection locking of two independent lasers using an optical phase locked

loop (PLL) configuration [7]. For verification of the spectral purity and stability of the optically

generated oscillator signal, phase and amplitude noise measurements have been performed. So

far, this has been done mainly at ‘lower’ frequencies in the microwave andmm-wave frequency

range. As an example, the phase noise of a mode-locked photonic oscillator in the microwave

region has been measured to be as low as �125dBc/Hz at 10 kHz offset from the oscillator

frequency.At higher frequencies in themm-wave regime the typical phase noise is of theorder of

�80 dBc/Hz to �95 dBc/Hz at 10 kHz offset. The phase and amplitude noise of a PO at THz

frequencies has not been thoroughly investigated yet but from initial experiments there is

evidence that the phase noise of photonic THz oscillators is comparable to the phase noise of all-

electronic Gunn oscillators and warm multiplier assemblies [8].

Further key specifications of a photonic oscillator, namely the frequency and power level of

the generated signal aswell as the possible tuning range, depend strongly on the performance of

the employed high-frequency photodetector. From Equation (4.8) we note that the output

power of the generated high-frequency oscillator signal is linearly dependent on the detector’s

responsivity at that frequency and the optical input power injected into the detector. Thus, for

generating high-power oscillator signals in the THz frequency range, the photodetector is

required to exhibit a reasonably high responsivity at THz frequencies and, secondly, it should

allow for a high optical input power. In other words, the detector should exhibit a large

saturation photocurrent.

In the next sectionwewill compare the various types of photodetectors in terms of their high-

frequency and high-saturation photocurrent performances. This general comparison will be

followed by a more detailed discussion on the high-frequency performance of so-called

travelling-wave photodetectors (TWPDs). It will be shown that TWPDs offer great potential

for high-power THz generation, that is they can be designed to exhibit a high responsivity at

high frequencies and a high-saturation photocurrent.

λ

Popt

OpticalOpticalmultimodemultimode

sourcesource

High-frequencyHigh-frequencyphotodetectorphotodetector

0λ

cfcf

Figure 4.5 Typical set-up of a photonic local oscillator consisting of a stabilized optical source for

generating an optical comb signal featuring a constant difference frequency fc between the various

longitudinal optical modes


4.4 Travelling-wave Photodetectors

The high-frequency photodetector is a key component for any photonic oscillator. The most

important requirement for this kind of application is a high responsivity at THz frequencies –

not necessarily a large 3 dB bandwidth. In the following subsection we will review recent

achievements in high-frequencyphotodetectors andwewill look at the physical effects limiting

the detector’s quantum efficiency, especially at THz frequencies. For a comprehensive review

on the bandwidth–efficiency product of photodetectors the interested reader is referred to

Kato [9].

The physical effects determining the detector’s high-frequency performances are usually

represented by time constants describing the dynamics of the photogenerated carriers. Most

important are the transit time and the carrier lifetime. Generally speaking, these two time

constants represent the average time it takes a carrier to travel between the electrodes and the

average time a free carrier exists before it finally recombines. Further, conventional lumped

photodetectors are limited by some internal and external RC time constants. At very high

frequencies ( > 100GHz) electrical wave propagation effects must also be considered since the

electrical wavelength is getting close to the device dimensions and further on microwave

propagation losses become more significant.

Looking at the various types of photodetectors, the conventional vertically illuminated

photodetectors such as the p–n or the p–i–n photodetector introduce a trade-off between

quantum efficiency or responsivity and high-frequency or bandwidth. In conventional photo-

detectors, light is coupled in through the upper layers of the device and is absorbed as it travels

through the structure. This fundamental absorption process generates electron–hole pairs

travelling under the influence of the applied electric field to the device contacts thus producing a

photocurrent. The frequency response of such conventional vertical photodetectors depends on

the transit time taken for the photo-induced carriers to reach the contacts. For short transit times

the absorbing layer of a conventional pn-PD or pin-PD needs to be thin; this, on the other hand,

results in a low efficiency and a large capacitance and thus the RC time constant is extremely

large preventing the conventional vertical PD from being used at very high frequencies.

An approach to reduce the RC time constant of the conventional PDs is to utilize very narrow

and closely spaced contact fingers which is usually done in metal–semiconductor–metal

(MSM) PDs. Obviously, due to the closely spaced contact fingers, the transit time is short and

also the device capacitance is small due to the small surface area. The short transit time and the

small capacitance together allow wide-bandwidth operation but the quantum efficiency of a

vertically illuminated MSM-PD is comparably small for reasonable applications.

A promising approach for achieving a high-efficiency is to utilize the uni-travelling carrier

(UTC) concept. In the UTC-PD the absorbing layer is p-doped and therefore only electrons

travel across the depletion layer, with holes disappearing quickly since they are majority

charge carriers. Due to the much higher drift velocity of electrons and due to the fact that

electrons can travel at overshoot velocity, the space charge effect is significantly reduced.

This principle allows high-saturation currents or higher optical input power. The bandwidth

of a UTC-PD is restricted by transit time effects mainly determined by the diffusion time of

electrons travelling in the p-doped absorbing layer and of course it is also limited by the RC

time constant. Recently, high-bandwidth UTC-PDs have been demonstrated and in terms of

the bandwidth–efficiency product, UTC-PDs exhibit much better performances than con-

ventional pin-diodes [9].


Tocircumventthetransit timelimitationonecanalsoemployamaterialwithaveryshortcarrier

lifetime. Besides the RC time constant the transit time determines the detector’s frequency

response provided that the carrier lifetime ismuch shorter than the transit time. For several years,

researchhasbeencarriedout togrowGaAslayersat lowtemperatures.Duetothelowtemperature

the grown GaAs layers contain numerous impurities which capture the free carriers and thus

reduce the carrier lifetime. Recently, low-temperature grown GaAs operating at around 850 nm

wavelength have been utilized for high-frequency signal generation in the THz range [10].

Edge-coupled or waveguide photodetectors can overcome some of the limitations discussed

above by allowing the optical signal to enter through the edge of the device. Thus the electrical

carrier transport is perpendicular to the propagation of light. In principle, this approach allows

for a long but narrow absorption layer having both a low transit time and a high efficiency. On

the other hand, the detector’s RC time constant becomes significantly large due to the long

absorbing layer, leading to the well-known lumped element RC time limitation.

All approaches discussed above lead to lumped elements since the detector’s responsivities

are determined by the total dimensions of the devices. These lumped element photodetectors,

however, can no longer cope with the requirements as the frequencies extend into the THz

regime. To overcome the RC time limitation so-called distributed or travelling-wave photo-

detectors (TWPDs) have been investigated since the early 1990s. In a TWPD the photo-

absorption process occurs in a distributed manner along the length of the device such that it

contributes to the overall electrical signal in the contact transmission line. Thus, travelling-

wave pin–PD (TWPDs), being two-port devices, are not limited by the RC time constant since

electrically the devices are not lumped elements with a concentrated capacitance but an

electrical waveguide (typically microstrip or coplanar) with a given characteristic impedance.

The bandwidth of a TWPD is therefore mainly determined by its transit time. Since it is not RC

time limited the travelling-wave concept can accomplish quite thin absorbing layers leading to

short transit times and high bandwidths. A major drawback is that the electrical characteristic

impedance of a TWPD needs to be matched to the external circuitry and, secondly, the optical

group velocity needs to match with the electrical phase velocity for achieving the highest

efficiencies. Although TWPDs exhibit a couple of challenges that need to be solved, they do

offer a great advantage for high-power THz signal generation, namely the prospect of

independently optimizing its quantum efficiency and saturation photocurrent. For this reason

various TWPD structures have been investigated by different research groups in the past. These

include p–i–n [11, 12], MSM [13], Schottky [14] and photo-transistor configurations [15], all

showing excellent performance at high frequencies. In the following subsectionwewill discuss

the various physical effects determining the high-frequency performance of a p–i–n TWPD in

more detail using a theoretical model that describes the vertical and longitudinal transport of

the photogenerated electrical carriers in a TWPD.

4.4.1 Drift-diffusion Model for p–i–n Photodetectors

A sketch of a high-speed travelling-wave photodetector (TWPD) employing a pin waveguide

structure is shown in Figure 4.6. The detector consists of an optical waveguide and an electrical

transmission line. The optical heterodyne signal launched into the optical strip loaded

waveguide is gradually absorbed resulting in a distributed current generation along the

detector’s length which contributes to the overall current propagating along the electrical

transmission line of the TWPD.


As can be seen from Figure 4.6, the p–i–nwaveguide structure of the detector consists of an

intrinsic region sandwiched between p- and n-doped semiconductor layers. In order to enable

optical waveguiding, the intrinsic core region needs to have a larger refractive index, that is a

smaller bandgap energy, compared with the p- and n-doped cladding layers. Further on, the

intrinsic region often consists of multiple layers instead of a single absorption layer. This is

mainly to reduce the modal absorption per unit length in order to allow for a long absorption

length. In our model we thus assume a waveguide core consisting of an intrinsic photon

absorbing layer with a thickness d0, and two adjacent nonabsorbing layers with thicknesses dipand din as illustrated in Figure 4.6.

In general, the photogenerated carrier transport in the p–i–n waveguide structure can be

described by the continuity equation. By neglecting the transversal carrier transport and

assuming harmonic time dependence, the one-dimensional (1D) complex continuity equations

for the electron and hole densities n(x) and p(x) are given by

jv � nðxÞ ¼ Dn � d2

dx2nðxÞ� d

dxnðxÞ � vnðxÞ½ � �Rc þGc; ð4:10Þ

jv � pðxÞ ¼ Dp � d2

dx2pðxÞ� d

dxpðxÞ � vpðxÞ

� ��Rc þGc: ð4:11Þ

The first term in the continuity equations describes the carrier diffusionwith the electron and

hole diffusion constants represented by Dn and Dp. These diffusion constants are functions of

carrier mobility and electric field and it has been shown in [10] that the carrier diffusion

constants become very small in value for high electric fields in excess of 20 kV/cm. Assuming

Figure 4.6 The left-hand figure schematically shows a waveguide photodetector with a coplanar

metallization. On the right, a typical cross section of a detector’s layer structure is shown. It consists of an

intrinsic core regionwith an absorbing (Wg < hf ) and two adjacent non-absorbing (Wg > hf ) waveguiding

layers


that the intrinsic layer(s) thickness of a high-frequency p–i–n detector iswell below500 nmand

also that a reverse voltage of a few volts is applied, high electric fields well above 20 kV/cm

occur. Thus, we can neglect carrier diffusion in the intrinsic region of the reverse biased

photodetector. The carrier recombination rate represented by Rc can also be neglected for our

purposes, since the carrier lifetime is in the order of a fewnanoseconds [11]which is about three

orders of magnitude larger than the average transit time of the investigated structure. Thus the

continuity equations (4.10) and (4.11) can be simplified to

jv � nðxÞ ¼ 1

q� d

dxðJnÞþGn; ð4:12Þ

jv � pðxÞ ¼ � 1

q� d

dxðJpÞþGp: ð4:13Þ

The first term in Equations (4.12) and (4.13) describes the carrier drift in the presence of an

electric field with vn(x) and vp(x) representing the electron and hole velocity, respectively. The

carrier generation rate is a function of the optical intensity at any point along the detector. We

can therefore infer that the generation rates of holes and electrons are equal and can be

expressed as

G ¼ Gn ¼ Gp ¼ G0 � expð� gopt � zÞ ð4:14Þ

with

G0 ¼ h � lh � c �aopt � Iopt: ð4:15Þ

Here h denotes the external quantum efficiency, gopt is the complex propagation constant of

the optical heterodyne input signal and Iopt represents the optical intensity. It should be noted

that the carrier generation rate is a function of the longitudinal coordinate z due to the optical

wave propagation determined by the complex heterodyne optical propagation constant gopt.Since diffusion can be neglected as discussed above, the electron and hole current densities

in the absorbing layer are given by

JnðxÞ ¼ � q � vnðxÞ � nðxÞ; ð4:16Þ

JpðxÞ ¼ � q � vpðxÞ � pðxÞ: ð4:17Þ

Here we assume that electrons and holes in the intrinsic region travel at constant saturation

velocities vn and vp, an assumption which is appropriate for reverse biased photodetectors with

strong internal electric fields. Introducing Equations (4.14) – (4.17) into Equations (4.12)

and (4.13) we derive two first-order differential equations with constant coefficients describing

the carrier densities in the absorbing layer:

dJn

dx¼ � jv

vn� Jn � q �G; ð4:18Þ


dJp

dx¼ þ jv

vp� Jp þ q �Gp: ð4:19Þ

To solve the above differential equations wemake use of the fact that electrons only travel in

the positive x-direction and thus the electron carrier density at x¼ 0 needs to be equal to zero.

Similarly, we can state that the hole current density is equal to zero at x¼ d0:

Jnðx ¼ 0Þ ¼ 0: ð4:20Þ

Jpðx ¼ d0Þ ¼ 0 ð4:21ÞByusing the two boundary values inEquations (4.20) and (4.21)we can solve the differential

Equations (4.18) and (4.19) and derive the electron and hole current densities in the absorbing

layer

Jn ¼ q �G � vnjv

e� jv � xvn � 1

� � ð4:22Þ

Jp ¼ q �G � vpjv

e� jv � d0 � x

vp � 1

� �ð4:23Þ

In a similar way, we can now proceed to find the current densities in the adjacent

nonabsorbing intrinsic layers. For simplicity, we only consider the hole current density in

the following derivation; the procedure for deriving the electron current density is similar. Due

to the fact that there is no photon absorption in these two layers the continuity equation for

holes – Equation (4.11) – becomes quite simple.

jv � p ¼ � 1

q� d

dxðJpÞ: ð4:24Þ

This leads to the following simple differential equation for the carrier density

d

dxðJpÞ ¼ jv

vp� Jp: ð4:25Þ

The required boundary values to solve Equation (4.25) are given by the consistency of the

current density at the boundary. From Equation (4.23) it follows that

Jpðx ¼ 0Þ ¼ q �G � vpjv

� e� jv � d0

vp � 1

� �ð4:26Þ

and by using Equation (4.26) we derive the hole current density in the nonabsorbing intrinsic

layer

Jp ¼ q �G � vnjv

� e� jv � d0vn � 1

h i� e� jv � x� d0

vn : ð4:27Þ


i0ðzÞ ¼ w

din þ dip þ d0� q �G0 � expð� gopt � zÞ

jv

�

vn � vn

jv� d0 � vn

jv� exp � jv � d0

vn

� ��

þ vp � vp

jv� d0 � vp

jv� exp � jv � d0

vp

� ��

þ vn � exp � jv � d0vn

� �� 1

� �� vn

jv� vn

jv� exp � jv � din

vn

� ��

þ vp � exp � jv � d0vp

� �� 1

� �� vp

jv� vp

jv� exp � jv � dip

vp

� ��

26666666666666664

37777777777777775

ð4:28Þ

The total current density is given by superimposing the electron and hole current densities in

the absorbing region (Equations (4.22) and (4.23)) and the electron and hole current

densities of the nonabsorbing regions (see Equation (4.27)). If we further proceed by

integrating the total current density along y, we gain the total photogenerated current per unit

length generated at any point z along the detector. This equation not only comprises the

carrier transport and generation within the intrinsic absorptive layer of the photodetector but

also the carrier transport through the adjacent non-absorptive intrinsic regions of the

waveguide core.

4.4.2 Transmission Line Model for Travelling-wave Photodetectors

In this section we will develop a transmission line model describing the contribution of the

distributed current source found above to the overall electrical wave propagating along the

electrical transmission line of the travelling-wave photodetector. The type of transmission line

formed in this TWPD is a slow-wave hybrid coplanar/microstrip waveguide [16]. Generally,

such transmission lines require ‘full-wave’ analysis for rigorous modelling. However, for our

purposes the quasi-TEM analysis using a quasistatic equivalent circuit model as shown in

Figure 4.7 satisfactorily describes the high-frequency properties of the detector’s transmission

line. Here, the photogenerated current per unit length is represented by the distributed current

source i0(z). R0 and L0 are the resistance and the inductance of the metal centre conductor per

unit length, respectively. R0S represents the semiconductor losses associated with transverse

current flow in the doped cladding layers and C0i and G0

i are the capacitance and the

conductance of the intrinsic core layer per unit length. For high frequencies in the THz

regimewe also need to consider the capacitance of the doped semiconductor layersC0S and the

outer air capacitance C00.

For further considerations employing a transmission line model, it is advantageous to

separate the active current source from the passive impedances. This is achieved by transform-

ing the distributed current source i0(z) into a form which is in parallel to all other passive

impedances of the equivalent circuit as shown in Figure 4.8. The former current source is

transferred to a distributed current source i00 which is in parallel with a completely passive

electrical transmission line of unit length represented by its characteristic impedance Z0 and


propagation constant gel. For the transformed distributed current source i00ðzÞ we obtain

i00ðzÞ ¼ i0ðzÞ � Z 0iðvÞ

Z 0iðvÞþ Z 0

SðvÞ : ð4:29Þ

The impedances shown in Figure 4.8 are not affected by this transformation and thus remain

unchanged. Therefore we can state that the characteristic impedance and the electrical wave

propagation constant are given by

Z0 ¼ffiffiffiffiffiffiffiZ 0

H

Y 0V

r; ð4:30Þ

gel ¼ffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiffiZ 0

H � Y 0V

p: ð4:31Þ

Now, we can proceed and define a transmission line model for the travelling-wave

photodetector which is shown in Figure 4.9. Here Ze represents the electrical impedance at

the input port of the detector’s transmission line and Za is the load impedance at the output port.

In order to calculate the total power delivered by the TWPD to the load impedance Za, we use a

superposition approach. First, we determine the photocurrent delivered by each current source

independently (the other sources are considered as open) and then we calculate the total

SSll

SSll

R ll LLll

ii ll (z(z ))

SSll

iill

00ll

iill

ZZ

ZZ

ZZ

HHll

YYVVll

iill

SSll

SSll

R

RR CCCC

CCGG

ll ll

))

SSll

iill

00ll

iill

HHll

YYVVll

iill

Figure 4.7 Equivalent circuit representing a unit-section of a travelling-wave photodetector

Z0,γel

dz

i´0(z)

Figure 4.8 Equivalent circuit representing a unit section of a TWPDwith a transformed current source

in parallel with a passive transmission line


photocurrent by integrating the contributions of all current sources. The photocurrent,

delivered to the load impedance Za by the current source i00ðzÞ located at z¼ zs, can be

calculated using the equivalent circuit shown in Figure 4.10 where all other current sources are

considered as open.

The electrical input reflection coefficient re and the output reflection coefficient ra are

respectively:

re ¼ ZE � Z0

ZE þ Z0; ð4:32Þ

ra ¼ Za � Z0

Za þ Z0: ð4:33Þ

Using these two reflection coefficients, we can further simplify the equivalent circuit as

shown in Figure 4.11, by transforming the input and the output impedances to the location of

the photocurrent source at z¼ zs.

The resulting transformed impedances are then given by

ZTe ¼ Z0 � 1þ re � e� 2g � zS

1� re � e� 2g � zS ; ð4:34Þ

dz dz dz

Ze Zai´0(0) i´0(Z) i´0(L)

Z0, γ Z0, γ Z0, γ Z0, γ

dz

IZa

....

.... ....

....

Figure 4.9 Transmission line model for a TWPD with input impedance Ze and load impedance Za

Figure 4.10 Equivalent circuit for the TWPD considering only the current source per unit length at the

position z¼ zs


ZTa ¼ Z0 � 1þ ra � e� 2g � ðL-zSÞ

1� ra � e� 2g � ðL-zSÞ ; ð4:35Þ

and by using these equations, we can determine the forward propagating photocurrent

generated by the current source at z¼ zs at the position of the load impedance ZS. This

photocurrent contribution is called i0ha and is given by

i0haðzSÞ ¼i00ðzSÞ � ð1þ ree

� 2g � zSÞ � ð1� rae� 2g � ðL� zSÞÞ � e� g � ðL� zSÞ

ð1þ ree� 2g � zSÞ � ð1� rae� 2g � ðL� zSÞÞ þ ð1� ree� 2g � zSÞ � ð1þ rae� 2g � ðL� zSÞÞ :

ð4:36ÞNow,we can determine the total photocurrent travelling to the load impedance by integrating

all contributions along the detector’s length

ihaðz ¼ LÞ ¼ðL

z¼0

i0haðzSÞ � dzS ; ð4:37Þ

and by considering the output reflection coefficient at the end of the detector’s transmission line

we can finally determine the total photocurrent at the load impedance and thus the total

electrical power delivered to the load impedance.

In order to numerically calculate the generated power the circuit parameters of the TWPD

equivalent circuit must be determined first. Generally, these parameters are frequency

dependent but for frequencies in excess of 20GHz the parameters L0, R0S, C

0i and G0

i are

considered to be constant. Only R0 increases with the square root of frequency due to the skineffect. As a good approximation C0

i can be determined by C0i ¼ «0«r�w/di. The constant

conductor resistance per unit length at frequencies below 10GHz is given byR0 ¼ rAu/(w�dmet).

For frequencies in excess of 10GHz R0 is considered to increase with the square root of the

frequency. The series resistance of the doped semiconductor layers for a 6mm wide rib

waveguide with 15mm separation between the centre and the ground electrode is typically of

the order of R0S � 0.25Wmm. The parallel conductance G0

i and the transversal inductance L0

can be determined from experimental S-parameter measurements. With all the equivalent

circuit parameters known, the complex characteristic impedance Z0 and the complex electrical

propagation constant gel of the TWPD transmission line can be calculated.

ZaTZe

T

i´0(zs)i´h(zs)

Figure 4.11 Equivalent circuit for the TWPD considering only the current source per unit length at the

position z¼ zs. Both the input and output impedances are transformed along the detector’s transmission

line to the location of the current source


Of course, in the above models we assumed some boundary conditions simplifying the real

physical effects inside the detector, leaving enough space for future optimizations or extensions

of the presented approach. Nevertheless, by using the frequency domain model developed here

we can study the most relevant intrinsic effects such as transit time limitations as well as

propagation effects such as microwave losses and the mismatch between the optical group

velocity and the electrical phase velocity. Furthermore, external effects such as the influence of

an impedance mismatch between the detector’s transmission line and the load impedance can

be investigated in detail and the contribution of the different effects on the total roll-off at high

frequencies can be identified. Therefore, although some simplifications were made, the model

yields a very good simulation as can be seen from Figure 4.12.

There is good agreement betweenmeasured and simulated datawith amaximumvariation of

about a few dB. This proves the accuracy and reliability of the analytical model. The total roll-

off of 50 dB for the full frequency span from DC to 1THz is due to the transit time effects and

intrinsic effects arising from carrier transport in the doped sections of the TWPD. In addition,

propagation effects such asmicrowave losses and velocitymismatch contribute to the total roll-

off. At frequencies in excess of 0.1 THz we found that the delivered power decreases with

frequency by about f-�3 for the investigated devices.

4.5 Terahertz Photonic Oscillators

The first promising photonic oscillators for continuous-wave THz generation were demonstrat-

ed in the mid-1990s, using LT-GaAs photodetectors [17]. State-of-the-art LT-GaAs photo-

detectors utilize a vertically illuminated MSM-PD on LT-GaAs either coupled to a log-spiral

antenna for wideband or coupled to a resonant dipole antenna for narrowband operation. Even

LT-GaAs based MSM-TWPDs have been developed and successfully employed for quasi

continuous-wave narrowband THz generation [18]. A major disadvantage associated with

LT-GaAs is the thermal failure due to the high thermal resistance of the LT-GaAs which limits

the maximum current density. Furthermore, the optimum optical wavelength for LT-GaAs

Figure 4.12 Simulated frequency response of a 50mm long and 6mm wide TWPD. Dots indicate

measured output power levels of the same device up to 110GHz


photodetetors is around 800 nm and thus advanced lasers and amplifiers developed for the

communication industry operating in the optical C-band cannot be employed.

The employment of photonic oscillators for continuous-wave THz signal generation using

1.55mmlasers is a relatively new approach and there have only been a few experimental results.

In the following, we will summarize the achieved results and we will explicitly describe the

constitution of compact photonic THz oscillators employing 1.55 mm photodetectors. The

presented photonic oscillators either yield free-space coupling by using resonant slot-antenna

structures for narrowbandor bow-tie antenna forwideband operation.Guidedwave coupling of

the generated oscillator signal using WR10 and smaller rectangular waveguides has also been

achieved.

In the photonic oscillators presented in the following paragraphs, a dual-mode optical

heterodyne input signal was used that was generated by two free-running and tuneable 1.55 mmDFB lasers. A subsequent erbium-doped amplifier (EDFA) was used for boosting the optical

power level. This concept allows one to easily sweep the frequency of the generated signal from

DC to THz frequencies, which is especially important for investigating wideband perfor-

mances of the POs.

For detecting the generated high-frequency oscillator signals in the millimetre-wave region

up to 220GHz, external single-diode harmonicmixers have been used.At higher frequencies in

the THz regime a number of detectors exist, including liquid helium cooled bolometers with a

typical noise equivalent power (NEP) of about 2 pW/Hz�1/2 or Golay cells with a typical NEP

of 100–200 pW/Hz�1/2. Although the Golay cell does not provide such a good NEP it does not

require any expenditure for liquid helium cooling since it operates at room temperature.

Therefore it is well suited for the experimental characterization of high-power POs. A typical

experimental set-up consisting of the photonic THz oscillator and a Golay cell as THz detector

is shown schematically in Figure 4.13.

4.5.1 Wideband Photonic Oscillators Employing Waveguide coupledTHz Transmitter

For specific applications, guided transmission in a rectangular waveguide ismore desirable than

free-space radiation of thegenerated THz signals and, consequently, efficient optical heterodyne

generation of guided THz waves has already been studied and demonstrated in [19–22], [25]

and [27] up to about 600GHz using WR10 waveguide integrated high-speed photodetectors.

tuneableLD

DFB-LD1560 nm

Polarizationcontrol

EDFA Photonictransmitter

Golaycell

λ0+ ∆λ

fRFc0

λ0∆λλ2

0

Figure 4.13 Photonic THz oscillator consisting of two tuneable 1.55mm DFB lasers, an EDFA and a

photonic transmitter employing a TWPD. The generated THz oscillator signal is detected using a Golay

cell


The following experiment, performed at the Universit€at Duisburg-Essen, Germany, demon-

strates ultra-wideband guided transmission up to1THzemploying a high-speed1.55mmTWPD

coupled to different rectangular waveguides (WR10, WR8 and WR5) [23]. For experimental

characterization the fabricated TWPDs have been connected using commercial coplanar to

waveguide transitions. The power levels of the generated oscillator signals have been measured

using a Golay cell as described in Figure 4.13 by quasi-optical coupling the THz power from the

waveguide into the Golay cell. The measured THz power level is shown in Figure 4.14 for an

ultra-wide frequency range up to about 1 THz. It should be pointed out that the lower cut-off

frequency at around 70GHz is given by the lower cut-off frequency of theWR10waveguide not

by the TWPD employed in the PO which can operate even at DC.

As can be seen in Figure 4.14, the maximum power level of about 100mW is achieved at

frequencies around 0.1 THz within the W-band. Here, the TWPD with an intrinsic region

thickness of d0¼ 100 nm generates about 5 dB more power than the TWPD with the 350 nm

thick intrinsic region which is due to the lower transit time penalty. It can further be observed,

from Figure 4.14, that the power decreases with frequency to the power of four. Similar results

were found by Huggard et al. in [20]. In their work, they fully-packaged a commercial

waveguide PD chip into a compact transmitter module with a WR10 waveguide output

(Figure 4.15). The measured frequency response as shown in Figure 4.16 reveals a similar

frequency dependence of the generated power level as shown in Figure 4.13with a lower cut-off

at 70GHz due to the WR10 waveguide. The maximum power level is about 100mW.

To investigate the power dependence on frequency further, the smaller waveguides (WR8

and WR5) were used since those waveguides exhibit significantly fewer modes that can

propagate at frequencies above 100GHz. Figure 4.17 shows the generated power level using

the same TWPD coupled to aWR8 and aWR5 waveguide. In addition, the cut-off frequencies

1000100Frequency (GHz)

-70

-60

-50

-40

-30

-20

-10

0

10

Gen

erat

ed (

sub

)mm

-wav

e p

ow

er (

dB

m) TW-PD/WR10

di=100nm

di=350nm~f~f -4-4

Figure 4.14 Ultra-wideband power generation employing rectangular-waveguide (WR10) coupled

TWPDs with different intrinsic region thicknesses of 100 and 350 nm


Figure 4.15 Fully packaged transmitter module developed by P. Huggard et al. at the Rutherford

Appleton Laboratory, UK, in cooperationwith theUniversity ofKent andNRAO,USA.Reproduced from

[20] by permission of Peter Huggard (� 2002 IEEE)

10-8

10-7

10-6

10-5

10-4

10-3

1000100

Det

ecte

d po

wer

(W

)

Frequency (GHz)

30 mW Optical powerestimate

10 mW Optical power

Figure 4.16 Ultra-wideband power generation employing a WR10 coupled waveguide photodetector.

The experiment has been carried out by P. Huggard at the Rutherford Appleton Laboratory, UK, in

cooperation with the University of Kent and NRAO, USA. Reproduced from [20] by permission of Peter

Huggard (� 2002 IEEE)


of the higher-order modes that can propagate in the twowaveguides are indicated by arrows in

Figure 4.17. A step-like response is observed with an almost flat response around the cut-off

frequencies of the higher-order modes. At lower frequencies the TWPD generates higher

power levels when coupled to a WR8 waveguide but at frequencies above 220GHz the power

level is about four times larger when the TWPD is coupled to a WR5 waveguide.

4.5.2 Wideband Photonic Oscillators Employing Broadband Antenna-coupled THz Transmitter

Although resonant type antennas exhibit a reasonably large bandwidth (e.g. to cover a single

astronomical band [12] in the ALMA telescope) it is also of great interest to develop an ultra-

wideband photonic transmitter that could eventually be employed not just for a single band but

for a number of astronomical bands or for spectroscopic THz imaging applications. For

developing an ultra-wideband photonic oscillator the high-speed TWPD in the transmitter part

of the PO needs to be integrated with a wideband antenna structure which exhibits a fairly

constant impedance within a large frequency range that can be matched to the detector’s

impedance. As an example, TWPDs have been integrated with bow-tie antenna structures as

shown in Figure 4.18 [26]. The inset shows a photograph of a fabricated chip. The length and

width of the TWPD and the opening angle of the bow-tie antenna are 116mm, 3.2mm and

u¼ 9.4�, respectively. The chip was also mounted on a hemispherical silica lens for improving

free-space coupling efficiency and in order to focus the generated THz oscillator signal. The

packagedmodules were investigated using the experimental set-up shown in Figure 4.13. Here

the generated powerwas quasi-optically radiated into theGolay cell without using any imaging

optics. Figure 4.19 shows themeasured THz power received by the Golay cell. Coupling losses

associated with the quasi-optical radiation into the Golay cell have not been excluded from the

measured results. As can be seen from Figure 4.19, the maximum power received by the Golay

cell is about 0.5 mW for a photocurrent of about 6mA. The generated power level is fairly flat

within a frequency range from 20GHz to 0.1 THz. Above 0.1 THz we observed that the power

Figure 4.17 Ultra-wideband (sub)mm-wavepowergenerationemployingaWR8andaWR5rectangular-

waveguide coupled TWPD


level approximately decreases with frequency to a power of three which is in accordance with

the simulations. Similar frequency dependence of awideband antenna integrated photodetector

was also found in [24].

4.5.3 Narrowband Photonic Oscillators Employing a Slot Antenna CoupledTHz Transmitter

To investigate the performances of a narrowband photonic oscillator and to demonstrate their

feasibility to pump the SIS junction of an astronomical receiver, a photonic 0.46 THz

Figure 4.18 Schematic of a TWPD monolithically integrated with a planar ultra-wideband bow-tie

antenna. The inset shows an SEM picture of a fabricated transmitter chip

Figure 4.19 Ultra-wideband (sub)mm-wave power generation employing a bow-tie antenna integrated

TWPD


transmitter has been fabricated. A sketch of the developed transmitter chip is shown in

Figure 4.20(a). In the transmitter, the TWPD was monolithically integrated with a planar full-

wave slot antenna resonant at 460GHz. A passive bias-T was also integrated on-chip

employing radial stubs as low-pass filters to allow for external DC-bias supply to the PD.

The SEM pictures in Figure 4.20(b) and Figure 4.20(c) show a top view of a transmitter chip

array and a single transmitter chip, respectively. The inset in Figure 4.20(c) shows an enlarged

view of thewaveguide PD (top) and the radial stub low-pass filter (bottom). The covered single

slot antenna, which cannot be seen in Figure 4.20, is located at the intersection between the

waveguide PD and the radial-stub filter and it is horizontally oriented to the optical waveguide

of the PD. The overall dimensions of a single transmitter chip are about 2.3� 1.7mm, and

about 300 transmitters have been fabricated from a single 2 inch InP substrate. The transmitter

chip was further mounted on a hemispherical silica lens with a diameter of 10mm as sketched

in Figure 4.20(d). The silica lens couples the antenna to free space, producing a near Gaussian

submm-wave beam, which can be re-imaged on any receiver optics (lens and horn). Finally, the

lens with the transmitter chip was packaged as can be seen from Figures 4.20(e) and 4.20(f).

To demonstrate the capabilities of the packaged THz transmitter module to pump an

astronomical receiver with an SIS junction an experiment has been undertaken using the

Figure 4.20 (a) Schematic of a photonic emitter consisting of a slot antenna integrated TWPD;

(b) photograph of a fabricated array of 0.46 THz transmitter; (c) SEM picture of a single emitter;

(d) schematic of an emitter mounted in the centre of a silicon ball lens; (e) and (f) photographs of the

fabricated modules


astronomical receiver as a mixer for down-converting the 0.46 THz signal generated by the

photonic oscillator. In the experiment, all receiver components are operated at liquid helium

temperature. At first, a 460GHz solid-state oscillator chain consisting of aGunn oscillator with

a subsequent tripler was used to pump the SIS junction of the receiver. The output power of the

solid-state oscillator was adjusted for optimum sensitivity (i.e. lowest noise temperature) of the

SIS junction and the correspondingDCbias curve of the SIS junctionwas recorded (dark line in

Figure 4.21). Hereafter, the solid-state oscillator signal was replaced by the optically generated

LO signal from the photonic oscillator module. Different DC bias curves of the SIS junction

Figure 4.21 DC current–voltage curves derived from the SIS junction of a 460GHz astronomical

receiver (photo) which was either pumped by a Gunn oscillator (black line) or by using the developed

photonic 0.46 THz oscillator at different photocurrent levels (grey curves)

Figure 4.22 THz power received by the SIS junction of a receiver with respect to the detector’s

photocurrent


were recorded as a function of laser input power level, that is as a function of the detector’s

photocurrent (grey lines in Figure 4.21). The inset in Figure 4.21 shows a photo of the employed

liquid helium cooled receiver which was used. As can be seen from Figure 4.21, at a

photocurrent of about 20mA the power generated by the PO is equivalent to the power

generated by the solid-state LO. Thus, the developed photonic transmitter is capable of

pumping the SIS junction of the receiver under optimum conditions. The total THz power

generated by the photonic oscillator module is shown in Figure 4.22 as a function of the

photocurrent in the TWPD. The total power generated by the TWPD follows the square-law

principle, as can be seen from Figure 4.23. No saturation effects are observed for photocurrents

up to 20mA.

References

[1] B. Leone et al., “Optical Far-IRwave Generation - An ESA review study”, Proceedings of the 14th International

Symposium on Space Terahertz Technology, Tucson, USA, April 2003.

[2] See for example “The Atacama Large Millimeter Array”, The ESO Messenger, no. 107, March 2002.

[3] P.H. Siegel, “Terahertz Technolgy”, IEEE Trans. On Microwave Theory and Techn., vol. 50, no. 3, March 2002.

[4] A. Hirata, M. Harada and T. Nagatsuma, “120-GHz Wireless Link Using Photonic Techniques for Generation,

Modulation, and Emission ofMillimeter-Wave Signals”, IEEE J. of Lightwave Technol., vol. 21, no. 10, October

2002.

[5] T. Yamamoto, H. Takara and S. Kawanishi, “Generation and Transmission of Tuneable Terahertz Optical Clock”,

International Topical Meeting on Microwave Photonics, Awaji Island, Japan, T2-2, pp. 97–100, Nov. 2002.

[6] P. Shen and P.A. Davies, “Millimetre Wave Generation Using an Optical Comb Generator with Optical Pase-

Locked Loops”, International Topical Meeting on Microwave Photonics, Awaji Island, Japan, T2-3, Nov. 2002.

[7] See for example L.A. Johansson and A.J. Seeds, “Millimeter-Wave Modulated Optical Signal Generation with

High Spectral Purity and Wide-Locking Bandwidth Using a Fiber-Integrated Optical Injection Phase-Lock

Loop”, IEEE Photon. Technol. Lett., vol. 12, no. 6, June 2000.

[8] M. Ishiguro et al., “A hybrid Option for the First LOs using Direct Photonic LO Driver”, ALMA memo 435,

September 2002.

[9] K. Kato, “Ultrawide-Band/High-Frequency Photodetectors”, IEEE Trans. On Microwave Theory and Techni-

ques, vol. 47, no. 7, July 1999.

-35

-30

-25

-20

-15

-10

-5

0

001011

Photocurrent (mA)

Gen

erat

ed m

m-w

ave

pow

er (

dB) fc = 100GHz Uncooled

Cooled T = -14°C

0

01

Figure 4.23 Generated output power as a function of DC-photocurrent


[10] See for example E.R. Brown, “THz Generation by Photomixing in Ultrafast Photoconductors”, Int. J. of High

Speed Electronics and Systems, vol. 13, no. 2, 2003.

[11] A. St€ohr, R. Heinzelmann, A. Malcoci and D. J€ager, “Optical Heterodyne Millimeter.Wave Generation Using

1.55mm Travelling-Wave Photodetectors“, IEEE Trans. on Microwave Theory and Techn., vol. 49, no. 10,

October 2001.

[12] V. Hietala, G.A. Vawter, T.M. Brennan and B.E. Hammons, “Traveling-Wave Photodetectors for High-Power,

Large-Bandwidth Applications”, IEEE Trans. onMicrowave Theory and Techn., vol. 43, no. 9, September 1995.

[13] J.-W. Shi, Y.-H. Chen, K.-G. Gan, Y.-J. Chiu, C.-K. Sun and J.E. Bowers, “High-Speed and High-Power

Performances of LT-GaAs Based Metal-Semiconductor-Metal Traveling-Wave Photodetectors in 1.3-mmWavelength Regime”, IEEE Photon. Technol. Lett., vol. 14, no. 3, March 2002.

[14] M. Alles, U. Auer, F.-J. Teude and D. J€ager, “Distributed velocity matched 1.55mm InP traveling-wave

photodetector for generation of high millimeterwave signal power,” IEEE Int. Microwave Symposium, MTT-S

Digest, Baltimore, USA, pp. 1233–1236, 1998.

[15] D.C. Scott, D.P. Prakash, H. Erlig, M.A. Bhattacharya and H.R. Fetterman, “High Power, High Frequency

Traveling Wave Heterojunction Phototransistors with Integrtaed Polyimide Waveguide”, IEEE Int. Microwave

Symposium, MTT-S Digest, Baltimore, USA, pp. 1237–1240, 1998.

[16] D. J€ager, “Slow-Wave Propagation Along Variable Schottky-Contact Microstrip Line,” IEEE Trans. on

Microwave Theory and Techn., vol. 24, no. 9, September 1976.

[17] E.R. Brown, K.A. McIntosh, K.B. Nichols and C.L. Dennis, “Photomixing up to 3.8THz in Low-Temperature

Grown GaAs”, Appl. Phys. Lett., vol. 66 (3), pp. 285–287, 1995.

[18] S. Matsuura and G.A. Blake, “A travelling-wave THz Photomixer based on Angle-Tuned Phase Matching”,

J. Appl. Phys. Lett., vol. 74, no. 19, pp. 2872–2874, May 1999.

[19] A. St€ohr, R. Heinzelmann, C. Kaczmarek and D. J€ager, “Ultra-Broadband Ka to W-band 1.55mm Travelling-

Wave Photomixer,” Electron. Lett., vol. 36, no. 11, 970–972, May 2000.

[20] P.G.Huggard, B.N. Ellision, P. Shen,N.J. Gomes, P.A.Davies,W.P. Shillue, A.Vaccari and J.M. Payne, “Efficient

Generation of Guided Millimeter-Wave Power by Photomixing,” IEEE Photon. Technol. Lett., vol. 14, no. 2,

197–199, February 2002.

[21] P.G. Huggard, B.N. Ellision, P. Shen, N.J. Gomes, P.A. Davies, W.P. Shillue, A. Vaccari and J.M. Payne,

“Generation of Millimetre and Sub-millimetre Waves by Photomixing in a 1.55mm Wavelength Photodiode”,

Electon. Lett., vol. 38, no. 7, 327–328, 2002.

[22] T. Noguchi, A. Ueda, H. Iwashita, S. Takano, Y. Sekimoto, M. Ishiguro, T. Ishibashi, H. Ito and T. Nagatsuma,

“Millimeter-Wave Generation using a Uni-traveling-carrier Photodiode”, Proceedings of the 12th International

Symposium on Space Terahertz Technology, San Diego, CA, USA, 2001.

[23] A. St€ohr, A. Malcoci, A. Sauerwald, I.C. Mayorga, R. G€usten and D. J€ager, “Ultra-Wide Band Traveling-Wave

Photodetectors for Photonic Local Oscillators”, IEEE J. Lightwave Technol., vol. 21, no. 12, December 2003.

[24] A. Hirata, T. Nagatsuma, R. Yano, H. Ito, T. Furuta, Y. Hirota, T. Ishibashi, H. Matsuo, A. Ueda, T. Noguchi, Y.

Sekimoto, M. Ishiguro and S. Matsuura, “Output Power Measurement of Photonic Millimetre-wave and Sub-

millimetre-wave Emitter at 100–800GHz”, Electron. Lett., vol. 38, no. 15, 798–799, July 2002.

[25] A. St€ohr and D. J€ager, “Ultra-Wideband Travelling-Wave Photodetectors for THz Signal Generation”, IEEE

LEOS Annual Meeting, Puerto Rico, Nov. 2004.

[26] A.Malcoci, A. St€ohr, A. Sauerwald, S. Schulz and D. J€ager, “Waveguide and Antenna Coupled Travelling-Wave

1.55mmPhotodetectors for Optical (Sub)Millimeter-WaveGeneration”, inMicrowave and Terahertz Photonics,

Proceedings of the SPIE, vol. 5466, ISBN 0-8194-5389-7, pp. 202–209, 2004.

[27] A. St€ohr and D. J€ager, “THz-Photomixers: An Overview”, in Millimeter-wave and Terahertz Photonics,

Proceedings of the SPIE, vol. 6194, ISBN 0-8194-6250-0, April 2006.


microwave photonics || photonic oscillators for thz signal generation

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