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Linac output charge monitor 1

Linac Output Charge Monitor notes

1. Charge monitor design

The linac charge monitor is intended to integrate the accelerator’s charge per pulse using the well‐established technique of using the electron bunch as a 1‐turn primary passing through the middle of a toroidal core wound with a given number of turns, n, which form the transformer secondary, of inductance L (Figure 1). The secondary is shunted with a capacitor, C, and a resonant tuned circuit is thus formed, acting somewhat like a child on a swing when given a ‘push’. The peak voltage, V, developed across the capacitor in this tuned circuit, resulting from the passage of a given charge, Q, through the toroid can be readily shown to be:

V = Q/n.C

Energy will then be transferred to and from the inductor and the circuit will oscillate with a frequency, F, given by the usual relation:

F = (42 LC)‐1/2

These oscillations will be damped by the core losses, the resistive load resistance, RL, and by the secondary’s resistance, Rs. Clearly, RL should be maximised and Rs should be minimised. Moreover, it is obvious that as n decreases, the charge sensitivity increases, but the inductance decreases as n2 and the resonant frequency F increases in inverse proportion to n. Similarly as C decreases, the charge sensitivity also increases but now the resonance frequency increases in inverse proportion to √C. In other words, high charge sensitivity can only be achieved if F is made as large as possible. However, F cannot be increased indefinitely if accurate integration of long pulses is to be achieved.

Perfectly accurate integration can in fact never be achieved; however when the electron pulse duration is short compared to a quarter cycle of F, an acceptable level of accuracy can be achieved. It can be shown1 that the

percentage error for an electron pulse of width is given by: % error ≈ 103 F2 2 / 6.

While it is desirable to make this error as low as possible, in our specific case, we only need to integrate

correctly pulse widths of up to 1 s (single shot pulses). The ‘next’ pulse width is 3.8 s (repetitive pulses) and here the pulse width is, in general, constant. This makes it straightforward to apply a calibration correction for these longer pulses in software.

This percentage error is plotted as a function of operating frequency in Figure 2. If we operated at a frequency of

10 kHz an error as low as 0.25% could be achieved. When dealing with pulses < 1 s, any frequency‐induced errors would be negligible. Even if we operated at a frequency of 20 kHz the error would be < 1%.

This simple explanation assumes that the damping factor is low, and that the exponentially decaying oscillations decay slowly. This requirement places constraints on the inductor core and its associated winding that are often hard to achieve. However, before tackling this, n is also restricted: a tuned circuit output voltage of >±10 V is undesirable as any preamplifier following the tuned circuit would then overload.

% integration error of 3.8 us pulse

0

0.2

0.4

0.6

0.8

1

1.2

1.4

1.6

10000 12000 14000 16000 18000 20000 22000 24000

Frequency (Hz)

% e

rro

r

Figure 2: Measurement (integration) error when

using a 3.8 s pulse and a toroid operating at different resonant frequencies.

L

C

nQ

V

calibration

L

C

nQ

V

calibration

Figure 1: A resonant inductive charge monitor, formed by an inductor, L, and a shunt capacitor, C. The voltage across C is proportional to the beam charge in the electron pulse (e‐). A 1 turn winding on the core can be used for calibration purposes.


The number of turns is thus likely to be in the range of 50‐400 turns if we need to measure charges exceeding

values of the order of 3.5 s x 0.2 A ≈ 0.7 C; this corresponds to the maximum linac output. For obvious

reasons, it would be convenient to make a 10 V output voltage 1 C. So the product n.C can be set to: n.C = 10‐7

As n increases, the tuning capacitance must decrease. However, as n increases, the inductance increases as n2 and F increases in proportion to n. For a constant sensitivity, F increases as √n. It is therefore desirable to place as many turns as possible on the core and reduce C as much as possible to operate at as high an F as possible. Three practical factors limit the extent to which n can be increased: (1) the inductor self‐capacitance increases; (2) the inductor core losses increase with F, though it is rather hard to estimated these from published core data; (3) the inductor resistance increases in proportion to n. In other words, if the core is wound with thinner, well‐spaced turns, the winding resistance increases and limits the tuned circuit Q‐factor. Similarly, if high permeability cores are used, these tend to have higher losses, again limiting the tuned circuit Q‐factor.

An associated limitation is the restriction placed by the beam area: our installation uses a beamline diameter of 50.8 mm, so the core internal diameter must be larger than this. T100 and T107 cores could be used: we chose

to use T107 cores. The T107 core format, has an external diameter of 1072 mm, an internal diameter of 651.3 mm and a thickness of 250.75 mm. Since a high inductance is always preferable, we chose to wind the inductor on two T107 cores places side by side. T107 cores are usually (i.e. easily obtained) as plain cores, i.e. not overpainted, so it is essential to eliminate the consequences of sharp edges by overwinding the core(s) with ‘transformer’ tape. We chose to use 3M transformer tape, part number “12 Tape 9mm” , supplied by OneCall (order code 213‐5637).

The internal diameter of the core dictates its inner perimeter: 200 mm. When the tape is added, its perimeter

reduces to 198 mm (inner diameter of 64.15 mm). Similarly the outer diameter could be as large as 110 mm

(worst case). The length of 1 turn can thus be taken to be <95 mm.

The winding resistance does of course play a role in obtaining a good Q‐factor. In practice, it is unlikely to be a limiting factor as core losses are likely to dominate. Inter‐turn winding capacitance is a bit of a red herring alas will be shown later; it is the dielectric constant of the core that defines the coil capacitance along with the contributions of other insulators used in the coil construction. The winding must of course be made mechanically rigid to result in a constant self‐capacitance. Long term variations due to e.g. radiation damage of insulators are harder to predict and it would be wise to reduce the self‐capacitance as much as possible.

Indeed there is no definitive method that can be found in the literature to calculate the self‐capacitance of wound ferrite ring cores. High dielectric constants are commonly found in MnZn ferrites (Generic composition for such ferrites is MnaZn(1‐a)Fe2O4) and this is likely to dominate.

The availability of large cores is limited and that of low loss cores even more so. T107 cores of 3F3 ferrite (Ferroxcube T107/65/25‐3F3; OneCall Farnell order code: 2103393) and 3F4 ferrite (Ferroxcube T107/65/25‐3F4 OneCall Farnell order code: 2103395) were selected as being appropriate. Although the AL factor for the 3F3 core is higher, 4.485 µH/turn, the losses also appear to be higher, although definitive data on this are not readily available form the manufacturer’s data sheets. The 3F4 material AL factor is 1.87 µH/turn but its losses are lower, again form extrapolation of manufacturer’s data. As will become obvious later, the inductance of coils made with 3F4 ferrite is far too low, and although coils wound on this material were experimented with, no data are presented here.

Different approaches were taken for winding the cores, as described below. We wanted to construct a device which was sensitive and able to detect very low pulse charges. Such low pulse charges will produce very low output voltages and some form of amplification close to the toroid then becomes essential. The conventional approach is to follow the toroid with a low noise, high input impedance preamplifier. While this approach is fine ‘on paper’, in practice, there are numerous stray magnetic and electric fields present in an accelerator installation and a balanced approach is thus desirable. Such a balanced approach can be realised by using a centre‐tapped coil followed by a balanced input amplifier (Figure 3). Moreover, the gain of this balanced amplifier should be made variable in order to cover a wide dynamic range.

L

C

nQ

V+calibration

V-

L

C

nQ

V+calibration

V-

Figure 3: A balanced resonant inductive charge monitor, formed by a centre‐tapped inductor and a shunt capacitor.


This can be achieved by using a high impedance FET instrumentation amplifier (INA 111, U1 in Figure 5). Ideally, the tuned circuit will ‘ring’ for a considerable time, a time determined by the tuned circuit ‘Q’ factor. A long oscillation time is desirable in order to allow measurement of the peak circuit voltage some time after the radiation charge pulse, so as to allow any accelerator generated noise to have died away. This is in fact the principle advantage of using a resonant charge monitor. However, a long decay time will make operation at high repetition rates impossible, since the oscillations must decay to close to zero before the next charge pulse. This compromise is readily solved by damping the tuned circuit quickly as soon as a measurement has been made.

However, this damping requires the use of a voltage‐controlled switch which inevitably has some capacitance between its control input and the switching element. Once the switch is released, some charge from the control input is thus inevitably passed to the tuned circuit, causing it to ring once again!

This problem can be overcome by using a soft, slow release that allows the shunting impedance to rise slowly. While this can be achieved with e.g. and FET operated as a voltage‐controlled resistance and an exponentially decaying shorting waveform, this is a tricky to implement in practice and the correct waveform will depend on the characteristics of the specific device used. This approach in the ‘digital’ age is cumbersome. However, when a balanced approach is used, as depicted in Figure 4, hard switching can indeed be used: any charge feedthrough introduced by the control input on one switch is counterbalanced by the same charge appearing on the other input of the balanced amplifier.

The arrangement in Figure 5 shows this arrangement: two optically isolated FET switches (U2, U3) are placed between ground and each of the inputs of the balanced amplifier. When the FETs are ‘on’ the tuned circuit is

damped by a 30 k resistance (2 x 15 k resistors). When they are ‘off’ the shunt resistance rises to a very large

value (>100 M). Logic level control of the reset function thus becomes possible.

The gain of the instrumentation amplifier is set by a resistance between pins 1 and 8. This resistance is made programmable through the use of two miniature reed relays. Gain selection of x1, x10 and x100 is provided with a single bipolar control line (the use of too many wires is always undesirable!) with current steering to the relay coils provided by D1 and D2 (with D3 and D4 providing the usual back‐emf protection). The gain of the INA111 is

given by: G = 1 + (50 k/RG), where RG is the resistance between pins 1 and 8. When the ‘x100’ reed relay is

energised, 504.95 is presented and the gain is 100.02; when the ‘x10’ reed relay is energised, 5554.455 k is presented and the gain is 10.0018. These gain ‘errors’ are well within errors caused by resistor tolerance.

Reset

Ctune

Cstray

Cstray

Reset

Ctune

Cstray

Cstray

Figure 4: A balanced resonance reset circuit arrangement. The tuned circuit ‘charge’ introduced by the ‘Reset’ line is distributed symmetrically through the feedthrough stray capacitances of the switches.

+Vcc

+

_

6

3

24

718

100 nF

100 nF

-Vee

5

100

TS4148TS4148 x100x10

x10 x100

0V for x1NEG for x10POS for x100

470 1

2

6

4H11F3

15

k

Test input

510 5.1 k

510 k 51 k

INA111470 1

2

6

4H11F3

15

k

Gain

-Vee

+Vcc

D1 D2

D3 D4

U1

U2

U3

100

TS4148 TS4148

100

10 nF

1 nF

100 pFOutput

Reset

Ctune

+Vcc

+

_

6

3

24

718

100 nF

100 nF

-Vee

5

100

TS4148TS4148 x100x10

x10 x100


470 1

2

6

4H11F3

15

k

Test input

510 5.1 k

510 k 51 k

INA111470 1

2

6

4H11F3

15

k

Gain

-Vee

+Vcc

D1 D2

D3 D4

U1

U2

U3

100

TS4148 TS4148

100

10 nF

1 nF

100 pFOutput

Reset

Ctune

Figure 5: A generic preamplifier circuit appropriate for use with a resonant toroidal charge monitor.


Of course, the system must eventually be calibrated in terms of charge sensitivity. The obvious way to do this is to provide a ‘one turn’ calibration winding through the toroid and inject a known current for known time. Things are not quite as simple as they sound, however. Since we live in the real world, we would somehow need to

terminate the line carrying the pulse; this termination would, in practice be 50 . So we could simply use a 50 resistor in series with the one turn primary and hope that the reflected load impedance would be negligible. When the toroidal coil turn number is large, this may indeed be true but, as will be seen later, it is advantageous to use a low turn number n. Moreover, this calibration system will end up damping the coil: the reflected

impedance will be 50 x n2 . Even with n = 200, the shunt impedance would be 2 M, a rather low value that would damp the tuned circuit. One could, in principle develop an arrangement which consists of a true current source: now the tuned circuit would no longer be damped. But remember we need to produce pulse width of

the order of 1 s…..so the rise and fall times would need be <<5 ns if the calibration could be ‘trusted’. The author of this document challenges anyone who could make such a current source: stray capacitances of most switched devices preclude high speed operation in the current mode.

The simplest (and best) way to achieve the calibration is to inject a known charge into a single turn primary test turn by using a calibrated voltage step into a known capacitor. This approach completely overcomes the reduction in tuned circuit Q when a current pulse of known duration is injected through a resistor. A reasonable

maximum voltage step is 10 V. If we want to generate a maximum charge of 1 C, we would need a capacitance of 100 nF. This capacitance is excessive: the time constant would be 5 s and there would thus be an additional error caused by the calibration. A 10 nF capacitor is thus the maximum that could be used. In practice, the use of a 1 nF capacitor is more appropriate, where the time constant would be negligible. It was therefore decided to use 3 capacitors (100 pF, 1 nF and 10 nF), only one of which could be inserted into the calibration circuit. This provides us with the following outputs at different preamplifier gains:

Input C = 100 pF Gain = 1

C = 1000 pF Gain = 1

C = 10 nF Gain =1

C = 100 pF Gain = 10

C = 1000 pF Gain = 10

C = 10 nF Gain = 10

C = 100 pF Gain = 100

C = 1000 pF Gain = 100

C = 10 nF Gain = 100

10 V step 1 nC Vout = 10 mV

10 nC Vout = 100mV

100 nC Vout = 1 V

1 nC Vout = 100mV

10 nC Vout = 1V

100 nC Vout = 10V

1 nC Vout = 1V

10 nC Vout = 10V

100 nC Vout = o/l

1V step 100 pC Vout = 1 mV

1 nC Vout= 10mV

10 nC Vout = 100mV

100 pC Vout = 10 mV

1 nC Vout = 100mV

10 nC Vout = 1 V

100 pC Vout = 100 mV

1 nC Vout = 1V

10 nC Vout = 10V

100 mV step 10 pC Vout = 0.1 mV

100 pC Vout = 1mV

1 nC Vout= 10mV

10 pC Vout = 1 mV

100 pC Vout = 10 mV

1 nC Vout = 100 mV

10 pC Vout = 10 mV

100 pC Vout = 100 mV

1 nC Vout = 1V

It is relatively straightforward to generate a known voltage step and we thus need to use close tolerance capacitors to complete the calibration. Polystyrene film capacitors are excellent for this purpose and are readily available in ±1 % tolerances. The purist may want to check the actual capacitor values independently, but a ±1% error is usually quite acceptable in most accelerator installations.

2. Toroid 1 construction

Version 1 of the toroidal charge monitor is constructed from toroidal ferrite cores, made from ferrite material Ferroxube 3F3. This is a MnZn type of ferrite with low losses and suitable for operation at frequencies of tens of

kHz. Two cores, are used, each core has an initial permeability of i = 2000 25%. They are available as part numbers T107/65/25‐3F3. They have an effective cross section area Ae = 514 mm² and an effective magnetic path length le = 259 mm. The nominal inductance factor AL for each core is 4.485 µH when the conductor is wound directly on the core; somewhat less if the conductor is spaced away form the core surface. With two cores we can reach AL values of the order of 9 µH. The cores were purchased from OneCall Farnell (Order Code:

2103393) at a cost £56.93 each. Each core weighs 700 g, so both strong hands and nimble fingers are essential. Robert G Newman’s hand satisfied these criteria.

The various steps in the construction of the toroidal coil are shown in Figure 6. This starts by insulating each of the cores with transformer tape, obtained as part “12 Tape 9mm” from 3M, as supplied by OneCall (order code 213‐5637), joining the two cores with the same tape, and winding with 2 x 103 turns of 0.8 mm diameter enamelled copper wire. The reason for the centre‐tapped arrangement is to allow balanced detection, as will be made clear later. This wire diameter and the turn number are such that it all fits within the inner perimeter

Around 31 metres in total is required to wind evenly 206 turns. The series resistance is very low, less than 1 . Once wound, the toroid assembly was varnished with Electrolube MR8008B varnish (obtained from OneCall, order code 958372 (£8.44 for a volume of 250 ml.)


3. Toroid 1 electrical testing

Just prior to varnishing, the assembly was tested by loading it with a 470 pF capacitor and a 10 M oscilloscope probe. A 1 turn primary was formed by a wire routed through the toroid centre, connected to a sine wave signal

generator through a 1 k series resistor. The secondary reflected resistance at the output was therefore

1 k x n2, i.e. 1 k x 2062 ≈ 42 M, which should be high enough to not significantly load the circuit. The resonance plot is shown in Figure 7.

The tuned circuit Q was thus 84, a respectable value at this frequency. Once the winding was completed (but before final mounting into its housing, the toroidal inductor had the following measured properties (measured on a TTi LCR400 bridge): 100 Hz test frequency L = 350 mH Q = 18 1 kHz test frequency L = 340 mH Q = 377 10 kHz test frequency L = 392 mH Q = 120

Figure 6: Starting with the basic uncoated core (A), it is insulated with transformer tape (B, C) and when both cores are thus prepared (D), they are tied together with thin strips of tape (E) and winding begins from the ‘half‐way’ centre tap (F). The completed winding (G) is varnished (H) and overwound with a final tape layer.

A B C

D E F

G H I

Figure 7: Results from a ‘quick’ experiment to determine the tuned circuit Q. The bandwidth was

138 Hz @ 11.65 kHz.

0

5

10

15

20

25

11 11.2 11.4 11.6 11.8 12 12.2 12.4

138 Hz


Next, various capacitors were placed in parallel with the probe and the resonance frequency was measured. Data were

obtained with a Tektronix probe that had a 10 M input impedance and a 8.5 pF input capacitance. C = 1 x 470 pF fres = 11.22 kHz Ctotal = 586 pF Lderived = 343 mH C = 2 x 470 pF fres = 8.43 kHz Ctotal = 1056 pF Lderived = 338 mH C = 3 x 470 pF fres = 7.02 kHz Ctotal = 1526 pF Lderived = 337 mH C = 4 x 470 pF fres = 6.13 kHz Ctotal = 1996 pF Lderived = 338 mH C = 5 x 470 pF fres = 5.51 kHz Ctotal = 2466 pF Lderived = 338 mH C = 6 x 470 pF fres = 5.06 kHz Ctotal = 2936 pF Lderived = 337 mH C = 7 x 470 pF fres = 4.702 kHz Ctotal = 3406 pF Lderived = 336 mH When 1/(fres)

2 is plotted against the known tuning capacitance, the stray capacitance can be determined, as shown in Figure 8. From these data, the stray capacitance can be estimated to be 116 pF. The probe capacitance of 8.5 pF is subtracted from that, yielding a value of 107.5 pF.

We can now work out the effective inductance:

1/F = 2√(LC); 1/F2 = 42 LC; L = 1/(42F2C) L = 340 mH

Similarly, when a short pulse waveform was fed to the 1 k resistor, the 1/e envelope decay time was found to

be 2.5 ms.


Version 2 of the toroidal charge monitor is constructed from the same ferrite material, Ferroxube 3F3, but only one core is used. This time the core is wound with 2 x 100 turns of 0.45 mm dia. Cu enamelled wire (26 swg), with a separation of 0.2 mm average between the turns. Rather than overwinding the core with paper tape, only the edges of the core were covered with the insulating tape to prevent damage to the wire enamel. The stages in the construction are shown in Figure 9.

5. Toroid 2 electrical testing

The data obtained from bridge measurements are shown below. The nominal inductance factor AL for the core is 4.485 µH, so that with 200 turns, a total inductance of 180 mH is expected. However, the tolerance on the AL figure is poor (±25%). So we seem to be above average (!) although bridge measurements are usually fraught with difficulties.

100 Hz test frequency L = 207.3 mH Q = 29.56 1 kHz test frequency L = 192 mH Q = 180 10 kHz test frequency L= 200.9 mH Q = 300

The wound core’s self resonant frequency was found to be 39.9 kHz. Once again, 1/(fres)2 was plotted against

load capacitance and the stray capacitance determined (including the 8.5 pF of probe capacitance). The results are plotted in Figure 10. From these data, the stray capacitance can thus be estimated to be 108 pF.

Figure 9. Stages in toroid construction: starting with the basic core, transformer tape is placed around the edges of the core

and the 2 x 100 turns are wound evenly, i.e. with even spacing between turns. The inner perimeter ≈200 mm; wire length of

one turn ≈ 95 mm. The total wire length is ≈ 18.4 metres and the total resistance is ≈ 2 .

Figure 8: Determination of inductor stray

capacitance: the stray capacitance was 116 pF.

y = 6.3005E‐12x + 7.3424E‐10

‐0.000000005

0

0.000000005

0.00000001

0.000000015

0.00000002

0.000000025

0.00000003

0.000000035

‐1000 0 1000 2000 3000 4000 5000

Tuning capacitance (pF)

1/f^2 (1/H

z^2)


C = 1 x 150 pF fres = 24 kHz Ctotal = 258 pF Lderived = 170 mH C = 1 x 330 pF fres = 18.15 kHz Ctotal = 438 pF Lderived = 176 mH C = 1 x 390 pF fres = 17.21 kHz Ctotal = 498 pF Lderived = 172 mH C = 1 x 470 pf fres = 15.52 kHz Ctotal = 578 pF Lderived = 182 mH C = 2 x 470 pf fres = 11.44 kHz Ctotal = 1048 pF Lderived = 186 mH C = 3 x 470 pf fres = 9.49 kHz Ctotal = 1518 pF Lderived = 185 mH C = 4 x 470 pf fres = 8.3 kHz Ctotal = 1988 pF Lderived = 185 mH We can now work out the effective inductance:

1/F = 2√(LC); 1/F2 = 42 LC; L = 1/(F242C) and the results are shown above, but the values obtained suggest that the interpolated self‐capacitance is perhaps somewhat incorrect.

The circuit Q, when loaded by a physical 330 pF capacitor (444 pF effective) is shown in Figure 11. The Q value is found to be 91. When the coil is optimally tuned with 500 pF effective, the Q rises to 106.

These results suggest that the self‐capacitance is related to the number of turns, and not to the capacitance between turns. This is consistent with data obtained by D W Knight (2008)2; where it is suggested that that inter‐turn capacitance is negligible. Other not quite so useful references3, 4 confirm this, though it is wort reading reference5. The performance of the toroid is shown in Figures 12‐16.

Figure 13: The first few cycles following

excitation with a 3.8 s pulse. Figure 12: Response when coil is loaded with Ccoil + 330 + 47 pF for 10 nC input: 100 mV.

Figure 14: Decay when loaded

by 10 M probe.

Figure 16: Decay when loaded

by 100 M probe.

Figure 15: Difference in decays

between 10 M (Y) & 100 M (W) probes.

Figure 11: Resonance plots when the coil is loaded by 444 pF effective (left) and 500 pF effective (right).

0

200

400

600

800

1000

1200

17 17.5 18 18.5 19 19.5 20

Frequency (kHz)

Relative

amplitude (AU)

0

200

400

600

800

1000

1200

16.5 17 17.5 18

163 Hz Q = 106

Frequency (kHz)

Figure 10: Determination of inductor

stray capacitance: intercept at 108 pF.

y = 7.084E‐06x + 7.686E‐04

‐0.005

0

0.005

0.01

0.015

0.02

‐200 300 800 1300 1800 2300 2800


1/f^2 (1/kHz^2)


The Q‐factor of a parallel tuned circuit is described by: Q = Rp (C/L)½. In this instance Q = 106, (C/L)½ = 5.27 x 10‐5

and the parallel resistance can thus be determined to be ≈ 2 M. Since this is obtained with a 10 M probe, the

true Rp = 2.5 M and the true Q is 132


Version 3 of the toroidal charge monitor is constructed again from the same ferrite material, Ferroxube 3F3, with the winding placed on a single core. This time the core is wound with 2 x 160 turns of 0.4 mm dia. Cu

enamelled wire (27 swg), with minimal separation between turns (typically <0.1 mm). Rather than overwinding the core with paper tape, only the edges of the core were covered with the insulating tape to prevent damage to the wire enamel. The stages in the construction are shown in Figure 17.

7. Toroid 3 testing

The data obtained from bridge measurements are shown below. The nominal inductance factor AL for the core is 4.485 µH, so that with 320 turns, a total inductance of 459 mH is expected. However, the tolerance on the AL figure is poor (±25%). Nevertheless the inductance seems reasonable although the Q was somewhat lower than expected. However, it is the Q of the tuned circuit which matters, not the inductor Q.

100 Hz test frequency L = 461 mH Q = 36.8 1 kHz test frequency L = 440 mH Q = 148 10 kHz test frequency L = 517 mH Q = 162.5


load capacitance and the stray capacitance determined (including the 8.5 pF of probe capacitance). The results are plotted in Figure 18. From these data, the stray capacitance can thus be estimated to be 108 pF including the 8.5 pF associated with the probe.

C = 0 pF fres = 23.45 kHz Ctotal = 108 pF Lderived = 427 mH C = 1 x 180 pF fres = 14.54 kHz Ctotal = 288pF Lderived = 416 mH C = 2 x 180 pF fres = 11.41 kHz Ctotal = 468 pF Lderived = 416 mH C = 3 x 180pF fres = 9.68 kHz Ctotal = 648 pF Lderived = 417 mH C = 4 x 180 pF fres = 8.56 kHz Ctotal = 828 pF Lderived = 418 mH C = 5 x 180 pf fres = 7.755 kHz Ctotal = 1008 pF Lderived = 418 mH

So this once again confirms that the self‐capacitance of the core does not depend on the number of turns.

Since we now have 320 turns instead of 200 turns, the load capacitance must be decreased in order to ensure that the correct sensitivity is obtained from n.C = 10‐7.

The load capacitance must thus be 312.5 pF. Since the total stray capacitance is 108 pF, the physical capacitor should have a value of 204.5 pF. This could be closely realised with a parallel combination of 82 and 120 pF (202 pF).

Figure 17. Stages in toroid construction: starting with the basic core, transformer tape is placed around the edges of the core and the 2 x 160 turns of 0.4 mm wire are wound evenly, with minimal spacing between turns. The inner perimeter ≈200 mm;

the length of one turn ≈ 95 mm. The total wire length is ≈ 18.4 metres and the total resistance is ≈ 4 .

y = 1.64742E‐11x + 1.78289E‐09

‐0.000000002

0

0.000000002

0.000000004

0.000000006

0.000000008

0.00000001

0.000000012

0.000000014

0.000000016

0.000000018

‐200 0 200 400 600 800


1/f^2 (1/H

z^2)

Figure 18: Determination of inductor stray capacitance: intercept at 108 pF.


Since the true inductor self capacitance is 99.5 pF and since the subsequent preamplifier differential mode capacitance is 6 pF, the total stray capacitance is 105.5 pF, we actually need a 207 pF tuning capacitor. This is better realised with a parallel combination of 56 pF and 150 pF (206 pF).

The Q of the tuned circuit is slightly reduced, to 81 (Figure 19). This somewhat lower than with the toroids previously

described but still acceptable. Of course the 10 M input impedance of the measuring probe limits the Q factor. Nevertheless, it is possible to derive the true Q factor by estimating the parallel resistance of the tuned circuit.

The Q‐factor of a parallel tuned circuit is described by: Q = Rp (C/L)

½. In this instance Q = 81, (C/L)½ = 2.74 x 10‐5 and

the parallel resistance can thus be determined to be ≈ 3 M. The intrinsic Rp due to core losses can thus be estimated to

be 4.29 M. The true Q is thus closer to Q = 4.29 x 106 x 2.74 x 10‐5 = 117.6. This new Q value can be used to determine the decay constant and thus the ringing time of the circuit. The

decay constant is given by = 2fres /2Q = 376, i.e. the 1/e decay time should be 2.66 ms.

8. Time to make a choice

So which of the three toroids is ‘better’? Toroid #1 is clearly not optimal: although two cores were used, the increased inductance, with a low turn number, does not offer any significant advantages over Toroid #3, wound with a higher number of turns. In the case of Toroid #2, the inductance is low and forces operation at a relatively high resonant frequency. Toroid # 3 thus seems to be optimal, offering a 3‐fold reduction in integration error for the longer pulses. However, Toroid #3 is only advantageous if its self‐capacitance remains constant during its service life. Toroid #2 is quite acceptable, but only if subsequent electronics are able to operate at >15 kHz. A secondary advantage of Toroid#3 is that the large number of turns makes the source impedance of a calibration turn appear to be significantly higher. This in turn minimises calibration error associated with Q reduction. On the other hand Toroid #2 does have a higher inherent Q.

It is clear from previous experiments that the stray capacitance is due to the ferrite’s permittivity. The bulk permittivity of MnZn ferrites seems to be in the region of 106 at low frequencies, (Ferroxcube data sheet) but there is no trustworthy information about the variation of permittivity with temperature (or with any other factors, other than operating frequency). It is therefore essential to somehow measure this temperature coefficient and to select an appropriate capacitor temperature coefficient to compensate (if at all required)

We could also place a larger number of turns and further reduce the tuning capacitor. With 400 turns, and inductance of 750 mH would be expected and a 250 pF total tuning capacitance would be necessary (e.g. 150 pF and 100 pF self‐capacitance), and resonance frequency of 11.5 kHz.

Then the factor (C/L)½ reduces to 1.83 x 10‐5, and assuming that a Q of 80 or so could be achieved, a

4.4 M Rp value would result. This would be even better, but does of course depend on minimal changes in the inductor self‐capacitance with temperature. So in a sense, we are back where we started: high inductance and permeability and low capacitance are to be preferred.

0

50

100

150

200

250

300

350

400

450

500

550

600

650

700

750

800

850

900

950

1000

1050

1100

1150

1200

1250

13.3 13.5 13.7 13.9 14.1 14.3 14.5 14.7 14.9

Frequency (kHz)

relative

amplitude (AU)

172 Hz

Figure 19. Resonance plots when the coil is loaded

by 204.5 pF.

14070 Hz

Figure 20: From: http://www.robkalmeijer.nl/techniek/ electronica/radiotechniek/hambladen/qst/1988/12/page33/index.html


However, it can be shown that the signal‐noise ratio of the resonant charge monitor is determined by6 :

∞ Q AL ½ fres/(kT + ½ q Iin Rp)

½

Where kT is the product of temperature and Boltzmann’s constant; q is the electronic charge, R is the effective load resistance and Iin is the peak preamplifier input current.

So a high Q is required as is a high operating frequency. From the signal‐noise point of view, the use of Toroid #2

would apparently result in being large by a factor of (106/81) x (17300/14070), i.e. a factor of 1.6. The fact that Iin is lower is compensated by the fact that Rp is higher. A S/N increase of 1.6 is worthwhile. This suggests that a low turn number is preferable, but then the operating frequency would then become excessive. For example, if we placed 100 turns, we would have an inductance of 45 mH and a tuning capacitance of 1000 pF, the frequency

would be a very high 23.725 kHz and the integration error for 3.8 s pulses of the order of 1.5%. The (C/L)½ factor would be 1.49 x 10‐4 and the core losses may dominate. Clearly it would be advantageous to operate at a high frequency, but not so high as to make the integration errors excessive. Something in between 100 and 200 turns is thus suggested.

The inter‐turn capacitance topic is an interesting one. Although is some of literature from the radio ham world describing the fact that inter‐turn capacitance does not play a role in determining the stray capacitance of a ferrite‐wound toroidal ring core, there is almost nothing readily available in the scientific literature. There are only two publications7, 8 that refer to this and one that deals with multilayer coils9. Furthermore, the temperature coefficient of ferrites is hardly discussed anywhere. There is some information available10 but this refers to commercially unknown ferrites and the powders used do not refer to the bulk dielectric properties.

A further issue of concern is the variation of core permeability with temperature. Both of these temperature coefficients require testing, e.g. comparing values at 20 degC and 37 degC.

Figure 21: Plot of change in initial permeability of 3F3 ferrite with temperature (top), (taken for Ferroxcube data sheet)

and variation of ferrite powder dielectric constant with temperature, (bottom), taken from 9.


9. Toroid #4

But we could still to a little better: more than 2 x 50 turns but less than 2 x 100, i.e. 2 x 70 turns. So a 4th toroidal coil was wound; never let it be said that we allow ourselves to be defeated! This was thus wound with 2 x 70 turns with 0.5 mm wire on the same 3F3 T107 core. The turns were spaced by at least 0.6‐0.7 mm between turns, but nevertheless, because the turn number was low, a lower proportion of the core area was

utilised, compared to previous windings (by 35%). The construction is shown in Figure 22.

The data obtained from bridge measurements are shown below. The nominal inductance factor AL for the core is 4.485 µH, so that with 140 turns, a total inductance of 88 mH is expected.

However, the tolerance on the AL figure is poor (±25%), so the fact that it ends up higher than expected is perhaps unsurprising. Although it is the Q of the tuned circuit which matters, not the inductor Q, the latter seems to be excellent.

100 Hz test frequency L = 112 mH Q = 29.3 1 kHz test frequency L =103.7 mH Q = 173 10 kHz test frequency L =104 mH Q = 400


load capacitance and the stray capacitance determined (including the 8.5 pF of probe capacitance). The results are plotted in Figure 23. From these data, the stray capacitance can thus be estimated to be 68 pF, including the 8.5 pF associated with the probe.

C = 0 pF fres = 61.56 kHz Ctotal = 68.6 pF Lderived = 97.4 mH

C = 33 pF fres = 50.76 kHz C total = 101.6 pF Lderived = 97.0 mH









C = 1000 pF fres = 15.66 kHz Ctotal = 1068.6pF Lderived = 96.7 mH

We can now work out the effective inductance:

1/F = 2√(LC); 1/F2 = 42 LC; L = 1/(F242C) and the results are shown above, but the values obtained suggest that the interpolated self‐capacitance is perhaps very slightly low, by 1.5 pF at most incorrect, i.e. it is more likely to be 69.5 pF.

Since we now have 140 turns, the correct sensitivity will be obtained with a total tuning capacitance of 714.3 pF. When 69 pF is subtracted from that, we find we need a total capacitance of 645 pF. Assuming the preamplifier has an input capacitance of 6 pF, we will need a 639 pF tuning capacitor (100 pF + 68 pF + 470 pF) nominal, excluding other strays. The operating frequency would then be close to 20 kHz.

However, these values will be modified slightly when the coil is varnished and the toroid is placed in it its final housing (see section 11), and indeed connected to its associated preamplifier. Once this is done, the following data are obtained, and the total extrapolated toroid self capacitance is found to be a somewhat higher 90.65 pF, as shown in Figure 24.

y = 3.79E+00x + 2.60E-10

0

5E-10

0.000000001

1.5E-09

0.000000002

2.5E-09

0.000000003

3.5E-09

0.000000004

4.5E-09

0 2E-10 4E-10 6E-10 8E-10 1E-09 1.2E-09

Tuning capacitance

1/F

req

^2

(1/H

z^2)

Figure 22: The toroid optimal for our purposes.

Figure 23: Determination of Toroid # 4

stray capacitance: intercept at 68 pF.


C = 0 pF fres = 54.5 kHz Ctotal = 90.65 pF Lderived = 94 mH

C = 27 pF fres = 48.05 kHz C total = 117.7 pF Lderived = 93 mH





C = 340 pF fres = 25.41 kHz C total = 430.7pF Lderived = 91 mH





Since we still need a 714.3 pF tuning capacitance, the physical capacitor to be added is 623.65 pF, which can be made up with a parallel combination of 470 pF + 150 pF +3.3 pF capacitors (total = 623.3 pF) of course capacitors with a better tolerance than ± 1% are almost impossible to obtain, so this combination is a reasonable compromise.

Fres will then be 19.5 kHz. So this seems to be the best coil so far. Will it be appropriate for our application? Remembering that the percentage error for different pulse width is given by %

error ≈ 103 F2 2 / 6, and that the resonant frequency is 19.5 kHz, the errors will be 0.0634% for a 1 s wide pulse and 0.915% for a 3.8 s wide pulse. Bearing in mind that all components that set the sensitivity will have a comparable tolerance, this is acceptable, provided that the system gains can be calibrated. The measured Q of the tuned circuit is shown in Figure 25 and was found to a very respectable Q = 138.

10. Toroid preamplifier

The complete preamplifier circuit is shown in Figure 26. The circuit is constructed on a small printed circuit board, shown in Figure 27. Although space is provided on the printed circuit board for tuning capacitors, in practice it was easier to add (and modify) these on two feed‐through terminals mounted on the coil electrostatic screen (see Figure 30). Connections to the following processing system are made through a 9 way D‐type connector, through a Molex wire‐to board plug/socket.

The calibration board, used to drive the single turn toroid winding is constructed on a separate board, where a

50 termination and 3 calibration capacitors are placed; one of the capacitors is selected at a time using links.

y = 3.616E+00x + 3.278E-10

0

5E-10

0.000000001

1.5E-09

0.000000002

2.5E-09

0.000000003

3.5E-09

0.000000004

4.5E-09

0 2E-10 4E-10 6E-10 8E-10 1E-09 1.2E-09

Tuning capacitance

1/F

req

^2

(1

/Hz^

2)

Figure 24: Determination of Toroid # 4 housed

stray capacitance: intercept at 90.65 pF.

Figure 25: Resonance plot of Toroid #4

@19.63 kHz

142 Hz

Figure 26: Circuit diagram of the toroid preamplifier

+Vcc

+

_

6

3

24

718

100 nF

100 nF

-Vee

5

100

TS4148TS4148 x100x10

x10 x100


4

3

5

7

6

8

2

1

+Vcc

- Vee

0 V

Reset

Gain

Signal HI

Signal LO

Signal HI

Yellow

Green

Violet

Blue

Grey

Red

100

4700 pF

10 F

+Vcc

-Vee

470 1

2

6

4 H11F3

15

470

pF

AO

T 3

.3 p

F

9 way D-type -7 way Tyco/Triad link cable

MonitorBNC

Tes

t in

put

5.1 k

510 k

INA111

Fres 19.5 kHz

470 1

2

6

4 H11F3

15 k

3

4

5

7

6

2

8

SC2

C1

R10

R9

R8R7

C5

R6R5R4

R3 R2

R1

C4

C3

C6, C7

NC NC

Signal LO 1

2 Range-Vee

+Vcc

D D2

D3 D4

U1

U2

U3

Orange

2 x 70 turns

T107 3F3

Link for stand - alone operation

TS4148 TS4148

100

10 nF

1 nF

100 pF

100 pF = LCR FSCEX 100PF 1% 1 nF = LCR FSCEX 1000PF 1% 250V

10 nF = LCR FSCEX 10000PF 1% 63V

150

pF

0 pF BNC SMB

LCR

FS

CE

X 1

50P

F 1

% 6

30V

LCR

FS

CE

X 4

70P

F 1

% 6

30V

Mul

ticom

p M

C0

805

N3R

3C5

00A

Relays: COTO 9007-12-01

9 w

ay D

type

56

00 p

F fi

ltere

d

S

Blue

Pink

Grey

Green

Yellow

Brown

White

100

510

5.1

10 F

2 x 1 k matched


11. Toroid screening and mounting on beamline

The toroid is mounted on the accelerator beam line by using an aluminium former complemented by two printed circuit board end‐cheeks which act as part of an electrostatic screen around the coil. A generous spacing is provided between this screen and the coil so as to minimise additional stray capacitance. The preamplifier and the calibration circuit boards are mounted onto one of the end cheeks by using right angle brackets (Schroff 60807‐181). The toroid is held onto an aluminium inner support by a Tufnol support ring. An aluminium

Figure 28: The calibration board

Figure 27: The preamplifier printed circuit board. Left: top layer; Middle: Bottom layer; Right composite view. The board dimensions are 25 x 89.1 mm. Symmetrical coupling of the shorting input charge into the balanced input by the FET switches is ensured by placing them on opposite sides of the board. The input area is free of the copper ground plane so as to minimise additional stray capacitance.


end flange is then screwed on to the threaded inner support so as to form supports for two FR4 board shield rings, one of which is gapped so as to prevent a shorted turn from forming once a brass foil is soldered onto the FR4 boards. Small holes for insulated feedthroughs (ITT CANNON 011‐2049‐040FB9) are drilled into the far shield to carry the connections from the coil and the calibration turn.

12. Bench performance

Toroid#4

Data for toroid #2 and Toroid #4 are presented here. Both toroids performed well but the devil is always in the details!

The first experiment performed was to evaluate the changes in resonant frequency and in Q as a function of

temperature. Unfortunately only two ‘temperatures’ were available to us: room temperature (23 degC) and incubator temperature (37 degC). Data provided for the 3F3 ferrite material from Ferroxcube (Figure 21)

suggest that a shift in the initial permeability of +1.2%/degC is expected. So we would expect the inductance to increase in proportion and the resonant frequency to drop as the square root of that value. In fact, we find that

the resonance frequency does indeed become lower as the temperature is increased, but only by ‐0.6 %/degC, as shown in Figure 31.

However, as the resonant frequency drops, so must the Q (since Q ∞ (C/L)½ ). Indeed we observe a drop in Q of

‐1.3%/degC, as shown in Figure 31.

Figure 29: Exploded and cross‐sectional views of the mounting arrangement of the toroid on the accelerator beam line.

9 mm 9 mm

31 m

m

29 mmCharge Monitor Shield Far

Charge Monitor Shield

Charge Monitor Shield Near

Tufnol coil support

Inner PTFE sheet

Dose Mon Inner Support

Dose Mon End Flange

Break in copper to

prevent shorted turn

Charge Monitor

Outer Shield

50.2 mm

117 mm

Figure 30: Photographs of the toroid housing during testing


A higher resonant frequency drop than that measured is expected, since fres ∞ 1/L½. It is unlikely that the tuning capacitor temperature coefficient would play a significant role (polystyrene capacitor temperature coefficient ‐150 ± 50 ppm/degC, ‐0.2%/degC) and in any case, this would be reflected in both the Q and resonant frequency measurements. This result is somewhat of a mystery, but it can be safely assumed that small changes in ambient temperature could be assumed to result in only small changes in performance. In practice, this really means that the absolute calibration can only be trusted to no better than 1% or so, when normal laboratory conditions are assumed.

For completeness a stray capacitance measurement was also performed at 37 degC and is shown in Figure 32. The stray coil capacitance is determined to be just below 96 pF (95.6 pF); this is a significant increase, from approximately 90.6 pF at 23 degC (the temperature coefficient of the test capacitors was 50 ppm / degC average. This should accurately measure whether capacitance or inductance changes ‘are to blame’.













C = 1030 pF fres = 14.67 kHz Ctotal =1125.64 pF Lderived = 105 mH

So we find that the inductance at 37 degC has increased from by 13 mH from 92 mH to 105 mH, i.e. by 14%, or pretty exactly by +1%/degC, i.e. in line with what is measured somewhat inaccurately on the permeability graph of Figure 21. This brings us any closer to explaining the Q change? Not really, since it should have been reduced by 0.71%/degC, i.e. from 155 to 135 if we just take the inductance into account. But the capacitance has also increased by 0.4 %/degC so the temperature coefficient of the Q change should have been 0.7%/degC and not

the measured ‐1.41%/degC and the temperature coefficient of the resonant frequency should similarly have

been ‐0.7 %/degC and not the ‐0.6%/degC. So for the time being, it seems that we can only ‘blame’ the excessive drop in Q on increased cores losses with temperature.

0

1

2

3

4

5

6

7

19 19.5 20 20.5 21

Frequency (kHz)

Relative

amplitude(AU)

0

0.5

1

1.5

2

2.5

3

3.5

4

4.5

5

17.5 18 18.5 19 19.5

Frequency (kHz)

Relative

amplitude (AU)

130 Hz 150 Hz

23 degC 37 degC

Figure 31: Change in resonant frequency with temperature for toroid #4. The left resonance plot is obtained at 23 degC

and the calculated Q is 152. The right resonance plot is obtained at 37 degC and the calculated Q is 122, i.e. a Q drop of

30 over 14 degC (‐1.41%/degC). Resonant frequencies are 19.84 kHz (23 degC) and 18.26 kHz (37 degC), i.e. a shift of

1580 Hz over 14 degC (‐113 Hz/degC or ‐0.6%/degC).

y = 4.130E+00x + 3.950E-10

0

5E-10

1E-09

1.5E-09

2E-09

2.5E-09

3E-09

3.5E-09

4E-09

4.5E-09

5E-09

0 2E-10 4E-10 6E-10 8E-10 1E-09 1.2E-09

1/F

req

^2

(1

/Hz^

2)

Tuning capacitance

Figure 32: Determination of inductor stray

capacitance: intercept at 95.6 pF.


All this once again reinforces the fact that an absolute accuracy of better than 1% can not really obtained with this type of monitor and this guides to some extent the approaches towards subsequent signal processing. On the more positive side, it suggests that an independent measurement of temperature (or resonant frequency) may go some way towards establishing a more absolute accuracy of measurement.

The increase in coil stray capacitance with temperature is perhaps even more worrying. Why does it change? Is this change a property of the ferrite (i.e. a change in bulk relative permittivity) or some subtle effect associated with the use of varnish to stabilise the windings? In any case this temperature coefficient will affect the calibration of the system; changes in frequency will have little effect on the calibration, and the subsequent processing of the signal will depend very much on whether variations in Q are important or not. But changes in the effective load capacitance are indeed critical. So this demonstrates that a low turn number (low inductance) and hence a high tuning capacitance are indeed desirable, i.e. Toroid 4 is preferred over Toroid #2. An improved ferrite material would be highly desirable, but cores with such a material are clearly not readily available.

It is therefore strongly suggested that regular calibrations of this measurement system are performed, unless the ambient temperature can be controlled. In our installation, the temperature variations over an 8 day period is shown in Figure 33, and a 3 week period in Figure 34. The variations do not exceed ±0.75 degC. Although small, such variations do require regular calibration

Now let’s turn to the signals provided by this system. Typical results obtained with Toroid #4 are shown in Figures 35‐42. These data were acquired using a TTi TG2000 signal generator and a pulse output 50 Ω source and 50 Ω load, 10 Hz repetition rate. Coupled to scope input 2 set to 1 MΩ and triggered on channel 2.

Figure 34: Variation in temperature in the linear accelerator vault over a 3 week period.

Figure 33: Variation in temperature in the linear accelerator vault over an 8 day period.

18

18.5

19

19.5

20

20.5

21

21.5

22

20

21

22

19

18

20.5

19.5

18.5

21.5

26/2

27/2

28/2 1/3

2/3

3/3

4/3

5/3

6/3


Toroid#2

For comparison, results obtained when Toroid #2 (2 x 100 turns), optimally tuned for a x1 sensitivity of 10 V/C are presented in Figures 43‐48. A good signal‐noise ration can be obtained down to 50 pC or so. These data were

obtained by injecting a 1 s pulse through a 1 k series resistor into the 1 turn calibration winding. Figures 49 and 50 were obtained by injecting a sinewave current into the 1 turn calibration winding and energising the reset input with a step or with a pulse.

Although the initial step response is superior, it is clear however that Toroid #4 is significantly superior. Toroid #4 was eventually fitted onto the linac beamline. Much of the initial high frequency ringing at the stat of the main oscillation using Toroid #4 can be attributed to poor common‐mode filtering of the various items of test equipment. A low pass filter at the output can easily filter out all this ‘hash’.

Figure 35: 1x gain, 1 nF capacitor for

calibration coil 10 V step (1 nC).

Figure 36: 1x gain, 1 V step into 1 nF calibration capacitor (100 pC).

Figure 37: 1x gain 0.1 V step into 1 nF calibration capacitor (10 pC).



Figure 40: Decay following 1 V step into 1 nF calibration capacitor (10x gain, 1 nC).

Figure 41: 1 nC step decay. Figure 42: Excitation at resonance, recovery following reset pulse, sinewave at resonant frequency input.


13. Fitting on the linac beamline

The installation process on the accelerator beamline was performed in early June 2015. Thee is not much space available between the linac’s output solenoid and a set of beam steering coil, as is obvious in Figure 50!

Figure 43: Toroid #2 with preamp; 5 nC input, gain x1, 330 pF tuning.

Figure 44: Toroid #2 with preamp; 5 nC input, gain x10, 330 pF tuning

Figure 45: Toroid #2 with preamp; 5 nC input, gain x100, 330 pF tuning.

Figure 46: Toroid #2 with preamp; damped oscillation, 5 nC input.

Figure 47: Toroid #2 with preamp; damped oscillation, 5 nC input.

Figure 49: Toroid #2 with preamp; recovery from reset step, input sine wave at resonance.

Figure 50: Toroid #2 with preamp; 1 ms reset pulse, input is sine wave at resonance.

Figure 48: Toroid #2 with preamp; 50 pC input, gain x100, 330 pF tuning; HF noise from pulse generator / RF earth loop.

Figure 50: Podgy fingers and tight spaces do not go together! Although it seems from the images above that B Vojnovic did all the work here, this was far from true. P Reynolds, RG Newman and IDC Tullis actually did most of it…..


It soon became apparent that there were two significant issues that had to be overcome. The first one was that switching motor controllers on a robot assembly, part of the linac experimental setup, were inducing significant interference in the toroid. The reason for this was that the ac power lines to this system were not adequately filtered and, to make matters worse, the switchers operated at a modulation frequency of 20 kHz, close enough the toroid’s 19.5 kHz resonant frequency that problems were inevitable. The solution was simple though rather awkward to install, as can be seen in Figure 52. Appropriate filters were installed on the ac mains inputs and happiness was restored: any remaining interference was well into the Johnson noise of he toroid preamplifier.

The next problem was not quite so readily ‘sorted’. This was caused by interference from the linac magnetron drive/RF system. Although the linac is housed in its own Faraday cage, at the highest toroid sensitivity, the remaining interference was causing the toroid to resonate. It is worth remembering that we are dealing with

signals of the order of <100 V at the toroid preamplifier input. Even with the balanced arrangement, enough interference was reaching the system to spoil the excellent toroid sensitivity. One of the problems was that the housing around the beamline had been fashioned out of aluminium, and to make matters worse, the housing was sand‐blasted. Our mechanical workshop had obtained a sand‐blaster and had been keen to ‘try it out’: ‘see what a rugged finish we can now obtain’! The notorious aluminium oxide present on the surface of aluminium reared its ugly head and it was found difficult to ensure that the outside case was properly shielded.

Figure 51: The completed Toroid #4 installed on the beamline, between two aluminium end‐cheeks. The anticipation of excellent performance was high at this point (Day 1)…..

Figure 52: Robot motor driver blues. It was clear that the wiring ‘mess’ (top left) had to be sorted out. But it was not designed to be sorted out! All kinds of contortions were necessary to get to offending structures. Had it not been for RG Newman’s nimble body, it is doubtful if it could have been so quickly sorted out.


So a replacement case was fashioned out of brass plates and shielding gaskets were fitted to make contact with the beamline and the outer cover, as shown in Figure 53. When the pairs are mated together, the end‐plates become circular, honest!

After a bit of re‐assembly, all was well; the copper shield was held in place with some large Jubilee clips to ensure that good electrical contact is made to the brass end‐plates, through some shielding braid. Ugly but effective, as shown in Figure 54.

So now we are set to check the performance of the monitor….at last! 14. Performance

Once all the screening modifications were in place and electron pulses were used to test the monitor response,

all was as expected (almost!). Starting with Figure 55 that displays the response of the monitor to a 170 nC electron pulse, we can see that negligible decay occurs in the first half cycle (25 s) of the output waveform. Subsequent processing circuitry could thus readily sample the negative half‐cycle, when almost no accelerator or radiofrequency noise would be present. It is noted that the waveforms shown in Figure 55 were low‐pass filtered a 200 kHz first order Butterworth filter prior to display. Data were acquired on a PicoScope® 3000 Series instrument multichannel input instrument.

Figure 53: Brass end‐plates used to improve shielding from interfering signals.

Figure 54: The completed toroidal charge sensor assembly; images from either side of the beamline. The monitor is sandwiched in between a focus maget and a pair of beam steering coils.

Figure 55: Response to 170 nC electron pulses, acquired at timebases of 50 s/division and 100 s/division, the latter showing the slight droop of the resonant signal even at times extending to sub‐milliseconds. Preamplifier gain was x1.

e‐ pulse Measure here

2 V

50 s

2 V

100 s


As expected, the resonant circuit damping comes into its own when repetitive firing of the linac is used, as shown in subsequent figures. The oscillation decay would not have reduced to zero without the clamping circuit and the ‘previous’ ringing would have interfered with the next pulse oscillation.

100 nC

200 nC

3.33 ms

1 ms

Figure 56: Response to 180 nC electron pulse, acquired at a timebase of 500 s/division with the accelerator deliberately set up to result in the production of pulses of unequal charge/pulse at a repetition rate of 300 Hz.

Figure 57: Response to 15 nC electron pulse, acquired at a timebase of 500 s/ division with the accelerator properly set up to result in the production of consistent charge/pulse at a repetition rate of 300 Hz. Preamplifier gain x10.

10 nC

20 nC

3.33 ms

1 ms


Once the amplifier gain is set to x100, noise starts to become apparent, at a level of 5 pC. The source of this noise has not yet been established, but it is not observed on the bench. The noise floor on the bench is observed to be equivalent to <1 pC. It was thought that a nearby turbmolecular pump was responsible but this proved not to be the case. It has to be remembered that linacs are electrically ‘dirty’ machines and numerous noise

frequencies are produced, with one of these, at 20 kHz will inevitably be enhanced by the toroid resonance. It should also be remembered the charges that the toroid is intended to measure: 100 pC corresponds to a peak

current of 28.6 A for 3.5 s or 1 mA for 100 ns. All these are pretty low outputs well below what our accelerator can routinely deliver in a stable fashion. It is thus fair to say that the charge monitor’s performance is perfectly acceptable.

The processing of signal from the toroid will be discussed elsewhere. The most straightforward approach is to invert the signal and determine (i.e. hold) its peak value. The peak detector is reset by the damping pulse: there is ample time to digitise the peak detector output dring the 2 ms actve period. The resulting digital values are then summed to yied a number proprtional to the total electron pulse charge delivered during the irradiation pulse sequence. For single shot work, the same approach can be used.

15. Discussion

How well does the performance compare to what other workers have achieved? One of the first publicationa using a resonant monitor is that by Zimek in 197711 that reported a minimum detectable chargeof 10 nC. A subsequent publication12 from that group described improvement down to 1 nC sensitivity using a toroid resonant at 100 kHz. We had previously described very sensitive13, 14 devices using a low number of turns (30 turns) on a ferrite core of verry high permeability. The resonance was damped very quickly and range‐switching was performed by changing tuning capacitor values. Although we could achiev sensitivities down to a few pC, a electrically ‘quiet’, Van de Graaff accelerator was used and the device was mainly aimed at measurements of short pulse lengths, down to 1.5 ns. The change in resonant frequency was acceptable but required multiple calibration factors to be used. Though more sensitive, this is not considered particualrly elegant. The design of charge monitors for very fast pulses is also readily possible15.

We are sure that better performance than that described here could be obtained. One obvious way is to improve magnetic screening. A better approach is to reduce the number of turns and use somewhat larger and

200 pC

400 pC

Figure 57: Response to 300 pC electron pulse, acquired at a timebase of 500 s/ division with x100 preamplifier gain.


higher permeability ferrites. Such cores however are not obtainable these days, at least not in small numbers. This is a shame but an economic reality. Maybe one day….

References

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2 www.g3ynh.info/zdocs/magnetics/appendix/Toroid_selfC.html.

3 g3rbj.co.uk/wp‐content/uploads/2015/08/Self‐Resonance‐in‐Toroidal‐Inductors.pdf Payne : Self‐Capacitance of Toroidal Inductors with Ferrite Cores.

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7 Weber, S.‐P., Schinkel, M., Hoene, E., Guttowski, S., John, W. and Reichl, H. “Radio frequency characteristics of high power common‐mode chokes” EMC Zürich 2005, Proceedings of the 16th International Zurich Symposium on Electromagnetic Compatibility. CD‐ROM Zürich, 2005 ISBN: 3‐9521199‐9‐7 pp.507‐510 International Zurich Symposium on Electromagnetic Compatibility (EMC) <16, 2005, Zürich.

8 Wang Shishan, Liu Zeyuan and Xing Yan “Extraction of parasitic capacitance for toroidal ferrite core inductor”Industrial Electronics and Applications (ICIEA), 2010 the 5th IEEE Conference on 15‐17 June 2010 Pages 451 ‐ 456 E‐ISBN: 978‐1‐4244‐5046‐6 Print ISBN: 978‐1‐4244‐5045‐9 INSPEC Accession Number: 11433954 Conference Location: Taichung Doi: 10.1109/ICIEA.2010.5517152.

9 Kanzi, K., Nafissi, H.R. and Kanzi, M. (2014) A straightforward Estimation approach for determining parasitic capacitance of inductors during high frequency operation. Journal of International Conference on Electrical Machines and Systems, 3, 339‐352.

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14 Vojnovic, B. (1987) Sensitive long pulse beam charge monitor for use with charged particle accelerators. International Journal of Radiation Applications and Instrumentation. Part C. Radiation Physics and Chemistry, 29, 6, 409‐413.

15 Simmons, R.H. and Ng J.S.T. (2007) AToroidal Charge Monitor for High‐Energy Picosecond Electron Beams. SLAC‐PUB‐12306 February 2007.

Acknowledgements This note was prepared by and B. Vojnovic, I.D.C. Tullis and R.G. Newman during 2015. B Vojnovic was responsible for system design, R.G. Newman designed the printed circuit board and I.D.C Tullis performed mechanical design and helped out with the measurements of the performance of various toroids.

We acknowledge the financial support of Cancer Research UK.

© Gray Institute, Department of Oncology, University of Oxford, 2015.

This work is licensed under the Creative Commons Attribution‐NonCommercial‐NoDerivs 4.0 Unported License. To view a copy of this license, visit http://creativecommons.org/licenses/by‐nc‐nd/4.0/ or send a letter to Creative Commons, 444 Castro Street, Suite 900, Mountain View, California, 94041, USA.

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