ipt lightng system

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 25, NO. 5, MAY 2010 1275

An AC Processing Pickup for IPT SystemsHunter Hanzhuo Wu, Student Member, IEEE, John T. Boys, and Grant Anthony Covic, Senior Member, IEEE

Abstract—This paper presents a new type of inductive powertransfer (IPT) pickup that directly regulates the power in ac form,hence producing a controllable high-frequency ac source suitablefor lighting applications. The pickup has significant advantagesin terms of increasing system efficiency, reducing pickup size, andlowering production cost compared to traditional pickups that alsoproduce a controlled ac output using complex ac–dc–ac conversioncircuits. The new ac processing pickup employs switches operatingunder zero-voltage-switching conditions to clamp parts of the res-onant voltage across a parallel tuned LC resonant tank to achievepower regulation over a wide load range. The operation of thepickup is analyzed and the circuit waveforms have been verified byexperimental results. A complete IPT system using the ac process-ing pickup was tested on a 500-W lighting system and an efficiencyof 96% was obtained when delivering 500 W to multiple resistivelight bulbs.

Index Terms—AC-AC power conversion, lighting, magneticfields, phase control, power electronics, resonant power conversion.

NOMENCLATURE

C2 Tuning capacitance.IL Pickup inductor current.ILc Pickup inductor current in clamp mode.ILr Pickup inductor current in resonant mode.ILpn In-phase component of the fundamental or harmonic

of the pickup inductor current.ILqn Quadrature component of the fundamental or har-

monic of the pickup inductor current.L2 Pickup inductance.Q2 Quality factor of secondary resonant circuit.R2 Load resistor.tc Time circuit stays in the clamp mode.tr Time circuit stays in the resonant mode.Vc Tuning capacitor voltage.Vcr Tuning capacitor voltage in resonant mode.Vcpn In-phase component of the fundamental or harmonic

of the tuning capacitor voltage.Vcqn Quadrature component of the fundamental or har-

monic of the tuning capacitor voltage.Voc Pickup induced voltage.φ Controlled phase delay.ω Angular frequency of primary track current.

Manuscript received August 2, 2009; revised October 26, 2009. Currentversion published May 7, 2010. Recommended for publication by AssociateEditor S. Williamson.

H. H. Wu is with the Department of Electrical and Electronic Engineer-ing, The University of Auckland, Auckland 0602, New Zealand (e-mail:[email protected]).

J. T. Boys and G. A. Covic are with the Department of Electrical and Com-puter Engineering, The University of Auckland, Auckland 1142, New Zealand(e-mail: [email protected]; [email protected]).

Digital Object Identifier 10.1109/TPEL.2009.2037002

Fig. 1. IPT system.

I. INTRODUCTION

INDUCTIVE power transfer (IPT) systems are widely usedin many applications to deliver power to both mobile and sta-

tionary loads without any physical contact [1], [2]. Such systemshave a number of advantages, as they are unaffected by dirt, ice,water, and other chemicals and, are thereby, environmentallyinert and maintenance free [3]. In addition, these systems canalso operate in very clean environments such as clean rooms.Since there is galvanic isolation when delivering electric power,the operating environment is kept very clean compared to ametallic bus–bar system, where unacceptable levels of carbonresidue are generated due to the frictional contact between thecarbon brushes and the metallic platform [4]. High-power appli-cations of this technology include continuous power transfer topublic transport systems [5], materials handling systems [1], andcontact-less battery charging of electric vehicles [3], [6]. Typicallow-power applications include contact-less battery charging ofcell phones [7], [8] and biomedical implants [9]–[11].

An IPT system comprises a resonant power converter operat-ing at very low frequencies (VLF) in the range of 10–100 kHzand maintains a constant track current in the order of 10–200 Ain a track loop, as shown in Fig. 1. One or more secondary pickuploads may be placed in proximity to the track and receive powerinductively. In each pickup, an inductor, comprising a magneticcore with a high-frequency winding, is magnetically coupled tothe track, and tuned for resonance at the track frequency usingcompensation capacitors [4]. A switch-mode controller controlsthe power received by the pickup coil, and thereby, regulates theoutput voltage to the desired value to drive the load.

Presently, most IPT systems produce a controlled dc outputfor recharging batteries or driving motors. In these applications,the most common secondary pickup controller rectifies the acpower, which is then regulated using a dc shorting (decoupling)switch. Such a pickup controller has the advantage of simplecontrol circuitry and the ability to operate over a wide load range[1], [6], [12]. The detailed operation of the pickup controller canbe found in [1]. In order to power high-frequency ac loads suchas fluorescent lights or stage lights, an extra resonant converter

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1276 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 25, NO. 5, MAY 2010

Fig. 2. AC–DC–AC conversion topology.

or dc–ac pulsewidth modulation (PWM) inverter is requiredto produce a controllable ac source. One method of achievingthis is simply cascading the IPT pickup controller with a push–pull converter, as shown in Fig. 2. However, the addition of asecond converter is not ideal because of the large number ofcomponents required, which in turn increase cost. In addition,the two-stage conversion process has losses in each stage, whichreduce efficiency. Likewise, a dc–ac PWM inverter also has ahigh component count and even more switching losses than theresonant converter, due to a hard switching operation at thesehigh frequencies.

Another technique is to use primary-side control to implementpower regulation of the IPT system. This method sends feed-back signals such as output voltage and current of the secondarypickup back to the primary converter via a wireless communica-tion channel. Generally, primary-side control has two possiblemethods of realization—frequency control [13], [14] or pri-mary current control [15], [16]. However, primary-side controlcan only be applied to applications, where a single secondarypickup is used. For lighting systems with multiple secondarypickups, power regulation on the primary-side cannot be used,since regulating power on one pickup will affect the operation ofother pickups, which may be operating at different power levels.

Other secondary-side-control techniques directly regulatepower on the ac side to deliberately tune or detune the reso-nant tank circuit by adding extra reactance. One method usedto realize a variable reactance component is to use a capacitorbank and switch capacitors in and out of the resonant circuitto control output power [17]. But this system requires a bulkycapacitor bank that increases both cost and size of the overallsystem. A different technique to realize a variable reactancecomponent is to use a magnetic amplifier to produce a variableinductor [18]. Although this may vary the ac power directly, theuse of a variable inductor in the nonlinear region of the B–Hcurve can limit the efficiency of the overall system. In addition,the variable inductor is expensive to manufacture because it hasto manage the high resonant current without fully saturatingand also take into account sensitivity issues in the system. Oneother method to produce a variable reactance component is byswitching a fixed inductor under soft-switching conditions witha certain duty cycle, and hence, producing an equivalent variableinductor [19]. However, this method lacks the ability to regulatepower over a wide load range, as the minimum output voltageit can deliver at no load conditions is the open-circuit voltage ofthe pickup. Moreover, all the pickups that use tuning/detuningcontrollers for power regulation reflect a poor power factor (PF)

Fig. 3. AC processing pickup.

to the primary track of the IPT system and the primary powersupply has to handle the poor PF.

In this paper, a simple ac-processing-pickup controller thatcan provide controlled ac power over a wide resistive load rangeis proposed. The pickup is used as a light dimmer to control thepower delivered to a bank of multiple incandescent light bulbswhile achieving high-frequency soft-switching conditions,contactless energy transmission, electric isolation, and highefficiency.

The organization of this paper proceeds as follows. Section IIdescribes the circuit operation of the proposed pickup. InSection III, an exact analytical analysis is given using a combi-nation of piecewise linearized operating states. The normalizedgraphs on the pickup operating characteristics are outlined inSection IV. A design procedure and the experimental resultson a 500-W lighting system are given in Section V. Finally,Section VI gives conclusions based on key contributions of thepaper.

II. CIRCUIT OPERATION

The proposed ac processing pickup is shown in Fig. 3. Here,capacitor C2 is tuned to inductor L2 at the frequency of theprimary track current (I1) to form a resonant tank. The diodes(D1 and D2) and switches (S1 and S2) form an ac switch. Fromstandard IPT theory, a pickup coil placed on the primary trackwill have an open-circuit voltage (Voc) induced in it as follows:

Voc = jωMI1 (1)

where ω is the operating angular frequency, M is the mutualinductance, and I1 is the primary track current.

To illustrate the circuit’s operation, Fig. 4 shows the one-period operation of the ac processing pickup as a sequence oflinear circuit stages with each corresponding to a particularswitching interval, as illustrated in Fig. 5. Vg1 and Vg2 are thePWM gate signals that are driving switches S1 and S2 at 50%duty cycle with the same switching frequency as the IPT trackfrequency. Consider the situation where waveforms Vg1 and Vg2are controlled with a phase delay φ relative to the phase of Voc toclamp parts of the resonant capacitor voltage, as shown in Fig. 5.In mode 1 (M1 , 0 < t ≤ t1), S1 is turned off and S2 is turned on.The series diode D2 blocks any current flowing through S2 , asit is reverse biased. Under this condition, capacitor C2 resonateswith pickup inductance L2 like a parallel resonant tank and the

WU et al.: AC PROCESSING PICKUP FOR IPT SYSTEMS 1277

Fig. 4. Operating modes of ac processing pickup.

capacitor voltage reaches a peak value and returns back to zero.When the capacitor voltage reaches zero, the circuit enters mode2 (M2 , t1 < t ≤ t2). Diode D2 and switch S2 start to conductand prevent the capacitor voltage building up in the negativedirection, as they ideally begin to conduct at zero volts, therebyclamping the output voltage at near zero. This causes S2 toclamp Vc for a phase known as the clamp phase (tc/ω) at thepoint where Vc changes from a positive to a negative voltage.In the beginning of mode 3 (M3 , t2 < t ≤ t3), S1 is turned onand S2 is turned off. Similar to M1 , the circuit operates like aparallel resonant tank and current flows into the load resistor. Inmode 4 (M4 , t3 < t ≤ T ), similar to M2 , the resonant cycle isterminated and the capacitor voltage is clamped. In this mode,the inductor current flows through the switch S1 and no currentflows through the load. After this mode, the circuit returns backto M1 , and thus repeating the switching process. In summary,the clamping action from the equivalent ac switch generates aphase shift between the open-circuit voltage and the capacitorvoltage waveform.

The ac processing pickup achieves zero-voltage-switching(ZVS) conditions. From Fig. 5, the resonant inductor currentstarts to flow through S2 at t1 , when there is no voltage acrossit, hence ZVS is achieved at turn on. When S2 is turned off att2 , the resonant capacitor in parallel with S2 forces the voltageacross S2 to increase slowly in the negative direction, while thecurrent through it decreases to zero. For most practical switches,the turn off is much faster than the rate of increase of the ca-pacitor voltage, so the dv/dt across the switch is relatively smalland ZVS is obtained at the switch-OFF condition. Switch S1operates in a similar manner and also achieves ZVS at turn on,while achieving a low dv/dt at turn off. Likewise, for diodes D1and D2 , low dv/dt is achieved at turn on, and ZVS is achieved atturn off. In summary, the switches and diodes in the ac process-ing pickup achieve soft switching. This gives the pickup desir-able characteristics such as low switching losses, low switchingstress, and reduced electromagnetic interference (EMI) levels.

III. PICKUP ANALYSIS

From the previous section, it can be seen that the phase shiftbetween Vc and Voc can be controlled by adjusting the controlled

Fig. 5. Operating waveforms of ac processing pickup.

phase delay φ. In this section, the phase delay φ is used in anexact analysis in the time domain to determine the character-istics of the circuit under steady-state operation. The basis ofthe analysis method is that the conditions existing in the circuit


Fig. 6. Waveform showing two operating states.

at the end of a particular switching period must be the initialconditions for the start of the next switching period, and theseconditions must be identical, thus allowing for steady-state res-onant operation.

The analysis procedure is greatly simplified based on thefollowing three assumptions.

1) The equivalent series resistance (ESR) of both capacitorC2 and inductor L2 are very small and are neglected. (Thisis because the resistive losses dissipated by the load areusually much larger.)

2) The switching action of the transistors and diodes areinstantaneous and lossless.

3) Capacitor C2 and inductor L2 are perfectly tuned forminga parallel resonant tank with the load.

Assuming the resonant tank is perfectly tuned

C2 =1

ω2L2. (2)

With reference to Fig. 6, the waveform can be separated intotwo operating states known as the resonant state and the clampstate.

A. Resonant State

During the resonant state, the capacitor voltage may be de-scribed as

d2Vcr

dt2+

1R2C2

dVcr

dt+

Vcr

L2C2=

Voc

L2C2sin(ωt + φ) (3)

where the subscript “r” of Vcr denotes the capacitor volt-age in the resonant mode. Considering the initial conditionVcr (t)|t=0 = 0 and (dVcr/dt)|t=0 = (−iL (0)/C2), the com-

plete solution of the (3) is

Vcr (t) =−Q2Voc cos (φ)

sin(θv )e−σt sin(ωf t − θv )

− Q2Voc cos(ωt + φ) (4)

where

Q2 =R2

ωL2; (5)

σ =1

2R2C2; (6)

ωf = ω

√1 − 1

4Q22; (7)

and

θv = tan−1(

ωf cos (φ)iL (0)/ (Q2

2Voc) − σ cos (φ) + ω sin (φ)

).

(8)In a similar way, considering the initial condition

iLr (t)|t=0 = iL (0) and (diLr/dt)|t=0 = (Voc sin φ/L2), thecomplete solution to the inductor current is as follows:

iLr (t) =iL (0) + β2

sin(θi)e−σt sin(ωf t − θi)

+Q2

2Voc

R2sin(ωt + φ) − Q2Voc

R2cos(ωt + φ) (9)

where

β2 =Q2Voc

R2(Q2 sin(φ) − cos (φ)) ; (10)

β3 =Q2Voc

R2(Q2 cos(φ) + sin (φ)) ; (11)

and

θi = tan−1(

−ωf (iL (0) + β2)−Voc sin(φ)/L2 + wβ3 + σ(iL (0) + β2)

).

(12)To investigate how long the circuit stays in the resonant state,

Vcr (t) = 0 can be substituted in (4), resulting in the followingexpression:

Vcr (tr ) = 0 (13)

where tr is the time the circuit operates in the resonant state.

B. Clamp State

During the clamp state, the inductor L2 and the voltage sourceis shorted and the current depends on Voc . By Kirchhoff’s volt-age law (KVL), the inductor current equation can now be writtenas

iLc(t) =Voc

L

∫ t

tz

sin(ωt + φ) dt + iL (tr ). (14)

Solving (14), the inductor current can be expressed as

iLc(t) = −Voc cos(ωt + φ)ωL2

+ i−L (15)


where

i−L = iL (tr ) +Voc cos(ωtr + φ)

ωL2. (16)

Because the resonant state and the clamp state are repeatedeach half cycle (with only a polarity change), the relationshipiL (0) = −iL (T/2) must hold. Hence, the capacitor voltage andinductor current may be represented as piecewise functions frommodes M1 to M4 as

Vc(t) =

Vcr (t) t0 ≤ t < t1

0 t1 ≤ t < t2

−Vcr (t) t2 ≤ t < t3

0 t3 ≤ t < T

(17)

iL (t) =

iLr (t) t0 ≤ t < t1

iLc(t) t1 ≤ t < t2

−iLr (t) t2 ≤ t < t3

−iLc(t) t3 ≤ t < T .

(18)

Fourier analysis can be performed on both the capacitor volt-age and inductor current waveforms to compute the harmonics.The in-phase and quadrature components of both the fundamen-tal and harmonics are given by

Vcpn =2ω

π

∫ π/ω

0Vc (t) cos (nωt) dt (19)

Vcqn =2ω

π

∫ π/ω

0Vc (t) sin (nωt) dt (20)

ILpn =2ω

π

∫ π/ω

0IL (t) cos (nωt) dt (21)

ILqn =2ω

π

∫ π/ω

0IL (t) sin (nωt) dt. (22)

By obtaining the in-phase and quadrature components ofthe fundamental inductor current amplitude with respect to thephase of Voc , the displacement PF (DPF) may be computedusing the formula

DPF = cos[φ + arctan

ILq1

ILp1

]. (23)

IV. PICKUP CHARACTERISTICS

The output voltage (or capacitor voltage) characteristics ofthe pickup is shown in Fig. 7 for different values of Q2 . Thenormalized output voltage is defined as the ratio of the outputvoltage over the open-circuit voltage. It can be seen that theoutput voltage asymptotically decreases as the controlled phasedelay φ increases from zero. The normalized output voltage canbe controlled from a maximum value of a parallel tuned pickup(Q2) to zero as φ increases for all load (Q2) conditions.

The normalized output current is shown in Fig. 8 for a range ofQ2 values. It can be seen that the output current of the pickup can

Fig. 7. Normalized rms output voltage versus controlled phase delay φ.

Fig. 8. Normalized rms output current versus controlled phase delay φ.

be controlled by φ as a controllable current source. Fig. 8 showsthat the output current stays approximately constant (or has verylittle variation) as the load resistance changes for pickups at highQ2 (5–10). Hence, this pickup demonstrates controllable currentsource behavior. For low Q2 values, the circuit is no longerconsidered as a current source, as the output current changeswith load (or Q2).

The output current–voltage characteristic is shown in Fig. 9.The current source behavior is again demonstrated, as the outputcurrent stays approximately constant for a given phase delay,irrespective of output voltage, as long as the output voltage isreasonably high.

The normalized DPF characteristic is shown in Fig. 10 atdifferent values of Q2 . For pickups with a Q2 above 4, it canbe seen that the DPF is nearly at unity under full-load con-ditions. The DPF drops rapidly for lower Q2 values even atfull-load conditions and this conforms with the traditional the-ory of parallel LC resonant circuits [1]. It can be seen that the


Fig. 9. Pickup output voltage current characteristics.

Fig. 10. DPF versus per-unit Power.

DPF decreases as the per-unit power decreases. The per-unitpower is defined as the ratio of the output power to the maxi-mum power of the pickup at each Q2 value. The DPF becomesmore lagging as φ increases to decrease output power. The lag-ging DPF corresponds to an increasing capacitive load reflectedonto the primary track, and the reflected var’s have to be sourcedby the primary power supply. Even if the pickup has low DPF atlower power levels, the primary power supply is not necessarilyoverstressed, as the overall power delivered has decreased sig-nificantly. Consequently, if a 500-W pickup with an efficiencyof 90% is assumed, the primary converter has to source 550 Wand 50 var’s to deliver 500 W to the load. To deliver 100 W,the primary converter only has to source 110 W and 200 varsat reduced DPF. As such, the overall power that needs to besourced has decreased significantly and the stress on the powersupply is lower despite reduced DPF.

Figs. 11 and 12 show the first four harmonics for the acprocessing pickup operating at a Q2 = 5 obtained from Fourier

Fig. 11. Harmonic components of output voltage as percentage of the maxi-mum fundamental value at Q2 = 5.

Fig. 12. Harmonic components of input current as percentage of the maximumfundamental value at Q2 = 5.

analysis for the capacitor (or output) voltage and inductor (orinput) current. In these figures, the amplitude of the harmonicsis expressed as a function of the fundamental component underfull-load conditions. It can be seen that the amplitudes of theharmonic components are relatively low as compared to thefundamental. The highest harmonic component for the outputvoltage and inductor current does not exceed 6.5% and 2% ofthe maximum fundamental component, respectively.

V. DESIGN PROCEDURE AND RESULTS

In this section, the design of an ac processing pickup for a500-W lighting system is described. The desired output voltageis 220 V and the equivalent load resistance of the light at themaximum power condition is 84 Ω. An asymmetrical S-shapedmagnetic inductor was chosen for the prototype pickup because


Fig. 13. Numerically calculated waveforms for (a) 100% power, (b) 50% power, and (c) 20% power.

TABLE IVOLTAGE AND CURRENT OF COMPONENTS AT RATED LOAD

of its higher output power with the same ferrite volume/lengthcompared to traditional magnetic pickup structures [20]. Thispickup has a measured Voc of 44.5 V and an inductance valueof 72.6 µH. The primary IPT converter uses an LCL topologyoperating at a fixed frequency of 38.4 kHz [21].

A. Design Procedure

The first step is to determine the tuning capacitance of thecircuit. From (2), the nominal tuning capacitance is 239 nF. Inthe prototype, the tuning capacitance is chosen to closely matchthe ideal nominal tuning capacitance. Using (5), Q2 of the circuitis 4.8 at maximum power.

Equations (17) and (18) are then used to solve for steady-stateoperating waveforms. Fig. 13 shows the calculated waveformsfor the circuit under 100%, 50%, and 20% output power. Fromthe top trace to the bottom trace, in descending order, the tracesare the capacitor voltage, inductor current, capacitor current, andS1 current, respectively. It can be seen that the capacitor voltageand inductor current waveform decrease as the controlled phasedelay is increased to decrease output power. To calculate themaximum rating conditions for the components, the controlledphase delay φ has to be set slightly above zero degrees in order toobserve the peak current through the switches and maximum rmsrating for the capacitor and inductor. The calculated peak andrms value of the voltage and current for the capacitor, inductor,and switch are listed in Table I. It can be concluded from Table Ithat the switches and diodes have to be rated for both 310 V and18 A at normal operation.

Fig. 14. Block diagram for controller.

B. Controller

The practical system setup and controller for the ac process-ing pickup are shown as a block diagram in Fig. 14. The phaseof Voc is measured using a separate phase sense coil L3 placedon the primary track to detect the phase of the track current,which is exactly 90 out of phase with the open-circuit volt-age (Voc ∝ I1 90). The controlled phase delay φ is set bya computer interface, while a microcontroller accordingly ad-justs the switch gate drive waveforms. The gate-control wave-forms are generated, as shown in Fig. 5. The pickup also op-erates with a closed-loop controller, where the output voltageis set to a desired value by the microcontroller. The micro-controller is configured to maintain the desired load voltageby adjusting φ in accordance with measurements of the outputvoltage.

C. Experimental Results

The ac processing pickup, as described earlier, was coupledto a small section of track (see Fig. 14) and used to drive an acload comprising a 500-W incandescent light bulb bank. Fig. 15shows the circuit waveforms for the ac processing pickup at


Fig. 15. Measured waveforms for (a) 100% power, (b) 50% power, and (c) 20% power with light bulb load.

Fig. 16. Output power versus φ.

100%, 50%, and 20% power when φ is set to 0, 49, and 75,respectively. From the top trace to the bottom trace, in descend-ing order, the traces are the capacitor voltage, inductor current,capacitor current, and switch current. The inductor current andcapacitor voltage are both sinusoidal having low distortion at100% power. The measurements as shown have very good cor-relation with the calculated waveforms in Fig. 13. The ampli-tudes of the measured waveforms are within 10% of the valuescalculated by (17) and (18). The capacitor and switch currentwaveforms each have square pulses either missing or present,as predicted by theory, with significant high-frequency compo-nents. Due to the 100-kHz bandwidth limitation of the currentprobe, the filtering effect results in oscillations in the capacitorand switch current, which do not exist in practice. Despite this,it is evident that the correlations between the calculated andexperimental waveforms are very good.

Fig. 16 shows that the output power can be controlled over awide load range by adjusting φ. Note that, although the controlrange is from 0 to 180, the phase controller only needs tochange between 0 and 120 to regulate power over the entireoperation range. The analytical results using (17) and (18) arealso plotted on the same figure for comparison purposes. Sincethe analytical analysis ignores the ESR losses in both the pickup

Fig. 17. DPF versus per-unit power.

inductor and tuning capacitor, and the losses in the switchesand diodes, the output power is higher than that obtained fromexperimental measurements. Despite this, the controller is quiteefficient and the difference between the two is not greater than10%.

Fig. 17 shows the DPF for both the calculated and measuredwaveforms. Similar to before, the DPF is measured at the ter-minals of the short length of track, as shown in Fig. 14. It canbe seen that both the measured and calculated DPF values arenear unity at high power levels. The measured PF is also addedfor comparison purposes. This PF is only marginally below theDPF at high output powers and starts to reduce faster than theDPF at lower output powers, as the harmonics in the inductorcurrent waveform increase. Although this pickup has low DPFand substantial harmonics at low power levels, the overall stressimposed on the primary power supply is relatively small at lightload.

An efficiency versus output power plot is shown in Fig. 18for both the overall IPT system and the ac processing pickupby itself. Referring to Fig. 14, the overall IPT system efficiencyis determined using measures of the dc input power to the pri-mary power supply and the ac output power from the secondarypickup. Similarly, the pickup efficiency is calculated using the


Fig. 18. Efficiency versus output power.

ac input power delivered to a short length of primary track thatthe secondary pickup is coupled upon and the ac output powerfrom the secondary pickup. The pickup efficiency measurementneglects the supply (LCL converter) power losses and gives amore meaningful measure of the conversion efficiency of thepickup itself. It can be seen that the efficiency of the pickup re-mains above 90% when the output power is more than 125 W tothe load. With a 500-W load, the efficiency of the ac processingpickup and the overall IPT system can reach as high as 96% and89%, respectively.

VI. CONCLUSION

This paper presents a new IPT pickup that has significantadvantages compared to traditional pickups that use ac–dc–acconversion topologies for producing a controllable ac outputvoltage. The output voltage of the ac pickup can be fully con-trolled while achieving ZVS conditions. At high Q2 value, theac pickup demonstrates controllable current source property,which may be desirable in lighting applications. Although thispickup has low DPF and substantial harmonics at low powerlevels, the overall stress imposed on the primary power supplyis relatively small. The ac processing pickup can be controlledover a wide load range for a 500-W lighting system and a max-imum efficiency of 96% was obtained.

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[18] J. U. Hsu, A. P. Hu, A. Swain, X. Dai, and Y. Sun, “A new contactlesspower pick-up with continuous variable inductor control using magneticamplifier,” in Proc. Int. Conf. Power Syst. Technol., 2006, pp. 1–8.

[19] A. P. Hu and S. Hussmann, “Improved power flow control for contactlessmoving sensor applications,” IEEE Power Electron. Lett., vol. 2, no. 4,pp. 135–138, Dec. 2004.

[20] G. A. J. Elliott, G. A. Covic, D. Kacprzak, and J. T. Boys, “A new concept:Asymmetrical pick-ups for inductively coupled power transfer monorailsystems,” IEEE Trans. Magn., vol. 42, no. 10, pp. 3389–3391, Oct. 2006.

[21] M. L. G. Kissin, C. Y. Huang, G. A. Covic, and J. T. Boys, “Detection ofthe tuned point of a fixed-frequency LCL resonant power supply,” IEEETrans. Power Electron., vol. 24, no. 4, pp. 1140–1143, Apr. 2009.

Hunter Hanzhuo Wu (S’05) received the B.E. de-gree (Hons.) in electrical and electronic engineeringfrom the University of Auckland, Auckland, NewZealand, in 2008, where he is currently working to-ward the Ph.D. degree.

His research interests include inductive (contact-less) power transfer systems and resonant powerconverters.


John T. Boys received the Ph.D. degree from theUniversity of Auckland, Auckland, New Zealand, in1962.

He was with SPS technologies for five years beforereturning to academia as a Lecturer with the Univer-sity of Canterbury, New Zealand. Since 1977, he hasbeen with the University of Auckland, where he iscurrently a Professor of electronics with the Depart-ment of Electrical and Computer Engineering, and isengaged in teaching and research. He has authoredor coauthored more than 100 papers in international

journals and holds more than 20 U.S. patents. His research interests includepower electronics and inductive power transfer.

Dr. Boy is a Fellow of the Royal Society of New Zealand and a DistinguishedFellow of the Institution of Professional Engineers New Zealand.

Grant Anthony Covic (S’88–M’89–SM’04) re-ceived the B.E. (Hons.) and the Ph.D. degreesfrom The University of Auckland, Auckland, NewZealand, in 1986 and 1993 respectively.

He was was a Full-time Lecturer in 1992, a SeniorLecturer in 2000, and an Associate Professor in 2007with the Department of Electrical and Computer En-gineering, The University of Auckland, where he iscurrently the Head of the power research cluster inthe Faculty of Engineering. He has authored or coau-thored more than 80 papers in international journals

and conferences and holds a number of patents in the field of inductive (con-tactless) power transfer (IPT). His research interests include power electronicsand IPT.

ipt lightng system

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