iet power electronics volume 6 issue 1 2013 [doi 10.1049%2fiet-pel.2012.0338] lin, c.-c.; wu, g.w.;...

www.ietdl.org

Published in IET Power ElectronicsReceived on 12th July 2012Revised on 21st September 2012Accepted on 29th October 2012doi: 10.1049/iet-pel.2012.0338

ISSN 1755-4535

Study of a non-isolated bidirectional DC–DC converterC.-C. Lin, L.-S. Yang, G.W. WuDepartment of Electrical Engineering, Far East University, Tainan City, TaiwanE-mail: [email protected]

Abstract: The study presents a non-isolated bidirectional DC–DC converter, which has simple circuit structure. The controlstrategy is easily implemented. Also, the synchronous rectifier technique is used to reduce the losses. The voltage gain of theproposed converter is the half and the double of the conventional bidirectional DC–DC buck/boost converter in the step-downand step-up modes, respectively. Therefore the proposed converter can be operated in wide-voltage-conversion range than theconventional bidirectional converter. The voltage stresses on the switches of the proposed converter are a half of the high-voltage side. In addition, the operating principle and steady-state analyses are discussed. Finally, a prototype circuit isimplemented to verify the performance of the proposed converter.

1 Introduction

Since the usage of the fossil fuel results in environmentalpollution, the clean energies become very important in theworld. In recent years, the renewable energy systems,including photo-voltaic systems, fuel-cell systems,wind-power generating systems, are developed rapidly.Because the renewable systems cannot provide a stablepower for user, the renewable energy systems and batterycan be utilised for the hybrid power systems. When therenewable energy systems cannot supply enough power forthe load, the battery must replenish insufficient power.Whereas the whole power of the renewable energy systemscannot be used completely by the load, the surplus energycan be used to charge the battery. Because the bidirectionalDC–DC converters can transfer the power between two DCsources in either direction, these converters are widely usedfor renewable energy hybrid power systems, hybrid electricvehicle energy systems and uninterrupted power supplies.The topologies of these converters have the isolated andnon-isolated types for different applications. The isolatedtypes include the flyback type [1, 2], forward-flyback type[3, 4], half-bridge type [5–7] and full-bridge type [8–10].These converters can achieve large voltage gain byadjusting the turns ratio of the transformer. Thebidirectional flyback converter has the advantages ofsimple structure and easy control. Since theleakage-inductor energy cannot be recycled, the switches ofthis converter suffer from high-voltage stresses. Thus, thisconverter is applied for low-power applications. In order torecycle the leakage-inductor energy and to reduce thevoltage stresses on the switches, the energy-regenerationtechniques are presented to clamp the voltage stressesand to increase the conversion efficiency [2–4]. Thenon-isolated types include the multi-level type,switched-capacitor type, cuk/cuk type, sepic/zetatype, buck-boost type, coupled-inductor type, three-level

30& The Institution of Engineering and Technology 2013

type and conventional buck/boost type. In multi-level andswitched-capacitor types, if large voltage gain must beprovided, more switches and capacitors are required[11–15]. Also, the control circuits of these types arecomplicated. For the cuk/cuk and sepic/zeta types, theconversion efficiency are low because these converters arecombined of two power stages [16, 17]. In addition, theseconverters cannot provide wide voltage-conversion range.The circuit structure of the buck-boost type is very simple,but this converter cannot operate in widevoltage-conversion range [18]. The coupled-inductor typescan achieve large voltage gain by adjusting the turn ratioof the coupled inductor [19, 20]. However, the circuitconfigurations are complicated. For the three-level type, thevoltage stresses on the switches are a half of theconventional bidirectional DC–DC buck/boost type[21–23]. Nevertheless, the voltage-conversion range in thisconverter is narrow. The conventional bidirectional DC–DCbuck/boost converter has simple circuit structure, as shownin Fig. 1a [24–26]. However, the step-up voltage gain islimited because of the effect of the switches and theequivalent series resistance (ESR) of the inductors andcapacitors. Thus, the conventional converter is not suitablefor wide voltage-conversion applications.Fig. 1b shows the proposed bidirectional DC–DC

converter, which has the merits of simple topology andcontrol strategy. The voltage-conversion range of thisconverter is wide, than the conventional bidirectional buck/boost converter. The operating principles and steady-stateanalyses in the step-up and step-down modes are describedin Sections 2 and 3, respectively. Some conditions areassumed as: (i) The ON-state resistance RDS(ON) of theswitches and the ESR of the capacitors are ignored. (ii) Thecapacitors CH1, CH2, and CL are large enough, andthe voltages across the capacitors can be treated as constant.(iii) The capacitance of the capacitors CH1 and CH2 areequal. Thus, VH1 = VH2 = VH/2.

IET Power Electron., 2013, Vol. 6, Iss. 1, pp. 30–37doi: 10.1049/iet-pel.2012.0338

www.ietdl.org

2 Step-down mode of the proposedconverter

In the step-down mode, the equivalent circuit of the proposedconverter is shown in Fig. 2. The pulse-width modulation(PWM) technique is used to control the switches S1 and S4.The switches S2 and S3 are used for the synchronousrectifiers. Fig. 3 shows some typical waveforms incontinuous-conduction-mode (CCM) operation. Theoperating principles and steady-state analyses are describedas follows:(1) Mode 1, [t0, t1]: The switches S1 and S3 are turned on

and the switches S2 and S4 are turned off. Meantime, theswitch S3 is used for the synchronous rectifier. Thecurrent-flow path of the proposed converter is shown inFig. 4a. The energy of the high-voltage side VH1 istransferred to the inductor L1, capacitor CL, and load RL.Thus, the voltage across the inductor L1 is given by

vIL1 =VH

2− VL (1)

The current through the inductor L1 is obtained as

iIL1 (t) = iL1 t0( )+ 1

L1

VH

2− VL

( )t − t0( )

(2)

Fig. 2 Equivalent circuit of the proposed converter in thestep-down mode

Fig. 1 Conventional and proposed bidirectional DC–DCconverter

a Conventional bidirectional buck/boost converterb Proposed bidirectional converter


(2) Mode 2, [t1, t2]: The switches S2 and S3 are turned onand the switches S1 and S4 are turned off. Meantime, theswitches S2 and S3 are used for the synchronous rectifiers.The current-flow path of the proposed converter is shown inFig. 4b. The energy stored in the inductor L1 is released tothe capacitor CL and load RL. Thus, the voltage across theinductor L1 is found to be

vIIL1 = −VL (3)

The current through the inductor L1 is derived as

iIIL1 (t) = iL1 t1( )− VL

L1t − t1( )

(4)

(3) Mode 3, [t2, t3]: The switches S2 and S4 are turned on andthe switches S1 and S3 are turned off. Meantime, the switch S2is used for the synchronous rectifier. The current-flow path ofthe proposed converter is shown in Fig. 4c. The energy of thehigh-voltage side VH2 is transferred to the inductor L1,capacitor CL, and load RL. Thus, the voltage across theinductor L1 is obtained as

vIIIL1 =VH

2− VL (5)

Fig. 3 Typical waveforms of the proposed converter with CCMoperation in the step-down mode


www.ietdl.org

The current through the inductor L1 is given as

iIIIL1 (t) = iL1 t2( )+ 1

L1

VH

2− VL

( )t − t2( )

(6)

(4) Mode 4, [t3, t4]: This operation principle is the same as themode 2. Thus, the voltage across the inductor L1 isdetermined as follows

vIVL1 = −VL (7)

The current through the inductor L1 is derived as

iIVL1 (t) = iL1 t3( )− VL

L1t − t3( )

(8)

By using the voltage-second balance principle on the inductorL1, one can obtain

∫DTs/2( )

0

vIL1 dt +∫(1−D)Ts/2( )

0

vIIL1 dtx +∫DTs/2( )

0

vIIIL1 dt

+∫(1−D)Ts/2( )

0

vIVL1 dt = 0 (9)

Fig. 4 Current-flow path of the proposed converter in thestep-down mode

a Mode 1b Modes 2 and 4c Mode 3


Substituting (1), (3), (5), and (7) into (9), the voltage gain isgiven by

Mstep−down =VL

VH

= D

2(10)

The voltage gains of the proposed converter and theconventional bidirectional converter are shown in Fig. 5. Ascan be seen, the voltage gain of the proposed converter is ahalf of the conventional converter.Some waveforms in boundary-conduction-mode (BCM)

operation are shown in Fig. 6. The peak value of theinductor current iL1 is found to be

IL1p =DTs2L1

VH

2− VL

( )(11)

By using the ampere–second balance principle on thecapacitor CL, the following equation can be obtained as

IcL = (1/2) DTs/2( )+ (1− D)Ts/2

( )[ ]IL1p2− ILTs

Ts= 0

(12)

Fig. 5 Voltage gains of the proposed converter and conventionalconverter in the step-down mode

Fig. 6 Some waveforms of the proposed converter with BCMoperation in the step-down mode


www.ietdl.org
Consequently, the following equation can be derived as
IL1p = 2IL = 2VL

RL

(13)

Then, the normalised inductor time constant is defined as

tLL ;L1

RLTs= L1fs

RL

(14)

Substituting (10), (11) and (14) into (13), the boundary of τLLis obtained as follows:

tLL,B = 1− D

4(15)

Using (15), the curve of τLL,B is plotted in Fig. 7. It is seen thatthe proposed converter in the step-down mode is operated inCCM at τLL > τLL,B.

3 Step-up mode of the proposed converter

In the step-up mode, the equivalent circuit of the proposedconverter is shown in Fig. 8. The PWM technique is usedto control the switches S2 and S3. The switches S1 and S4are used for the synchronous rectifiers. Fig. 9 shows sometypical waveforms in CCM operation. The operatingprinciples and steady-state analyses are described as follows:

Fig. 7 Boundary condition of the proposed converter in thestep-down mode

Fig. 8 Equivalent circuit of the proposed converter in the step-upmode


(1) Mode 1, [t0, t1]: The switches S2 and S3 are turned onand the switches S1 and S4 are turned off. The current-flowpath of the proposed converter is shown in Fig. 10a. Theenergy of the low-voltage side VL is transferred to theinductor L1. The capacitors CH1 and CH2 are stacked todischarge for the load RH. Thus, the voltage across theinductor L1 is given by

vIL1 = VL (16)


iIL1 (t) = iL1 t0( )+ VL

L1t − t0( )

(17)

(2) Mode 2, [t1, t2]: The switches S1 and S3 are turned on andthe switches S2 and S4 are turned off. Meantime, the switch S1is used for the synchronous rectifier. The current-flow path ofthe proposed converter is shown in Fig. 10b. The energies ofthe low-voltage side VL and inductor L1 are series to releasetheir energies to the capacitor CH1. The capacitors CH1 andCH2 are stacked to discharge for the load RH. Thus, thevoltage across the inductor L1 is found to be

vIIL1 = VL −VH

2(18)

Fig. 9 Typical waveforms of the proposed converter with CCMoperation in the step-up mode


www.ietdl.org

The current through the inductor L1 is given as

iIIL1 (t) = iL1 t1( )+ 1

L1VL −

VH

2

( )t − t1( )

(19)

(3) Mode 3, [t2, t3]: The operation principle is the same as themode 1. Thus, the voltage across the inductor L1 is derived as

vIIIL1 = VL (20)


iIIIL1 (t) = iL1 t2( )+ VL

L1t − t2( )

(21)

(4) Mode 4, [t3, t4]: The switches S2 and S4 are turned on andthe switches S1 and S3 are turned off. Meantime, the switch S4is used for the synchronous rectifier. The current-flow path ofthe proposed converter is shown in Fig. 10c. The energies ofthe low-voltage side VL and inductor L1 are series to releasetheir energies to the capacitor CH2. The capacitors CH1 andCH2 are stacked to discharge for the load RH. Thus, thevoltage across the inductor L1 is found to be

vIVL1 = VL −VH

2(22)


iIVL1 (t) = iL1 t3( )+ 1

L1VL −

VH

2

( )t − t3( )

(23)

Fig. 10 Current-flow path of the proposed converter in the step-upmode

a Modes 1 and 3b Mode 2c Mode 4


By using the voltage–second balance principle on theinductor L1, the following equation is derived as

∫DTs( )/2( )

0

vIL1 dt +∫1−D)Ts/2( )

0

vIIL1 dt

+∫DTs/2( )

0

vIIIL1 dt +∫1−D)Ts/2( )

0

vIVL1 dt = 0

(24)

Substituting (16), (18), (20), and (22) into (24), the voltagegain is given by

Mstep−up =VH

VL

= 2

1− D(25)

Fig. 11 shows the voltage gains of the proposed converter andthe conventional bidirectional converter. One can see that thevoltage gain of the proposed converter is double of theconventional bidirectional converter.

Fig. 12 shows some waveforms in BCM operation. Thepeak value of the inductor current iL1 is given by

IL1p =DTsVL

2L1(26)

By using the ampere-second balance principle on thecapacitor CH1, one can obtain the following equation

IcH1 =(1/2) (1− D)Ts/2

( )IL1p − IHTs

Ts= 0 (27)

Thus

IL1p =4IH

1− D= 4VH

(1− D)RH

(28)

Then, the normalised inductor time constant is defined as

tLH ;L1

RHTs= L1fs

RH

(29)

Fig. 11 Voltage gains of the proposed converter and conventionalconverter in the step-up mode


www.ietdl.org

Substituting (25), (26) and (29) into (28), the boundary of τLHcan be determined as follows

tLH,B = D(1− D)2

16(30)

Fig. 12 Some waveforms of the proposed converter with BCMoperation in the step-up mode


The curve of τLH,B is depicted in Fig. 13. If τLH is larger thanτLH,B, the proposed converter in the step-up mode is operatedin CCM.

4 Experimental results

In order to illustrate the performance of the proposedconverter, a prototype circuit is implemented in thelaboratory. The electric specifications and circuitcomponents are chosen as VH = 200 V, VL = 24 V, fs = 50kHz, Po = 200 W, CH1 =CH2 = CL = 100 μF. The switchesS1 – S4 are selected to be IXTQ96N20P. In the step-downmode, the voltage gain M is equal to 0.12. SubstitutingM = 0.12 into (10), the duty ratio D is derived as 0.24.Substituting D = 0.24 into (15), the boundary normalised

Fig. 13 Boundary condition of the proposed converter in thestep-up mode

Fig. 14 Circuit diagram of the proposed converter with control circuit in the step-down mode


www.ietdl.org

inductor time constant τLL,B is obtained as 0.19. One sets thatthe proposed converter is operated in CCM from 10% of thefull load, namely RL = 28.8 Ω. When τLL is larger than τLL,B,the proposed converter is operated in CCM. Using (14), theinductor L1 is found to be

tLL = L1fsRL

= L150 k

28.8. 0.19

L1 . 109 mH

In the step-up mode, the voltage gain M is equal to 8.33.Substituting M = 8.33 into (25), the duty ratio D is derivedas 0.76. Substituting D = 0.76 into (30), τLH,B is obtained as0.00273. One sets that the proposed converter is operated inCCM from 10% of the full load, namely RH = 2000 Ω.When τLH is larger than τLH,B, the proposed converter isoperated in CCM. Using (29), the inductor L1 is given by

tLH = L1fsRH

= L150 k

2000. 0.00273

Fig. 15 Experimental waveforms of the proposed converter in thestep-down mode

a vgs1, vS1, and iS1b vgs2, vS2 , and iS2c VL and iL1


L1 . 109 mH

Therefore the inductor L1 is selected to be 110 μH.The circuit diagram of the proposed converter with control

circuit in the step-down mode is shown in Fig. 14. Someexperimental waveforms in the step-down and step-upmodes are shown in Figs. 15 and 16, respectively. FromFigs. 15a, b, 16a and b, one can see that the voltagestresses on the switches S1 and S2 are equal to VH/2. Also,the waveforms of the switch current iS1 and iS2 agree withthe analysis of operating principle. Fig. 15c shows thewaveforms of the low-voltage side VL and inductor currentiL1 . It can be seen that the low-voltage side VL is wellregulated at 24 V and the proposed converter is operated inCCM in the step-down mode. The waveforms of thehigh-voltage side VH1, VH2, VH, and inductor current iL1 areshown in Fig. 16c. One can see that the high-voltage sideVH is well regulated at 200 V and VH1 = VH2 = VH/2. Themeasured efficiencies of the proposed converter are shownin Fig. 17. As can be seen, the measured efficiencies arearound 92.3–94.8% in the step-down mode and are around91.2–94.1% in the step-up mode.

Fig. 16 Experimental waveforms of the proposed converter in thestep-up mode

a vgs1, vS1 , and iS1b vgs2, vS2 , and iS2c VH1, VH2, VH, and iL1


www.ietdl.org

5 Conclusions

A nonisolated bidirectional DC–DC converter is presented inthis paper. The topology of this converter is modified from theconventional buck/boost converter. The control strategy ofthis converter is implemented easily. The proposedconverter can be operated in wide voltage-conversion rangethan the conventional converter. In addition, the voltagestresses on the switches are a half of the high-voltage side.From the experimental results, one can see that thewaveforms agree with the operating principle andsteady-state analyses. Also, the efficiencies are around92.3–94.8% in the step-down mode and are about91.2–94.1% in the step-up mode.

6 References

1 Venkatesan, K.: ‘Current mode controlled bidirectional flybackconverter’. Proc. IEEE PESC, 1989, pp. 835–842

2 Chen, G., Lee, Y.S., Hui, S.Y.R., Xu, D., Wang, Y.: ‘Actively clampedbidirectional flyback converter’, IEEE Trans. Ind. Electron., 2000, 47,(4), pp. 770–779

3 Huber, L., Jovanovic, M.M.: ‘Forward-flyback converter withcurrent-doubler rectifier: analysis, design, and evaluation results’,IEEE Trans. Power Electron., 1999, 14, (1), pp. 184–192

4 Zhang, F., Yan, Y.: ‘Novel forward–flyback hybrid bidirectionalDC–DC converter’, IEEE Trans. Ind. Electron., 2009, 56, (5),pp. 1578–1584

Fig. 17 Measured efficiency of the proposed converter

a Step-down modeb Step-up mode


5 Li, H., Peng, F.Z., Lawler, J.S.: ‘A natural ZVS medium-powerbidirectional DC–DC converter with minimum number of devices’,IEEE Trans. Ind. Appl., 2003, 39, (2), pp. 525–535

6 Peng, F.Z., Li, H., Su, G.J., Lawler, J.S.: ‘A new ZVS bidirectionalDC–DC converter for fuel cell and battery Application’, IEEE Trans.Power Electron., 2004, 19, (1), pp. 54–65

7 Lin, B.R., Huang, C.L., Lee, Y.E.: ‘Asymmetrical pulse-widthmodulation bidirectional DC–DC converter’, IET Power Electron.,2008, 1, (3), pp. 336–347

8 Mi, C., Bai, H., Wang, C., Gargies, S.: ‘Operation, design and control ofdual H-bridge-based isolated bidirectional DC–DC converter’, IETPower Electron., 2008, 1, (4), pp. 507–517

9 Zhao, C., Round, S.D., Kolar, J.W.: ‘Full-order averaging modelling ofzero-voltage-switching phase-shift bidirectional DC–DC converters’,IET Power Electron., 2010, 3, (3), pp. 400–410

10 Naayagi, R.T., Forsyth, A.J., Shuttleworth, R.: ‘High-powerbidirectional DC–DC converter for aerospace applications’, IEEETrans. Power Electron., 2012, 27, (11), pp. 4366–4379

11 Khan, F.H., Tolbert, L.M., Webb, W.E.: ‘Hybrid electric vehicle powermanagement solutions based on isolated and nonisolated configurationsof multilevel modular capacitor-clamped converter’, IEEE Trans. Ind.Electron., 2009, 56, (8), pp. 3079–3095

12 Monge, S.B., Alepuz, S., Bordonau, J.: ‘A bidirectional multilevelboost–buck DC–DC converter’, IEEE Trans. Power Electron., 2011,26, (8), pp. 2172–2183

13 Peng, F.Z., Zhang, F., Qian, Z.: ‘A magnetic-less DC–DC converter fordual-voltage automotive systems’, IEEE Trans. Ind. Appl., 2003, 39, (2),pp. 511–518

14 Lee, Y.S., Chiu, Y.Y.: ‘Zero-current-switching switched-capacitorbidirectional DC–DC converter’, IEE Proc. Inst. Elect. Eng. Electr.Power Appl., 2005, 152, (6), pp. 1525–1530

15 Ko, Y.P., Lee, Y.S., Chao, W.H.: ‘Analysis, design and implementationof fuzzy logic controlled quasi-resonant zero-current switchingswitched-capacitor bidirectional converter’, IET Power Electron.,2011, 4, (6), pp. 683–692

16 Jose, P., Mohan, N.: ‘A novel ZVS bidirectional Cuk converter for dualvoltage systems in automobiles’. Proc. IEEE IECON, 2003, pp. 117–122

17 Kim, I.D., Paeng, S.H., Ahn, J.W., Nho, E.C., Ko, J.S.: ‘Newbidirectional ZVS PWM sepic/zeta DC–DC converter’. Proc. IEEEISIE, 2007, pp. 555–560

18 Caricchi, F., Crescimbini, F., Capponi, F.G., Solero, L.: ‘Study ofbi-Directional buck-boost Converter topologies for application inelectrical vehicle motor drives’. Proc. IEEE APEC, 1998, pp. 287–293

19 Wai, R.J., Duan, R.Y.: ‘High-efficiency bidirectional converter forpower sources with great voltage diversity’, IEEE Trans. PowerElectron., 2007, 22, (5), pp. 1986–1996

20 Wai, R.J., Duan, R.Y., Jheng, K.H.: ‘High-efficiency bidirectional dc–dcconverter with high-voltage gain’, IET Power Electron., 2012, 5, (2),pp. 173–184

21 Jin, K., Ruan, X., Yang, M., Xu, M.: ‘A hybrid fuel cell power system’,IEEE Trans. Ind. Electron., 2009, 56, (4), pp. 1212–1222

22 Grbovic, P.J., Delarue, P., Moigne, P.L., Bartholomeus, P.: ‘Abidirectional three-level DC–DC converter for the ultracapacitorapplications’, IEEE Trans. Ind. Electron., 2010, 57, (10), pp. 3415–3430

23 Jin, K., Yang, M., Ruan, X., Xu, M.: ‘Three-level bidirectional converterfor fuel-cell/battery hybrid power system’, IEEE Trans. Ind. Electron.,2010, 57, (6), pp. 1976–1986

24 Lee, W.C., Yoo, C.G., Lee, K.C., Cho, B.H.: ‘Transient currentsuppression scheme for bi-directional DC/DC converters in 42 Vautomotive power systems’, J. Power Electron., 2009, 9, (4),pp. 517–525

25 Samosir, A.S., Yatim, A.H.M.: ‘Implementation of dynamic evolutioncontrol of bidirectional DC–DC converter for interfacing ultracapacitorenergy storage to fuel-cell system’, IEEE Trans. Ind. Electron., 2010,57, (10), pp. 3468–3473

26 Do, H.L.: ‘Nonisolated bidirectional zero-voltage-switching DC–DCconverter’, IEEE Trans. Power Electron., 2011, 26, (9), pp. 2563–2569


iet power electronics volume 6 issue 1 2013 [doi 10.1049%2fiet-pel.2012.0338] lin, c.-c.; wu, g.w.;...

Documents