Experimental Verification of DC/DC Converter with Full-Bridge Active Rectifier

Andrei Blinov\ Volodymyr Ivakhn02, Volodymyr Zamaruev2, Dmitri Vinnikovl, Oleksandr Husev3 ITallinn University of Technology, Estonia; 2National Technical University "Kharkiv Poly technical Institute," Ukraine;

3Chernihiv State Technological University, Ukraine andrei.blinov@ieee.org

Abstract- The half-bridge converter with a phase-shifted active rectifier bypasses the traditional hard-switched half-bridge converter with a diode rectifier regarding to high switching losses and parasitic oscillations in the inverter. The presented galvanically isolated step-down DC/DC converter with an improved control algorithm for the full-bridge active rectifier features reduced energy circulation and switching losses. The performance can be improved under wide input voltage and load variations. The advantages of the converter were verified with a 1 kW converter prototype and the test results were in full agreement with the expected waveforms. The presented steady­state operation principle and mathematical analysis of the converter based on the simulation and experimental results can be used as design guidelines for component and parameter estimation in practical applications.

I. INTRODUCTION

The phase-shifted synchronous rectifier concept is a well­known method to reduce the ringing, increase the efficiency and achieve ZVS (zero voltage switching) of converter switches. The other advantages are the possibility of using non-dissipative capacitive snubbers in the inverter and constant frequency operation, allowing for simple control of the converter. Generally, these converters comprise a half- or a full-bridge inverter, a high-frequency transformer and a rectifier [1] [2]. The rectifier part could be classified as: full­bridge, central-tapped and current-doubler [3] .

In converters with a synchronous rectifier the output voltage is generally controlled by the varying delay time between the turn-on of the switches in the rectifier and the turn-on of the IGBTs in the inverter. At the beginning of each half-period, the transformer current will have the same direction as in the previous one and will only change it when

C2

I 1------ -------� I

iT7) DTi � L _____ _______ J T

I

-------, 1------ --------1 I

!TBJ DB! ..iT ' , , , l..-_____ _ ______ 1 I I

the other switch pair in the rectifier turns on. Therefore, at the beginning of each half-period the current will flow through the freewheeling diode of the IGBT module to be turned on next, allowing the ZVS of both inverter transistors. This switching algorithm allows non-dissipative capacItlve snubbers to be used in the inverter and the inductive ones in the rectifier. The role of the latter ones could be performed by the transformer leakage inductance. The disadvantage of such control algorithms is the presence of the intervals of energy return from the load to the power supply (time intervals of the opposite sign of the current and voltage of the primary transformer winding during the delay time). If the energy return is not possible, there will be an increase of the input voltage of the inverter and a deviation of the midpoint potential of the capacitor input voltage divider. Moreover, energy circulation corresponds to the generation of the reactive power, resulting in the reduction of the converter power factor and the efficiency due to increased conduction losses. In the case of high input voltage (large periods of energy return), the effects of these drawbacks could be inacceptable. One of the ways to reduce such effects is to increase the capacitance value of the input voltage divider, however, that leads to an increase in the dimensions and cost of the converter.

The current paper presents an analytical and experimental evaluation of the converter with a full-bridge phase-shifted active rectifier based on reverse blocking (RB) switches (Fig. 1), reported by the authors in [4]. The aim of the topology described is to minimise switching losses, energy circulation and intervals of energy return from the load to the power supply.

52

Urect Uout

53

Fig. 1. Investigated half-bridge converter circuit with controlled RB switches at the secondary side.

978-1-4673-2421-2/12/$31.00 ©2012 IEEE 5179

II. NOVEL CONTROL METHOD

To overcome the disadvantages of a conventional phase­shifted synchronous rectifier, the control algorithm of the rectifier switches could be modified, at the same time keeping the advantages of the reference phase-shifted control algorithm. A similar concept for the full-bridge converter was first introduced in [5] . The proposed algorithm provides phase-shifted control whereas practically no energy is returned into the power supply. This is achieved by introducing two additional switching states of the rectifier switches. The voltage and current waveforms of the converter are presented in Fig. 2 with the associated switching states shown in Fig. 3. The following events during the switching half-period could be distinguished: to-t] - transistor TT and switches Sf, S3 are conducting

(Fig. 3a). The voltage of the transformer primary winding is +U;nI2, and at the output, the voltage of the rectifier is + Uno_so Switch S4 could be turned off with ZCS.

trt2 - transistor TT is turned off. The transformer voltage and Urect change their sign as the snubber capacitors are recharged (Fig. 3b).

trt3 - the snubber capacitors are recharged and the transformer primary voltage reaches -U;nI2. The freewheeling diode DB opens, starting the energy return interval and the transistor TB could be opened with ZVS from now on. The output voltage of the rectifier is now -UTr-s (Fig. 3c). Until the moment t3 the processes do not differ from the corresponding ones in the conventional phase-shifted synchronous rectifier.

tTt� - switch S2 is now opened (Fig. 3d). The voltage across the transformer secondary drops to zero and the transformer current decreases gradually, as the load current transfers from Sl to S2 with dildt limited by the circuit equivalent inductance.

trt5 - the energy return interval is over, the negative voltage -Ul'r-s is applied to the transformer secondary and switch Sf, which can be turned off now with ZCS. Only small magnetising current is flowing through the transformer primary. The load current freewheels throughS3, S2, Ljand the load (Fig. 3e).

LJ

53

(a) Lr

(b)

(e)

t5-t6 - switch S4 is opened (Fig. 3t). The voltage across the transformer secondary drops to zero and the transformer and the TB current increase gradually, as the load current transfers from S3 to S4 with dildt limited by the circuit equivalent inductance.

The processes are then repeated with the difference that the transistors and the primary side diodes replace each other, so do switches Sf, S2 and S3, S4.

+UTU

frr_s

-UTr_s UTT

frT

UTB

ITB

151 152 51 52 51 t 154

53 153 t Urect t

.",-to/ t�/··t2,· -b: ',t4 \\:t}"".t6 I ta I ttr1 I trev I ttr2 fFR4 ttr3 1

Fig. 2. Generalised operation principle of a converter with a phase-shifted active full-bridge rectifier using the modified control algorithm

LJ Lr

52

Cj Cr

53

(c) Lt

C{ I)

(f) Fig. 3. Operation modes of the converter with the modified phase-shifted active full-bridge rectifier.

5180

The control of the output voltage can be achieved by varying the current freewheeling duration ts (time interval tT t6). The energy return interval (time interval trt3) could be constant and should be kept as short as possible in order to maintain high power factor and reduce conduction losses.

III. DESIGN CONSIDERATIONS

This section provides guidelines for the selection of the parameters for the experimental converter with the phase­shifted active full-bridge rectifier (AFBR). In order to simplify the calculations it is assumed that the opponents are lossless, the input capacitors and the transformer magnetising inductance are large enough and therefore the input voltage ripple and the magnetising current are negligible. For the introduced modified control algorithm, six PWM channels are used. The inverted control signals of S1 and S2 are shifted by the ratio Dy relative to the tum-off of inverter switches (Fig. 4). The inverted control signals for S3 and S4 are shifted by the ratio Ds relative to S1 and S2. Rectifier channels have a constant duty cycle and the output voltage is regulated by varying the ratio Ds. The values of Dr and D, are defined as follows:

D -t3 -tl

Y- T '

D = t6 -t3 = ts S T T

(1)

(2) The active state duty cycle Do of the converter can be

calculated by

D = � = T,w - 2 . t, - 2 . tid - 2 . trel,

a T 2.T,w (3)

The output voltage can be expressed by the following equation:

(4)

and the amplitude voltage across the transformer secondary is

t .. � TT !! 1,-------, TB J I

51 ! 1 52 11

[� t

�

C: I t

� t 53 " 1 , �

TT

TB

51

52

53

(5)

t .. � I I [� t

� J I 1 C:

I t

� 11 t �

54 10 I I: 54 -++---t----'----li c--J c=J t JI

I I c=�t h t

U.a, -+iiU--+-' ----1Urr----'-----+u.--.�t u •• ,' ---!H'U---+- ---1r-r------'--+r--� ..

0 � ro u t t."

• • T"

C Tow

(a) (b)

I

Fig, 4, Proposed control algorithm realisation for the converter with a phase­shifted active rectifier at maximal (a) and minimal (b) input voltages

The average and maximum collector currents of inverter transistors are calculated by

_ N, I'll ripple IC',! .) -IC'+ - '---' ,max

. N 2 p

(6)

(7)

where LJIripple is the filter inductor Lf peak-to-peak current ripple.

The inverter switches require a certain dead time td to recharge snubber capacitors to maintain the ZVS condition, hence the following expression must be satisfied:

> _ _ 2_' L_!!.!.,!in_' _C,,--s

td -tlrl-IC(max)

(8)

According to this equation, the most demanding point is at the minimum load and the maximum input voltage.

Certain time is required for the current of the rectifier switches to fall to zero during the natural commutation, therefore the rectifier switches should operate with a duty cycle higher than 0.5 to maintain the ZCS condition and exclude the situation when only one transistor in the rectifier is turned on. The additional time required can be equal for all the rectifier switches and is estimated from flr2:

(9)

where Lb is the equivalent inductance, which is mainly represented by the leakage inductance of the transformer secondary winding.

To estimate the parameters of the output filter inductor it is assumed that the current of the inductor is continuous:

where kL is the relative current ripple of the output filter inductor :

10

g .. � 8 -5

-10 50

600

450

� 300

f 150

-150 50

100

100 Time (J-ls)

150

150

200

200

25

20

g 15

� 10

� 5

-5

120 80

� 40 ::, 0 � -40 >

-80 -120

50

trev

50

100 150 200

Sl transistor part

100 Time (Ils)

150 200

(a) (b) Fig, 5, Simulated voltage and current waveforms of the TT IGBT module (a) and S I switch (b)

5181

0,8

J' 0,6

---, 0,4 ;;;,0

0,2

° ° 0,1 0,2 0,3 0,4

D, Fig. 6. Simulated output voltage regulation characteristics (D,,=0.025).

10 25

20

$ 15

� <'

0 ';: 10 =

8 1: 5 -5 8

-10 -5 50 100 150 200 50 100 150 200

500 120

400

E 300 80

E . � 200 � 40

� 100 J3 � -

-100 -40 50 100 150 200 50 100 150 200

Time O.1s) Time (Ils) (a) (b)

Fig. 7. Experimental voltage and current waveforms of the TT IGBT module

(a); SI transistor (b) (U;n=350 V;f,w=lO kHz, 800 W load).

kr = jjJripple ' Uout (11) , �JUt The capacitance of the output filter capacitor can be approximated as:

jjJripple . ta _ ta . kL . Pout (12) Cf = _...!...!...._-. t1U ripple kU . U(;u{ where L1 UOllf is the peak-to-peak output voltage ripple and

ku is the relative voltage ripple.

ku = t1UOU{ , (13) Uout

The proposed control concept was simulated using PSIM software. The simulation parameters were selected to the data presented in Table I. As shown in Fig. 5, the simulation results were in full agreement with the estimated waveforms. Fig. 6 presents the relationship between the normalised output voltage (Uoll/UTr-,) and the freewheeling state duty ratio D,.

IV. EXPERIMENTAL VERIFICATION

To validate the proposed novel control algorithm for an AFBR, a small-scale prototype with the output power of I kW was assembled. Six independent PWM channels were used. Two channels with small dead time were used to drive IGBTs in the inverter and four channels were used to control the rectifier. As mentioned, rectifier switches should have reverse blocking capability. In the prototype, MOSFETs with series connected diodes were used. In practical applications these switches could be replaced by reverse-blocking IGBTs or fast thyristors for reduced power dissipation during the on-

TABLE I PARAMETERS AND COMPONENTS OF THE EXPERlMENTAL PROTOTYPE

Parameter Symbol Value/Type Input voltage, V Uin 300 . . . 500 Output voltage, V UOUI 30 Switching frequency, kHz f,w 10 Transformer turns ratio NiND 0.28 Output filter capacitance, uF C( 220 Output filter inductance, mH Lr 1.5 Inverter switch IT, TB BSM75GBl20DLC Rectifier switch SJ-S4 IXFX 48N60P Rectifier diode ON MBR40250 Output power, W Pout 1000

state. The main parameters and components are presented in Table I.

During the first tests the converter was operating without snubbers and the experimental waveforms are presented in Fig. 7. As shown, the test results completely correspond to the estimated waveforms. Since the application of capacitive snubbers allows turn-off losses of the inverter IGBTs to be reduced, the operation of the experimental prototype was tested with different snubber capacitors. The required capacitance value could be roughly estimated by [6]

C "" letf

s 4.U ' In

(14)

where ttis the interval starting when the collector current falls from 90% to 10% of the turned-off current.

The theoretical IGBT tum-off losses with two parallel connected snubber capacitors assuming linear current and voltage slopes can be calculated by

U .lc2 .tf2

E - In off - ---':':'::""2-4=--"- (15)

The measured waveforms are shown in Fig. 8. As it could be observed, with the application of a snubber the dUcr/dt is reduced to 500 AI/l-s with a 10 nF snubber capacitor and to 330 AI/l-s with a 22 nF snubber capacitor from 2000 AI/l-s in the case of hard-switched tum-off. With the implementation of a snubber, the duration of the tail current remained approximately the same, but its amplitude increased, since the voltage applied to drive out the remained charge carriers from the drift region was lowered. The turn-off loss reduction with a snubber connected is substantially lower than that estimated from the theoretical equations due to the tail current increase with a lower dUcridt. Nevertheless, even with a 10 nF snubber the total turn-off losses were significantly reduced (by 30-35%) in comparison to turn-off without a snubber and roughly by 50% from the losses expected during hard­switched tum-off (Fig. 9). However, in terms of energy losses the effect of using a larger snubber capacitor becomes less obvious, especially at lower voltages.

The reduction in the IGBT tum-off power loss by help of capacitive snubbers has been reported to be dependent on the IGBT technology and the junction temperature. Typically it is lower than expected from theoretical estimations due to increased current tail duration and it should be experimentally

5182

12 600

10 Voltage

500

8 400

§: 6 300 E � 4 2 00 �

100

0 0

-2 -1 00 0 2 3 4 5

Time (Ils) (a)

12 600

10 Current Voltage

500

8 400

§: 6 300 E � 4 200 " ..:::

100

0 0

-2 -100 2 3 4 Time (Ils)

(b)

12 600

10 Voltage

500 Current 8 400

§: 6 300 E ..;; 4 200 �

100

0 0

-2 -100 0 2 3 4

Time (Ils) (c)

Fig. 8 Measured turn-of I waveforms of the II IGBI module: no snubber (a), I OnF snubber (b) and 22nF snubber (c) (Uin=500 V; f",=1 0 kHz, 1000 W load).

0.9

0.8

� 0.7

5 0.6

� 0.5 "

0.4

0.3

--e- No snubber ____ 10 nF � 22 nF

350 380 410

U;"M 440 470 500

Fig. 9. Tum-off energy losses of the TT IGBT module with different snubber capacitors.

estimated for every particular application to obtain a precise result. The reduction in the IGBT turn-off power loss by help of capacitive snubbers has been reported to be dependent on the IGBT technology and the junction temperature. Typically it is lower than expected from theoretical estimations due to increased current tail duration and should be experimentally estimated for every particular application to obtain a precise result. Around 50% reduction of turn-off losses of soft­switched PT- and NPT-IGBTs is obtained [7] . Similar results were reported with FS- and SPT-IGBTs [8] . However, no significant reduction in switching losses with a capacitive

snubber at high power operation was observed with trench IGBTs [6] .

V. CLOSED LOOp CONTROL SYSTEM

The main task of the proposed topology is to ensure stable output voltage despite the input voltage or converter load variations. In this case both the static error in the steady-state mode and the dynamic of the transient process are important.

Controlled Rectifier

Fig. 10. Digital control system of the DCfDC converter

Fig. 1l. Simplified digital control system of the DCIDC converter.

The ability to compensate both the input voltage and the load variations is a very important dynamic parameter of the control system. The digital control system and the gating signal for the discussed converter are shown in Fig. 10. The PI regulator determines the active state duty cycle Da taking into account the minimum value of the phase shifting that is required for energy return interval. After subtraction the freewheeling state duty cycle Ds is obtained as a main regulator parameter in the control system.

Depending on the fluctuations of the input voltage the freewheeling state duty cycle D, is changing. That determines the gain factor of the converter. For example, with a decrease in the output voltage the error signal increases, which leads to a decrease in the zero state duty cycle D,. According to Eq. (4) the gain factor of the converter increases and, as a result, voltage on the output is still the same.

The dynamic behaviour of the proposed converter is defined by passive components and parameters of the PI regulator. Usually differential component can evoke instability of the system and was set to zero. In order to adjust regulator parameters a simplified closed loop transfer function was proposed. Taking into account the discretion of the converter, the structure of a closed loop control system is shown in Fig. 11.

The output voltage of the converter is compared with the reference voltage and the resulting error is sent to the input of the PI regulator. The inverter, transformer and rectifier can be represented as the united continuous linear part (CLP) having the gain factor equal to

K - UDC2 - 2 · n L

- - . [Jf)e

(16)

5183

,-----------, 600 3.5 ,-----------, 600 3.: Current SOD Voltage Current

500 2.5 J----. 2.5 Iwo--,...'" 400 400 E

300 � :s 2 ;----1 300 :0' :s 2 J----r ...:.,51.5 200 �5 .... '"

1.5 J-.--"'\.."-1

0.5

0.05 0.1 0.15 0.2

100 1 200

0.5

0.05 0.1 0.15 0.2

100

30 ,------------, 60 30 ,------------, 60

25 50 25 Current 50

20 �--'""''''-..... ,.. ... -..-__ -1 40 � g 15 -V:; 30 �Q 20

�cu;;.m:;;n�t __ ", ..... -""\ 40

:s 15 30 � J 10 20

10

J 10 Voltage

20 ';;;);

10

+--�-���--+o +--�-��-�o 0.05 0.1

Time (s) 0.15 0.2 0.05 0.1

Time{s) (a) (b)

0.15 0.2

Fig. 12. Converter input (top) and output (bottom) voltages and currents measured during input voltage variations.

The transfer function of the regulator in the s-space: 1 Kl'f =KI'+- ' (17)

T'P where Kp is the proportional gain and T is the time

constant. Further, the transfer functions of control for the opened and

closed system, respectively, could be expressed in a general form:

KO(S)=( Kp+ -1_ .KL J . lies , l T . p lies + Ls

Kc(s) = Ko(s) . 1 + Ko(s)

(18)

(19)

Regulator parameters were specified during the experimental investigation. The integrator time constant is equal to T= 0.04 s, the proportional part is equal to Kp = 0.00005. The control system operation is shown in Fig. 12. Despite significant input voltage variations, the output voltage delivered was stabilised.

VI. CONCLUSIONS

The introduced modified control algorithm for the half­bridge converter with an active rectifier allows ZVS of the inverter switches and ZCS of the rectifier switches. At the same time, the parasitic oscillations after the tum-off of the inverter IGBTs are completely avoided. The leakage inductance of the transformer acts as the tum-on snubber for rectifier transistors and turn-off losses of the inverter transistors can be significantly reduced using capacitive snubbers. Unlike resonant converter solution, the proposed system requires only minor modifications to the hard­switched topology, it does not need any additional bulky passive components or a complex frequency-control algorithm and allows the energy circulation to be reduced during the operation.

The presented steady-state operation principle and mathematical analysis of the converter can be used as design guidelines for component and parameter estimation in practical applications. The operation of the converter was verified with the experimental prototype and the open-loop

test results were in full accordance with the expected waveforms.

Further, the converter operation was analysed with different values of snubber capacitors across inverter IGBTs and with the implemented voltage-based control system. According to the results, the presented step-down DC/DC converter topology with a phase-shifted active rectifier could be one of the most promising candidates for high-power conversion systems due to its reduced switching losses and a wide regulation range.

Focus in the further research will be on the possibilities of conduction loss reduction by using RB IGBTs in the rectifier and on the estimation of the overall converter efficiency.

ACKNOWLEDGEMENT

This research work has been supported by Estonian Ministry of Education and Research (Project SFO 1400 16sll), Estonian Science Foundation (Grant ETF8020) and Estonian Archimedes Foundation (project "Doctoral School of Energy and Geotechnology II").

REFERENCES

[I] Yingqi Zhang; Sen, P.c. ; "A new ZVS phase-shifted PWM DC/DC converter with push-pull type synchronous rectifier", Canadian Conference on Electrical and Computer Engineering, IEEE CCECE'03, voU, pp. 395- 398 voU, 4-7 May 2003

[2] Hong Mao; Abu-Qahouq, J.A. ; Shiguo Luo; Batarseh, I. ; "Zero­voltage-switching (ZVS) two-stage approaches with output current sharing for 48 V input DC-DC converter", Nineteenth Annual IEEE Applied Power Electronics Conference and Exposition, APEC'04, voL2, pp. 1078- 1082 voL2, 2004

[3] Chiu, H.J. ; Lin, L.W. ; Mou, S.C. ; Chen, C.C. ; "A soft switched DC/DC converter with current-doubler synchronous rectification", The 4th International Power Electronics and Motion Control Conference, IPEMC'04, voL2, pp.526-531 VoL2, 14-16 Aug. 2004

[4] Blinov, A; Ivakhno, Y.; Zamaruev, Y.; Vinnikov, D. ; Husev, O. "A Novel High-Voltage Half-Bridge Converter with Phase-Shifted Active Rectifier", IEEE International Conference on Industrial technology, ICIT'2012, pp.967-970, 19-21 March 2012

[5] Moisseev, S. ; Soshin, K. ; Sato, S. ; Gamage, L. ; Nakaoka, M. ; "Novel soft-commutation DC-DC power converter with high-frequency transformer secondary side phase-shifted PWM active rectifier", lEE Proceedings - Electric Power Applications, vo1.l51, no.3, pp. 260- 267, 8 May 2004

[6] Naayagi, R.T. ; Shuttleworth, R. ; Forsyth, AJ. ; "Investigating the effect of snubber capacitor on high power IGBT turn-off," 1st International Conference on Electrical Energy Systems, ICEES'II, pp.50-55, 3-5 Jan. 2011

[7] Byeong-Mun Song; Huibin Zhu; Jih-Sheng Lai; Hefner, AR. , Jr. ; "Switching characteristics of NPT- and PT-IGBTs under zero-voltage switching conditions", Conference Record of the Industry Applications Conference, Thirty-Fourth IEEE lAS Annual Meeting, voU, pp.722-728, 1999

[8] Fujii, K. ; Koellensperger, P. ; De Doncker, R.W. ; "Characterization and Comparison of High Blocking Voltage IGBTs and IEGTs Under Hard­and Soft-Switching Conditions", IEEE Transactions on Power Electronics, voL23, no.I, pp.I72-179, Jan. 2008

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