3.36-/5.24-GHz Concurrent Dual-Band Oscillator For WiMAX/WLAN Applications

Brijesh Iyer, Anirudh Kumar and Nagendra P. Pathak

Radio Frequency Integrated Circuit (RFIC) Research Group

Department of E&CE Engineering, Indian Institute of Technology Roorkee-247667-India

bijuiyer@gmail.com, anirudh07ece@gmail.com,nagppfec@iitr.ernet.in

Abstract — This paper reports a concurrent dual-band oscillator using two HEMT transistors and a single dual-band matching network. The device simultaneously oscillates at frequencies 3.36 GHz and 5.24GHz. Concurrency of the oscillator has advantageous as it provides more functionality, high data rate and system compactness in a single hardware. The measured characteristic of the proposed prototype shows anoutput power of -9.99dBm at 3.36 GHz and -15.58dBm at 5.24 GHz. It exhibits a phase noises with 10 MHz offset as -102.86 dBc/Hz and -93.80 dBc/Hz respectively at designated bands.

Index Terms — Concurrent, dual-band, harmonics, matching network, oscillator, phase noise, resonator.

I. INTRODUCTION

An oscillator is a key constituent for any communication, navigation, and measurement system. It provides the critical clocking function for high speed digital systems; generates electromagnetic energy for radiation, enables frequency up and down conversion when used as local oscillators, and are used as a reference source for system synchronization. To cope with the rapid growth of wireless applications, the transceivers must support multiple standards with minimal hardware. Current trend in radio frequency integrated circuit design is to develop a compact, low power and multi-standard devices based on hardware sharing. For development of a dual-band transceiver in WLAN frequencies, the primary hurdle is the implementation of a concurrent dual-band oscillator [1]. Various architectures [2-5] have been proposed in the literature for the implementation of dual-band oscillators. Some of the relevant reported configurations are: use of switchable tuning between the desired frequency bands, connecting two single resonator based oscillators in parallel, using a wideband oscillator operating in dual mode (i.e.450-900MHz and 900-1800MHz) or use of a series switched resonator based dual-band oscillator.

The simplest approach is a parallel architecture having different track for each individual frequency. The advantages of this architecture are low interference and possibility of separate optimization of each stage. However due to bulky circuit, the layout and PCB design becomes difficult. In the present article, a modification in the parallel architecture has been achieved by employing a concurrent dual-band matching network to operate simultaneously over the desired bands. The proposed dual-band concurrent oscillator has been designed

by using two active devices, one each for the individual frequency, a single dual-band concurrent matching network and separate resonators for the individual frequency band. The proposed architecture is advantageous over [2] and [5] as it operates concurrently without any switching mechanism. Unlike conventional parallel mode operation, proposed architecture uses a concurrent dual-band matching network. The geometry of the designed oscillator has been described in section II whereas the results are discussed in section III.

II. GEOMETRY OF THE OSCILLATOR

The oscillator is realized using hybrid microstrip integrated technology (HMIC). All surface mount devices (SMDs) are mounted on the commercial NH9320 substrate, which is a Poly-Tetra-Fluoro-Ethylene (PTFE)/glass/ceramic dielectric. The substrate is characterized by the dielectric constant ( r) of 3.2 and the substrate height (h) of 1.524mm. Matching networks consist of only distributed elements of microstrip transmission line (TL) sections. Fig. 1 gives the prototype of the proposed concurrent dual-band oscillator.

The active device used in the design is NE4210S01 from NEC Corporation. The device is super low-noise high electron mobility (HEMT) FET. The drain and gate direct current (DC) bias circuits of the oscillator consists of a quarter-wavelength, high-impedance TL stub. The high-impedance line provides

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DC path for their respective supply voltages, while, acting as open circuit for alternate current (AC) signals in the circuit, thereby, avoiding unwanted AC coupling. A radial stub is employed to have the short circuit at the point of connection with quarter wave line for providing high impedance at the operating frequencies. Coupling capacitors of 100nF are employed at the output ends of the transistor to couple only RF power, while, blocking DC signals. Table I provides the bias points for the NE4210S01 transistor.

It is found that the transistor is unstable at selected bias points. Here, any value for reflection coefficient at the the output side ( ) can be selected such that the input reflection coeeficient can be maximised.To maximize the input reflection coefficient, can be choosen such that

.The selected reflection coefficients are summarised as-

The concurrent dual-band matching network is designed using microstrip lines by calculating reflection coefficients at the load end of the transistor with DC bias network. The matching network employs series TL sections and stubs to transfer complex impedance seen at transistor’s terminals to 50 at the port. Two stubs are connected in shunt to the main line and are open-circuited.

Fig. 2 shows the impedance transformer structure that has been used for dual-band matching. The drawback of infeasible characteristic impedances is mitigated by considering four ideal lengths of series TL sections and stubs as design parameters. Hence, designer can arbitrarily set characteristic impedances of all TL sections. Such consideration not only allows dual-band matching of unequal complex impedances but also feasible fabrication of microstrip TL sections. Considering, the normalized impedance with respect to Z0 as zA, zB and zL, the transmission line theory leads to following design relations:

( )( )

1

11

1 .cottan1tan

−

−+

+= Sj

LjYLjY

zB

BA β

ββ

(1)

( )( )

1

22

2 .cottan1tan

−

−+

+= SjLjYLjY

zL

LB β

ββ (2)

LLL jXRz += (3)

Based on derived equations, a MATLAB code is developed to provide all possible solutions for feasible length parameters for dual-band impedance matching. Required impedances can be calculated from the reflection co-efficient values.

Using the complex impedances as target load, design parameters for output matching network is obtained through the MATLAB code. The code is written to solve the design equations (1) to (3) for concurrent dual-frequency complex impedance matching through the conventional double open-ended shunt stubs structure. Inputs to the program are the two design frequencies along with corresponding load refection coefficients. Moreover, substrate parameters, such as, dielectric constant, height, etc., are also provided in order to take their effects into account while performing computations at the two frequencies of interest [6]. Based on reflectioncoefficients, corresponding optimized lengths of matching network used in the fabrication prototype are given in Table III. In the design of layout, widths of all microstrip line are kept constant at 3.64 mm, which corresponds to characteristic impedance of 50 for the chosen substrate.

A concurrent dual-band matching network is devised and placed at the drain side of the individual transistors. Further, the input impedance at desired frequencies is estimated. This is required to design the resonator at individual frequency bands. Table IV provides the negative resistance used for the resonator design.

TABLE ISUMMARY OF BIAS POINTS

Symbol Parameter Value Vds Operational Drain to Source voltage 1.1 V Ids Operational Drain to Source current 10 mA Vgs Operational Gate to Source voltage -0.5 V

TABLE IIREFLECTION COEFFICIENTS

Frequency 3.36 GHz 0.87∠24.525o

5.24 GHz 0.72∠110.259o

TABLE III DIMENSIONS OF MATCHING NETWORK

L1(mm) L2(mm) S1(mm) S2(mm) 21.30 22.80 11.80 14.80

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Based on the input impedance obtained, the length of the resonator is selected such that real part of the input impedance of resonator is practically equal to one third of the negative resistance and imaginary part will be complex conjugate of it. Table V summarizes the input impedance for the resonator design.

The resonators are realized by using open stub TL and estimated based on the equation (4) and (5).

• For positive reactive part

90tan0

1 += −

ZX Sθ (4)

• For negative reactive part

= −

SXZ01tanθ (5)

The theoretical values obtained from the equations (4) and (5) are further optimized to obtain the desired impedance for the design of the resonator. The optimized resonator dimensions used in the prototype fabrication at the individual frequencies are summarized as-

III. RESULTS AND DISCUSSION

The layout of the oscillator is placed in the Advanced Design System (ADS) schematic as a component to carryout co-simulation with the active device and power supply. It is set to include all the parasitic and stray losses that might occur in the circuit. Further, the prototype of the proposed concurrent dual-band oscillator is fabricated on NH9320 substrate using photolithography technique. Fig.3 shows the fabrication of the oscillator prototype. The dual-band oscillator characteristics are measured on spectrum analyzer (Agilent FieldFox RF analyser-N9912A-6GHz/R&S FSP:

9KHz to 30GHz). Fig.4 depicts the measurement setup for the proposed oscillator. The fabricated dual-band oscillator is connected to the spectrum analyzer through 50 cable connector. Cable is having 0.80dBm insertion loss.

Fig.5 shows the measured output power spectrum. The output power spectrum (Fig.5b) shows the presence of oscillator peaks at 1.703, 3.422 and 5.128 GHz. The presence of 1.703 GHz frequency in the output is due to the mixing of two sources at 3.422 and 5.128 GHz. An output power of -9.99dBm and -15.58dBm at frequencies 3.36GHz and 5.24GHz has been obtained, respectively.

TABLE IV NEGATIVE RESISTANCE

Frequency INZ3.36 GHz -67.539-j101.544 5.24 GHz -9.286-j134.575

TABLE VINPUT IMPEDANCE FOR RESONATOR DESIGN

Frequency INZ3.36 GHz 22.513+j101.544 5.24 GHz 3.095+j134.575

TABLE VI RESONATOR DIAMENTIONS

Frequency W(mm) L(mm) 3.36 GHz 3 2.3 5.24 GHz 1 15

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Some spikes were observed near higher frequency band in the initial measurement results shown in Fig. 5(a). Fig. 5 (b) shows a better measurement result which has been taken by replacing the previously used cables and connectors. It is clear that spikes present in Fig 5 (a) were due to the use of faulty cable and connectors.

Fig.6 shows the phase noise measurement. At 3.36 GHz a harmonic suppression of approximately 13dBm with a phase noise of -102.86dBc/Hz at an offset of 9.8 MHz is obtained. At 5.24 GHz, approximately 18dBm harmonic suppression is obtained along with a phase noise of -93.80 dBc/Hz at an offset of 10MHz. This has proved the concurrent dual-band nature of the proposed design

IV. CONCLUSIONS

Design and characterization of concurrent dual-band oscillator for simultaneously operating at 3.36-/5.24GHz employing two HEMT transistors and a single dual-band matching network have been presented. It can be used for the WiMAX, Wireless LAN, and HiperLAN communication applications.

ACKNOWLEDGEMENT

This work has been partially supported by CSIR India through its grant No.22 (0622)/13/EMR-II.

REFERENCES

[1] M.Feng, S. Shen, D.C.Caruth, and J. Huang, "Device technologies for RF front-end circuits in next-generation wireless communications," Proceedings of the IEEE, vol. 92, no.2, pp.354-375, February 2004.

[2] A. Kral, F. Behbahani, and A. Abidi, “RF-CMOS oscillators with switched tuning,” Proceedings of the IEEE Custom Integrated Circuits Conf., pp.555-558, May 1998.

[3] J. Lee, S. Lee, H. Bae, and S.-H. Kim, “A Concurrent dual-band VCO with dual resonance in single resonator,” IEEE Topical Meeting on Silicon Monolithic Integrated Circuit in RF System,pp.135-138, January 2007.

[4] U. L. Rohde and A. K. Poddar, “Multi-Mode wideband voltage controlled oscillators,” 13th IEEE International Conference on Electronics, Circuits and Systems (ICECS-2006), pp.184-187, December 2006.

[5] V. Sharma, R. Yadav, and N. P. Pathak, “Series switched resonator based dual-band oscillator,” XXXth URSI General Assembly and Scientific Symposium, pp.1-4, August 2011.

[6] C.Garg, V. Sharma, and N.P. Pathak, “Double shunt stub impedance matching network based concurrent Dual-WLAN-Band amplifier,” Procedings of National Conf. on Recent Trends on Microwaves Techniques and Applications, Jaipur-India, July 2011.

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