iee proceedings - microwaves antennas and propagation volume 143 issue 2 1996 [doi...

8/13/2019 IEE Proceedings - Microwaves Antennas and Propagation Volume 143 Issue 2 1996 [Doi 10.1049_ip-Map-19960260] Langley, J.D.S.; Hall, P.S.; Newham, P

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Balanced antipodal Vivaldi antenna for widebandwidth phased arraysJ.D.S.LangleyP.S. HallP.Newham

Indexing terms: Vivald antenna, Phased a rrays, ide bandlimited arrays, Stripline antenna

Abstract:The Vivaldi antenna, a form of taperedslot radiator, has been shown to produce goodperformance over a wide bandwidth, limited onlyby the traditionally used slotline to microstripfeed transition. The authors present a newantenna, the balanced antipodal Vivaldi, whichincorporates an ultrawide bandwidth transitionand overcomes the poor polarisation performanceof the antipodal form. Good performance over a1 to 40 frequency range has been obtained. Theuse of the antenna in a linear phased array hasalso been investigated using elements constructedon high permittivity substrate. Wideband wideangle scanning with good cross-polarisation levelsis obtained.

1 IntroductionMulti-octave performance phased arrays are importantfor a number of applications, including dectronic war-fare and multiple mode radar systems. Wide bandwidtharray action is obtained primarily through the use ofwide bandwidth array elements, although arrays incor-porating clusters of elements covering sections of thedesired bandwidth have been reported [l]. Such ele-ments should have, in addition to wide bandwidth,symmetrical beamwidths to optimise scanning andshould be compact to allow sufficiently small elementspacing to prevent grating lobe formation at the maxi-mum operating frequency. An additional requirementis that the element should allow integration with trans-mitheceive modules constructed using a printed circuittransmission medium such as microstrip.

There are several ways of creating a wide bandwidtharray element. The ridged horn [2] exhibits bandwidthsof up to two octaves with a highly symmetric beam andgood power handling capability. The spiral antenna [l]has bandwidths in excess of four octaves but requires awide bandwidth balun. Log periodic antennas [3] have

EE, 1996TEE Proceedings online no. 19960260.Paper first received 24th August 1995 and m revised form 18thDecember1995J.D.S. Langley and P. Newham are with GEC Marconi Defence SystemsLtd., The Grove, Warren Lane, Stanmore, Mid& HA7 4LY, UKP.S. Hall is with the Univensty of Birmingham, Edgbaston, BkmhghamB15 27T, UK* Formerly wlth the University of Birmingham,UK

been used in HF or VHF skywave radar to give wideangle scanning. However, none of these elements are ofa suitable form for circuit integration. The tapered slotantenna, however, can be fabricated using printed cir-cuit techniques and is thus ideal for circuit integration.The slot antenna can be fabricated in either triplatestripline or microstrip. The stripline version, known asthe tapered notch [4], is generally fabricated with anexponential taper. All other types of tapered slot arefabricated on microstrip and incIude the Vivaldi, withan exponential taper [5] , he linear taper, broken lineartaper and constant width slot antennas [6]. All theseantennas exhibit low cross-polarisation characteristicsin the principle planes, however in the diagonal planethe CO- to cross-polarisation ratio decreases rapidlyaway from boresight [6]. This group of antennas is nowwidely used not only in phased arrays but also in radioastronomy, remote sensing, multiple beam satellitecommunications and spatial power combining tech-niques.In this study both the tapered stripline notch [4] andthe Vivaldi [5] antennas have been tested using identicalelliptical tapers, these antennas being fed by striplineand microstrip respectively. Nearly identical perform-ance is noted in our studies, in terms of gain,beamwidths and cross-polarisation, while references [4]and [5] suggest differences in operation. However onedifference which is pertinent to phased array operationis that the Vivaldi antenna has an open feed line whichcan radiate and perturb the radiation pattern.Although both these elements can have equalbeamwidths and can in principle be directly connectedto an integrated circuit, the slotline to feedline transi-tion limits the bandwidth and requires considerableingenuity to give broadband performance.

This paper describes the development and perform-ance of a new tapered slot antenna element (firstdescribed in [7]) that overcomes the transition problemto produce an ultra-wideband element for circuit inte-gration. Performance in a small linear phased array isalso presented and performance discussed.2 Batanced antip odal Vivaldi antennaIf the feed transition is made a collinear extension ofthe slot, then the bandlimiting effect is removed givingvery wide bandwidth operation. In the antipodalVivaldi [S, 91 shown in Fig. 1, a smooth transitionbetween twin line and microstrip is used. The metallisa-tion on either side of the substrate is flared in oppositedirections to form the tapered slot. Fig. 2 shows the

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input return loss of the antipodal antenna compared toa Vivaldi of similar taper characteristics and substratematerial. It is clearly seen that band limitation causedby the Vivaldi transition is removed and widebandaction is indeed obtained. The lower frequency limit isnow determined by the cut-off mechanism of the flare,namely that at the lowest operating frequency the aper-ture is half a wavelength wide. However the antipodalnature of the antenna gives rise to very high levels ofcross-polarisation particularly at high frequencies asFig. 3 shows, due to the skew in the slot fields close tothe throat of the flare.

groundplane twin ine dielectric

E field

microstripFig.1 Diagrammatic view o antpodal Vivaldiantenna0

-51 01 5

m -20-25-30-3540 2 4 6 8 10 12 l r 16 18 20

frequency,GHzFig.2 Input return loss~ Vivaldi_ _ _ _ antipodal Vivaldi(Physical details of antennas; substrate thickness = 1.58mm, E = 2.32, width =40mm; flare length = 30mm, width at aperture = 1 5 m , shape elliptical; tri-plate stripline width = 3mm; microstrip line width 5mm, transition flare length= 30mni, spacing between flares = 20mm, flare shape elliptical, major to minoraxis ratio = 3.33).

-- I /-35i6 7 8 9 b 1; 1 2 13 l 1 5 l'6 i7 l b

frequency,GHzFig.3 Cross-polarisation~ balanced antipodal Vivaldi. . antipodal Vivaldi_ _ _ _ Vivaldi(Details as Fig. 2)

To overcome this high cross-polarisation we haveadded a further layer of metallisation to form a bal-anced antipodal Vivaldi as shown in Fig. 4. The result-ant electric field in the slot region is now oriented98

parallel to the metallisation whilst the output transmis-sion medium is triplate stripline. Elliptical radiatingtapers were chosen for this antenna because previouswork with the Vivaldi and tapered notch antennasshowed that this particular taper gave similar E and Hbeamwidths. The notch length-to-width ratio was set at2:l to give gain of between 5 and lOdBi as found inour previous Vivaldi studies.

balancedqroundplanes dielectricsmetalisationflaredslot--

V

stripline resultantFig.4 Dia g rm mt ic v iew o balanced antipodal V ivaldi antennaE field

-5-1 0-1 5m -20-25-30

-40 3 5 L 1 4 6 8 10 12 14 16 18 20frequency,GHzFig 5 Input return lossBalanced antipodal Vivaldi: measured~ _ _ _heory(Details as Fig. 2

A l 2120100 / ' 10

k \80 , I 8

o v , , T 06 7 8 9 10 11 12 13 14 15 16 17 18frequency,GHzMeasured beamwidths and gain of balanced mtpodal Vivaldiig.6antenna_ _ _ ~-planeH-plane

~ gain(Details as Fig 2)

The input return loss, Fig. 5 is similar to thatobtained for the antipodal Vivaldi while the cross-polarisation, Fig. 3 , is improved and is typically below-20dB for this particular E~ = 2.32 substrate antenna.Fig. 6 shows that E and H beamwidths are approxi-mately equal and constant over a 6 to l8GHz band-width. Fig. 16 also shows that the gain varies from 5 to1ldBi. Radiation patterns are shown in Figs. 7 and 8.IEE Proc -Microw Antennas Propag , Vol 143, No 2, April 1996


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Although Fig. 2 only extends to 20GHz, for compari-son, a larger antenna has been shown to operate from1 to 40GHz [lo].

-50 1-90 -60 -30 0 30 60 90azimuth angle,degFig 7antenna: E-plane~ co-polarisation (measured)ross-polarisation (measured). . E-plane co-polarisation (measured) with asymmetric flares

~ co-polarisation (computed)_ _ _ _ _ cross-polarisation (computed)(Details as Fig. 2; asymmetric flare antenna shown inset, major to minor axisratio = 5.711

Radiation patterns at lOGHz or balanced antipodal Vivaldi

-500 30 60 90 120 150 180elevationangle, degRudmtzon pattern us in Fig. 7 -planeig.8It can be seen from Fig. 7that there is a squint ofabout 15 in the E-plane radiation pattern. This squintappears to be independent of both frequency and per-mittivity. It is believed that this squint is due primarilyto the unequal propagation velocity experienced by thecurrents on each side of the slot due to their differentgeometries. Measurements of the antenna aperturefields confirmed the presence of phase asymmetriestogether with some amplitude asymmetry. Variousmethods were tried to reduce the squint including cut-ting away some of the substrate, reducing the distancebetween the radiating flares and the transition flares,the addition of balancing flares and shorting pinsbetween the two outer flares. However none of thesereduced the squint but the use of asymmetric flaresallowed the introduction of some asymmetry into thepatterns which was found to offset the squint as shownin Fig. 7.The inset shows the shape of the flares usedin this example.The tapers in the transition are of a similar ellipticalform with length and spacing between tapers chosen tobe greater than half a wavelength at the lower operat-ing frequency. We have not performed optimisation onthese transitions and it may well be that shorter anten-nas could be developed.

IEE Proc.-Microw. Antennas Propag., Vol. 143, No. 2, April 1996

3 AnalysisThe balanced antipodal Vivaldi antenna has been ana-lysed using the finite difference time domain method.Thiele [l 11has recently shown the utility of this methodfor the Vivaldi antenna and we have used a similartechnique. The balanced antipodal antenna is modelledin a 79 by 215 by 29 cell volume with h/20 cell size atthe highest frequency. The spacing between metallisa-tion layers is modelled by two cells and the excitation isapplied to the triplate stripline end by equal electricfields within the 4 by 5 cells representing the geometri-cal area of the stripline. Both Gaussian pulse andramped sinusoidal excitation have been used. Far fieldsare obtained by a frequency domain near to far fieldtransform based on a measurement volume surround-ing the antenna spaced away from the structure by 3cells. Taper curvature is represented by a staircaseapproximation. Computed results for input return lossand radiation patterns are shown in Figs. 5 , 7 and 8respectively. The input return loss calculation predictsboth the low frequency cut-off associated with maxi-mum aperture width and the fast ripple due to slightreflections from the dielectric to air interface. While thecalculated radiation patterns for this antenna showgood agreement with measured results, both for co-and cross-polarisation.Surface currents on the metallisation layers can alsobe computed using this method. As previously noted[121 the current is primarily confined to the metallisa-tion edges within the radiating flares and to the regionof the strip in the transitions. Only a small amount ofcurrent exists on the flares of the transitions and thissuggests that there will be only small perturbations tothe radiation patterns from unwanted radiation in theseareas. The other notable feature of this result is thelarge current standing wave on the structure close tothe radiating region on the flare.

microstrip o stripline ransition

,screened monolithic circuits/J:labsorbing allY . ~ ._ _ - - - - - - ----

Fig.9strip to stripline transition ( I O ]Integrat ed antenn a circuit modu le concept ncluding the micro-

4 Stripline to microstrip transitionTo enable direct integration with a microwave inte-grated circuit a stripline to microstrip transition wasdesigned so that the complete antenna-circuit modulecould take the form shown in Fig. 9. Although such amodule has not been made within the scope of thisstudy a double-sided transition has been constructedand tested [lo]. The bottom ground plane is continuouswhilst the upper one is electrically connected to thelower using via holes. Both the top ground plane andsubstrate are elliptically tapered while the strip width istapered in the same fashion to ensure 50Q impedancein each medium. The transition was built on E = 10.5substrate for integration with the antennas made on thesame substrate described in the next Section. The loss

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for two transitions is less than 2dB with a -12dBreturn loss across the desired 6-18GHz band (for a sin-gle transition) [lo].5 Arrays of balanced antipodal Vivald i antennasAlthough the balanced antipodal Vivaldi antenna isintended to be used in a dual polarised array, only itsperformance in E-plane arrays has been demonstrated.To incorporate this wideband element into a scanningarray, the elements must be placed at I2 at the high-est frequency, where ho is the free space wavelength.Thus at the bottom end of a 3:l band these elementswill be spaced by h016. It is well known that taperedslot antennas exhibit a low frequency cut-off whichoccurs when the maximum flare width at the apertureis hJ2, where hsis the wavelength in the slot. Althoughthis may not occur in very large phased arrays due tohigh mutual coupling at the lowest frequency, it willoccur in the small arrays considered here and the fol-lowing design method is therefore appropriate. Assum-ing an effective dielectric constant of cS in the slot thesetwo conditions are met when

1)0 A s A06 2 &2giving E = 9. The substrate dielectric constant toachieve this could be approximately derived from uni-form slot theory but its value is constrained both bythe materials available and the likely materials to beused in the microwave integrated circuit. The valuechosen for the initial demonstrator was 10.5, with anadditional array being made on E = 6 material whileusing separation of alternative elements in the H-planeto prevent grating lobes. The silhouette of the E = 10.5array is shown in Figs. 10and 11 with that for the E~ =6 array being similar. The seven element array was pro-duced using one manufacturing process, to work over a3 to 9GHz range.

Fig.10 Silhouette of 7-element E-plane array, outer conductor__ outline of dielectric showing semi-circular extensions(array details, substrate thickness = 0.64mm, E = 10.5; radiating flare length =24mm, width at aperture = 17mm, shape elliptical; stripline width = 0.2mm;transition flare length = 14mm, spacing between flares = 4mm, flare shapeelliptical, major to minor axis ratio = 3.33)

. . . . . . . . . . . . .Fig.11 Silhouette of array as in Fig. 10, inner conductor

Initial results for a single element using a networkanalyser in both frequency and time domain moderevealed a substantial reflection from the dielectricedge at the flare aperture. Shaping of a dielectricextension beyond this aperture was found to reducethis reflection, with a semicircular extension as indi-cated in Figs. 10 and 11, giving optimum perform-ance. This was then used on all array elements, forboth the 10.5 and 6 permittivity substrates.

The beamwidths for the E = 10.5 elements are ingeneral very large and therefore the E-plane asymme-try noted in the low permittivity elements is notobserved. This is due to surface wave interaction andthe fact that the radiation occurs from the front of thedielectric extension. In the E, = 6 case the trappedwaves are reduced and some element asymmetry isobserved in the E-plane. The principle plane cross-polarisation levels in these higher permittivity ele-ments are similar to those found in the low permittiv-ity versions (5 -20dB). However the cross-polarisationin the diagonal planes is generally better in the higherdielectric constant medium. This is due to the factthat the physically smaller elements have shorter lon-gitudinal current paths and therefore an increase ineffective cancellation occurs.

0

-10

dB -20

-30

-40-90 -70 -50 -30 -10 10 30 50 70 90azimuthangle,degFig 123GHz~ 0 scan20 scan40 scan60 s c a n

Measured scan patterns or the array shown in Figs 10 and 11,~ _ _ __ _ _ _

-90 -70 -50 -30 -10 10 30 50 70 90a z i m u t h angie,degFig.136 GHz~ 0 scan20 scan40 scan60 scan

Measured scan patternsfor the array shown in Figs 10and 11,_ _ ~ ~_ _ _ _

6 Array performanceMeasured radiation patterns for the 7-element E-planelinear array with E = 10.5 elements are shown in Figs.12-14. Well formed beams are obtained in general.However some gain reduction is noted for the 60 scanat all frequencies due to the element beamwidth ofapproximately 100 . Fig. 15 shows the measured gain

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at the scan angles presented in Figs. 12-14, whileFig. 16 shows the cross-polarisation levels which arebelow -15dB across the 3 to 9GHz bandwidth.

0

-10

dB-20

-30i-40-90 -70 -50 -30 -10 10 30 50 70 90

azimuthangle,degFi I4___ 0 scan20 scan40 scan60 scan

Measured scan patterns or the array shown in Figs 10 and 119 8 H z- _ _ _- _ _ -

1~

dBi

O b3 4 5 6 7 8 9frequency, GHzMeam redgain fo r 7-element E-plane array e, = 10.5)ig 15___ 0 scan20 scan40 scan60 scan

- _ _ _- _ _ _

0,---10L-1 5

dB -20-2 5

I3 4 5 6 7 8 9frequency.GHzFig.16

__ 0 scan20 scan40 scan60 scan

Cross-polarisation or 7-element E-plane array E, = 10.5)- _ - -- _ _ _

Measured inter-element coupling for the E = 6.0(triangular lattice) E-plane array is below -20dB formost of the 3 to 9GHz band, with levels of -13dBbelow 3.5GHz. This is an improvement in coupling ofIEE Proc-Microw. Antennas Propag., Vol. 143,No. 2, April 1996

4dB across the band when compared with the E =10.5 array elements. Both the larger element spacingand the reduced amount of surface wave propagationin the E = 6.0 array help contribute to this improve-ment. However, because the E-plane element asym-metries are more evident in the E = 6.0 array, thescan patterns are not as well formed as in the E =10 5 array. Similar gain is obtained from both thesearrays while the cross-polarisation levels from the E= 6.0 array (-20dB across the 3 to 9GHz band-width) are in general 5dB better than for the E = 10 5array.Wideband, wide angle scanning has been achievedusing these antennas. However these design tech-niques are perhaps only applicable to small phasedarrays, as used in ECM, ECCM and DF systems. Inlarge phased arrays (number of elements > 100) it islikely that the use of dielectric materials would not benecessary, as the high levels of mutual coupling expe-rienced in such an array would enable operation withelements working well below their individual cut-offfrequency [131.7 ConclusionsThe limitations on the bandwidth of the Vivaldiantenna due to the slotline to microstrip transitionhave been overcome whilst preserving low cross-polar-isation by the development of the balanced antipodalVivaldi antenna. The new antenna allows simple inte-gration with microwave integrated circuit transmit/receive modules using an additional stripline to micro-strip transition which on E = 10.5 substrate has beenshown to have a loss of less than IdB. An antenna onE = 2.32 substrate has been shown to have a band-width in excess of 40:1, whilst over a 3:1 bandwidth,cross-polarisation below -20dB is obtained. Radiationpatterns are in general well controlled but an E-planesquint is noted which can in principle be compensatedfor using asymmetrical flares. Performance on E =10.5 substrate and to some extent on E = 6 is affectedby the dielectric-air interface at the flare aperture andthis mismatch has been reduced with the introductionof a semicircular substrate extension.

Two 7-element E-plane arrays of these balancedantipodal Vivaldi elements have been constructed, oneon E = 10.5 and the other on E = 6.0 with a triangu-lar lattice structure to avoid the formation of gratinglobes at the high frequency end of the band. Wide-band, wide angle scanning has been achieved withthese arrays while maintaining suitable cross-polarisa-tion levels.8 AcknowledgmentsThis work was supported under an EPSRC CASEstudentship by GEC Marconi Defence Systems Ltd,UK and the authors acknowledge the company's per-mission to publish.

ReferencesSHIVELY, D.G., and STUTZMAN, W.L.: 'Wideband arrayswith variable element sizes', IEE. Proc. Microwaves, Antennas andPropagation, 1990, 187 (4), pp. 238-240MONSER, G.J.: 'Design considerations for broadband phased-array elements beyond two octaves', Int. Con Milita ry Micro -waves, June 1986, pp. 392-396

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3 TITTENSOR, P.J., and ORTON, R.S.: Calibration of a multi-octave phased array, IEE Znt. Con$ Antennas and Propagation,ICAP 91, pp. 790-7934 LEWIS, L.R., FASSET, M., and HUNT, J.: A broadband strip-line array element, IEEE Syrnp. Antennas and Propagation,Atlanta, USA, 1974, pp. 335-3375 GIBSON, P.J .: The Vivaldi Aerial, 9th European MicrowaveConference, Brighton, UK, September 1979,pp. 101-10590, 13-15 November 1990, pp. 253-264LANGLEY$J.D.S., HALL, p.s., and NEWHAM, p.:ultrawide-bandwidth Vivaldi antenna with low crosspolarisation,GAZIT, E.: Improved design of the Vivaldi antenna, IEE Proc.Microwaves, Antennas and Propagation, 1988, 135, (2), pp. 89-92

9 FOURIKIS, N., LIOUTAS, N., and SHULEY, N.V.: Paramet-ric study of CO and cross polarisation characteristics of taperedplanar and antipodal slotline antennas, ZEE Proc. Micicrowaves,Antennas and Propagation, 1993, 140 (l), pp. 11-2210 LANGLEY, J.D.S., HALL, P.s.3 and NEWHAM, p.1 Bal-anced antipodal Vivaldi antenna for multi-octave bandwidthphased arrays, Int. Con JINA 94 , 8-10 November 1994, pp.585-588flared horn antennas and arrays, IEEE Trans. Antennas andPropagation, 42 (S), pp. 633-64112 JANASWAMY, R.: An accurate moment method model for thetapered slot antenna, IEEE Trans. Antennas and Propagation,

13 : Phased array workshop discussions, IEEE Int. Conf. AP-S,June 1995

6 SCHAUBERT, D.H.: Endfire slotline antennas, rnt. conf INA THIELE, E.T., and~TAFLoVE, A.: FD-TD Of Vivaldi

Electron. Lett. , 1993, 29 (23), pp. 2004-2005 1989,37, (12), pp. 1523-15288

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