hybrid field generator controller for optimised … · hybrid field generator controller for...
TRANSCRIPT
HYBRID FIELD GENERATOR
CONTROLLER FOR OPTIMISED
PERFORMANCE
by
Christopher Teboho ‘Moleli
B-Tech: Electrical Engineering
A research dissertation submitted in compliance with the
requirements for the degree
Magister Technologiae: Electrical Engineering
in the
Faculty of Engineering
Port Elizabeth Technikon
Promoter: Mr. A. Roberts
M Dip Electrical Engineering (Heavy Current)
Co-promoter: Dr. H. A. Van der Linde
PhD: Electrical Engineering
Submission Date: 10/12/2003
i
Acknowledgements
The following persons are sincerely acknowledged for their support that
contributed to the successful completion of the research project:
Mr. A. Roberts and Dr. H. A. Van der Linde for their continued
academic guidance.
Faculty of Engineering, for providing test instruments and
equipment to meet the project requirements.
ii
Declaration
I Christopher Teboho ‘Moleli hereby declare that:
The work in this dissertation is my own original work;
All sources used or referred to have been documented and
recognized; and
This dissertation has not been previously submitted in full or partial
fulfillment of the requirements for an equivalent or higher
qualification at any other recognized education institution.
___________
C. T. ‘Moleli Date 10/12/2003
iii
Abstract
Battery charging wind turbines like, Hybrid Field Generator, have become
more popular in the growing renewable energy market. With wind energy,
voltage and current control is generally provided by means of power
electronics. The paper describes the analytical investigation in to control
aspects of a hybrid field generator controller for optimized performance.
The project objective is about maintaining the generated voltage at 28V
through out a generator speed range, between 149 rpm and 598 rpm. The
over voltage load, known as dump load , is connected to the control circuit
to reduce stress on the bypass transistor for speeds above 598 rpm.
Maintaining a stable voltage through out the speed range, between 149rpm
and 598rpm, is achieved by employing power electronics techniques. This
is done by using power converters and inverters to vary the generator
armature excitation levels hence varying its air gap flux density. All these
take place during each of the three modes of generator operation, which
are: buck, boost and permanent magnet modes.
Although the generator controller is power electronics based, it also uses
software to optimize its performance. In this case, a PIC16F877 micro-
controller development system has been used to test the controller
function blocks.
iv
Table of contents
Page
ACKNOWLEDGEMENTS .............................................................. I
DECLARATION .......................................................................... II
ABSTRACT............................................................................... III
TABLE OF CONTENTS .............................................................. IV
LIST OF FIGURES .................................................................... IX
LIST OF SYMBOLS ................................................................. XIII
ABBREVIATIONS ..................................................................... XV
DEFINITION OF CONCEPTS ................................................... XVII
CHAPTER 1 ............................................................................... 1
Introduction ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
1.1 Controller overview .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
1.2 Problem Statement .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
1.3 Sub-problem Statement .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2
1.4 Hypothesis.. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
1.5 Project Delimitations... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3
1.6 Project Outline ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
1.7 Project Significance ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
1.8 Methodology ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
v
CHAPTER 2 ............................................................................... 8
Project literature ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
2.1 Literature overview .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
2.2 Hybrid Field Generator characteristic background ... . . . . . . . . . . . . . . . . . . 9
2.2.1 Permanent magnet mode ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
2.2.2 Boost mode... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
2.2.3 Buck mode ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
2.3 Charge control .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
2.3.1 Rectification ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
2.3.2 DC voltage and current regulator .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
2.3.3 Charge load protection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
2.4 Excitation control .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
2.4.1 DC/DC converter .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
2.4.2 Exciting current control .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
2.4.3 Current protection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
2.5 Charge and excitation control system .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
2.5.1 Micro-controller selection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
2.5.2 Signal conditioning ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
2.5.3 Interfacing... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
2.6 General aspects of the project literature ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
CHAPTER 3 .............................................................................. 39
Controller specifications and considerations ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
3.1 Overview .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
3.2 Generator specifications ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
3.2.1 Generator tests .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
3.3 Battery bank specifications ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
3.4 Components selection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
3.4.1 Discrete components .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44
3.4.2 Passive components .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 45
vi
3.4.3 Integrated circuits .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
3.5 Transformer and choke ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51
3.6 Sensors .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
3.6.1 Hall-effect sensors .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
3.6.2 Optical and electromagnetic isolators .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54
CHAPTER 4 .............................................................................. 57
Charge voltage regulation design ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57
4.1 Overview .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57
4.2 Uncontrolled rectification ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 57
4.2.1 Controller protection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62
4.3 Over voltage load ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63
4.4 Charging voltage regulation ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 68
4.4.1 Overload protection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70
4.5 Battery protection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 72
CHAPTER 5 .............................................................................. 74
Excitation circuit design ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74
5.1 Overview .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 74
5.2 DC/DC converter .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 77
5.2.1 Transformer Selection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 92
5.3 Exciting current control .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 94
5.3.1 Exciting current polarity changer .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 98
CHAPTER 6 ............................................................................ 100
Control Circuit design ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100
6.1 Overview .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100
6.2 I/O circuits .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 100
6.3 Switching circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111
6.4 Micro-controller .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 114
vii
6.5 Operation control modes ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 115
6.5.1 Analog and digital input signals execution ... . . . . . . . . . . . . . . . . . . . . 117
6.5.2 Operation mode selection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 118
CHAPTER 7 ............................................................................ 122
Conclusion ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 122
7.1 Conclusion ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 122
7.2 Recommendations ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 126
7.3 Future work ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 127
List of references ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 129
Application Notes ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 133
Abstracts .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 133
CD-ROMs .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 134
APPENDIX A .......................................................................... 135
Flow charts and blocks code ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 135
APPENDIX B .......................................................................... 158
Hybrid field generator performance characteristics tests results .. . . 158
Generator test results .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 158
APPENDIX C .......................................................................... 176
Tests boards-data ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 176
Line voltage harmonics ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 176
Load current influence on ripple voltage at zero exciting current ... 177
Switched exciting voltage and current waveforms ... . . . . . . . . . . . . . . . . . . . . . . . . 185
viii
APPENDIX D .......................................................................... 186
Schematics test results .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 186
APPENDIX E........................................................................... 204
Project Test boards and schematics .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 204
ix
List of figures Page
Figure 1.1 Project Block Diagram.... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Figure 2.1 Permanent magnet mode generator speed vs generated voltage
... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Figure 2.2 Average permanent magnet air gap flux density distribution
over a five teeth pole. .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Figure 2.3 Boost mode speed vs voltage at different exciting current
levels .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Figure 2.4 Boost mode average air gap flux density at different exciting
current levels .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Figure 2.5 Buck mode speed vs voltage at different exciting current levels
.. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Figure 2.6 Buck mode average air gap flux density at different exciting
current levels .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Figure 2.7 Charge control block diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Figure 2.8 Three-phase AC/DC converter .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Figure 2.9 Six-pulse bridge rectifier .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Figure 2.10 Excitation control block diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Figure 2.11 Gapped and un-gapped core magnetic characteristic curves23
Figure 2.12 Line noise filtering ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Figure 2.13 Screened transformer ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Figure 2.14 Continuous waveform .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Figure 2.15 Discontinuous waveform .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Figure 2.16 Frequency response of three common types of fi lters .. . . . . . . . . 30
Figure 2.17 Charge and excitation control system .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Figure 2.18 Line current sensor block diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
Figure 2.19 Speed sensor block diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
Figure 2.20 Opto mechanical speed sensor .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
Figure 2.21 Speed disk mounting ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
x
Figure 2.22 Line currents and speed sensors power supply and rotor
excitation current control power supply ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
Figure 3.1 Generator tests setup ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
Figure 3.2 IR2153 functional blocks ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
Figure 3.3 (a) Inverting op-amp (b) non-inverting op-amp ... . . . . . . . . . . . . . . . . . . 47
Figure 3.4 (a) Differential op-amp (b) Voltage follower ... . . . . . . . . . . . . . . . . . . . . . 48
Figure 3.5 Instrumentation op-amp ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50
Figure 3.6 LOC11X .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56
Figure 3.7 LOC11X V i n vs Vou t . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56
Figure 4.1a Line voltage (Ch 1) and line current (Ch 2) waveforms ... . . . . 58
Figure 4.1b Line current 5A waveform.... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59
Figure 4.2 Open circuit excitation characteristics curves at 24V line ... . . 61
Figure 4.3 Ripple voltage waveform at 0A load current .. . . . . . . . . . . . . . . . . . . . . . . . 61
Figure 4.4 Ripple voltage at 2A load current .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61
Figure 4.5 Ripple voltage waveform at 2A load current and 2A excitation
current .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 61
Figure 4.6 Ripple voltage waveform at 2A load current and 4A excitation
current. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 62
Figure 4.7 Voltage protection circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 63
Figure 4.8 MOSFET switching voltage levels .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65
Figure 4.10 Battery bank voltage regulator.. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 70
Figure 4.11 Battery bank voltage monitor .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 73
Figure 5.1 Static air gap flux density ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 75
Figure 5.2 Dynamic air gap flux density ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 76
Figure 5.3 Rotor (armature) excitat ion schematic diagram .... . . . . . . . . . . . . . . . . . 78
Figure 5.4 IR2153 RT vs frequency curves ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 79
Figure 5.5 DC/DC converter drive signals, signal 1 for Q1 signal 2 for Q2
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 79
Figure 5.6 DC/DC converter transformer primary signals with respect to
battery bank positive terminals. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 80
Figure 5.7 DC/DC converter transformer secondary signal .. . . . . . . . . . . . . . . . . . . 80
xi
Figure 5.8 DC/DC signal overshooting and ringing ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . 81
Figure 5.9 Open circuited rotor back emf waveform .... . . . . . . . . . . . . . . . . . . . . . . . . . . 82
Figure 5.10 Freewheeled back emf waveform .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 82
Figure 5.11 Zener clamped freewheeled back emf waveform.... . . . . . . . . . . . . . . 83
Figure 5.12 rotor circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 84
Figure 5.13 Exciting current control circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 84
Figure 5.14 Continuous mode waveforms ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 87
Figure 5.15 Discontinuous mode waveforms ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 89
Figure 5.16 Continuous mode waveform at 30% duty cycle ... . . . . . . . . . . . . . . . . 91
Figure 5.17 PWM generator block diagram.... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 94
Figure 5.18 Characteristic behavior of the PWM generator .. . . . . . . . . . . . . . . . . . . 95
Figure 5.19 PWM duty cycle levels at different DC levels. . . . . . . . . . . . . . . . . . . . . 96
Figure 5.20 PWM generator schematic diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 96
Figure 5.21 PWM generator signal waveforms ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 98
Figure 5.22 Exciting current polari ty changer .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 99
Figure 6.1 Three terminal voltage regulator with by pass transistor .. . . . 102
Figure 6.2 ADC voltage protection ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 102
Figure 6.3 Active low pass filter .. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103
Figure 6.4 60Hz active low pass fi lter characteristics. . . . . . . . . . . . . . . . . . . . . . . . . 104
Figure 6.8 Speed sensor schematic diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 106
Figure 6.9 Speed pulses waveforms ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 106
Figure 6.10 Line current sensor characteristics curve ... . . . . . . . . . . . . . . . . . . . . . . . 108
Figure 6.11 Line current sensor schematic diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . 109
Figure 6.12 4N25 current and voltage characteristic curves ... . . . . . . . . . . . . . . 109
Figure 6.13 Hall-effect current sensor characteristic curves ... . . . . . . . . . . . . . 110
Figure 6.14 Voltage detector schematic diagram .... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 111
Figure 6.15 Load disconnect switch ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 112
Figure 6.16 2N3904 VB E vs VC E . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 113
Figure 6.17 Relay drive circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 114
Figure 7.1 On load excitation curves at 24V line voltage and at 600rpm
.... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 124
xii
Figure 7.3 Hybrid field control station network ... . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 128
xiii
List of Symbols
a : armature
A : ampere
Ac : core cross sectional area
Av : amplifier gain
Cp : pulse count per second
CH : speed disk holes
D : duty cycle
DX : diode ( note)
e : instantaneous voltage
E : maximum voltage
f : frequency
G : gate
Hz : hertz
i : instantaneous current
I : current
k : kilo
L : inductance
mT : milli tesla
M : mega
N : revolutions per minute
p : pole pair
P : power
Ph : phase
Q : transistor
R : resistor
Sp : speed
t : t ime
T : period
V : volt
xiv
W : watt
Wa : window area
X : reactance
Z : impedance
π : PI
τ : t ime constant
°c : degree celcius
Φ ex : excitation flux
Φ F : field coil flux
Φ L : leakage flux
Φ m : permanent magnet flux
dT/dt : temperature rate of change
di/dt : current rate of change
Note: x refers to a subscript. Example D1, D2
xv
Abbreviations
AC : alternating current
AC/DC : alternating current to direct current converter
BJT : bipolar junction transistor
CMRR : common mode rejection ratio
CT : t iming capacitor
DC : direct current
DCX : direct current voltage level (note)
D/A : digital to analog
DC/DC : direct current to direct current converter
emf : electromagnetic force
EMI : electromagnetic interference
FET : field effect transistor
Ho : high output
Icg : charge current
IC L : controlled load current
IcTot a l : total input charge current
IC : integrated circuit
IO : input output
IR : international rectifier
Isc : short circuit current
IV : current voltage
L-disc : load disconnect
LED : light emitting diode
Lo : low output
LOC : linear optocoupler
MOSFET : metal oxide field effect transistor
Neg : negative
O.C. : open circuit
xvi
Ov : over voltage load
pfm : peak frequency modulation
pwm : pulse width modulation
Pos : positive
Pot : potentiometer
rpm : revolution per minute
RFI : radio frequency interference
Rlyx : relay (note)
RT : t iming resistor
sd : shutdown
UVLO : under voltage lockout
Vb a t : battery bank voltage
Vb mi n : minimum battery bank voltage
Vb mx : maximum battery bank voltage
VC E(SAT) : collector to emitter saturation voltage
Vcg : charge voltage
V i n : input voltage
VGS :gate to source voltage
Vline : l ine voltage
Vo : output voltage
V re f : reference voltage
V ro t or : rotor voltage
V t r c : tracking voltage
VSD : variable speed drive
Note: x refers to the position of the relay. For an example: Rly1, first
relay.
xvii
Definition of concepts
Hybrid Field Generator: Combination of a permanent magnet field
generator and wound field generator.
Hybrid Field Generator: The power controller that stabilizes the
power controller generated power by controlling the
generator excitation.
Performance data sheets: Documentation of tests results.
Bit addressable : A micro-controller whose single bit can be
micro-controller addressed without bit masking.
Low power components : Components that consumes less power.
On board power losses : Power that is dissipated by power
components on-board.
Region of operation : Operation region of high efficiency.
Safe region : Region where the controller is operated
safely.
MOSFET driver : MOSFET switching integrated circuit
MPLAB : It is a software-developing tool used to
write PIC micro-controller programs.
xviii
Uncontrolled rectifier : One type of AC-to-DC converter that does
not need a triggering pulse in order to
operate.
Controlled rectifier : Another type of AC-to-DC converter that
needs to be triggered before it operates.
1
Chapter 1
Introduction
1.1 Controller overview
Wind turbines have become widely used for the generation of electricity.
A disadvantage of wind turbines is that they do not generate constant
voltage as required by most electrical appliances. This is essential
because the generated voltage is directly proportional to the wind speed.
The generated voltage is maintained at the rated voltage over a wide range
of wind speeds with the aid of an electronic power controller.
A novel generator development for a wind turbine application is the
Hybrid Field Generator. The generator characteristics are a combination
of those of the permanent magnet generator and the wound rotor
generator. The Hybrid Field Generator can therefore operate as a
permanent magnet generator or a field controlled generator.
The Hybrid Field Generator operates in three modes: permanent magnet
mode, buck mode and the boost mode. The operation mode is determined
automatically by the captured energy level. Varying the excitation of the
field coils during buck mode and boost mode, controls the generated
voltage and this is achieved by making use of an electronic power
controller.
The power controller monitors the generated voltage and the charging
current to the battery bank. It ensures that sufficient current is supplied to
the load. It also maintains the energy dissipated by the battery bank and
the DC load within the rated values. During strong winds, when the
2
generator speed is over 598rpm, the captured wind energy is more than it
is needed. At this point, the excess energy is transferred to excess energy
load (over voltage load), normally referred to as a dump load. As a
result the generator is always kept running under load instead of shutting
the hybrid system down and leaving the generator running under no load.
This is to prevent uncontrolled acceleration of the generator, mainly when
strong wind is blowing, which will result in over-speeding.
Although the battery bank is one of the loads, it also ensures that the load
voltage remains stable regardless of the generated voltage fluctuations
within limits. The battery bank acts as a power source for the controller
electronic circuit. It also serves as an excitation field coils power source
during buck and boost modes.
The power controller applied in this context uses a micro-controller to
continuously monitor the generated voltage. It compares the voltage with
the reference voltage set within the software and thereafter generates
control signals that select operation mode. The controller also has an
electronic voltage monitor that compares the generated voltage to the set
PWM reference voltage to determine excitation pwm duty cycle during
boost mode and buck mode.
1.2 Problem Statement
The electronic power controller for the Hybrid Field Generator discussed
in the introduction does not operate at optimum efficiency.
1.3 Sub-problem Statement
The Hybrid Field Generator required an electronic controller that would
carry out the control functions efficiently and effectively. For effective
3
power transfer the circuit has to dissipate as little power as possible to
ensure optimum performance over a wide wind speed range.
Although the charge circuit seemed to be the circuit solely responsible for
the controller performance, other circuits such as monitoring circuit and
excitation circuit, also contributes positively towards the performance. To
ensure optimum performance, the circuits power dissipation has to be as
low as possible.
To increase control abilities of the power controller, the monitoring
circuit (micro-controller) has to have analog inputs and outputs as well as
digital inputs and outputs.
The excitation circuit DC/DC converter MOSFETs driver had to be self-
oscillating to minimize the driver components count and optimize DC/DC
efficiency. Therefore, the driver ensures converter stability and reliability
hence an improved converter performance.
1.4 Hypothesis
The need for an electronic power controller has been established for the
effective control of the hybrid field generator. Based on the results
determined by an analytical investigation of the technical design
parameters of the electronic power controller, the controller performance
will be optimised producing maximum efficiency.
1.5 Project Delimitations
The project’s emphasis is on the Hybrid Field Generator and its power
controller. The power controller protects the generator from overloads and
4
short circuits. It would also provide deep discharge and overcharge
protection to the battery bank.
1.6 Project Outline
The objective of the project was to set the standards with regards to where
and how the power controller is to be operated to optimize its
performance. The controller design considerations were also taken in to
account and will be discussed in the following sections. The following
topics were taken into consideration:
Charge control circuit to stabilize the charging voltage at
28VDC for different levels of the charging current.
Control and monitoring circuit to continuously monitor the
controller inputs and outputs, and thereafter generates control
signals.
Controlled exciting current injector circuit and controlled
DC/DC converter circuit which generates the exciting current.
5
Generator
Chargecontrol
Control &Monitoring Unit
ControlledExciting Current
Injector
Battery Bank
ControlledDC/DC
Converter
DC loadcontrol
Load
Figure 1.1 Project Block Diagram
1.7 Project Significance
The Hybrid Field Generator is a Variable-speed generator and as a result
the generated voltage fluctuates in direct proportion to the wind speed.
Although the generated voltage would tend to fluctuate, a charge control
circuit is used to keep it constant over a wide range of wind speeds. This
is achieved by continuous monitoring of the generated voltage and by
varying the generator air gap flux density.
The power controller ensures the improved generator reliability, lower
operation power losses and lower maintenance costs. The objective is not
only to maintain the generated voltage and to keep the generator operating
safely, but also to maximize the generator service period.
The generated voltage is dependent on the wind speed, therefore the
power controller controls forward exciting current when there is
6
insufficient wind speed and vice versa (Van der Linde, 2001, pp.63-100).
The controller does not only stabilize the generated voltage but also
protects the generator against overloads and short circuit. It also provides
protection against overcharging and deep discharging of the battery bank
as was mentioned earlier. This is necessary as it prevents the battery bank
from being damaged.
1.8 Methodology
The following procedure was followed in the project research:
A literature survey was done and related information was
accumulated from relevant information sources.
The generator was tested for performance although there
were some limitations imposed by the test equipment during
air gap flux density tests; limitations like temperature change
with respect to time (dT/dt) at maximum speed and maximum
load. The tests results were analyzed and documented.
Study characteristics of switching components like
MOSFETs and BJTs. Sample circuits will be designed, built
and tested on the project board as per section and the results
will be analyzed accordingly.
Some circuits will also be simulated with circuit-maker. This
will be done as a further study on the circuit performance.
The project schematics, as per controller part, will be drawn.
The project PCBs will manufactured and populated for
further testing of the power controller main parts. This will
7
be done to facilitate the controller performance data
acquisition for further analysis.
The PIC 16F877 development system board will be populated
and will be used as a sample micro-controller board for the
project. The project communication control command
software functions will also be tested and modified
accordingly.
The tests results will be complied and documented
8
Chapter 2
Project literature
2.1 Literature overview
The control of the voltage generated by the generator is achieved by
adjusting the generator’s air gap flux density during the buck, boost and
permanent magnet modes of operation. This is because the generated
voltage is directly proportional to the flux density. The generator
specifications and its magnetic response curves under boost mode, buck
mode and permanent magnet mode described in (Van der Linde, 2001,
pp.63-100) are similar to the ones discussed in this document. The
response curves were drawn after performing number of tests on the
generator. Refer to appendix B.
The power controller employed power and digital electronics as well as
software commands to continuously monitor the generator DC output and
the battery bank voltage. It also generates control signals during
generator’s three modes of operation:
Permanent magnet mode
Buck mode
Boost mode
Power electronics and digital electronics control techniques employed in
the project are discussed in the literature. The techniques are classified
according to the following subsystems:
Charge Control
Control and monitoring
Excitation control
9
2.2 Hybrid Field Generator characteristic background
The Hybrid field generator operates in the same manner as a permanent
magnet synchronous generator. The output of the generator is rectified
with either a controlled or an uncontrolled full-wave bridge rectifier. The
generator air gap flux density is controlled by a wound field coil.
Depending on the generator speed the field coil is either excited in buck
or boost mode to keep generated voltage constant.
2.2.1 Permanent magnet mode
When the generator is running at 149 rpm, the permanent magnet flux will
be sufficient to produce 24VAC . That is, ideally
ex = m 2.1
Where Φ ex is excitation flux
Φ m permanent magnet flux
This is true if the leakage flux is not considered. The speed could also
extend beyond the synchronous speed, 750 rpm (50 Hz), in this
application. Because the generator charges a 24V battery bank, the
permanent magnet mode is suitable for 149 rpm (Van der Linde, 2000,p
63).
Figure 2.1 shows the relationship between the generator speed in rpm and
the generated voltage, DC voltage. It is clear that when speed increases
above 149rpm, the generated voltage also increases to levels high than the
desired voltage level. Conversely, the voltage level decreases with the
decreasing speed.
10
Permanent mode V/Speed curve
0
100
200
300
0 500 1000 1500 2000
Speed (rpm)
DC
Vol
tage
(V)
Voltage
Figure 2.1 Permanent magnet mode generator speed vs generated voltage
Average permanent mode flux density Curve
0.00
2.00
4.00
6.00
8.00
10.00
12.00
0 1 2 3 4 5 6
Pole teeth position
Flux
den
sity
(mT)
0 A
Figure 2.2 Average permanent magnet air gap flux density distribution
over a five teeth pole.
2.2.2 Boost mode
During boost mode the flux density was increased by 65% over permanent
magnet flux density level at maximum exciting current, 5A.
An increase in forward exciting current resulted in a proportional increase
in the generated emf.
11
With reference to figure 2.3, it is clear that the generated voltage is
directly proportional to the exciting current and the generator speed. This
shows that although the generated voltage decreases with a decrease in
speed it can still be maintained by increasing the exciting current. An
increase in the exciting current causes a proportional increase in the air
gap flux density. See figure 2.4.
ex = (m + F ) - L 2.2
Where Φ F is field coil flux
Φ L is leakage flux
Boost V/Speed Curves (O.C)
0
100
200
300
400
500
600
0 500 1000 1500 2000
Speed (rpm)
Vol
tage
(V)
0 A1 A2 A3 A4 A5 A6 A7 A
Figure 2.3 Boost mode speed vs voltage at different exciting current
levels
Figure 2.3 shows that besides an increase in speed the generated voltage
also increases with the increasing exciting current. This is because the air
gap flux density is directly proportional to the exciting current.
12
Average Boost mode flux density
0.002.004.006.008.00
10.0012.0014.0016.0018.00
0 1 2 3 4 5 6
Pole teeth position
Flux
den
sity
(mT) 0A
1 A2 A3 A4 A5 A
Figure 2.4 Boost mode average air gap flux density at different exciting
current levels
During the boost mode, the field coils flux flows in the same direction as
that of the permanent magnet, therefore the active air gap flux density is
the sum of the fluxes.
2.2.3 Buck mode
During buck mode the generated voltage is kept constant at 24V
regardless of increased speed. This is achieved by reducing the exciting
current. The exciting current polarity is also reversed in the process.
Under these conditions the flux produced by the field coils flows in the
opposite direction to that of the permanent magnet, hence a reduced air
gap flux density. Therefore
ex = m – (F + L) 2.3
13
Buck V/Speed Curves (O.C)
0
100
200
300
0 500 1000 1500 2000
RPM (rpm)
Vol
tage
(V)
0 A-1 A-2 A-3 A-4 A-5 A-6 A-7 A
Figure 2.5 Buck mode speed vs voltage at different exciting current levels
Average Buck mode flux density
0.00
2.00
4.00
6.00
8.00
10.00
12.00
0 1 2 3 4 5 6
Pole teeth position
Flux
den
sity
(mT) 0A
-1 A-2 A-3 A-4 A-5 A
Figure 2.6 Buck mode average air gap flux density at different exciting
current levels
It was found that in buck mode flux density could be reduced by 60%
below the permanent magnet flux density.
2.3 Charge control
As the power controller is expected to be more efficient, an uncontrolled
rectifier is used for AC/DC conversion. The uncontrolled rectifier is
14
simple and it does not need a control circuit, as it is a case with a
controlled rectifier.
With reference to figure 4.9, Q1 occasionally transfers excess voltage to
the over voltage load to ensures that the bypass transistor is operated
below its VC E (SAT) .
The charge regulator maintains the output voltage at the rated voltage,
and provides the generator with short circuit and overload protection. It
also prevents the battery bank from being over charged or deep
discharged. It has the ability to reduce the AC component that is usually
superimposed on the charging current and voltage. The AC component
normally causes the battery bank to heat up and eventually starts gassing
(Lander, 1993, pp. 207-210).
The choice of power components was based on the merits of one
component over the other for a specific application such as MOSFETs as
opposed to BJTs with emphasis on power dissipation and operating
frequency bandwidth. MOSFETs dissipate less power during conduction
because their forward resistance is smaller than that of BJTs. They can
also handle high frequencies better than BJTs.
LM723 was selected over other linear voltage regulators because of its
popularity in DC power supplies. This is further discussed in section 4.4.
The blocking diode and the freewheeling diode are introduced to protect
the charge control circuit from the back EMF (Ahmed, 1999, p.157). The
blocking diode also protects the charge control circuit form the battery
bank discharge. This takes place when the charge voltage level is below
that of the battery bank.
15
Figure 2.7 Charge control block diagram
Figure 2.7 shows how the generated voltage is processed to charge the
battery bank. The first stage is the generator stator. The stator is a star
configured source that delivers the generated voltage to the uncontrolled
three-phase bridge rectifier. The rectified voltage is then fed to an over
voltage load control circuit. After this stage follows the 28V voltage
regulation circuit and the back emf protection diodes which are blocking
and freewheeling diodes.
2.3.1 Rectification
The rectification forms the first part the controller. It converts the AC
voltage from the generator into a DC voltage. The AC voltage waveform’s
magnitude and frequency are not stable. They fluctuate in sympathy with
the wind speed.
16
According to Faradays law, the voltage induced in each of the stator coils
is be expressed as:
ea = ωNΦ p sin ωt 2.3
= Ea max sin ωt
Because the generator is a balanced star configured generator, the induced
voltages are displaced by 120° from each other (Sen, 1997, p.215). That
is, if ea is at 0,° eb is at +120° and ec is at -120° if positive phase
sequence is followed. They are also expressed as:
ea = Ea max sin (ωt) 2.4
eb = Eb max sin (ωt +120°) 2.5
ec = Ec max sin (ωt - 120°) 2.6
Owing to the fact that the generator configuration is balanced, all the
induced voltages are equal in magnitudes and so are the maximum
voltages. The latter is expressed as:
Emax = 4.44fNΦ pKW 2.7
The line voltages: VAB , VB C and VC A are also equal and are expressed as
VAB = √3 E sin ωt . 2.8
Figure 2.8 is the graphical representation of the line voltages and the DC
voltage after line voltage rectification. The ripple voltage is smaller
because the line voltages are rectified with a six-pulse diode rectifier and
therefore a reduced AC component exists in the charging DC voltage.
17
The ripple fluctuates between 1.414 V s and 1.225 V s where V s is the RMS
value of the line Voltage. An average output voltage is expressed by
Vo (avg) = 1.654 Vm 2.9
Where Vm is maximum phase voltage, therefore in terms of line voltage it
is given as
Vo (avg) = 0.955 V lm 2.10
Where V lm is the maximum line voltage. According to Ohms law an
average output current is given by:
Io (avg) = Vo (avg) / R (Ahmed, 1999, pp. 200-202). 2.11
Figure 2.8 Three-phase AC/DC converter
18
Uncontrolled rectification
One of the wind turbine system’s objectives is to capture and store most
of the energy in a cost effective manner. An uncontrolled rectifier
presents the system with a simple and cost effective AC/DC power
conversion. This is because it does not need a drive circuit and has
minimum power loss, because there are only six converting components.
See figure 2.9.
3phgenerator
Voltageregulator
ABC
Figure 2.9 Six-pulse bridge rectifier
Controlled rectification
A controlled rectifier’s main drawback is the production of radio
frequency interference (RFI) and generation of harmonics (Ahmed, 1999,
p. 151). This rectifier needs a drive circuit that introduces complexity in
rectification mainly in rectifying the generator AC voltage. The other
problem is that the generator is a wind driven generator. The result is that
the frequency of the generated AC voltage is not periodic. Therefore, an
on-delay timing circuit becomes very complex.
2.3.2 DC voltage and current regulator
The voltage regulator is the key component of the charging circuit. The
regulator ensures that the charging voltage does not follow the input
voltage fluctuations. It maintains the voltage stable at 28V, which is used
19
to ensure that the charging current flows from the regulator to battery
bank as well as to other DC loads connected to it . The voltage level
accommodates the battery bank full charge voltage level. The voltage
level is 27.24V for a 24V battery bank at 25oc (Clive, 1991, p. 208) . Table
3.1shows lead acid cell voltage.
The regulator circuit incorporates current limiting to provide over load
and short circuit protection to the charging circuit. This is also protecting
the generator. In most cases current limiting with fold back characteristics
is preferred since it restricts the short circuit and overload current to a
low value, less than the short circuit current and the overload current.
Since the input voltage can be as high as 125VDC , an over voltage load is
used to reduce stress on the regulator circuit bypass transistor. This way
the circuit is protected from high voltages that lead to high power
dissipation on the bypass transistor, which could result in the transistor
failing.
Foldback current limiting
The charge current protection employed is foldback current limiting. This
technique provides both over current and short-circuit protection to the
charge circuit. It is very dominant in linear power supplies (Billings,
1999, p. 113). It is capable of limiting current to less than overload
conditions when there is a short circuit or an overload, hence power loss
reduction on the linear series components of the circuit.
20
2.3.3 Charge load protection
Although the charge circuit maintains the charging voltage, it also
provides the battery bank with an overcharge and deep discharge
protection. It also eradicates under voltage and over voltage conditions
from the 24V sensitive DC loads by isolating them from the battery bank
when the condition exists.
2.4 Excitation control
This sub-system deals with the generator air gap flux density control,
which is performed by adjusting the exciting current during buck and
boost mode (Van der Linde, 2001, pp. 63-100). The DC/DC converter is a
push-pull converter. It is used because of its advantages over other
converters; one of them being the fact that it uses transformer coil
bidirectionally. The converter is driven from a self-oscillating FET driver,
IR2153D, to switch current in an alternating manner through the
transformer primary coil.
The exciting current is adjusted with a step down DC/DC chopper during
the forward and reverse excitation. Refer to section 5.2 for further
discussion on step down DC/DC chopper operation.
The exciting current is generated from the DC/DC converter that is driven
from the battery bank. Its magnitude is determined by the duty cycle of
the PWM signal that drives the step down DC/DC chopper. It is then fed
to the polarity changer that changes the current direction through the
generator rotor coil. See figure 2.10.
21
Figure 2.10 Excitation control block diagram
2.4.1 DC/DC converter
A power electronic converter employing switching devices such as
MOSFETs is used to control rotor excitation of the generator. This
includes the associated control and interfacing circuits (Rahman, 2002, p.
4). It allows fast control of exciting current and voltage. The control of
the converter takes the efficiency and reliability of the converter into
account.
The power converter’s power supply is a fixed DC/DC converter. In order
to maintain the generated voltage, the air gap flux density of the generator
is either increased or decreased by exciting the rotor. Therefore, the DC
current has to be controllable. The excitation is inversely proportional to
the rotor speed and the generated voltage.
The power converter configuration used in this application is a class A
type buck chopper. The chopper is driven from a switched PWM
generator. The configuration uses gate turn-off devices such as MOSFETs.
22
The DC/DC converter is a push-pull type. The DC converter in this
application uses power semiconductor switches that are operated in the
switched mode. When they are turned off, they block the supply voltage
across them with no current flow, hence dissipating zero watts. This is
due to
P = I * V (w) 2.12
where P is power, I is current and V is voltage. Therefore, when I is equal
to zero amperes P will equal zero watts. When they are turned on, the
voltage drop across them is very low for IRF9640 when VG is equal to
12V. Refer to figure 4.8. When VG is 12V Vd s is approximately 0.011V.
At full load, at 10A, the power dissipation is equal 0.11watts. Therefore
they do not consume significant power in both the on and off state.
An overlap of the switches turn-on and turn-off transients is carefully
considered in selecting the switching frequency devices. This is so to
ensure that power losses due to switching are lower compared to the
output power.
Converter transformer
Ferrite is an ideal material for inverters transformers and inductors
operating in the range of 20 kHz to 3 MHz. This feature makes it an
excellent material for the named devices. The material has high
permeability and very high resistance to eddy currents. Therefore, eddy
current losses are negligible. However, it is not recommended for high
current applications because it has lower saturation flux compared to that
of laminated and powdered iron material (Pressman, 1989, pp. 240-241).
23
A push-pull inverter is efficient because it makes bi-directional use of a
transformer core, thus providing an output with low ripple and low noise.
Although the inverter is efficient, it suffers from flux imbalance that in
most cases leads to failure. Therefore, the choice of switching devices is
very important. The flux imbalance in the inverter becomes less serious
when MOSFETs are used as switching devices.
The inverter transformer core is gapped to avoid saturation under a DC
bias condition. This is because the saturation causes transistor failure in
push-pull topology when transistors have unequal switching
characteristics (Magnetics). The core can then handle larger volt-second
inequality (Pressman, 1998, p 49).
When the gap is placed in series with the magnetic flux lines, it t ilts the
slope of the hysteresis loop, keeping the point where the loop crosses the
zero-gauss (0-G) level (called coercive force Hc) fixed (Pressman, 1998, p
49). See figure 2.11.
Figure 2.11 Gapped and un-gapped core magnetic characteristic curves
24
EMI filters
The switch mode power supply generates excessively high frequency
noise, RFI and EMI, which hampers the performance of the micro-
controller. The noise can be reduced with an EMI filter. The filter is
inserted between the switch mode power supply and the control circuits of
the controller. The commonly used EMI filters are:
• Common mode noise filter
• Differential noise filter
Common mode noise filter Differential noise filter Figure 2.12 Line noise filtering
Windings are connected in series with input power lines in common mode
noise filters. The windings are such that the flux set by one winding
cancels the one set by the other winding, keeping the power lines free of
noise.
Screening
Although gapping reduces residual magnetic effects, it radiates EMI and
RFI that causes interference in the other parts of the circuitry. The
radiation effects are reduced by screening the transformer with a thin
copper screen that is 1% percent of the rated output power depending on
the air gap (Billings, 1999, pp. 1.46-1.48)
The air gap for ETD core is usually filled with a plastic shims in the
center and outer legs.
25
The shims reduce RFI generation in the transformer as the gaps are
constant but RFI is reduced further when the center leg is ground down to
twice the shim thickness (Pressman, 1998, pp. 49-50).
Figure 2.13 Screened transformer
2.4.2 Exciting current control
This section describes figure 5.3 . A DC/DC converter is the first stage of
the exciting current control. It converts 24VDC to 48VDC . The converter is
driven with self-oscillating PWM IC, IR2153 , for the DC conversion. The
converter output is fed to the PWM chopper whose main task is to vary
the exciting current in proportion to the generated voltage deviation from
the control reference voltage. The deviation determines the driving signal
duty cycle. See figure 5.21.
The polarity of the exciting current is also determined by the nature of the
voltage deviation. The positive deviation occurs when the generated
voltage is greater than the control reference voltage. During this time
26
buck mode is selected and the exciting current polarity is reversed so that
the air gap flux density is reduced. Conversely, the boost mode is selected
and the flux density is increased.
DC buck chopper
A DC/DC converter is used to convert a fixed battery bank voltage to a
DC voltage that is adjusted with a DC buck chopper for the rotor
excitation. The chopper can either be driven from a PWM or PFM circuit.
A DC chopper varies generator rotor exciting current to levels that best
suit the excitation requirements. This happens at every instant of change
of generated voltage above or below the generated voltage reference
voltage. As a result, air gap flux density is varied accordingly.
Vex = tON * VS 2.13 (tON + tOFF)
= tON * Vs
T
Therefore
Iex = Vex Ra
PWM is used to vary the excitation voltage at switching frequency of
2KHz.
The chopper is operated in two modes, continuous and discontinuous
mode under both buck and boost mode of the generator.
27
During continuous mode the period T is very small compared to the
chopper time constant τ, that is T << τ. The time constant is a ratio of the
armature (rotor) inductance to the armature resistance.
τ = La 2.14 Ra
In continuous mode, the current Ia continuously flows into the rotor. This
feature makes continuous mode preferable in armature excitation.
Discontinuous mode is undesirable because of the breaks in the Ia .
Normally this mode comes into play when TON is approximately equal or
greater than τ.
TON
TTOFF
Imax
Imin
Ia
iin
iD
Continuous mode
Va
Figure 2.14 Continuous waveform
Discontinuous mode occurs as a result of a small armature inductance.
This can be avoided by increasing the armature inductance. The
inductance is increased by adding more armature coil turns such that
2πfLa >> Ra (Rahman, 2002, pp. 4-17).
28
The minimum required inductance is determined as
La = TOFF * Ra 2.15 2
TON
T TOFF
Imax
Ia
iin
iD
Discontinuous
Discontinuous mode
Figure 2.15 Discontinuous waveform
When the rotor coil inductance is increased, its current peak-to-peak
ripple reduces. The other way is by increasing the switching frequency so
that the period T is much smaller than the time constant τ.
If the rotor (armature) inductance is high, the excitation current Iex ripple
is approximately Zero. Then the current can be given by
Ia = Va 2.16 Ra
where Iex = Ia
In addition, the armature inductor voltage is expressed as
Ea = L dia 2.17 dt
29
2.4.3 Current protection
The excitation current protection is achieved by continuous monitoring of
the exciting current, so that the converter can be shutdown when the
current exceeds the rated current. In addition to this, a rated current fuse
is also included in the control circuit.
2.5 Charge and excitation control system
This sub-system forms the core of the hybrid field wind energy system. It
monitors the inputs and outputs of the power control system and it
generates control signals accordingly. The micro-controller interfacing
circuits are discussed in chapter 6. The circuits isolate the micro-
controller from the noisy analog circuit and also protect the micro-
controller input and output pins from voltages higher than 5V. The
circuits also provide the micro-controller with noise protection and
interference protection to ensure an effective and efficient operation. See
section block diagram, figure 2.17.
Active filter design
In analog signal processing, filtering plays an important role. It is used to
get rid of unwanted frequencies, commonly known as noise. Practical
filters are normally of the second or higher order. There are three
configurations of active filters, namely
• Butterworth filter which presents flattest response characteristics
• Chebyshev filter which presents steeper roll-off with less flat
response characteristics
• Bessell filter which presents rapid roll-off with a number of
ripples
30
Figure 2.16 Frequency response of three common types of filters
The filters present the following advantages to the signal translating
circuit:
• Provide gain to overcome signal attenuation in the filter.
• Present high input impedance to prevent excessive loading of the
driving circuit and present low output impedance to prevent the
filter from being affected by the load that it is driving.
• The gain can be easily adjusted over a wide range of frequencies
without changing the desired response (Boylestad, pp. 683-687).
• Sharp cutoff characteristics and high-level attenuation of unwanted
signals.
They are compact and cheaper than those that use passive components.
31
Figure 2.17 Charge and excitation control system
2.5.1 Micro-controller selection
With reference to figure 2.17, nine inputs are monitored with a micro-
controller. The inputs are read from three main parts of the system, which
are:
• Stator line current and voltage circuit,
• Rotor excitation circuit and
• Charge and monitoring circuit.
All the line currents and speed are measured at the stator side of the
generator. They are depicted as Ir , Iy, Ib and Sp respectively. The four
inputs are linked optically to the micro controller.
32
The exciting current Ir t , l ike the rotor voltage V r t are on the rotor side of
the generator. Ir t is magnetically linked to the microcontroller with a hall-
effect sensor and V r t is optically linked with a linear opto-coupler. This
way, the microcontroller is isolated from the rotor
The generated voltage, charge current and the battery voltage are read
directly from the charge circuit. This is due to the circuit being supplied
from the same source with the micro-controller, unlike the inputs from the
stator and the rotor.
When the inputs are read and executed, the control command signals are
sent to: DC/DC converter, excitation polarity changer, fault load connect
and disconnect (Con and Dis) and DC load connect and disconnect
respectively.
The description in the preceding paragraph and figure 2.17 shows that a
micro-controller with at least 8-analog input channels and 5-digital output
may be used. A single channel can still be used at the cost of execution
time. This is because the analog inputs are multiplexed before being read.
Digital inputs requirement is determined by the method used to read
speed. That is if an 8-bit counter is used to count the generator speed
pulses an 8-bit inputs port would be required for speed. Alternatively, the
micro-controller could be used to read speed pulses directly from the
speed sensor.
2.5.2 Signal conditioning
In section 2.5.1, 8-analog inputs are read with the micro-controller but
prior to the actual reading, the signals are conditioned so that their
magnitudes fall within the 0 to 5V voltage range. The conditioning
process is performed with linear operational amplifiers.
33
The linearity of the signals eliminates a need for a complex mathematical
function within software commands. Therefore, the analog inputs can be
expressed as:
y = mx + c 2.18
Where y is the tailored analog output that corresponds with the actual
parameter, m defines the gradient of the y, dy/dx. x is the analog input
and c is the constant value that exists in the input signal when the
measured quantity is zero. This value signifies the value of y when x is
zero. It is set to zero with a zero adjust.
Figure 2.18 Line current sensor block diagram
Figure 2.18 shows an example of a signal conditioning circuit block
diagram. From the current sensor, the current signal is fed to a low pass
filter and then an opto coupler. The signal is then sent to a differential
amplifier. After the amplifier, the signal is converted from analog to
digital value that can be understood by the microcontroller.
2.5.3 Interfacing
As the generator speed is the only input that involves timing, two methods
of speed sensing are discussed in the following paragraphs.
34
Opto-electronic speed sensor
This type of sensor operation principle is based on counting the number of
either positive half cycle, negative half cycle or both half cycles of an AC
voltage waveform. This takes place after the AC signal has been half
rectified or fully rectified. Under a fully rectified signal condition, the
counter gives the count that is twice the actual number of cycles
completed. Therefore, two to get the actual number of cycles completed
divides the count.
The generator speed measured in hertz with the opto-electronic speed
sensor. The generator speed can be expressed in RPM as:
N = f * 60 2.19 p
Where: N is revolutions per minute (rpm)
f is generator frequency in hertz (Hz)
p is number of pole pairs
Each negative half cycle generates a pulse across an opto isolator. These
pulses are read with a digital counter that converts them into binary data.
This can be used to generate control signals within the micro-controller.
The sensor has a pulse count limit of 60 Hz set within hardware as
indicated in figure 6.4. Further details about this sensor are discussed in
the following sections.
Figure 2.19 Speed sensor block diagram
35
Opto-mechanical speed sensor
With reference to figure 2.20, the speed disk is an optical encoding disk.
The disk has holes around its outer edge. On either side of the disk, there
is an infrared LED and an infrared sensor. The spaces between the holes
in the disk break the light beam from the LED, so that the infrared sensor
picks up the light pulses as the rotor shaft rotates. The rate of pulsing is
directly proportional to the speed of the rotor. The pulses are read with
the micro-controller that turns them into binary data. An added advantage
of the opto-mechanical isolator is that it does not need secondary power
supply compared the opto-electronic sensor.
In this case, the generator speed is determined by dividing the pulses
count per second with the number of disk holes. That is
f = Cp /CH (Hz) 2.20
N = f * 60/p (rpm) 2.21
Where Cp is the pulse count per second
CH is the number of disk holes
36
+5V +5V
Pulses
Rotor Shaft
Speed disk
Optical Speed detector
Shaft
Disk
R1 R2
Figure 2.20 Opto mechanical speed sensor
Opto-mechanical (stray light)
It was found that almost every opto-mechanical sensor suffers from
optical noise (Gauvin, Freniere). The noise is normally caused by bright
sources of light, light that manifests itself in two ways: ghost images and
scattered light that reaches the infrared receiver and introduces false
pulses. In this application, a stray light source is an arcing between the
generator brushes and the slip rings. This is due to the generator
vibration.
Stray light effects can be reduced by shaping a speed disk as indicated in
figure 2.21 and by using non-reflective material.
37
Figure 2.21 Speed disk mounting
2.6 General aspects of the project literature
PIC micro-controller and the 8051 micro-controller families development
systems, like MPLAB, are found online from Microchip and from Intel
respectively. The micro-controllers are easily affected by noise and
interferences, like EMI, and these hamper them from performing at
optimum efficiency. The microcontrollers have to be protected against
noise and interferences for a better performance. The high efficiency and
accuracy in micro-controllers can be achieved by using linear amplifiers
and linear signal translating circuits such as an instrumentation amplifier.
Most of the switching, in power controllers, is done by means of
MOSFETs. VGS of the MOSFETs is limited to voltages up to 20V
therefore MOSFETs should have gate to source voltage protection to
counter act negative effects of gate to source over voltage. The effects
such as gate to source over voltage lead to MOSFET failure.
38
According to the generator tests and specifications, the exciting current is
limited to 5A maximum (Van der Linde, 2001, pp. 69-71). It is therefore
important to consider current protection for the project. Foldback current
limiting presented a continuous load connection to the generator as its
short circuit and overload currents are lower than the load current
(Loveday, 1992, pp.83-87).
Figure 2.22 Line currents and speed sensors power supply and rotor
excitation current control power supply
The controller has three independent circuits that are electrically isolated
from each other. See figure 2.17. The circuits are linked optically and
magnetically to bring about communication between them and the micro
controller.
Although the line currents and the generator speed sensing as well as
rotor excitation circuit are powered from a DC/DC converter, batteries
can be used as an alternative.
39
Chapter 3
Controller specifications and considerations
3.1 Overview
The controller specifications and considerations are determined by the
generator specifications and its optimum operation requirements. Due to
these requirements, the generator was tested for more information about
its hardware as well as its performance.
3.2 Generator specifications
Generator Specifications:
Power 1.25 kW
Number of phase 3 Ph star configuration
Generator voltage 24 VAC => 32.414 VDC
Load current 7.404AAC => 10ADC
Design speed 264 rpm
Cut-in speed 149 rpm
Cut-out speed 548 rpm
Field: Coil resistance RC 3.367 Ω
Current 5 ADC
Coil power loss 84.175 W
(Van der Linde, 2000, pp.46-47)
40
Measured rotor and stator quantities:
Rotor: Coil inductance La 40.9 mH
Coil resistance Rc 3.5 Ω
Field circuit resistance RFC (Ra) 4.18 Ω
Brushes resistance, slip rings resistance and contacts resistance are a
difference between RC and RFC .
Therefore, their resistance is 0.680 Ω .
Stator: Coil inductance per phase Ls 4.75 mH
Coil resistance per phase R s 0.611 Ω
3.2.1 Generator tests
The generator tests discussed in this section were performed to determine
some of the hardware quantities, such as rotor inductance, that were not
specified in (Van der Linde, 2001, p 46). The following tests describe the
tests performed on generator:
• Rotor DC test and AC test to determine rotor actual resistance and
inductance respectively.
• Stator DC test and AC test to determine stator actual resistance and
inductance respectively.
The following test were performed under permanent magnet mode, buck
mode and boost mode of the generator excitation at various speeds except
for static test:
• Open circuit test and short circuit test
• Static air gap flux density distribution as per pole
• Dynamic air gap flux density
• Load tests at mid range speed, 374rpm, and at 600rpm
• Synchronous load test
41
• Open and closed rotor terminals back emf tests with blocking diode,
freewheeling diode and zener diode for voltage clamping. The test
was also performed without the zener diode.
Refer to appendix B and C for the test results.
DC/ACcurrentsource
A B
Primmover
VSD3 ph
mains
vA
vv
A A
Wind generator
Stator
Rotor
R
Y
B
RL
A DC/ACcurrent sourceB
Y
C
Figure 3.1 Generator tests setup
Rotor and stator hardware tests
The fist rotor test was the DC test. Switch A was closed for this test. A
small DC voltage was applied across rotor terminals so that one-ampere
flowed into the rotor. So according to Ohm’s law
R = V 3.1 I
Where V is applied voltage
I is the current flowing into the rotor
R is the rotor resistance
Therefore R ro t or = 4.18Ω (V s = 4.2V)
42
The similar DC test was performed for the stator test although R, in this
case, was the sum of two stator-coils resistance. For example, when DC
was applied between line Y and B, as shown in figure 3.1, R was the sum
of Rco i l Y and Rco i l B . Therefore, the stator R per coil was
R = R = 0.611Ω (V s = 9.61V) 2
The rotor AC test was performed in a similar manner to the DC test except
that an AC source was used. The current flow was once again set to one
ampere. So, because the rotor impedance Z expression was:
Z = V = √(R ro t or2 + XL ro t or
2) 3.2 I XL ro t or = √(Z2 – R ro t or
2) 3.3
Therefore Lro t or = XL ro t or*ω-1 = 41.195mH (V s = 13.6V)
The stator AC test was performed in a similar manner to that of the rotor.
Because the stator coils were in series, they were treated in the same way
as the resistors in series. Each coil inductance was equal to half the two
stator coils inductance. The stator inductance per coil Ls t a t o r was
Ls t a t o r = L = 4.75mH (V s = 9.6V) 3.4 2
Open circuit test was performed with switch A closed at all levels of
exciting current at different speeds. The generated voltage was measured
after a six-pulse bridge rectifier with switch B open. Similarly, the short
circuit was performed with switch A, switch B and switch C closed. The
short circuit current was also measured after the bridge rectifier. The test
results indicated that the generated voltage and the load current were
directly proportional to the generator speed and the exciting current.
43
The load tests were also performed in the similar manner to those of the
open and short circuit except that switch C was left open. It was found
that the generated voltage and the load current were indirectly
proportional to each other.
3.3 Battery bank specifications
Battery bank specifications were determined from table 3.1. Because a
24V battery had 12 cells, the maximum voltage range was 27V – 27.24V
over temperature range of 25°c - 29°c respectively.
Table 3.1 Recommended battery cell voltages
3.4 Components selection
The component selection was based on the merits of individual
components towards the controller hardware for optimized performance.
Components with minimal power losses were preferred, mostly the
temperature compensated components. These types of components assured
a stable and reliable hardware and linear response for accurate hardware
monitoring.
44
3.4.1 Discrete components
Switching components such as MOSFETs are susceptible to failure in
switching converters. This is because they are subjected to severe
switching stress due to rapid change in voltage and current. Although
MOSFETs have increasingly replaced BJTs in switching converters, they
still need to be operated within their safe operating areas to ensure
reliability and optimum converter performance (Ang, 1995, p. 320).
MOFETs are preferred in switch mode power supplies and push pull
inverters, because they do not have storage time, as is the case with BJTs.
More importantly, the on-state voltage in MOSFETs increases with the
increasing temperature. Thus the runaway condition would not occur. The
fact that the MOSFET on-state VGS voltage is proportional to temperature,
it provides feedback, which tends to correct the flux imbalance (Pressman,
1998, pp. 44-45).
BJT drive circuits are complex as BJTs are current and voltage controlled
devices unlike MOSFETs. MOSFETs are voltage-controlled devices and
they need simple drive circuits (Ahmed, 1999 p. 39) although their VGS
has to be maintained at 12VDC by means of a zener diode.
Schottky diodes were preferred over junction diode for their fast
switching speed. The diodes turn-off time made them suitable for high
frequency applications such as DC/DC converters. The converter
switching frequency is 100 kHz and PWM frequency is 2 kHz.
Switching loses
Switching losses at high frequencies in a power transistor, BJT, could
contribute to more than half of the total power loss. This could be because
45
of overlapping of the collector current and collector-to-emitter voltage
during turn on and turn off process of the BJT. The power loss during this
process would be the sum of power during delay time and the rise time
(Pressman, 2002, pp 4-17, 60-62)
IGBTs have an advantage over both MOSFETs and BJTs because they are
a combination of the two named transistors. Thus, they can serve as either
MOSFETs or BJTs. However, they have a very limited frequency range,
10 – 50 kHz, (Ang, 1995, p. 351). Owing to this fact, MOSFETs were
chosen over the IGBTs because their range extends into Megahertz. This
was because the converter frequency was chosen to be 100 kHz.
3.4.2 Passive components
Fuses
Most applications call for protection against fault conditions that in most
cases destroy semiconductor switches like those of the IR2153. Fusing is
the simplest protective option although it is often ineffective at
preventing damage to the switches, and is not resettable. This is a
problem with applications where very fast response is required. However,
protection is sometime achieved by a rapid turn off of the FETs.
Resistors
Trimpots, cermet type, were found to be reasonably high quality
components with good stability. They are very dominant in applications
where small signals are amplified like in measuring instruments that are
microprocessor or controller based.
46
Metal film resistors also play an important role in bringing quality,
stability and accuracy in a signal translating circuit. It is because they
have very low tolerance, equal or less than +1%, and very low
temperature coefficient, + 50 ppm, as well as extremely low current noise
level (RS Components, 2003, p. 801).
3.4.3 Integrated circuits
Figure 3.2 IR2153 functional blocks
IR2153D is an upgrade of self-oscillating control IR ICs and it has
enhanced electrical performance and functionality. The IC has a built in
feature, under voltage detect UVLO, that ensures that the gate drive
outputs, HO and LO are both low when bias becomes too marginal for
comfortable gate drive to the output transistors. UVLO also ensures a
repeated start-up sequence and control bias current required for various
IC elements.
Most of the control ICs like IR2153D, are susceptible to electrical noise
that impairs their performance. However, the noise is reduced with a
47
decoupling capacitor and a reservoir capacitance from Vcc to com (IR
design tips).
IR2153 also provides a shutdown mode that is used to shutdown its
operation when there is a fault. This mode is also used to stop the
conversion process. In addition to this, the mode acts as excitation master
switch. Refer to section 5.2 for application description.
Operational amplifier
Operational amplifier, op-amp, is one of the key components in analog
design circuits. It is used dominantly in signal translating circuits. A
required op-amp configuration is determined by the application where it is
to be used.
The following op-amp configuration are used in the signal translating
circuits:
Inverting amplifier
Non inverting amplifier
Differential amplifier
Voltage follower
Instrumentation amplifier
R1
R2
R3
ViVo
Com
(a)
R1
R2
Vo
Com
Vi
(b) Figure 3.3 (a) Inverting op-amp (b) non-inverting op-amp
48
Inverting amplifier’s gain is determined by
Av = -R2 3.5 R1
where R2 is a feedback resistor. This amplifier has no common mode error
and temperature drift is reduced by making
R3 = R2 3.6 R1
See figure 3.3 (a).
Non-inverting amplifier’s gain Av is equal to
Av = 1 + R2 3.7 R1
In addition, it has a small common mode error.
See figure 3.3 (b)
R1
R2
R3
R4
dVi
Vo
Com
(a)
Vi
Vo
(b) Figure 3.4 (a) Differential op-amp (b) Voltage follower
A differential amplifier has a common mode problem that can be reduced
by matching the resistance ratios.
49
That is:
R1R4__ = __R2R3__ 3.8 (R1 + R4) (R2 + R3)
and its gain is determined by
Av = R4 3.9 R1
See figure 3.4 (a)
Differential amplifier’s output voltage is proportional to the product of its
gain and the difference between two voltages at its inverting and non-
inverting input terminals (Carr, 1991, p. 249). That is
Vo = Av (V1 – V2) 3.10
Where Av is the amplifier gain.
Differential amplifier circuit is very economical. It needs one IC like LM
741. Although it has the ability to reject common-mode voltages when
resistor ratios are matched, it is not suitable for high gain applications.
This is due to its low input impedance (Carr, 1991, p 250).
An instrumentation amplifier is used to alleviate differential amplifier
problems.
In the case of instrumentation amplifier, R2 and R3 have to be equal to
avoid voltage gain error.
50
Its gain is represented as follows
Av = 1 + 2R3 3.11 R1
provided R2 = R3 = R4 = R5 = R6 = R7 . See figure 3.5.
Besides solving a differential amplifier technical problem, an
instrumentation amplifier presents very high input impedance and high
common mode rejection.
R5
R6
R7
R1
R2
R3
R4
VodVi
Com
Figure 3.5 Instrumentation op-amp
Figure 3.4 (b) shows how an op-amp is configured to operate as unity gain
amplifier, commonly known as voltage follower. An op-amp in this
configuration has an improved output current.
Voltage regulator ICs
Voltage regulators play an important role in keeping voltage constant as
per controller circuit hardware section. Three terminal voltage regulator
ICs such as LM78xx and LM79xx are an on-board mount type and
therefore design with these ICs is simple.
51
Three terminal voltage regulator ICs have all the important features of
series regulators as well as overload protection built into a package. A
fixed voltage regulator circuit is simpler than that of an adjustable voltage
regulator circuit. However, the former circuit is limited to the
manufacturer’s preset voltage level whereas the latter allows the user to
set the voltage level to suit the application.
Fixed voltage regulators have the following advantages that ensure a cost
effective and reliable regulation:
• The ease of use due to few external components requirement
• Reliable operation
• Built in overload protection
• Internal thermal drip
Refer to section 6.2 for further discussion on fixed voltage regulators and
section 4.4 for discussions on high power voltage regulation. The high
power voltage regulation is built around LM723 because it was found to
be common in high power voltage regulation systems.
3.5 Transformer and choke
Ferrite is an ideal material for transformer core operating in the frequency
range 20kHz to 3MHz. The core is gapped to avoid saturation under DC
bias conditions.
ETD core is used in high frequency applications due to their low cost,
ease of assembly and winding. The core is readily available for a variety
of hardware wiring. It gives adequate space for the large size wires. This
allows air to flow through the transformer window. This feature keeps the
assembly cooler.
52
The ETD core center post is round and thus windings have shorter path
length around it , 11% shorter than that of the square post. Therefore, this
reduces winding losses by 11% and enables the core to handle higher
output power (Magnetics, Page 4.6)
Transformer power handling is determined by its WaAc product where
Wa is a core window area
Ac is an effective core cross-section area
The WaAc defined by the power output relationship is obtained by starting
with Faraday’s law:
E = 4BN f Ac * 10 8 (for square wave) 3.12
WaAc = EAW * 10 8 = Po C * 10 8 3.13 4Bfk 4Bfk
k = NAw 3.14 Wa
Where:
E = applied voltage (rms)
B = flux density in gauss
Ac = core area in cm2
N = number of turns
f = frequency
Aw = wire cross sectional area in
cm2
Wa = window area in cm2
C = current capacity in cm2 /amp
k = winding factor
I = current (rms)
Pi = input power
Po = output power
e = transformer efficiency
53
Switching power transformer
Switching power transformers are called either: buck, boost, converter or
inverter depending on the application. These are specified for high
efficiency, small size, and low weight applications. In this application, it
is a converter and it operates from a 24V DC power source that is
switched at 100 kHz. The switched DC power is seen as a square wave AC
at the transformer as used in a push pull configuration. See figure 5.6.
The configuration is efficient as it makes bi-directional use of the
transformer core windings. Therefore provides an output with low ripple.
3.6 Sensors
3.6.1 Hall-effect sensors
Hall-effect sensors offer a non-contact sensing, a high degree of accuracy
and the ability to measure DC and AC currents. Owing to the fact that a
hall sensor presents non-contact sensing, there are no electrical power
losses.
The hall sensor sensitivity is proportional to the number of current
carrying conductor turns around the sensor core.
Hall effect open-loop current sensors have the following advantages over
resistive current sensing techniques:
The sensor has temperature compensation circuitry for stability and
accuracy.
It presents non-contact current sensing hence, there are no electrical
power losses.
The open loop sensors are also preferred in battery-powered circuits
due to their low operation power requirements (Bell, 2001. p 2).
54
3.6.2 Optical and electromagnetic isolators
Opto-couplers
Since mechanical switches introduce a series of narrow pulses in
electronic circuits every time when pressed, this effect contributes
positively towards noise problems in the circuits. Besides isolating the
controller from the electronic circuit, opto-couplers provide the circuit
with a bounce free switching and can only allow signal flow in one
direction.
Opto-couplers use light flux between an LED and a phototransistor to
couple digital signals and analog signals from one circuit to the other.
The coupler current transfer ratio is improved by adding a second
transistor in a darlington pair configuration.
Therefore
Ic = BIb (before adding an extra transistor) 3.15
Ic = (B1B2) * Ib (darlington pair) 3.16
Where B1 is the opto-coupler current gain
B2 is the extra transistor current gain
Although opto-couplers are popular in electronic circuits not all of them
maintain a linear response when the internal infrared diode junction
temperature increases. Nevertheless, the linear opto-coupler (LOC110)
family solves the in-linearity sad-back.
55
LOC110 assures accuracy and linear response from the input to the
output. Unlike 4N25, it has extra phototransistor that operates as a
feedback mechanism to control the diode drive current effect of
compensating for the diode’s non-linear time and temperature
characteristics (Clare, p.2). See figure 3.6.
The common methods in which LOC110 is utilized as an isolating
amplifier are photoconductive and photovoltaic configuration.
Photoconductive configuration is best suited for high frequency
applications. This is because its bandwidth is up to 200kHz. The linearity
and drift characteristics of this configuration are comparable to those of
an 8-bit D/A with ±1 bit error (Clare, p 4). This error is mitigated when
photovoltaic configuration is utilized.
Photovoltaic configuration presents the best linearity, lowest noise and
drift performance. Linearity of up to a 14-bit D/A is achieved in this
configuration although its bandwidth is limited to 40kHz. In spite of the
limitation, the configuration is still favored in the controller application.
This is because only DC analogue signals are processed. LOC110
phototransistors in this mode are 0V biased to eliminate voltage
dependence of a photogenerator and its non-linearity. This way the
linearity improves further (Clare, p 4).
56
Figure 3.6 LOC11X
Figure 3.7 LOC11X V i n vs Vou t
Relay
Relays play an important role where a certain degree of isolation is
required between the controller and the controlled circuit like the fault
load. This becomes a critical factor, especially when the two circuits are
operated from different power sources. Refer to section 6.3.
57
Chapter 4
Charge voltage regulation design
4.1 Overview
The charge voltage regulation played an important role in the controller
hardware. It ensured that the charging voltage and current were
maintained within limits, 28V and 10A respectively. It also protected the
generator against overloads.
Other important aspects were an over voltage load, battery bank
overcharge protection and deep discharge protection. It also protected DC
loads from over-voltage and under-voltage supply. All the protection
techniques ensured robustness of the controller charge unit.
4.2 Uncontrolled rectification
The literature indicated that an uncontrolled rectifier is more cost
effective compared to the controlled rectifier in this application.
Therefore, uncontrolled three-phase bridge rectifier was used to rectify
the generated voltage.
The generated line voltage and the line current waveforms were not pure
sinusoidal waveforms. See figure 4.1a. The waveforms indicated that the
generated line voltage and the line current had some harmonics. The
harmonics were considered as noise to the controller. Therefore, their
effects on the fundamental waveform are discussed in the following
paragraphs.
58
With reference to the line voltage harmonics chart in appendix C, odd
harmonics were dominant in the line voltage. The eleventh harmonic was
4.2% of a fundamental harmonic. Its period was 2.283ms.
Although even harmonics were dominant in the line current, the 19 t h
harmonic was 0.78% of the fundamental current harmonic. Refer to line
current harmonics chart in appendix C. All the information and the
waveforms were captured with Tektronix wave star software for
oscilloscopes.
TT
TT
1 >1 >1 >1 >
2 >2 >2 >2 >
1) Ch 1: 50 Volt 5 ms 2) Ch 2: 10 mVolt 5 ms
Figure 4.1a Line voltage (Ch 1) and line current (Ch 2) waveforms
The waveform displayed in Figure 4.1b indicates that each diode of the
six-pulse bridge rectifier conducted for 120° . That was because each
diode conducted for 60° in each of the two line voltages cycles, like in
VAB and VAC .
59
The waveform showed that average diode current was one-third of the
average load current. See figure 2.8 for the VAB and VAC .
The current could be expressed as:
ID av = IL av/3 4.1
In addition, the RMS value of the bridge diode was
ID R M S = IL av/√3 4.2
T
2 >2 >2 >2 >
2) Ch 2: 1 Volt 5 ms
Figure 4.1b Line current 5A waveform
Figure 4.1b shows line current waveform when the six-pulse bridge
rectifier was connected.
Because the generator is a 1.25kW generator, at 10A load the generated
voltage would be 125V. Although the generated voltage could be much
higher than the charge voltage, a desired charging voltage would still be
maintained at 28V.
60
This was because the excess voltage was diverted to an over voltage load.
Refer to section 4.2.
Figure 4.2 shows that 24V was maintained over a wide range of speed,
from 149rpm to 600rpm. During the time when the line voltage was
maintained the dump load was off. At the time when speed was above
600rpm, the line voltage increased to magnitudes greater than 24V.
Therefore, the dump-load voltage was a difference between the generated
DC voltage and the steady state voltage of 32V.
At speed higher than 600rpm, the generate voltage started to increase.
That was because the exciting current would not be increased any further.
The generator operated in the boost mode for 22.27% of the speed range.
Therefore, the generator reverse exciting current flowed through the rotor
for 77.73% of the speed range while maintaining 24V line voltage. During
boost mode the forward exciting current increased exponentially at a
higher rate than in the buck mode, and so did the power losses.
At 24V the ratio of power losses to the controller input power was higher
than when the voltage was increased to 48V. When the battery voltage
was doubled to 48V, the efficiency of the charge control was increased by
9.31% from 85.89% at full load current (10A). The operation speed range
was increased by 66% although the cut-in speed was changed to 250rpm.
See table 4.1.
The operation speed range was determined by the exciting current.
Therefore, because of an increase in full exciting current a wider speed
range was achieved. A disadvantage of the current increase was the
exponential increase of the armature loses.
61
Open circuit Excitation IV curves at 24V V-line
0
10
20
30
40
50
60
100 200 300 400 500 600 700 800
Speed (rpm)
Vol
tage
s (V
)
-8-6-4-20246
Exc
iting
cur
rent
(A)
Vline (V) Vdc (V) Vrotor (V) Iexc (A)
I => boost mode region II => buck mode region III => Excess voltage region Figure 4.2 Open circuit excitation characteristics curves at 24V line
TT1 >1 >1 >1 >
Figure 4.3 Ripple voltage waveform at 0A load current
T1 >1 >1 >1 >
Figure 4.4 Ripple voltage at 2A load current
T1 >1 >1 >1 >
Figure 4.5 Ripple voltage waveform at 2A load current and 2A excitation
current
CH 1 10V 1mS
CH 1 10V 1mS
CH 1 10V 1mS
62
TT1 >1 >1 >1 >
Figure 4.6 Ripple voltage waveform at 2A load current and 4A excitation
current.
After rectification, some voltage ripple waveforms were also captured and
analyzed. Refer to figure 4.3 up to 4.6. The analyses proved that an
increase in load current and in an exciting current caused a reduction in
ripple voltage magnitude. Although high currents increased the power
losses on board, they reduced an AC component in the charging voltage.
See ripple data in appendix C. The AC component would cause heating of
the battery bank and consequently gassing would take place.
4.2.1 Controller protection
The controller protection was carried out in two ways: by monitoring the
generated DC voltage within hardware and within software. Within
hardware, the protection was achieved by comparing the input charge
voltage tracking reference (V t r c) and the protection reference voltage
(V re f).
With reference to figure 4.7, the comparator generated a positive error
signal when V t r c was greater than V re f . This happened when V i n turned out
to be greater than a desired voltage level, 35V. The error forward biased
D2 and switched Q1 on. Alternatively, a “Low” (approximately 0V) at S4
turned Q1 on, and relay 1 (Rly1) became energized. Refer to section 6.5.
When the relay was energized, the fault load became connected to the
generator and the charge circuit was isolated form the generator. The
process was reversed when V i n was less or equal to 35V.
CH 1 10V 1mS
63
Voltageregulator
Over voltageload control
+15V
+5V
S4Vtrc
Vref
D1
Q1
R1
R2
R3R4
Pot1
Rly1
VinVg
+_
Figure 4.7 Voltage protection circuit
4.3 Over voltage load
An over voltage load (Ov) was an auxiliary DC load to the generator. See
figure 4.9. It only became part of the circuit when VDC exceeded 32V.
That was because Q1 turned off and forced charging current to flow
through Ov. As the current flowed through Ov, voltage VOv developed
across Ov and as a result, power dissipation occurred. The dissipated
power was not seen as a loss to the system. It could be used to reduce
costs in heating systems e.g. water heating system. The voltage across Ov
appeared as an excess voltage hence why the charging current was
diverted to Ov. Therefore
Vg = VOv + VDC 4.3
Where VDC was a sum of all voltage drops across series components and
thus
VDC = VQ4 + VR 2 2 + VD7 + VC g 4.4
In addition, VQ4 was forward VC E of the bypass transistor Q4
VR 2 2 was charge current sensing resistor voltage
64
VD7 was blocking diode D7 forward voltage.
When Vg = VDC
V re f > VDC t rc
the comparator generated a positive error signal whose potential was
divided between R3 and R4 and as a result, Q2 was turned on. That was
because VR 4 was greater or equal to Q2 VB E and would be expressed as:
VR 4 = VQ2 VB E = Vo * R4 = 0.6V (electrical characteristics) 4.5 (R4 + R3)
When Q2 was on, DZ clamped Q1 VGS to 12V such that VR 2 is:
VR 2 = Vg - VDz 4.6
When VDC > 32V
V re f < VDC t rc
and the comparator generated a negative signal that turned off Q2 . When
Q2 was off, there was no current flow through both R1 and R2 hence VR 1
was zero and Q1 was turned off. During that time, VOv was greater than
zero. That holds truth because according to Ohms law VOv was a product
of ROv and the charging current.
Figure 4.8 showed switching characteristics of international rectifier P-
channel MOSFET, IR9640. When the MOSFET gate to source voltage was
12V, its drain to source voltage was approximately zero. During that time,
the MOSFET was fully on. The power MOSFET could effectively be
turned on at voltages higher than it threshold voltage, 4V.
65
IRF9640 Vgs vs Vds
0.0001
0.001
0.01
0.1
1
10
100
0 1 2 3 4 5 6 7 8 9 10 11 12
Vgs (v)
Vds
(v)
Figure 4.8 MOSFET switching voltage levels
Because Ov was in series with Q4 , R2 2 and D7 , i t also dissipated power
that was proportional to a square of a charge current. The power was also
a product of the current and VOv . It was not regarded as a loss because it
could be stored, most commonly in water in a form of heat energy for
future use. Therefore, the only main power losses on board were those of
other series components.
The fact that power in DC circuits was a product of current and voltage
product, the losses were expressed as:
P los s = IC g (VQ4 + VR 2 2 + VD7) 4.7
= 46W
when IC g = 10A (maximum charge current)
VQ4 = 3V (forward voltage (Loveday, 1992. p 63))
VR 2 2 = 1V (R2 2 = 0.1)
VD7 = 0.65V (diode forward voltage)
66
P i n = Pou t + P los s 4.8
= (IC gVC g) + P los s
= 326W
Therefore the minimum charging efficiency was
η = Pou t * 100 = 85.89% (without POv) 4.9 P i n
When Pou t was 280W.
+Vref
Dz R1
R2
R3
R4
R5
R6
Q1
Ov
Vg Vin
VDC
VDC trc
Q2
Vo
Vg
Figure 4.9 Over voltage load drive circuit
For higher voltage battery banks, like 48V, the charging voltage could be
maintained at 54V and the line voltage at 42V. In this case, the input
charging power would be 567.24W and the output power would be 540W
at full load. Therefore, the efficiency would be increased to 95.2%.
At maximum exciting current 104.5W was dissipated into the rotor and
that brought the system efficiency down to 53.8% and 76.86% for 24V and
48V battery banks respectively.
67
The hybrid system efficiency included the stator power losses, 3I2R,
which was 100.49W at full load. That brought down the efficiencies to
22.6%, 55.8% and 79.93% for 24V, 42V and 92.55V respectively.
Although the controller efficiency was optimized, the system efficiency
still had to be improved. It could be improved by changing the generator
to a brush-less generator. That would reduce the generator excitation
power losses by 16.27% and improve the controller efficiency by 19.21%
at 24V.
The fact that the generated voltage dropped linearly with increasing load
current, the current could be used to reduce the generated voltage to a
desired level before excitation process. In that way the speed range would
be increased further. That would be achieved by including a controlled
load that would be used to increase and maintain load current at full load
current. The load current would be a sum of the controlled load current
and the charge current. It could be expressed as:
IL = IC L + IC g 4.10
Power dissipation on the controlled load could still be treated in the same
way as that of an over voltage load.
Beside the improvement in the controller efficiency, at 42V line voltage
the speed range could be increased by 66%. Refer to table 4.1.
68
24V Battery bank 48V Battery Bank
Speed in RPM 149rpm – 598rpm 250rpm – 1000rpm
Charging efficiency
without excitation 85.89% 95.2%
Charging efficiency
with excitation 53.8% 76.86%
Table 4.1 24V and 48V charging systems comparison table
The fact that VOv was outside the voltage regulation window, it increased
with increasing excess voltage ∆VOv above the desired input charging
voltage and vice versa. Ideally, IC g was equal IC t ot a l. A difference between
the two currents was very small due to very high shunt resistors like: R5
and R6 .
Therefore, the total charging efficiency could be expressed as
η t o t a l = Pou t * 100 = (IC g (VC g + ∆VOv)) – P los s) * 100 P i n (IC g Vg)
Determine Ov, Ov = VOv max IC g
4.4 Charging voltage regulation
Linear voltage regulator LM723 was used to maintain charging voltage
VC g at 28V irrespective of fluctuations in an unregulated input voltage,
V i n C g . A 12V zener diode was used to fix regulation reference voltage at
11.2V.
69
With reference to figure 4.10, R4 and R5 were used to determine the
reference where by it was expressed as:
V re f = VR 5 = VDz * R5 = 11.2V 4.11 (R5 + R4)
where VDz was 12V
It would have been simpler to just short circuit LM723 pin 5 and pin 6
and thus V re f would ideally be equal to VDz. That would be when the
internal reference voltage, 1.2V is discarded. Therefore, that implied that
then the reference would be 13.2V.
At switch on, VC g increases from zero volts level to the charging voltage
level, 28V. To ensure that 28V level was not exceeded with further
increase in the input voltage, the regulator internal comparator (Loveday,
1992, p. 99) used a potential divider network to monitor VC g to generate a
regulation signal. The regulation signal was positive to allow charging at
the correct voltage level. The signal only changed state to negative when
VC g exceeded 28V.
Figure 4.10 shows how R3 and VR were connected to a regulator. They
were connected such that VVR kept track of VC g . When VC g was equal to
28V, VVR was also equal to V re f . Every time when VVR turned to be either
greater or less than V re f , the comparator would generate the regulation
error signal that was either negative or positive respectively. The signals
determined the state of the bypass darlington-pair transistors. The
negative signal turned off the transistor pair and the positive signal turned
the transistors on.
70
R3 was assumed to be 20k and VR was determine as:
VR = VVR * R3 (VVR = V re f) 4.12 (VC g – VVR)
VR = 13.31k (when VC g = 28V)
= 6.09k (when VC g = 48V)
Refer to LM723 data sheets in appendix D.
4.4.1 Overload protection
Overload protection employed foldback current limiting technique. The
foldback current limiting was found to be a suitable protection for the
application. It prevented excess heat dissipation on the series components
when the output was short-circuited and when there was an overload.
During the short circuit and overloading of a charge circuit, the knee
point current was exceeded. The charging voltage dropped to zero when
the output was short-circuited. At short circuit, Isc was limited to a value
lower than the rated charging current.
1 2 3 4 5 6 7
891011121314LM723
Rsen
R1
R2
R3R4
R5VR
Dz
Vin VCgQ1Q2
Figure 4.10 Battery bank voltage regulator
71
V sen was given as
V sen = IC g R sen – (VC g + IC g R sen). R1 /(R1 + R2) 4.13
Therefore IC g = V sen(R1 + R2 + VC g R1) 4.14 R sen R2
And thus
Ik = 0.6(R1 + R2) + VC g R1 4.15 R sen R2
Where Ik was the knee point current and V sen was 0.6V.
When the output was short circuited charge voltage VC g was zero and
Ik = Isc .
Therefore
Isc = 0.6 (R1 + R2) 4.16 R sen R2
(Loveday, 1992, p. 84)
When VC g = 28V, IC g = Ik = 10A, R sen = 0.1Ω
R2 = 71.5 R1
R1 = 1k and R2 = 71.5k 75k (1/4W series metal film resistors).
The Ik = 9.813A and Isc = 6.08A.
During short circuit, Ik was reduced to 61.96 % of its maximum value.
The power dissipation on the series components also was reduced
proportionally.
72
That was because in DC circuit
P = IV = I2R. 4.17
4.5 Battery protection
The battery bank was protected from overcharging and deep-discharge by
means of a voltage window comparator. The comparator monitored the
bank voltage continuously to ensure that the required voltage level limits
were not exited (Loveday, 1992, p. 140).
Battery bank window comparator
With reference to figure 4.11 an output changed state, from (High) to
(Low), when Vb a t was outside a defined battery bank voltage limits (Vb mx)
and Vb mi n). That was true when S2 was Low. DC load could be
disconnected with a micro-controller via S2. The limits were set within
software hence a similar operation like the one described in the next
paragraphs, could be achieved with a micro-controller. However, with the
micro-controller the load could be disconnected irrespective of whether
Vb a t was within limits or not. That was achieved by setting S2 High within
software. The output (L-disc) stayed Low while S2 was High. DC load
was disconnected from the battery bank when L-disc was Low and was
connected back when L-disc was High.
When Vb a t was greater than Vb mx , (A) was High and (B) was Low. When A
was High diode A was forward biased and diode B was reverse biased.
Diode B was reverse biased because op-amp B was current sinking and
there was no current flow because the diode B state. That way (C) would
be High.
73
Vb mi n < Vb a t > Vb mx C = High L-disc = Low (S2 = Low)
When Vb a t was less than Vb mx and was greater than Vb mi n both A and B
were Low. Therefore both A and B diodes were reverse biased. Then
leakage currents flow through the diodes. During that time, C was Low
and L-disc was High.
Vb mi n < Vb a t < Vb mx C = Low L-disc = High (S2 = Low)
+
+
Vbmx
Vbmin
Vbat L-disc
A
B
S2
C
Vbmin Vbmx
Ouput (L-disc)
Battery bank voltage (V)
OFF OFFON
Figure 4.11 Battery bank voltage monitor
During the time when Vb a t was less than both Vb mi n and Vb mx , A was Low
and B was High and thus B diode was forward biased and A diode was
reverse biased. As C was the sum (A+B=C) of A and B, it was High when
either A or B was High. Therefore, L-disc was Low.
74
Chapter 5
Excitation circuit design
5.1 Overview
This chapter is focused on excitation of the rotor with the main accent on
the exciting current and the induced emfs. The emfs were induced because
of a rotating flux in the generator. Therefore, the excitation power supply
had to overcome back emf before controlled excitation could take place. A
freewheeling diode allowed continuous current flow into the rotor due to
the back emf.
The induced line emf was directly proportional to the generator speed and
the air gap flux density. So, the fact that the flux density was proportional
to an exciting current and the speed the emf could be expressed as
ea = Emax sin ωt
and the rms value of the Emax was
E rms = 4.44 f NΦ p KW
Where f was the frequency in hertz
Φ p was the flux per pole
N number of rotor series turns
KW was the winding factor
(Sen,1997, pp. 215-216)
75
Static operation
This section entails studying the effects of exciting current on the rotor.
Static exciting flux test per-pole was performed. The test results proved
that an increase in exciting current caused a proportional increase in the
exciting flux density in the generator air gap. Conversely, a decrease in
the exciting current caused a proportional decrease in the generator air
gap flux density. See figure 5.1.
Static air gap flux density
0.002.004.006.008.00
10.0012.0014.0016.0018.00
-6 -4 -2 0 2 4 6
Exciting current (A)
Air
gap
flux
dens
ity (m
T)
Pt1 AvPt2 AvPt3 AvPt4 AvPt5 Av
Figure 5.1 Static air gap flux density
The flux density responses indicated in figure 5.1 showed clearly that the
air gap flux density could be varied to either boost or buck the generator
operation, and as a result, vary the line voltages accordingly.
Dynamic operation
In the preceding section, it was proven that the air gap flux density
changed proportionally with the exciting current. The similar flux density
response also existed in the dynamic operation. It was confirmed that a
76
change in exciting current had a direct influence in the generated voltage.
Therefore, the exciting current not only had a proportional influence on
air gap flux density but also had a similar influence on the generated
voltage. See figure 5.2.
Dynamic air gap flux density
0
2
4
6
8
10
12
-6 -5 -4 -3 -2 -1 0 1 2 3 4 5 6
Exciting current (A)
Flux
den
sity
(mT)
149 rpm
264 rpm
374 rpm
598 rpm
750 rpm
Figure 5.2 Dynamic air gap flux density
During dynamic operation, air gap flux density was increased by 69.98%
above permanent air gap flux density. Conversely, the flux density was
reduced by 57.63%. That was the exciting flux density was 42.97% of the
permanent exciting flux density.
77
5.2 DC/DC converter
Exciting current was supplied from a 24V battery bank with a DC/DC
converter. The converter did not only convert DC but also isolated the
exciting current circuit form the battery bank.
DC/DC converter converted the 24V of the battery bank in to a 48 VDC
source that supplied a PWM buck chopper. The chopper controlled the
armature exciting current. The converter was operated at 100 KHz to
reduce its transformer size and filter size. It gave lower ripple voltage on
the output. The basic switch mode power conversion had evolved from
basic PWM converter to soft-switching converter (Bordry, Dupaquier)
because of reduced switching losses.
Switching converters, in particular buck and boost converters were known
to be the worst EMI generators due to their pulsating input current. That
was due to a switching action of their semiconductor switches. However,
the EMI was reduced with an EMI filter and by making use of soft
switching converter (Bordry, etal).
Soft switching advantages were:
• Reduced switching losses.
• The improved reliability due to reduced high voltage and current
stresses.
• A limited frequency spectrum, which was an advantage with respect
to EMI and losses in passive components.
• A reduction in size of the components resulting from higher
switching frequency.
Figure 5.3 was a DC/DC converter schematic diagram. On the input stage,
it showed a 12V driven PWM self-oscillating MOSFET driver IC, IR2153,
with external components.
78
An opto isolator phototransistor ensured that transistor Q sd shutdown
when opto-diode was on. RT and CT were the timing components.
HO
LO
VB
VS
VCC
RT
CT
Com
12 V
PWMGen
Excitationpolaritychanger
Ra
La
Ea
Dz
Dbd
Dfd
Q1
Q2
Q3
Qsd
opt 24V
Inductance (48uH / coil) Figure 5.3 Rotor (armature) excitation schematic diagram
IR2153 shutdown when VC T was less than the threshold voltage, Vcc / 6.
During that time both outputs, HO and LO , were set low within minimal
delay.
IR2153D presented zero switching losses. It eliminated cross conduction
by providing enough dead time during alternate switching of MOSFETs,
Q1 and Q2 . See figure 5.5. During that time both HO and LO outputs were
low. Dead time was fixed inside the IC. In addition to the mentioned
advantage, the dead time helped to maintain zero voltage switching and
soft switching (IR design tips).
The switching frequency was determined as:
fs = 1 5.1 (1.38 CT (RT + 75Ω)
79
CT and RT could be determined form figure 5.4 for the operation
frequency of 100kHz as used by Magnetics to test P type ferrite
transformer cores.
Figure 5.4 IR2153 RT vs frequency curves
T
T
1 >1 >1 >1 >
2 >2 >2 >2 > 1) Ch 1: 5 Volt 2.5 us 2) Ch 2: 5 Volt 2.5 us
dX: 1.17 us X: 5.17 us
Figure 5.5 DC/DC converter drive signals, signal 1 for Q1 signal 2 for Q2
80
T
T
1 >1 >1 >1 >
2 >2 >2 >2 >
1) Ch 1: 20 Volt 2.5 us 2) Ch 2: 20 Volt 2.5 us
Figure 5.6 DC/DC converter transformer primary signals with respect to
battery bank positive terminals.
T1 >1 >1 >1 >
1) Ch 1: 20 Volt 2.5 us Figure 5.7 DC/DC converter transformer secondary signal
The transformer efficiency at full excitation was 80%. To achieve the full
excitation (104.5W) the battery bank delivered 130.62W.
81
The fact that general-purpose diodes have poor response at high
frequencies, like 100kHz, schottky diodes were used to rectify DC/DC
transformer secondary voltage.
T
1 >1 >1 >1 >
1) Ch 1: 10 Volt 500 ns
dX: 800 ns X: -50.0 ns
Figure 5.8 DC/DC signal overshooting and ringing
Output pulse shape
At switch ON the voltage rose rapidly from zero volts, overshoots to a
peak voltage and then settled back to normal pulse height, which was the
pulse voltage. The pulse voltage was equal to a DC power supply voltage.
A rapid pulse voltage rise time resulted in some oscillations and ringing
at the pulse maximum voltage. Those appeared before the voltage settled
back to normal pulse voltage. See figure 5.8. The oscillations became a
serious problem when switching MOSFET pair was not matched. Refer to
DC/DC converter waveforms in appendix C.
The pulse rise time was defined as the time required for the voltage to rise
from 10% to 90% of the peak voltage. Instantaneous change in voltage,
82
dv/dt, was a slope of the voltage rise in volts per microseconds. It was
approximately 80% of the peak voltage divided by the rise time
(ABB,1998, p 11). Therefore, it could be expressed as:
dv = Peak voltage * 0.8 5.2 dt t r
T
1 >1 >1 >1 >
Figure 5.9 Open circuited rotor back emf waveform
The waveform in figure 5.9 was captured across an open circuited rotor
terminals. The waveform showed that the exciting flux also induced emf
into the rotor coils. Likewise the generated voltage, back emf increased
with increasing generator speed. It also increased with increasing load
current. It was therefore evident that a blocking diode and a freewheeling
voltage clamping diode were required to protect the exciting circuit
against back emf. The diodes assured reliability of the circuit. Refer to
appendix C for back emf analyses data.
T1 >1 >1 >1 >
Figure 5.10 Freewheeled back emf waveform
83
With a freewheeling diode included in the exciting current circuit, the
negative half of the back emf was clipped off. The negative overlap was
due to the freewheeling diode forward voltage. See figure 5.10.
TT1 >1 >1 >1 >
Figure 5.11 Zener clamped freewheeled back emf waveform
When a zener diode was connected in parallel with a freewheeling diode,
as indicated in figure 5.3, the waveform in figure 5.11 appeared across the
rotor terminals. The waveform showed clipped positive pulses. The
maximum magnitude of the pulses was determined by the zener diode. It
was high enough to drive 5A into the rotor. The zener diode not only
limited the exciting voltage but also protected both the blocking and the
freewheeling diode from being driven into avalanche by the positive
pulses of the back emf.
DC buck chopper
A DC buck chopper was used to adjust the exciting current to the required
levels in order to adjust air gap flux density of the generator. The chopper
was found to be more energy efficient compared to a DC boost chopper
(Beak et al. , 2001). The project served as one of many applications of the
power electronics choppers. (Ang, 1995, p. 357)
A buck chopper DC input was pulsed to an armature in a manner that it
increased and decreased the exciting flux density in direct proportion to
the armature current. Duty-cycle D determined average voltage across the
84
armature terminals. During the chopper off state, the exciting current
continued to flow into the armature because of the induced back emf. The
emf forward biased the freewheeling diode. Therefore, the diode
conducted back emf current back into the armature. Therefore during that
time the armature current (exciting current) Ia was equal to the diode
current Id .
EaLaRcRloss
Ra
Figure 5.12 rotor circuit
PWM
Ra
La
Ea
Dfd
DbdQ
Vin
Figure 5.13 Exciting current control circuit
The field circuit total resistance, as shown in figure 5.12, was a sum of
the brushes resistance, slip-rings resistance, contacts resistance and the
field coil resistance. The first three resistances caused excitation power
loss. They were defined as R los s , therefore the excitation power loss was
expressed as:
Ploss = Ia2*Rloss = 17W
85
The field coil resistance (RC) determined the actual excitation power. The
excitation power was expressed as:
Pex = Ia2*RC = 87.5W
With reference from figure 5.13, the required input power was determined
by Ra as:
P i n = Ia2*Ra = P los s + Pex = 104.5W
Therefore the field circuit efficiency was
η = Pex* (P i n) -1*100 = 83.73%
It was assumed that DC/DC buck chopper would operates in two modes,
which were continuous current conduction and discontinuous current
conduction. The two modes would come in to perspective respectively
when
Tof f < Td s Td s = (-dia / dt) * Tcg 5.3
and
Tof f > Td s .
where Td s was discharge time and Tcg was charge time
Voltage and current continuous mode
With reference to figure 5.3, when Q3 was switched on, the armature
current would be increased exponentially to Ia max due to the armature
86
inductance. Similarly, when Q3 was switched off, the current would
decayed exponentially to Ia mi n . Ia could not decay to 0A because T s was
much smaller than the armature time constant La/Ra . The turn on and turn
off, Ton and Tof f , respectively would be given by:
Ton = DT s 5.4
and
Tof f = (1 – D)T s 5.5
where
T s = 1/fs , ω = 2πf 5.6
During Ton , Db d would be forward biased and D fd reverse biased therefore
Ia would flow into the armature and VLa would developed because of La
dia/dt. During Tof f similar characteristics would exist because of decaying
Ia . During the decay time, VLa polarity would change and thus had D fd
forward biased. When D fd was forward biased, IDfd would be equal to Ia .
VLa = Ea where Ea is an armature emf
Figure 5.14 is a graphical representation of the preceding description of
how an armature current would respond in continuous mode.
87
TON
TS
TOFF
Imax
Imin
Ia
iin
iD
Vi - Vo
Vo
VL (avg)
Continuous mode
TON = DTSTOFF = (1 - D)TS
Vo
Figure 5.14 Continuous mode waveforms
The average voltage across the generator armature terminals would be
Va = V s DT s = DV s (V) 5.7 T s
During interval 0 ≤ t ≤ Ton
V s = Ra ia + L (dia/dt) + Ea 5.8
An average current through a generator armature coil would be
ia = (V s – Ea) (1 – e – t / τa) + Ia mi ne – t / τa 5.9 Ra
88
τa = La / Ra
During interval Tof f ≤ t ≤ T s
Va = 0 = Raia + La (dia/dt ′) + Ea 5.10
where
t ′ = t – Ton = t - DT s 5.11
Ia = -Ea/Ra (1 – e – t ′ / τa) + Ia max e - t ′ / τa 5.12
In steady state, voltage across La would be zero because VLa = Ldia/dt and
di/dt = 0v
Therefore
Va = DV s = RaIa + Ea 5.13
Ea = K ′E φƒ ω = DV s – IaRa 5.14
Ia = (DV s - Ea) / Ra 5.15
(Rahman, 2002. pp. 9-10)
89
TON
TS
TOFF
Imax
Ia
ia
iD
Vi - Vo
Vo
VL (avg)
Discontinuous mode
tr
TON = DTSTOFF = (1 - D)TS
t
Ea
Vo
Figure 5.15 Discontinuous mode waveforms
Discontinuous mode would take place when armature current becomes
zero during switching period T s . Then Q3 would be off and a freewheeling
diode would be on. Va would then be equal to back emf across the
armature terminals. Refer to figure 5.15.
An armature current would drop to almost zero amperes at time tϒ
(corresponding to the conduction angle ϒ). That would be due to the back
emf induced into the armature by the rotating flux in the generator. The
armature voltage would therefore be:
Va = V s for 0 ≤ t ≤ T s
≈ 0V for DT s ≤ t ≤ tϒ
= Ea for tϒ ≤ t ≤ T s
90
The average armature voltage would be
Va = DV s + (1- (tϒ / T s)) 5.16
During 0 ≤ t ≤ DT s
With discontinuous mode armature current would start form zero because
Ia mi n would be zero. Therefore
ia = V s – Ea (1- e – t / τa) 5.17 Ra
During freewheeling Ia would be
ia = -Ea (1 – e - t ′ / τa) + Ia max e - t ′ / τa 5.18 Ra
And at tϒ ia i t would be approximately equal to zero.
An armature inductance La and resistance Ra determine the conduction
time constant τa for a period during which ia flowed through the armature.
τa = La/Ra
The time constant formula shows that the higher the inductance the lesser
the chances of discontinuous armature current (Rahman, 2002, p. 10). The
inductance also provided current smoothing. It limited Ia peak-to-peak
ripple value to lower losses that lead to the generator derating (Beak,
Buddingh, 2001, p.1162). Therefore the minimum inductance required to
ensure continuous mode when Ton < Tof f was determined by
La mi n = Tof f * Ra/2 5.19
91
because the chances of discontinuous mode would be high when
Ton ≈ τ = 250us then La minimum would be:
La mi n = 522.5uH
Cosequently, it was obvious that the generator would only operate in the
continuous mode. The armature inductance was much higher than the
minimum required inductance. It was 40.9mH.
Armature power losses were expressed as:
P = Ia2 Ra + Ea Ia 5.20
EaIa represents power that includes mechanical power, friction and
windage (Rahman, 2002, p. 11)
T
TT1 >1 >1 >1 >
2 >2 >2 >2 >
1) Ch 1: 100 mVolt 250 us 2) Ch 2: 5 Volt 250 us
Figure 5.16 Continuous mode waveform at 30% duty cycle
Figure 5.16 showed the armature current ripple waveform, top waveform,
and the transistor voltage waveform, bottom waveform. The current
waveform indicated that although the duty cycle was 31.39% at 1.23A, the
armature response was still that of continuous conduction. This was
92
because the armature inductive reactance, 2πfLa was much greater than
the armature resistance. TS was also much smaller than the time constant
τa .
5.2.1 Transformer Selection
The choice of a DC/DC converter transformer was such that the losses
would be kept as low as possible. It was done to ensure high efficiency
and reliability of the converter. Although the transformer efficiency was
assumed to be 90%, (Magnetics, p 4.3), it was limited by the transformer
temperature rise and the core saturation.
Ferrite core transformer offered low core losses and high saturation flux
density when it was operated below ±2000 gauss (Magnetics, p 4.4). This
ability made the ferrite transformer suitable for high power and high
temperature operations. From core material point of view, materials with
negative temperature coefficient were more suitable because their
maximum saturation flux density decreased with temperature rise. For
example, P material core losses decreased with increasing temperature up
to 70°c and an R material up to 100°c (Magnetics. p 4.2).
Although core losses increased at frequencies above 20kHz, they could be
reduced by operating with the core flux level lower than ±2000g. Refer to
flux density vs frequency curve in appendix D for reduced gauss as per
frequency level.
The ETD59 transformer core was used for DC/DC converter as it was
decided. That was because the core and wound bobbin were readily
available. The physical dimensions and magnetic data tables in appendix
D were used to determine the core WaAc .
93
Although the transformer was readily available, its windings had to be
tested to see if they meet minimum DC/DC transformer test requirements.
The core typical power handling capacity was 2500watts and operation
flux density was 900 gauss at 100kHz. Refer to ferrite core selection table
in appendix D. The primary turns Np and secondary turns N s were
determined as follows:
Np = Vp * 108 = 6.7 turns 5.21 4BAwpf
Where Vp was 24V and V s was 20.2V at 5A Is
N s = V s * Np = 5.6 turns 5.22 Vp
Ip = Po = 4.838A 5.23 eVp
Where Po was the excitation power, 104.5W, and the efficiency η (e) is
0.9. N s is multiplied 1.1 to allow for losses (Magnetics. p 4.9).
Because of transformer windings test, it was found that the transformer
could still be used except that its ratio was 1:2. Therefore, the ratio
limited the PWM signal to 50% duty cycle. This was because the
maximum required Va was 20.2V to obtain ±5A armature excitation. Refer
to open circuit test at 24V line voltage table in appendix B.
Due to the transformer ratio test V s would ideally be 48V and Ip would be
10A.
94
5.3 Exciting current control
The exciting current control was based on PWM technique whose duty
cycle was proportional to dv/dt of the generated DC voltage. Figure 5.17,
was a block diagram of the exciting current control circuit. The PWM was
generated because of comparison between the ramp signal and the
generated voltage difference, dvDC to the pre-set voltage reference. The
difference was proportional to the generated voltage and the wind speed.
Therefore, the duty cycle would vary with a change in the voltage
difference.
The PWM ramp signal was generated with a 555 timer and the signal was
then fed to a non-inverting amplifier. The amplifier magnified the ramp
signal as shown in figure 5.17 . In that way, the ramp could be adjusted to
suit the PWM requirements of the current control.
With generated voltage difference, an instrumentation amplifier was used
as it presented easy gain adjust with reduced CMRR. Either positive or
negative voltage difference could be obtained from the amplifier. The
positive difference was fed to a non-inverting amplifier whilst the
negative was fed to the inverting amplifier. In that way, a comparator
only received positive voltage difference that compared to the ramp signal
easily to generate pulses.
RampGenerator
Noninvertingamplifier
Inst.Dif.
Amp.
Vg >
Vg <
Non inv.amp
Inv.amp
Signalblocking
diode
PWMpresetlimit
Excitationbuck and
boostchopper
+
SignalComparator
Vg+
+
+
Vref
Figure 5.17 PWM generator block diagram
95
Figure 5.18 shows a graphical representation of voltage difference
inversion as compared to a ramp signal. According to the foregoing
description, a negative voltage difference was inverted whilst dv/dt was
left negative. Therefore, it meant that the PWM translating circuit
operated in the first and second quadrant.
+ dVdc
- dVdc
Non inverting
Inverting
dv/dt-dv/dt
DC 3
-DC 3
100%100%
0%
Figure 5.18 Characteristic behavior of the PWM generator
Figure 5.19 showed how varying duty cycle is achieved at different
voltage levels. DC1 level sets the duty cycle D to 25% of PWM period.
That is only 25% of the input power is deliver to the output. DC2 level set
D to 50% and DC3 level set D to 75%.
The duty cycle also had a pre-set limit. The limit was dependent on the
requirements of the application. The PWM duty cycle limiter was
positioned just before the chopper to make limit setting easy and simple.
However, it would still be possible to place it before the comparator,
although the circuit would be complex.
96
25%50% 75%
0% 100%
DC 1DC 2DC 3
Ramp signal
PWM signal
+dVdc
Figure 5.19 PWM duty cycle levels at different DC levels.
CTL5
GN
D1
OUT 3
RST4
THR6
TRIG2 DISC 7
U1
R1
R2
Vin1
GN
D2
Vout 3RG1
Vin1 GN
D2
Vout 3
RG2
C1
C2 C3
C4
+B
-B -10
GND
VCC
-B
GND
R3
R4
R5
VR1
Q1
C5
C6
VCC
GND
GND
VCC
VR2
R6
3
21
411
U2A
5
67
U2B
10
98
U2C
12
1314
U2D
VCC
-10R7
R8
U5
Q2
GNDR9
VR3
VCC
GND
R22
VR4
VR5
R10
R11
R12
R13 R14
R15
R16
R17
R18
R19
R20
R21 3
21
411
U3A
5
67
U3B
10
98
U3C
12
1314
U4D
3
21
411
U4A
5
67
U4B
GND
GNDGND
-10
VCC
VCC
-10
LK
LK
-B
+15V
VgP
+15V
+PWM
-PWM
R8a
D1
D2
1
2
J1
DZ1
Figure 5.20 PWM generator schematic diagram
The PWM generator was divided into two sections: ramp generator and
DC level generator. The ramp signal and the DC level signal were
compared to generate the PWM signal.
The ramp generator was build around the LM555 timer. LM555 timer is a
highly stable device for generating accurate ramp oscillations. It was for
97
this reason that the LM555 timer was selected for generating a linear ramp
signal. The ramp signal was amplified with a non-inverting amplifier and
was then fed to a voltage follower. After the voltage follower, the signal
was compared with a DC level. The comparator output went high every
time when DC level was greater than the ramp signal. The ramp period T
was determined as
T = 2/3 Vcc VR1 (R3 + R4)*C5 = 0.5mS 5.24 R3 Vcc – VB E (R3 + R4)
Therefore its frequency was
F = T -1 = 2KHz 5.25
The DC level was generated at a voltage difference between the generated
voltage ratio and a preset voltage reference. The difference was amplified
with an instrumentation amplifier. It was also amplified with an inverting
and non-inverting amplifier afterwards. That depended on whether the
difference was negative or positive. The inverting amplifier inverted the
negative difference to positive difference as shown in figure 5.18.
When the ramp signal level was greater than DC signal level, the
comparator U2C generates negative error signal that reverse biased U5
LED. Therefore coupling light flux was zero and U5 transistor stayed off.
Because the transistor was off the output pulse level was low. Conversely,
a positive error signal was generated and then an LED was forward
biased, hence the transistor was turned on and the output pulse level was
high. Because of the comparison, square pulses were generated. See figure
5.21. The DC level determines the pulse on time. Therefore, the pulse on
time was proportional to the DC level.
98
A final stage of the PWM generator was the PWM limiter. The limiter
compared the ramp signal level with a preset voltage level. The level was
set such that it was less than Vcc. At every instant of time when a ramp
signal exceeded the reference comparator U2D generates a negative error
signal that turned off transistor Q2 and thus U5 LED stayed off
irrespective of U2C output. Jumper J1 could be used to overwrite Q2
effects on the LED.
Figure 5.21 PWM generator signal waveforms
5.3.1 Exciting current polarity changer
The polarity of an exciting current was changed with a double-pole-
change over relay. The relay home position was in reversed excitation
mode. That was indicated in figure 5.22. When a relay coil was energized,
the relay poles change their positions from a1 to a2 and b1 to b2 . When the
99
poles positions changed the exciting current polarity changed from
forward mode to reverse mode.
DC chopper
Armature(rotor)
Relay coil
a1
b1
a2
b2
+
Figure 5.22 Exciting current polarity changer
Positioning of a freewheeling diode and the clamping diode was such that
the change in polarity did not affect their purpose in the exciting current
circuit. For instance, when the diodes were placed after the relay, they
would short circuit the exciting current when its polarity changed from
reverse to forward mode. Therefore, the diodes were placed before the
relay. See figure 5.3.
100
Chapter 6
Control Circuit design
6.1 Overview
The control circuit is a very integral part of the controller hardware. This
is because it continuously monitors an operation of each section of the
controller hardware. It reads analog and digital signals generated from
other hardware parts and compares them, within software, to the preset
control references. In addition, there after generates control command
signals. So, read and write control commands as well as interface circuit
characteristics are discussed in this chapter.
6.2 I/O circuits
Input and output circuits brought about communication between a micro-
controller and the rest of the controller hardware. Their main function was
to translate analog and digital signals between the micro-controller and
other circuits. The function of these circuits was to ensure that read signal
magnitudes were kept within safe operation limits of the micro-controller.
The control circuits, for both analog and digital, were powered from on-
board power supply. 5V, 12V and 15V power supply circuit operations
are explained in the following paragraphs.
With reference to figure 6.1, V i was the battery bank voltage, which was
24V. The three power supplies were chain connected, beginning with the
101
15V power supply. They were the same in the way they operated except
that their input and output voltage levels were different.
Three terminal voltage regulators were used to supply 5V, 12V and 15V.
Their current handling abilities were increased with a high current bypass
transistor, Q2 . Q2 protected the regulator from back emf when V i was less
than VO . That was due to the fact that Q2 base-emitter junction presented
feedback path. R2 was connected in series with the voltage regulator such
that VR 2 was proportional to the regulator input current Ireg . Therefore
VR 2 = IregR2 6.1
The value of R2 was determined to ensure that Q2 turns on at any stage
when Ireg tended to exceed a predetermined Ireg . Q2 turns on when
VR 2 ≥ Q2 Vb e = 0.58V (BD436 data sheets page 2)
and output current IO was a sum of Q2 current IQ2 and Ireg . Because Q2 was
outside regulation protection loop, it required its own current limiting, Q1
and R1 . Where R1 is designed to turn Q1 on when
VR 1 ≥ Q1 Vb e = 0.65V (2N3906 data sheets page 2)
This occurred when IQ2 was equal and greater than a predetermined
current value, and for the fact that VR 1 was proportional to IQ2 .
VR 1 = IQ2R1
When Q1 was on, VR 2 dropped to about Q1 Vce sa t , which was 0.2V (2N3906
data sheets page 2). That was because Q1 short circuited R1 . At that
moment, Q2 turned to switch off and thus VR 1 dropped to a value less than
102
Q1 Vb e, hence Q1 switched off and VR 2 was equal to Q2 Vb e though it was
dependent on Ireg . This process repeated itself while IQ2 turned to exceed
its preset value. Therefore, in this way IQ2 was kept within limit.
R2 value was kept the same in all three voltages regulation but R1 value
was determined by the required output current when Ireg was maintained.
RegVo
Vi
0V
R1
R2 C1 C2 C3 C4
Q1
Q2
Figure 6.1 Three terminal voltage regulator with by pass transistor
One other important aspect in I/O circuit was ADC input voltage level
protection. It was achieved by using two diodes as it was indicated in
Figure 6.2. When Van a log i n was greater than Vcc, D1 became forward
biased and thus pulling VADC i n to Vcc. Therefore a micro-controller input
pin was protected from over-voltage. When Van a log i n was below 0V, D2
became forward biased and thus pulled VADC i n to 0V. Because of ADC in
pin protection, Van a log i n could only fluctuate between Vcc and 0V.
R1
C1
D1
D2
Analog in ADC in
Vcc
0V
Figure 6.2 ADC voltage protection
103
Active low pass filter
Because generated voltage was not free of noise, a micro-controller as one
the noise sensitive components was protected against the noise by
introducing active low pass filter circuits in the signal translating circuits.
The filter was a high performance filter. It was built with an op-amp as
active element. It needed few external resistors and capacitors. The filter
circuit was compact, because it did not need inductors, and therefore it
cost less than a passive filter. See figure 6.3.
R1 R2
R3
R4
C1
C2
Vo
Com
Vi
Figure 6.3 Active low pass filter
An op-amp in active filter provided gain so that there was no insertion
loss over the pass band. It also provided good isolation because it has
high input impedance and low output impedance.
A type of filter shown in figure 6.3 was a Sallen-Key filter, and it
consisted of two RC networks in its input circuit and the feedback loop.
Its roll-off was twice as much as of a single pole whose roll-off was
-20db/decade, so the filter roll-off was –40db/decade. R1 and R2 , C1 and
C2 are matched respectively, so the filter cut off frequency was
determined by
FC = __1__ = 60Hz 6.2 (2πRC)
104
A disadvantage was the frequency limitations of an op-amp. General-
purpose op-amps can only be used to filter audible frequencies. In this
application, the disadvantage could be ignored as the cutoff frequency
was within audible limits. See figure 6.4. The figure showed a response
curve that was very similar to that of Butterworth filter. R3 and R4
determined the filter damping factor that was given by
DF = R3 / R4 +1= 1.556 6.3
Low Pass Filter VF Curves
0
0.5
1
1.5
2
2.5
3
3.5
0 30 60 90 120 150 180 210Frequency (Hz)
Vol
tage
(V)
Voltage
Figure 6.4 60Hz active low pass filter characteristics.
Generator speed sensor
A generator speed sensing could have been simpler if the generated
voltage ripple was not dependent on the charging current (load current)
0.707Vmx
105
and the exciting current. An opto-electronic sensor was used to determine
the generator speed from the generator line voltage.
Figure 6.8 was the generator speed sensor schematic diagram. By
monitoring line voltage (V-line), a pulse was generated every time a
revolution was completed. Thus, every time a negative half cycle of the
voltage started a pulse was generated.
Terminals, +RL and +BL, were connected across red line and blue line
such that a fraction of V-line appeared across R2 6 . VR 2 8 appeared to be a
half rectified waveform that was fed through into the circuit by the
differential amplifier. The input pins of a differential amplifier were
protected against high voltage with a zener diode Dz2 . The waveform was
compared with a preset reference voltage that was set within hardware
with VR7. At every instant when the waveform magnitude exceeded the
reference voltage, the comparator generated a pulse as indicated in figure
6.9. The pulses were then fed optically through to a digital counter that
converted them into digital data that could be understood by a
microcontroller.
The counter was synchronized with a micro-controller within software so
that number of pulses read in one-second equals number of hertz. This
was achieved by sending a low pulse to the counter pin 1. The pin is an
active low pin and it cleared the counter outputs every time it received a
low pulse. Then the counter started counting the speed pulses, and after
one second, the pulse count was read with the micro-controller. The
process was repeated every time before reading the speed count.
An octal high voltage high current darlington transistor arrays, ULN2803
was used to interface a counter to a micro-controller.
106
The arrays had freewheeling clamp diodes for transient suppression. This
feature presented an added advantage of using ULN2803. Refer to
ULN2803 data sheet in appendix D.
V-line sensor Low pass filter and the pulse generator
1 2 3 4 5 6
R25
R26
R27
R28
R29
R30
R31
R32 R33
VR6
3
21
411
U6A
5
67
U6B
10
98
U6C
12
1314
U6D
C10
C9
R34
R35
R36
R37R38
VR7
COM
COM +4.5 COM
+4.5
-4.5
+BL
+RL
U7
+5V
-B
PLC
R23
R24
C7
C8
+4.5
-4.5
COM
D3
DZ2
DZ3
Figure 6.8 Speed sensor schematic diagram
T
T
1 >1 >1 >1 >
2 >2 >2 >2 >
1) Ch 1: 2 Volt 10 ms 2) Ch 2: 2 Volt 10 ms
Figure 6.9 Speed pulses waveforms
107
Generator line current sensor
Although hall-effect current sensors had no power losses and do not need
secondary power source, they were not easy to source as well as linear
opto couplers for optical voltage coupling.
As an alternative to the hall-effect current sensors, current sensing
resistors were used. The sensing resistors were connected in series in the
generator lines. According to Ohm’s law
V = I*R
The voltage developed across the sensing resistor was proportional to the
current flowing in the line. This voltage was used to determine the line
current.
Figure 6.11 described how a line current was detected. As mentioned in
the preceding paragraph the line current was measured in terms of
voltage, terminals +BL and –BL are connected across the sensing resistor.
The voltage was picked up with a low cost, low power, true RMS-to-DC
converter AD736 configured in a differential input mode. Refer to figure
6.11 for the configuration characteristics.
108
Line current sensor characteristic curve
01020304050607080
0 1 2 3 4 5 6 7 8 9 10
Line current (A)
Vol
tage
(mV
)
sen-voltage (mV)
Figure 6.10 Line current sensor characteristics curve
The output of the AD736 configuration was fed to a non-inverting
amplifier with a voltage follower to drive an opto-coupler.
When AD736 output voltage increased, the opto-coupler diode current IF
increased proportionally with the increasing output voltage. See figure
6.10. Consequently, the coupler collector current increased and thus set
up voltage drop across resistor VVR 1 6 . When VVR 1 6 increased the transistor
VC decreased. See figure 6.12. VVR 1 6 was then amplified with unity gain
differential amplifier. The amplifier resistor ratio conformed to the
optimization requirement of the amplifier as stated in section 3.4.3.
The amplified potential difference across VR16 also conformed to
potential divider rule and therefore could be expressed as:
VVR 1 6 = (V s – Vc) * VR16 6.4 (VR16 + R6 8)
109
Line current sensor L-pass filter Non inverting amp Unity dif. Amp.
3
26
15
74 U12
C13
C16
COM+4.5
+4.5
-4.5
-4.5
1234 5
678U13
+BL
-BL
10
98
U16C
R61R62
R63
R64
R65
R66
VR14 VR15
3
21
411
U19A5
67
U19B
10
98
U19C
C21
C22
+4.5
-4.5COMCOMU20
COM
R67
R68 R69
R70
VR16+15V
-B
IB
R71 -B
Figure 6.11 Line current sensor schematic diagram
Opto IV curves
0
2
4
6
8
10
12
14
16
0 5 10 15 20
I fwd (mA)
V (V
)
Vc I fwd Ic
Figure 6.12 4N25 current and voltage characteristic curves
Owing to the fact that there were some problems with sourcing the linear
opto-coupler LOC110, an opto-isolator 4N25 was tested for linearity. The
test was done to determine whether 4N25 could be used in place of
LOC110. As indicated in figure 6.12, 4N25 presented a linear response
from 2mA to about 10mA IF. When IF was increased further above 10mA,
the increase in IF did no longer generate a proportional increase in VVR 1 6 .
110
The graph indicated that 4N25 LED saturation started at 10mA. Therefore
when it was used in a signal translating circuit it could not be driven into
saturation because the linear response was critical.
Because hall-effect current sensors could not be sourced, it was decided
that the one at hand be used to determine the rotor current. Figure 6.13
showed that an increase in rotor current caused a proportional increase in
the hall sensor output voltage. As a result, a linear response was obtained.
The response obviated a need for complex mathematics within software.
According to y = mx + c the crucial parameter was m, dy/dx.
This parameter defined a relationship between a measured current and the
sensor output. A third parameter, c, was eliminated within hardware with
an op-amp offset adjust, or by setting analog inputs reference so that it
was equal to the analog signal offset. This way, response curve started
from the graph’s point of origin. Therefore y = mx.
BB-100 current test
00.10.20.30.40.50.60.70.80.9
1
0 1 2 3 4 5 6
Load current (A)
V-o
ut (V
)
Figure 6.13 Hall-effect current sensor characteristic curves
With reference to figure 6.14 generated voltage was read as a fraction of
the actual generated voltage. The read value was then amplified with an
111
instrumentation amplifier to voltage level that represented the generated
voltage accurately. The voltage could therefore be expressed as:
Vg = +Vg *dPT1/(R1+PT1+R2) 6.5
R3
+Vg
- VB
PT1
R1
R2
+
-
U1Vo
Vg
Figure 6.14 Voltage detector schematic diagram
6.3 Switching circuit
Interfacing circuit linked one circuit to the other. For instance, it l inked
the generator stator and rotor circuits to the micro-controller. The
interfacing circuit’s main task was to translate analog signals and digital
signals between the micro-controller and controlled parts of the circuit.
An interface circuit performed a number of tasks in the controller
circuitry, and those were:
Buffering data temporarily
Isolating micro-controller from the noisy parts of the controller
circuitry to prevent damage to the micro-controller.
Multiplexing input data
Converting analog inputs to digital inputs
Timing adjustment to ensure synchronized data exchange.
112
Dz R1
R2
R3
R4
Q1
S2
Vbat limits detector
Q2
Figure 6.15 Load disconnect switch
Figure 6.15 showed two switching resistor networks, R1 and R2 for
switching Q1, and R3 and R4 for switching Q2 . When Q2 was on, R1 set up
Q1 switching voltage and when that voltage was greater than Q1 threshold
voltage, Q1 started conducting. The voltage was clamped within limits
with a zener diode DZ. When DZ clamped VR 1 at 12V, R2 served as a DZ
current limiting resistor. To ensure that Q2 turned on and off when an OR
gate output was High and Low respectively, a second potential divider
resistor network, R3 and R4 , was used. The network was such that VR 4 was
equal to Q2 VB E. It was approximately equal 0.85V when the OR gate
output was High, 5V, and Q2 VC E was nearly zero volts. See figure 6.16.
On the other hand, when the gate output was Low, VR 4 turned Q2 off. That
was because the VR 4 was less than the required VB E to keep Q2 on. At that
moment, Q2 VC E was approximately equal to the supply voltage. See figure
6.15.
113
2N3904 Switching voltage test
0.01
0.1
1
10
100
0.01 0.11 0.21 0.31 0.41 0.51 0.61 0.71 0.81 0.91
V BE (V)
V C
E (V
)
Figure 6.16 2N3904 VB E vs VC E .
Isolation techniques
A micro-controller was optically and magnetically linked to the generator
stator and rotor circuit. That was done to ensure safe and reliable
operation of the micro-controller. The two common isolation techniques
were those where relays and opto-isolator were utilized. Relays and opto-
isolators present good isolation between low voltage circuits and high
voltage circuits. In addition, sensitive control circuits were protected
from inadvertent voltage and current spikes generated in the high voltage
circuits.
An electromagnetic switch, relay, was still preferred for isolation between
the generator circuits and the micro-controller. Because relay coils
generated back emf when they were switched off, a freewheeling diode
was used to protect the relay driver. The relay also had frequency
limitations that determined its life expectancy. According to (Loveday,
1992, p 218) electromagnetic switches were suitable for frequencies
below 20Hz.
114
Dv-B
oard
Vcc +V
R1
U1
0V
Load
D1
Q1
Figure 6.17 Relay drive circuit
6.4 Micro-controller
Micro-controllers have ability to store and run unique programs in a
similar manner to that of the computer. For instance, they could be
programmed to perform functions based on the predetermined situation.
Therefore, they defined the intelligence of the control systems.
The programmable nature of the micro-controller enabled it to perform
complex tasks. To get it to change tasks one had to arrange for a change
in software. The micro-controller continuously reads inputs from the
sensors, computes errors and then outputs control commands to drive
external circuitries, one of which was the DC/DC converter.
Micro controller’s ability to control external devices made it very useful
where an automatic control of electronic circuit was required. The
controller eliminated the need for analog to digital converters ADC and
digital to analog converters DAC as it had them built in (Carr, 1991,
p420). One such controller was PIC16F877. It was decided that a
PIC16F877 development system board be used to test software command
functions.
115
6.5 Operation control modes
Operation control modes were selected by setting digital output port bits
either “High” or “Low” depending on an error generated during execution
of the software command functions. The next paragraph describes the
output port bit status.
Digital output control port
MBS LSB S8 S7 S6 S5 S4 S3 S2 S1 Converter DC load disconnect Iex current polarity changer Over voltage load Speed counter enable Analog input select bits
Bit status definition
Bit State Definition
S1: 1 Turns OFF DC / DC converter
0 Turns ON DC /DC converter
S2: 0 Disconnects DC loads from the battery bank
1 Connects DC loads to the battery bank
S3: 0 Selects positive exciting current polarity
1 Selects negative exciting current polarity
S4: 0 Disconnects fault load from the generator and connects
charge control circuit.
116
1 Connects fault load to the generator and disconnects fault
load
S5: 0 Enables generator speed counter
1 Disables generator speed counter
Bit S6, S7 and S8 were used to select an analog quantity to be read when
a single mode operation was selected. The bits status was changed by
writing a hexadecimal number to the LED port, therefore the required
control state was set, for example, passing 2 to the LED port as
Write_LEDS(0X02);
resets the speed counter for a start of a new count. It was mentioned that a
digital counter was used to convert the speed pulses in to digital data that
the microcontroller could understand. The number of pulses read in one-
second represented the generator speed in hertz. The following expression
showes how the speed count was read:
Sp = Read_Switches();
where Sp was the speed variable and Read_Switches() was a function that
read the digital port state. This was because the digital counter was
interfaced to the microcontroller via the 8-bit switches-port.
Select variable bits are described below.
S8 S7 S6 Q Schematic-symbol Selected analog 0 0 0 Vg A DC generated voltage 0 0 1 Irt B Rotor Current 0 1 0 Ireg C Regulator current (charge current) 0 1 1 Vbat D Battery bank voltage 1 0 0 Vrt E Rotor voltage 1 0 1 Ir F Red line current
117
1 1 0 Iy G Yellow line current 1 1 1 Ib H Blue line current
6.5.1 Analog and digital input signals execution
Analog input signals were multiplexed and read in sequence with channel
zero. An average of 50 samples of each analog input signal was taken to
assume an accurate reading. The following lines describe how the analog
input was read:
read_analog() byte i=0; set_adc_channel(0); //Select channel 0 adc_value = 0; //Zero variable for(i=0;i<50;i++) //Setup 50 times loop for 50 ADC samples adc_value = adc_value + Read_ADC_10(); //Add all readings //to variable delay_us(12); //time delay between samples taken //See table 11.1 note 1 in //datasheet of the PIC16F877 p117 // (PIC16F877 Development System) adc_value = adc_value/50; // Assigns adc_average value to // adc_value. Analog signals were continuously read during each mode of operation
although the address bit status was unique for each mode. The bit status
was such that the mode operation conditions were left unchanged when
both analog and digital inputs were read. The LED port bits were bit
masked to make inputs reading simple without changing mode
requirements. See Bit-masking tables in appendix A.
The generator speed was read as a digital input signal to switches port of
the development board. In the same way, the speed was read during all the
118
modes without changing the modes operation conditions. The following
lines show how the speed was read after the speed counter enable pulse
was sent.
Write_LEDS(0x02); // enables generator speed counter Write_LEDS(0x00); delay_ms(1000); // one second delay Sp=Read_Switches(); // Reads speed in hertz Write_LEDS(0x11); // selects Vg & disconnects DC load
6.5.2 Operation mode selection
The operation modes were selected because of continuous monitoring of
generated voltage Vg and generation of appropriate error signals to suit
the selected mode. Control bits in a lower nibble of the LED port were
used to retain the state of the selected mode during execution. The higher
nibble was for inputs selection.
Operation conditions as per mode:
Permanent magnet mode
DC/DC converter was kept off
Exciting current polarity changer was kept off
The balanced star fault load was kept disconnected
Disconnect voltage sensitive DC loads from the battery bank when
its voltage was out of range, 24V – 28V.
Boost mode
DC/DC converter was kept on
Exciting current polarity changer was kept on
The balanced star fault load was kept disconnected
119
Disconnect voltage sensitive DC loads from the battery bank when
its voltage was out of range, 24V – 28V.
Buck mode
DC/DC converter was kept on
Exciting current polarity changer was kept off
The balanced star fault load was kept disconnected
Disconnect voltage sensitive DC loads from the battery bank when
its voltage was out of range, 24V – 28V.
Fault mode
The fault mode was mainly focused on the system protection: line
currents, over speed, exciting current and charge current protection. It
ensured that the line currents were balanced as an imbalance caused
uneven loading of the generator. In this mode, permanent mode settings
are adopted as they presented reduced power losses as opposed to other
modes.
The following lines of code evaluate line currents balance by first
working out what each current should be, and creating a ±5% tolerance
window. That was because the possible line currents imbalance error was
found to be 3.3%. See performance data results in appendix C.
Il=(Ireg*(1000/955))*(707/1000); // Determine line current Ilmx=Il+(Il*(5/100)); // I-line max = I-line + 5% Ilmn=Il-(Il*(5/100)); // I-line min = I-line - 5% sel=0; // initialize current select variable This part of the code evaluated the line currents and determined whether
they were within the ±5% tolerance range. If one line current was not
within range, the line current variable will be set High or otherwise Low
as indicated in the following case definitions.
120
for(sel=0;sel<3;sel++) switch(sel) case 0: if(!((Ilmn<Ir)&&(Ir<Ilmx))) // Is Ir not within tolerance range? Red=1; // If yes note fault in red line else // like wise with other lines REd=0;break; // break indicates end of case case 1: if(!((Ilmn<Iy)&&(Iy<Ilmx))) Yellow=1; else Yellow=0;break; case 2: if(!((Ilmn<Ib)&&(Ib<Ilmx))) Blue=1; else Blue=0;break; At the end of line currents inspection, the line current’s variables, Red,
Yellow and Blue states were checked whether they were equal. If they
were not equal, the balanced star fault load would be connected to the
generator with the charge circuit disconnected. Then the new line current
average would be determined with the read line current values as shown in
the following lines of code. In the same way as in the preceding paragraph
the ±5% tolerance window was created and the currents were inspected for
the second time. Then the operation would be locked in the fault mode if
the currents were not balanced.
while(!((Red==Yellow)&&(Yellow==Blue))) // Wait untill lines are not balanced // Generator is connected to balanced star sel=0; // network so all the line currents are expected Fault_ana(); // to have close relationship, 5% tolerance. Ilav=(Ir+Iy+Ib)/3; // Determine average line current Ilavmx=Ilav(1+(5/100)); // Determine max average line current Ilavmn=Ilav(1-(5/100)); // Determine min average line current sel=0; The next lines of code continuously monitored the line currents for an
imbalance.
121
for(sel=0;sel<3;sel++) switch(sel) case 0: if(!((Ilavmn<Ir)&&(Ir<Ilavmx))) // if Ir is not within range //and if yes,note the fault. Like wise Red=1; else // with other lines. Red=0;break; case 1: if(!((Ilavmn<Iy)&&(Iy<Ilavmx))) Yellow=1; else Yellow=0;break; case 2: if(!((Ilavmn<Ib)&&(Ib<Ilavmx)) Blue=1; else Blue=0;break; sel=0; // resets select variable
122
Chapter 7
Conclusion
7.1 Conclusion
The controlled hybrid field generator test results as well as the control
test boards results have proved that introducing a controller for the
generator will optimize the generator performance. The literature survey
also indicated that utilizing low power and temperature compensated
components exhibits reduced on board power losses, and such components
ensure reduced drift in signal translating circuit response hence an
improved linearity and performance.
With an over voltage load included in the input stage of the charging
circuit a wider rage of speed, 149rpm up to 815rpm (10Hz to 54Hz), is
covered whilst still maintaining charging voltage at 28V.
Because an over voltage load was not within the regulation window, its
voltage was not regulated. It reduced stress on charging voltage regulation
components. This was because at speeds above 598rpm, the input charging
voltage started to increase to levels higher than 32.41VDC , resulting in an
excess voltage being diverted to the load. A limiting factor in this case
was the voltage ratings of input stage components like: bridge rectifier
diodes maximum blocking voltage and smoothing capacitor’s rated
voltages, which were 200VDC . That is, excitation could resume at any
desired voltage level below the components rated voltages.
In the case of the hybrid field generator, 24VAC l ine voltage, at 149rpm,
was maintained to keep the charging voltage fixed at 28VDC before
123
excitation. The excitation in this case started at speeds below and above
149rpm, and at voltages above and below 32.41VDC , where 32.41VDC is a
corresponding average voltage when line voltage is 24VAC .
Based on a DC/DC converter test data in appendix C, it is evident that a
1:1 transformer would be suitable for the converter because the generator
rotor needed 22V for maximum excitation whilst the battery bank voltage
is 24V. This would also halve the transformer’s primary copper losses as
opposed to the 1:2 transformer. In addition to reduced losses, the 1:1
transformer would improve the battery bank durability because of reduced
discharge rate.
To ensure an efficient power conversion, switching MOSFETs had to be
matched. Matching the MOSFETs reduced chances of conversion failure
and ringing effects in the converter circuit.
A zener diode and a blocking diode in the converter output stage reduced
effects of back emf mainly when the converter was shutdown because the
effects escalate during the shutdown period. Those were directly
proportional to the generator speed and the load current.
A relationship between the fixed 24V at 600rpm, the load current and the
exciting current is shown in figure 7.1. The 24V was maintained over the
load current range (0A to 10A). At every instant of time when load
current increased, the exciting current decreased resulting in a stable
voltage and vice versa. A decrease in the exciting current due to the
increase in the load current resulted in the reduced armature power losses.
In this way, the excitation control efficiency was improved. That also
implied that the controlled generator loading could be used to stabilize
the generated voltage prior to the armature excitation process resumption.
124
The generator loading process would stop when the full load current is
reached. Then the excitation process would start.
Full load current can therefore be represented as follows:
Ifu l l load = 10A = Ireg + Icon t ro l led load i n g 7.1
Excitation IV curves at 24V V-line 600rpm
01020304050607080
0 2 4 6 8 10 12
DC Load current (A)
Vol
tage
s (V
) AC
&
DC
-6-4-2024681012
Cur
rent
s (A
)
Vline (V)Vdc (V)Vrotor (V)Iline (A)Idc (A)Iexc (A)
Figure 7.1 On load excitation curves at 24V line voltage and at 600rpm
It was found that the efficiency of the system at full load could be
improved by setting the load current to be directly proportional to the
generator speed. For example, if the current at 149rpm could be restricted
to 2A the generator stator copper losses would be reduced by 92.7%. The
controller on board losses would be reduced to 25% of the maximum
losses.
The generated voltage level appeared to have dropped by 7.7% after a
series of repeated generator tests. See figure 7.2. This was because the
magnetism of the permanent magnets embedded in the rotor was reduced.
This was clearly indicated by the generated voltage formula:
125
E = 4.44NΦ fK
where N, f and K are constants. Therefore, a drop in the voltage could be
due to a drop in exciting flux density. The demagnetizing effect could be
reduced by replacing the embedded magnets with the magnets that can
retain their magnetism after the buck and boost modes of operation.
O/C votage characteristics over a range of tests (750rpm)
70
80
90
100
110
120
0 2 4 6 8
Test NO:
Vol
tage
(AC
& D
C) i
n V
olts
V-DC V-Line
Figure 7.2 Open circuit voltage characteristics at 750rpm over a range of
voltage tests
Using a microcontroller also plays an important role in that it
continuously compares parameters like; generated voltage and line
current, with their corresponding references, which are set within
software. It also makes modifications of the controller parameter
references easier than when they are to be done in hardware.
Analog inputs to the micro-controller may be read independently, each
input assigned a channel, or be multiplexed. In the latter case, one
channel is used to read the inputs and only one voltage reference is set for
the inputs. This may be a problem if the inputs response graphs do not all
cross the y-axis at the same point. See figure 2.1 and figure 6.10.
126
However, if a problem of this nature arises the former case may be used
although the reference voltage is required for each channel.
Because the generator characteristics were studied and the controller
specifications could be tailored successfully, a controller to suit the
optimized performance of the hybrid field generator may be designed.
7.2 Recommendations
The generator performance test and the control signal test-board test
results analysis lead to the following recommendations for the controller
hardware design purpose:
• Use of temperature compensated components like hall-effect current
sensors and linear opto-coupler LOC110, and Low temperature
coefficient resistors like metal film type resistors should be used in
the hardware design of the controller. These types of components
ensure very low drift in linearity due to temperature rise.
• Signal translating circuitry has to be powered from a dual power
supply. This guarantees easy control over offset voltages that are
associated with operational amplifiers.
• The generator speed sensing needs to be performed with the disk
method because this method presents a more accurate speed
detection. It can also read a fraction of a revolution e.g 149.3rpm
whereas the electronic one presents ±10% error at cut-in speed
(149rpm) and ±2.5% error at cut-out speed (598rpm).
• Power dissipation in the generator leads to temperature rise in the
generator, therefore the temperature will also have to be taken into
127
account during the design and development of the controller. This
component can lead to the failure of the generator if not mitigated.
• The DC/DC converter transformer needs to be designed and
developed to suit the generator excitation.
7.3 Future work
Present day technology focuses on human machine interface. Therefore, a
further improvement in the development of hybrid field generator control
systems also has to be channeled in the same direction. Figure 7.3 shows a
supervisory control system that would enable the system operator to view
the system performance. It would also make system fault diagnosis and
maintenance easier.
Referring to chapter 7, the causes of line current imbalance could be
located by comparing the line currents before and after connecting the
fault load. If the currents imbalance could be maintained even when the
load is connected, then the problem would be with the generator or
otherwise the charging unit. The fault location could be carried out to the
extent where it could be known whether the problem is with the red,
yellow or blue line by just monitoring the line currents.
128
Generatorexcitation and
power controller
24VBattery bank DC loads
Hybrid fieldgenerator
Operator station
ControlledloadingSynchronizing unitGrid
terminal
Figure 7.3 Hybrid field control station network
129
List of references
ABB. (1998). Effects of AC Drives on Motor Insulation USA: ABB
Industrial Systems Inc
Ahmed, A. (1999). Power Electronics for Technology. New Jersey:
Prentice-Hall-Inc.
Ang, S. S. (1995). Power Switching Converters. New York: Marcel
Dekker, Inc
Beak, J ., Budding, P., & Scaini V. (2001). Reusing and Rerating Older
Rectifiers with New DC/DC Choppers. IEEE Transactions on industry
applications, VOL. 37, NO. 4, pp. 1160 – 1166.
Bell, F. W. (2001).Current Sensors Catalogue USA: Sypris
Carr, J . T. (1991). Microcomputer interfacing New Jersey: Prentice hall
Cripps, M. (1989. Computer interfacing. Great Britain: Routledge
Bergquist, C. (2001). Guide to PICMICRO Microcontrollers. USA:
Prompot.
Billings, K. H. (1999). Switchmode Power Supply Handbook.(2n d ed)
Mexico: McGraw-Hill Inc.
Byers T. J . (1991). Printed Circuit Board Design with Microcomputers.
New York: McGraw-hill.
130
Clive, D. S. (1991). Modern Battery Technology. England: Ellis Horwood
Ltd
Datta, S. K. (1985). Power Electronics and Controls. USA: Prentice-Hall
Daniel, H. S. (1981). Transducer Interfacing Handbook. USA: Analog
Devices.
Erickson, R. W. & Maksimovic D. (2001). Fundamentals of Power
Electronics (2n d ed). USA: Kluwer Academic Publisher.
Hickman, I. (1990). Analog Electronics. Great Britain: Newnes.
Hnatek, R. E. (1989). Design of Solid State Power Supplies (3 rd ed). New
York: Van Nostrad Reihold.
James, M. R. (1997). Microcontroller Cookbook PIC & 8051. Great
Britain: Newnes.
Karki, J . (2000). Active Low Pass Filter Design USA: Texas Instruments
Lander, C. W. (1993). Power Electronics (3 rd ed). England: McGraw-Hill
Loveday, G. C. (1992). Designing Electronic Hardware. United Kingdom:
Longman Scientific & Technical.
Morley. A, Hughes. E, Bolton. W. (1994). Principles of Electricity (5 t h
ed). United Kingdom: Longman.
Muhammad, H. R. (1993). Power Electronics Circuit , Devices, and
Applications (2n d ed). New Jersey: Prentice Hall Inc.
131
Pressman, A. I. (1998). Switching power supply design (2n d ed). USA:
McGraw-Hill
Rahman, F. (2002). ELEC4216/9231: PWM Converter Drives England:
University of South Wales
Talor, B.E. (1993). Power Mosfet Design. England: John Willey & Sons
Ltd.
Tihanyi, L. (1995). Electromagnetic Compatibility in Power Electronics.
Florida USA: J. K. Eckert & Company. Inc.
Van Der Linde, H. A. (2001). Development of a Hybrid Field Generator
for Wind Turbine Applications. Unpublished Phd Thesis. University of
Hertfortshire, United Kingdom.
Development tools [online].(2002, February 19): Internet [cited 2002-02-
21]. Available from the Internet URL.
http://www.microchip.com/1000/pline/tools/picmicro/icds/mplabicd/detail
ed/index.htm
Developer [online]. (2000, November 14): Internet [cite 2002-02-21].
Available from the Internet.
http://developer.intel.com/sites/developer/index.htm?iid=Homepage+Sites
_Developer&#IntelTop
Online Abstracts and Reports [online]. (2002, January 31): Internet [cited
2002-02-21]. Available from the Internet.
http://www.sandia.gov/Renewable_Energy/wind_energy/topical.htm
132
Inverter technology [On line]. (2003): Internet [cited on 16-09-2003].
Available form internet URL
www.rvtechstop.com/articles.htm
IR design tips[online]. (2002). Internet [cited on 10-10-2002]. Available
from internet URL
www.irf.com/technical-info
NMTC wind turbine research[on line]. Internet NWTC webmater, (cited
on 12-2-2002) Available form internet
www.nrel.gov/wind/components2.html
Olmstead. R.(2000)Level shifting nixels need for dual power supply [cited
on 27-03-2003]. Available form internet URL
www.e-insite.net/ index asp?layou
Phillips G. Adlam F. (2001) PIC16F877 Development system [online]:
Internet [cited 2003]. Available from the internet URL
http://www.petech.ac.za/elec_project
Wind Generator Air 403 [online]: Internet [cited 2002-02-21]. Available
from the Internet URL
http://www.empowerco.com/windgen.htm
Second Wind Phaser [online]. (2002, January 11): Internet [cited 2002-02-
21] Available from the Internet.
http://www.secondwind.com/phaserhome.htm
INFOWEB [online]: Internet [cited 2002-04-10]. Available from Internet.
http://www.powerdesigners.com
133
Reid. B (2002) Proven Engineering Products Ltd [online]: Internet [cited
2003-01-24] Available from Internet: URL
http://www.almac.co.uk/proven/PAGES/STORAGE.Htm
Application Notes
Adams, J . AN-1024a: Flyback Transformer Design for the IRIS40XX
series, USA: International Rectifier
Conel, W., & Mclymal, T. AN-109:Push-Pull Converter Design using a
Coremaster E2000Q core, USA: Coremaster International, Inc
Gilbert, J ., Dewey, R. (2002). AN27702A: Linear Hall-Effect Sensors.
Massachusettes: Allegro Microsystems, Inc
Clare. Application note AN-107: Internet [cited on 2002-08-21] Available
from the internet.
http://www.clare.com
Magnetics. Butler, PA. Section 4: Power Design: Internet [cited 2003-9-
30] Available from the internet
http://www.topmagnetics.com/Magnetics_design_application_notes.pdf
Abstracts
Bordy, F., & Dupaquier, A. High Current, Low Voltage power converter
for LHS. Switzerland: SL Division CERN
Gauvin, M. A., & Frenniere, E. R. Reducing Stray light in Opto-
mechanical System. USA: Lambda Research Corporation
134
CD-ROMs RS Cataloque [CD-ROM]. (2003). SA: RS Components SA.
SPICE Simulator for the evaluation of ON Semiconductor SMPS Solutions
[CD-ROM]. (2002). US: Intusoft.
135
Appendix A
Flow charts and blocks code
Read Analog Value Functionread_analog()
Start
Initialize Variable
set_adc_channel(x)
initialize acd variable
For int i<50
READ_ADC( )
Assign sum of read values to ADC variable
Assign read values average to ADC variable
End
<50
=>50
136
Reverse / Forward / Permanent Mode Functions Flow chartRM_read_analog() / FM_read_analog() / PM_read_analog()
Start
Initialize selection variableinteger sel
For sel. < 8
If !(24<Vbat<28)
Read speed (Sp) in Hz &Convert Sp to RPM
Read speed (Sp) in Hz &Convert to RPM
Select generatedvoltage with load con.
Select generatedvoltage with load Disc.
read_analog()
Assign read analogvalue to Vg
If !(24<Vbat<28)
Select Irt withload Disc.
Select Irt withload con.
read_analog()
Assign read analogvalue to Irt
If !(24<Vbat<28)
Select Ib withload Disc.
Select Ib withload con.
read_analog()
Assign read analogvalue to Ib
End
case 0
case 1
case 7
switch (sel)
All cases block sequences are similar except that each case selectsa variable to be read. The case sequence of variables for "Fault_ana()" is different from those of other functions, refer to the fault software.
Fault variables read function
Not
Within
Not
Within
Not
Within
<8
=>8
(Fault_ana())
137
Fault Analysis and Operation Function
Fault_ana()
Start
While !(149<Sp < 598)
Fault_ana()
Determine I-line from Ireg(load Current)
Check whether I-lines are within 5% tolerance range
While not within range connectbalanced star network
Determine I-lines averagefrom sum of the line currents
Fault_ana()
Check whether I-lines are within 5% tolerance rangeand display fault message as per current
Not
End
This function analyses the currents and consequently display fault location. It first checkswhether line currents are within 5% tolerance range. If not, it connects a balanced starnetwork and checks whether the currents are within 5% tolerance range. If yes, the fault ison the controllers side and if not it is on the generator's side for that particular line.
Within
NotWithin
138
Main Function
Start
InitializePIC16F877
Setup port A
Setup ADC
Write To LEDS
Select operationmode
While powerON
End
ON
OFF
Select Operation Mode
Start
Permanent mode
If 149<speed <598rpm
If 28<G-voltage <35V
Display PM mode
Permanent mode
If G-voltage< 28V
Display FM mode
Forward mode
Display RM mode
Reverse mode
Fault operation
End
>35V
Yes
NoYes
No
<28V
Opr_Msel()void main (void)
139
/********************************************************************* Compiler: CCS C compliler www.ccsinfo.com Editor: Microchip MPLAB www.microchip.com Filename: HFG_TstC.c Description: The Hybrid Field Generator (HFG) control function simulation blocks
were tested with PIC16F877 development system board. The test was focused on the sequential handling of the functions as per control event occurrence.
Test Device: PIC16F877 development system (Phillips. Adlam. 2001) Author: C.T. 'Moleli Date (2003) *********************************************************************/ #include "D877_INI.C" #include "LCD.C" #include "KEYPAD.C" int sel; unsigned int Sp=374,Vg=35,Irt=3,Ireg,Vbat=26,Vrt,Ir=4,Iy=4,Ib=4,Il, Ilav,Ilavmx,Ilavmn,Ilmx,Ilmn,Red,Yellow,Blue,adc_value=0; /********************************************************************* The function reads and returns a selected analog input value *********************************************************************/ read_analog() byte i=0; set_adc_channel(0); //Select channel 0 adc_value = 0; //Zero variable for(i=0;i<50;i++) //Setup 50 times loop for 50 ADC samples adc_value = adc_value + Read_ADC_10(); //Add all readings //to variable delay_us(12); //time delay between samples taken //See table 11.1 note 1 in //datasheet of the PIC16F877 p117 adc_value = adc_value/50;
140
/********************************************************************* The function measures Speed during calibration *********************************************************************/ Get_Speed() Write_LEDS(0x03); //The Write_LEDS() commands set a pulse Write_LEDS(0x13); //to triger speed counter Sp=Read_Switches(); //Read speed in hertz /******************************************************************** The function allows an operator to calibrates the variables during run-time. The calibration block immediately switches to the permanent magnet except for the rotor current and rotor voltage. ********************************************************************/ Calibrate() while(Read_Keypad()=='*') // Hold cursor in calibration lcd_clrscr(); //Clear LCD module display delay_ms(100); while(Read_Keypad()!='*') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Calibration ON"); //Write message to LCD lcd_gotoxy(0,1); Lcd_putc("Press * to Exit"); switch(Read_Keypad()) case '0':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') //Wait utill # is pressed lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Generator Speed"); //Write message to LCD Get_Speed(); lcd_gotoxy(0,1); printf(lcd_putc,"Speed = %2u",Sp ); lcd_gotoxy(11,1); lcd_putc("Hz"); lcd_clrscr(); //Clear LCD module display break; case '1':lcd_clrscr(); //Clear LCD module display
141
while(Read_Keypad()!='#') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Generated Volt"); //Write message to LCD Write_LEDS(0x13); //Selects Vg read_analog(); Vg=adc_value; lcd_gotoxy(0,1); printf(lcd_putc,"Voltage = %2u",Vg); lcd_gotoxy(13,1); lcd_putc("V"); lcd_clrscr(); //Clear LCD module display break; case '2':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Exciting Current"); //Write message to LCD Write_LEDS(0x33); read_analog(); Irt=adc_value; lcd_gotoxy(0,1); printf(lcd_putc,"I ex = %2u",Irt); lcd_gotoxy(11,1); lcd_putc("A"); lcd_clrscr(); //Clear LCD module display break; case '3':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') lcd_gotoxy(0,0); lcd_putc("Charging Current"); //Write message to LCD Write_LEDS(0x53); read_analog(); Ireg=adc_value; lcd_gotoxy(0,1); printf(lcd_putc,"I Charge = %2u",Ireg); lcd_gotoxy(14,1); lcd_putc("A"); lcd_clrscr(); //Clear LCD module display break; case '4':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("B. Bank Voltage"); //Write message to LCD
142
Write_LEDS(0x73); read_analog(); Vbat=adc_value; lcd_gotoxy(0,1); printf(lcd_putc,"V B Bank = %2u",Vbat); lcd_gotoxy(14,1); lcd_putc("V"); lcd_clrscr(); //Clear LCD module display break; case '5':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Rotor Voltage"); //Write message to LCD Write_LEDS(0x93); read_analog(); Vrt=adc_value; lcd_gotoxy(0,1); printf(lcd_putc,"V Rotor = %2u",Vrt); lcd_gotoxy(13,1); lcd_putc("V"); lcd_clrscr(); //Clear LCD module display break; case '6':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Red Line Current"); //Write message to LCD Write_LEDS(0xb3); read_analog(); Ir=adc_value; lcd_gotoxy(0,1); printf(lcd_putc,"I Red = %2u",Ir); lcd_gotoxy(11,1); lcd_putc("A"); lcd_clrscr(); //Clear LCD module display break; case '7':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Yellow L Current"); //Write message to LCD Write_LEDS(0xd3); read_analog(); Iy=adc_value; lcd_gotoxy(0,1);
143
printf(lcd_putc,"I Yellow = %2u",Iy); lcd_gotoxy(14,1); lcd_putc("A"); lcd_clrscr(); //Clear LCD module display break; case '8':lcd_clrscr(); //Clear LCD module display while(Read_Keypad()!='#') lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Blue L Current"); //Write message to LCD Write_LEDS(0xf3); read_analog(); Ib=adc_value; lcd_gotoxy(0,1); printf(lcd_putc,"I Blue = %2u",Ib); lcd_gotoxy(12,1); lcd_putc("A"); lcd_clrscr(); //Clear LCD module display break; lcd_clrscr(); //Clear LCD module display /********************************************************************* The function reads analog input values during reversed rotor excitation, while it is still monitors a battery bank voltage. It disconnects DC loads when the battery bank voltage is outside a desired voltage range (24V to 28V). *********************************************************************/ RM_read_analog() lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc(" Reversed "); //Write message to LCD lcd_gotoxy(0,1); lcd_putc("Excitation"); Calibrate(); sel=0; for(sel=0;sel<8;sel++) switch(sel)
144
case 0: if(!((24<Vbat)&&(Vbat<28))) // check whether battery bank voltage is // within limts Write_LEDS(0x00); Write_LEDS(0x10); delay_ms(1000); Sp=Read_Switches(); Write_LEDS(0x10); // selects Vg & disconnects DC load else Write_LEDS(0x02); Write_LEDS(0x12); delay_ms(1000); Sp=Read_Switches(); Write_LEDS(0x12); // selects Vg (Generated DC voltage) // with DC load connebted delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vg=adc_value; // assigns adc_value to Vg break; case 1: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x30); // selects Irt & disconnects DC load else Write_LEDS(0x32); // selects Irt (Rotor/excitation current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Irt=adc_value; // assigns adc_value to Irt break; case 2: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x50); // selects Ireg & disconnects DC load else Write_LEDS(0x52); // selects Ireg (Voltage regulator current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ireg=adc_value; // assigns adc_value to Ireg break; case 3: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x70); // selects Vbat & disconnects DC load else Write_LEDS(0x72); // selects Vbat (Battery bank voltage) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vbat=adc_value; // assigns adc_value to Vbat break;
145
case 4: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x90); // selects Vrt & disconnects DC load else Write_LEDS(0x92); // selects Vrt (Rotor Voltage) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vrt=adc_value; // assigns adc_value to Vrt break; case 5: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xb0); // selects Ir & disconnects DC load else Write_LEDS(0xb2); // selects Ir (Red line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ir=adc_value; // assigns adc_value to Ir break; case 6: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xd0); // selects Iy & disconnects DC load else Write_LEDS(0xd2); // selects Iy (Yellow line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Iy=adc_value; // assigns adc_value to Iy break; case 7: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xf0); // selects Ib & disconnects DC load else Write_LEDS(0xf2); // selects Ib (Blue line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ib=adc_value; // assigns adc_value to Ib break; /********************************************************************* The function reads analog input values during forward rotor excitation, while it is still monitors a battery bank voltage. It disconnects DC loads when the battery bank voltage is outside a desired voltage range (24V to 28V). *********************************************************************/ FM_read_analog() lcd_gotoxy(0,0); //Set LCD cursor to position
146
lcd_putc("Forward "); //Write message to LCD lcd_gotoxy(0,1); lcd_putc("Excitation"); Calibrate(); sel=0; for(sel=0;sel<8;sel++) switch(sel) case 0: if(!((24<Vbat)&&(Vbat<28))) // check whether battery bank voltage is // within limts Write_LEDS(0x04); Write_LEDS(0x14); delay_ms(1000); Sp=Read_Switches(); Write_LEDS(0x14); // selects Vg & disconnects DC load else Write_LEDS(0x06); Write_LEDS(0x16); delay_ms(1000); Sp=Read_Switches(); Write_LEDS(0x16); // selects Vg (Generated DC voltage) // with DC load connebted delay_ms(3000); // delay before taking a reading read_analog(); // reads selected analog input Vg=adc_value; // assigns adc_value to Vg break; case 1: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x34); // selects Irt & disconnects DC load else Write_LEDS(0x36); // selects Irt (Rotor/excitation current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Irt=adc_value; // assigns adc_value to Irt break; case 2: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x54); // selects Ireg & disconnects DC load else Write_LEDS(0x56); // selects Ireg (Voltage regulator current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input
147
Ireg=adc_value; // assigns adc_value to Ireg break; case 3: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x74); // selects Vbat & disconnects DC load else Write_LEDS(0x76); // selects Vbat (Battery bank voltage) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vbat=adc_value; // assigns adc_value to Vbat break; case 4: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x94); // selects Vrt & disconnects DC load else Write_LEDS(0x96); // selects Vrt (Rotor Voltage) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vrt=adc_value; // assigns adc_value to Vrt break; case 5: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xb4); // selects Ir & disconnects DC load else Write_LEDS(0xb6); // selects Ir (Red line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ir=adc_value; // assigns adc_value to Ir break; case 6: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xd4); // selects Iy & disconnects DC load else Write_LEDS(0xd6); // selects Iy (Yellow line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Iy=adc_value; // assigns adc_value to Iy break; case 7: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xf4); // selects Ib & disconnects DC load else Write_LEDS(0xf6); // selects Ib (Blue line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ib=adc_value; // assigns adc_value to Ib break;
148
/******************************************************************** The function reads analog input values during no rotor excitation, while it is still monitors a battery bank voltage. It disconnects DC loads when the battery bank voltage is outside a desired voltage range (24V to 28V). ********************************************************************/ PM_read_analog() lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("P Magnet "); //Write message to LCD lcd_gotoxy(0,1); lcd_putc("Excitation"); Calibrate(); sel=0; for(sel=0;sel<8;sel++) switch(sel) case 0: if(!((24<Vbat)&&(Vbat<28))) // check whether battery bank voltage is within limts Write_LEDS(0x01); // enables generator speed counter Write_LEDS(0x11); delay_ms(1000); // one second delay Sp=Read_Switches(); // converts generator speed in Hz to RPM Write_LEDS(0x11); // selects Vg & disconnects DC load else Write_LEDS(0x03); // enables generator speed counter Write_LEDS(0x13); delay_ms(1000); Sp=Read_Switches(); //converts generator speed in HZ to RPM Write_LEDS(0x13); // selects Vg (Generated DC voltage)
// with DC load connebted delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vg=adc_value; // assigns adc_value to Vg break; case 1: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x31); // selects Irt & disconnects DC load
149
else Write_LEDS(0x33); // selects Irt (Rotor/excitation current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Irt=adc_value; // assigns adc_value to Irt break; case 2: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x51); // selects Ireg & disconnects DC load else Write_LEDS(0x53); // selects Ireg (Voltage regulator current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ireg=adc_value; // assigns adc_value to Ireg break; case 3: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x71); // selects Vbat & disconnects DC load else Write_LEDS(0x73); // selects Vbat (Battery bank voltage) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vbat=adc_value; // assigns adc_value to Vbat break; case 4: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x91); // selects Vrt & disconnects DC load else Write_LEDS(0x93); // selects Vrt (Rotor Voltage) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Vrt=adc_value; // assigns adc_value to Vrt break; case 5: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xb1); // selects Ir & disconnects DC load else Write_LEDS(0xb3); // selects Ir (Red line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ir=adc_value; // assigns adc_value to Ir break; case 6: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xd1); // selects Iy & disconnects DC load else Write_LEDS(0xd3); // selects Iy (Yellow line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Iy=adc_value; // assigns adc_value to Iy break;
150
case 7: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xf1); // selects Ib & disconnects DC load else Write_LEDS(0xf3); // selects Ib (Blue line current) delay_us(12); // delay before taking a reading read_analog(); // reads selected analog input Ib=adc_value; // assigns adc_value to Ib break; /******************************************************************** The function reads analog input values during the fault period, while it is still monitors a battery bank voltage. It disconnects DC loads when the battery bank voltage is outside a desired voltage range (24V to 28V). ********************************************************************/ Fault_ana() Write_LEDS(0x79); // selects battery bank voltage & disconect the load read_analog(); Vbat=adc_value; for(sel=0;sel<6;sel++) switch(sel) case 0: if(!((24<Vbat)&&(Vbat<28))) // Checks whether Vbat is within
// limilts Write_LEDS(0xb9); // selects Red line current with load
// disconnected read_analog(); Ir=adc_value; else Write_LEDS(0xbb); // selects Red line current with load
// connected read_analog(); Ir=adc_value; break; case 1: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xd9); // selects Yellow line current with
151
// load disconnected read_analog(); Iy=adc_value; else Write_LEDS(0xdb); // selects Yellow line current with //load connected read_analog(); Iy=adc_value; break; case 2: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0xf9); // selects Blue line current with load // disconnected read_analog(); Ib=adc_value; else Write_LEDS(0xfb); // selects Blue line current with load // connected read_analog(); Ib=adc_value; break; case 3: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x09); // selects Speed with load
//disconnected Write_LEDS(0x19); // selects Generated voltage delay_us(12); Sp=Read_Switches(); // Converts Hz to RPM read_analog(); // read Generated voltage Vg=adc_value; else Write_LEDS(0x0b); // selects Speed with load connected Write_LEDS(0x1b); // selects Generated voltage delay_ms(1000); Sp=Read_Switches(); // converts Hz to RPM read_analog(); // read Generated voltage Vg=adc_value; break; case 4: if(!((24<Vbat)&&(Vbat<28)))
152
Write_LEDS(0x79); // selects Vbat with load // disconnected read_analog(); Vbat=adc_value; else Write_LEDS(0x7b); // selects Vbat with load connected read_analog(); Vbat=adc_value; break; case 5: if(!((24<Vbat)&&(Vbat<28))) Write_LEDS(0x39); // selects Irt with load disconnected read_analog(); Vbat=adc_value; else Write_LEDS(0x3b); // selects Irt with load connected read_analog(); Vbat=adc_value; break; /******************************************************************* The function monitors an exciting current. It connects fault load and shutdown the DC/DC converter when the exciting current exceeds 5A. *******************************************************************/ Rotor_Current() lcd_clrscr(); //Clear LCD module display while(Irt>5) lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("Over Excitation"); //Write message to LCD Calibrate(); delay_us(12); Fault_ana();
153
/******************************************************************* The function minitors the line currents. It connects fault load and shutdown the DC/DC converter when the currents are not +/- 5% equal to each other. *******************************************************************/ Fault_Opr_Current() lcd_clrscr(); //Clear LCD module display while(!((Red==Yellow)&&(Yellow==Blue))) // Wait utill lines are balanced // Generator is cconnected to balanced star sel=0; // netwok so all the line currents are expected Fault_ana(); // to have close relationship, 5% tolerance. Ilav=(Ir+Iy+Ib)/3; // Determine average line current Ilavmx=Ilav+(Ilav*(5/100)); // Determine max average line current Ilavmn=Ilav-(Ilav*(5/100)); // Determine min average line current sel=0; for(sel=0;sel<3;sel++) switch(sel) case 0: if(!((Ilavmn<Ir)&&(Ir<Ilavmx))) // if Ir is not within range
// and Red=1; // if yes,note the fault. like wise else // with other lines. Red=0;break; case 1: if(!((Ilavmn<Iy)&&(Iy<Ilavmx))) Yellow=1; else Yellow=0;break; case 2: if(!((Ilavmn<Ib)&&(Ib<Ilavmx))) Blue=1; else Blue=0;break; lcd_gotoxy(0,0); //Set LCD cursor to position lcd_putc("L Current Imbal"); //Write message to LCD delay_ms(1000); sel=0; // resets select variable
154
/******************************************************************** The function monitors the line currents for the line imbalance ********************************************************************/ Test_Opr_Current() Il=(Ireg*(1000/955))*(707/1000); // Determine line current Ilmx=Il+(Il*(5/100)); // I-line max = I-line + 5% Ilmn=Il-(Il*(5/100)); // I-line min = I-line - 5% sel=0; for(sel=0;sel<3;sel++) switch(sel) case 0: if(!((Ilmn<Ir)&&(Ir<Ilmx))) // Is Ir not within tolerance range? Red=1; // If yes note fault in red line else // like wise with other lines Red=0;break; case 1: if(!((Ilmn<Iy)&&(Iy<Ilmx))) Yellow=1; else Yellow=0;break; case 2: if(!((Ilmn<Ib)&&(Ib<Ilmx))) Blue=1; else Blue=0;break; Rotor_Current(); Fault_Opr_Current(); /******************************************************************** The function minitors the HFG speed when HFG speed is not within range. It connects fault load and shutdown the DC/DC converter. *********************************************************************/ Fault_Opr_Speed() while(!((10<Sp)&&(Sp<40))) // Checks whether the speed is within range and // wait for as long as it is not. lcd_gotoxy(0,0); //Set LCD cursor to position
155
lcd_putc("Speed Range"); //Write message to LCD Calibrate(); delay_ms(1000); sel=3; Fault_ana(); Test_Opr_Current(); /******************************************************************** The function monitors HFG speed and the generated voltage. It then selects an operation mode accordingly. ********************************************************************/ Opr_Msel() Test_Opr_Current(); if((10<Sp)&&(Sp<40)) // Checks whether speed is within limits if((30<Vg)&&(Vg<35)) PM_read_analog(); else if(Vg<30) FM_read_analog(); else if(Vg>35) RM_read_analog(); Calibrate(); else Fault_Opr_Speed(); Calibrate(); /******************************************************************** The function initializes the PIC16F877 development system and the lcd display module for HMI. ********************************************************************/ void main (void) D877_Init(); //Initialise PIC16F877 development system
156
lcd_Init(); //Initialise LCD display setup_port_a(RA0_RA1_ANALOG_RA3_REF); //Enable channels 0 and 1 - RA3 is //used as reference input setup_adc(ADC_CLOCK_INTERNAL); //Use internal RC clock for ADC //Conversion time is minimum 72uS PM_read_analog(); //Update variables while(1) Calibrate(); Opr_Msel();
157
Operation modes bit masking tables S3 S2 S1 Ct E Dmp Pol L D Con Permanent Forward Reverse Variable MSB LSB mode mode mode 7 6 5 4 3 2 1 0 L L D L L D L L D Speed 0 0 0 0 0 0 1 1 0X03 0X01 0X06 0X04 0X02 0X00 Vg 0 0 0 1 0 0 1 1 0X13 0X11 0X16 0X14 0X12 0X10 Irt 0 0 1 1 0 0 1 1 0X33 0X31 0X36 0X34 0X32 0X30 Ireg 0 1 0 1 0 0 1 1 0X53 0X51 0X56 0X54 0X52 0X50 Vbat 0 1 1 1 0 0 1 1 0X73 0X71 0X76 0X74 0X72 0X70 Vrt 1 0 0 1 0 0 1 1 0X93 0X91 0X96 0X94 0X92 0X90 Ir 1 0 1 1 0 0 1 1 0XB3 0XB1 0XB6 0XB4 0XB2 0XB0 Iy 1 1 0 1 0 0 1 1 0XD3 0XD1 0XD6 0XD4 0XD2 0XD0 Ib 1 1 1 1 0 0 1 1 0XF3 0XF1 0XF6 0XF4 0XF2 0XF0 Fault mode bit masking table S3 S2 S1 Ct E Dmp Pol L D Con Fault Variable MSB LSB mode 7 6 5 4 3 2 1 0 L L D Speed 0 0 0 0 1 0 1 1 0X0B 0X09 Vg 0 0 0 1 1 0 1 1 0X1B 0X19 Irt 0 0 1 1 1 0 1 1 0X3B 0X39 Ireg 0 1 0 1 1 0 1 1 0X5B 0X59 Vbat 0 1 1 1 1 0 1 1 0X7B 0X79 Vrt 1 0 0 1 1 0 1 1 0X9B 0X99 Ir 1 0 1 1 1 0 1 1 0XBB 0XB9 Iy 1 1 0 1 1 0 1 1 0XDB 0XD9 Ib 1 1 1 1 1 0 1 1 0XFB 0XF9
158
Appendix B
Hybrid field generator performance characteristics tests
results
Generator test results
Generator Characteristics At Mid-Range Speed ( 374 rpm ) Excitation V_ Load I_load V_ Line I_ Line Torque Temp Power in DC Power ( A ) DC DC AC AC Nm Dgr. Cel. W W
0 57.1 0 42.6 0 0.8 21 31.332 00 52.2 1 39.8 0.74 2.4 21 93.996 52.20 48.1 2 37.72 1.48 3.4 21 133.162 96.20 45.7 3 36.05 2.24 4.5 21 176.243 137.10 42.4 4 33.75 2.98 5.6 21 219.325 169.60 39.7 5 31.95 3.71 6.3 21 246.741 198.50 36.95 6 29.77 4.49 7.1 21 278.073 221.70 33.67 7 27.63 5.21 7.3 21 285.906 235.690 30.17 8 25.27 5.95 8.5 21 332.904 241.360 26.1 9 22.43 6.69 8.6 24 336.821 234.90 20.64 10 18.64 7.41 8.8 25 344.654 206.4
0 39.7 5 31.95 3.71 6.3 24 246.741 198.51 43.7 5 34.97 4.1 7.4 24 289.822 218.52 47.5 5 37.93 4.47 9.2 24 360.320 237.53 51.1 5 39.9 4.81 10.1 24 395.568 255.54 55 5 42.8 5.18 11.6 24 454.316 2755 58.1 5 45.4 5.49 13.3 25 520.897 290.56 61.3 5 47.7 5.79 14.9 27 583.561 306.57 64.5 5 50.2 6.1 16.9 28 661.892 322.5
0 39.7 5 31.95 3.71 6.3 24 246.741 198.5
-1 35.23 5 28.13 3.3 5.6 24 219.325 176.15-2 31.03 5 24 2.91 4.5 24 176.243 155.15-3 26.65 5 21.33 2.49 3.5 24 137.078 133.25-4 22.35 5 18.25 2.07 2.7 25 105.746 111.75-5 18.06 5 14.96 1.68 2 25 78.330 90.3-6 13.95 5 11.82 1.26 1.5 26 58.748 69.75-7 9.51 5 8.44 0.88 1.1 28 43.082 47.55
159
Fwd Ex. Torque Power Curves @ 374rpm
0
200
400
600
800
0 5 10 15 20
Torque (Nm)
Pow
er (W
)
Pow er in W DC Pow er W
Torque & Current Curves @ 374rpm
0
2
4
6
8
10
12
14
16
18
0 5 10 15
DC Current (A)
Torq
ue (N
m)
No Ex I_DC Fwd Ex @ 5A I DCRev Ex @ 5A I DC
N0 Excitation Load IV Curve @ 374rpm
0
10
20
30
40
50
60
0 2 4 6 8 10 12
Current (A)
Vol
tage
(V)
Torque & Power Curves @ 374rpm
050
100150200250300350400
0 5 10
Torque (Nm)
Pow
er (W
)
Pow er in W DC Pow er W
160
V I Excitation Curves @ 374rpm
0
10
20
30
40
50
60
70
0 5 10 15
DC Current (A)
DC
Vol
tage
(V)
No Ex V_ DCFwd Ex V_DC @ 5A DCRev Ex V_DC @5A_DC
161
Static air gap flux density measured with digital millitesla meter per pole Pole 1
Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5
0 -0.0001 -8.43 -10.09 -10.75 -9.74 -7.96 -1 5.59 -7.52 -8.97 -9.55 -8.44 -6.87 -2 8.97 -6.71 -8.11 -8.44 -7.59 -6.09 -3 12.83 -5.82 -6.95 -7.29 -6.62 -5.46 -4 16.45 -5.15 -6.01 -6.25 -5.69 -4.63 -5 21.2 -4.12 -4.85 -5.05 -4.61 -3.82
1 -4.61 -9.13 -11.33 -11.9 -10.82 -8.79 2 -8.97 -10.13 -12.47 -13.15 -11.88 -9.59 3 -12.75 -11.05 -13.64 -14.43 -13.04 -10.48 4 -16.76 -12.06 -14.97 -15.78 -14.12 -11.41 5 -20.7 -12.59 -15.98 -17.21 -15.43 -12.29
Pole 2
Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5
0 -0.0001 8.44 10.25 10.88 9.78 8.07 -1 5.25 7.52 9.22 9.84 8.85 7.19 -2 9.92 6.38 7.97 8.34 7.62 6.21 -3 13.39 5.63 6.97 7.34 6.73 5.45 -4 17.34 4.79 5.94 6.21 5.64 4.62 -5 21.5 4.08 5.02 5.23 4.78 3.87
1 -4.88 9.01 11.31 11.89 10.92 8.35 2 -9.27 9.97 12.49 13.21 11.75 9.71 3 -13.15 11.02 13.57 14.49 13.02 10.62 4 -17.4 12.03 15.01 15.91 13.98 11.7 5 -21.4 12.43 16.06 17.27 15.48 12.54
Pole 3
Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5
0 -0.0002 -8.21 -10 -10.29 -9.19 -7.65 -1 5.79 -7.13 -8.61 -8.84 -8.15 -6.84 -2 9.51 -6.31 -7.64 -7.78 -6.96 -5.75 -3 13.39 -5.31 -6.43 -6.54 -5.91 -4.85 -4 17.37 -4.31 -5.33 -5.42 -4.87 -4.06 -5 21.5 -3.05 -3.87 -4.31 -3.82 -3.17
1 -5.05 -8.31 -10.27 -10.48 -9.56 -7.23 2 -9.13 -9.23 -11.36 -11.71 -10.41 -8.01 3 -13.06 -10.18 -12.33 -12.91 -11.52 -9.98 4 -17.09 -11.1 -13.89 -14.25 -12.62 -10.92 5 -21.1 -12.01 -15.23 -15.61 -13.75 -11.87
162
Pole 4
Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5
0 -0.0002 6.92 8.76 8.85 7.96 6.64 -1 5.42 5.87 7.31 7.46 6.75 5.39 -2 9.63 4.95 6.08 6.17 5.54 4.51 -3 13.61 3.96 5 5.04 4.48 3.68 -4 17.8 3.08 3.8 3.84 3.49 2.75 -5 21.8 2.16 2.64 2.77 2.54 2.08
1 -5.03 7.94 9.92 9.98 9.04 7.56 2 -9.16 9.08 11.23 11.36 10.11 8.56 3 -13.58 10.32 12.81 12.84 11.45 9.59 4 -17.51 11.34 13.88 14.38 12.66 10.89 5 -21.6 12.02 15.84 16.14 14.11 11.95
Pole 5
Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5
0 0.0002 -7.11 -8.54 -9.12 -8.41 -7.04 -1 5.66 -6.02 -7.14 -7.63 -7.12 -5.71 -2 9.86 -4.97 -6.1 -6.31 -5.82 -4.79 -3 13.82 -4.02 -4.79 -5.05 -4.71 -3.8 -4 17.55 -3.21 -3.71 -3.99 -3.58 -2.95 -5 21.9 -2.24 -2.58 -2.67 -2.56 -2.11
1 -4.84 -8.09 -9.72 -10.34 -9.43 -7.86 2 -9.23 -9.11 -10.96 -11.76 -10.81 -8.83 3 -13.14 -9.89 -12.33 -13.21 -12.05 -9.91 4 -17.29 -11.01 -13.81 -14.82 -13.51 -11.04 5 -21.6 -12.26 -15.43 -16.68 -15.21 -12.25
Pole 6
Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5
0 -0.0003 7.47 8.87 8.91 7.84 6.57 -1 5.63 6.33 7.38 7.54 6.71 5.58 -2 9.38 5.35 6.46 6.55 5.77 4.76 -3 13.7 4.36 5.21 5.31 4.76 3.67 -4 17.68 3.58 4.17 4.24 3.77 2.92 -5 21.4 2.75 3.19 3.31 2.93 2.49
1 -4.83 8.23 9.91 9.97 8.98 7.48 2 -9.35 9.33 11.18 11.51 10.08 8.27 3 -13.25 10.35 12.49 12.65 11.35 9.15 4 -17.26 11.43 13.59 14.08 12.34 10.15 5 -21.6 12.36 15.32 15.39 13.59 11.32
163
Pole 7 Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5
0 -0.0002 -7.42 -9.18 -9.48 -8.63 -7.25 -1 6.08 -6.39 -7.85 -8.14 -7.29 -6.15 -2 9.88 -5.61 -6.79 -7.04 -6.32 -5.33 -3 13.96 -4.69 -5.71 -5.86 -5.29 -4.41 -4 17.99 -3.92 -4.65 -4.8 -4.29 -3.59 -5 22.2 -3.03 -3.57 -3.76 -3.41 -2.94
1 -4.8 -8.35 -10.32 -10.42 -9.69 -8.15 2 -9.91 -9.41 -11.54 -11.92 -10.81 -8.86 3 -13.25 -10.3 -12.54 -12.98 -11.8 -9.64 4 -17.65 -11.36 -13.95 -14.4 -12.91 -10.67 5 -21.9 -12.03 -15.45 -15.89 -14.29 -11.72
Pole 8
Idc Vdc Pt1 Pt2 Pt3 Pt4 Pt5
0 -0.0003 7.05 8.67 8.98 8.24 6.69 -1 6.23 6.05 7.43 7.71 6.92 5.76 -2 9.83 5.29 6.51 6.71 6.03 4.98 -3 14.33 4.34 5.26 5.48 4.92 4.23 -4 17.95 3.74 4.41 4.52 4.08 3.43 -5 22.3 2.87 3.36 3.48 3.19 2.69
1 -5.15 7.82 9.71 10.16 9.25 7.62 2 -9.91 8.86 10.97 11.51 10.38 8.62 3 -13.61 9.76 12.08 12.64 11.43 9.42 4 -17.96 10.75 13.33 13.98 12.66 10.07 5 -22.2 11.76 14.54 15.49 13.94 10.98
Average pole flux density in milli Tesla Iex Pt1 Av Pt2 Av Pt3 Av Pt4 Av Pt5 Av
0 7.63 9.30 9.66 8.72 7.23-1 6.60 7.99 8.34 7.53 6.19-2 5.70 6.96 7.17 6.46 5.30-3 4.77 5.79 5.99 5.43 4.44-4 3.97 4.75 4.91 4.43 3.62-5 3.04 3.64 3.82 3.48 2.901 8.36 10.31 10.64 9.71 7.882 9.39 11.53 12.02 10.78 8.813 10.36 12.72 13.27 11.96 9.854 11.39 14.05 14.70 13.10 10.865 12.18 15.48 16.21 14.48 11.87
164
Average permanent mde flux density Curve
0.002.004.006.008.00
10.0012.00
0 1 2 3 4 5 6
Pole teeth poistion
Flux
den
sity
(mT)
0 A
Average Buck mode flux density
0.00
2.00
4.00
6.00
8.00
10.00
12.00
0 1 2 3 4 5 6
Pole teeth position
Flux
den
sity
(mT) 0A
-1 A-2 A-3 A-4 A-5 A
165
Average Boost mode flux density
0.002.004.006.008.00
10.0012.0014.0016.0018.00
0 1 2 3 4 5 6
Pole teeth position
Flux
den
sity
(mT) 0A
1 A2 A3 A4 A5 A
Buck & Boost static average flux density
0.00
2.00
4.00
6.00
8.00
10.00
12.00
14.00
16.00
18.00
0 1 2 3 4 5 6
Pole teeth positions
Flux
den
sity
(mT)
0 A-1 A-2 A-3 A-4 A-5 A1 A2 A3 A4 A5 A
166
Static air gap flux density
0.00
5.00
10.00
15.00
20.00
-6 -4 -2 0 2 4 6
Exciting current (A)
Air
gap
flux
dens
ity (m
T)
Pt1 AvPt2 AvPt3 AvPt4 AvPt5 Av
Open circuit dynamic flux density
0
2
4
6
8
10
12
-6 -5 -4 -3 -2 -1 0 1 2 3 4 5 6
Exciting current (A)
Flux
den
sity
(mT)
149 rpm
264 rpm
374 rpm
598 rpm
750 rpm
167
Open Circuit test ( V_dc ) Iexc Speed In RPM Cut_in Mid_range Cut_out Sync Above Sync Speed 149 374 598 750 1000 1200 1500 1785
0 23 56 91 115 152 187 231.7 2771 26.6 63 101 127 171 207 257.2 3082 28 69 111 140 187.5227.5 283 3383 31 75 121 152 203.8244.7 307 3684 33 81 131 164 220264.8 331.6 3985 35 87 141 176 236.9283.8 355.5 4286 38 93 150 187 250.7302.2 379.4 4557 40 98 159 200 267322.3 402 4850 23 56 91 115 152 187 231.7 277
-1 21 51 83 104 139.8 167 208.8 250-2 18 45 72 90 122.5146.7 182.9 218-3 16 35 62 78 105.2126.4 155.5 187-4 13 32 52 66 88.8104.5 130.9 157-5 10 26 42 52 70.8 85 105.6 127-6 8 20 32 40 54 64.5 79.5 93-7 6 14 23 28 38.6 44.5 54 58
Open circuit buck & boost voltage curves
0
100
200
300
400
500
600
0 500 1000 1500 2000
Speed (rpm)
Vol
tage
(V)
0 A1 A2 A3 A4 A5 A6 A7 A-1 A-2 A-3 A-4 A-5 A-6 A-7 A
168
Short Circuit test ( I_dc ) Excitation Current Speed In RPM Cut_in Mid_range Cut_out Sync 149 374 598 750
0 10 12 12.5 12.51 11 13 13.5 142 11.5 14.7 15 153 13 16 16.5 16.54 14.7 16.5 17.5 17.55 16 18 19 196 17 19.5 20.5 20.57 21 22 220 10 12 12.5 12.5
-1 7 10.7 11 11.5-2 7.5 9.5 9.7 9.5-3 6.5 7.8 8 8-4 6 6.8 6.5 6.5-5 5 5.5 6 5.5-6 4.5 4 4.5 4.5-7 2.5 3.5 3.5 3.5
Short circuit buck & boost current curves
0
5
10
15
20
25
0 200 400 600 800
Speed (rpm)
Cur
rent
(A)
0 A
1 A
2 A
3 A
4 A
5 A
6 A
7 A
-1 A
-2 A
-3 A
-4 A
-5 A
-6 A
-7 A
169
Excitation power loses ( I*I*R) Ex Current Armture Power in (A) resistance in (W)
0 4.18 0 1 4.18 4.18 2 4.18 16.72 3 4.18 37.62 4 4.18 66.88 5 4.18 104.5 6 4.18 150.48 7 4.18 204.82 8 4.18 267.52 9 4.18 338.58
10 4.18 418
Armature Excitation Power loses
0100200300400500
0 2 4 6 8 10 12Exciting current (A)
Pow
er lo
ses
(W)
Maintaining 24V on the lines(Open circuit test) Iexc (A) Vline (V) Vdc (V) Vrotor (V) Flux Speed Torque
4.55 24 33 18.1 3.91 149 0.8 1.55 24.8 33.7 6.82 3.83 200 0.8
-0.78 24.2 32.8 4.11 3.55 250 0.8 -1.23 24.2 32.8 5.88 3.47 264 0.8 -2.18 24.1 32.7 9.42 3.29 300 0.8 -3.15 24.1 32.7 12.95 3.03 350 0.8 -3.64 24 32.4 14.79 2.94 374 0.8 -3.97 24 32.5 16.12 2.83 400 0.8 -4.47 24.1 32.6 18.02 2.73 450 0.8 -4.88 24.4 33 19.74 2.58 500 0.8 -5.21 24 32.5 21.3 2.37 550 0.8 -5.54 23.9 32.4 22.4 2.25 600 0.9
-5 31.2 42.3 20.3 2.79 650 0.9 -5 33.3 45.3 20.4 2.82 700 0.9
-4.98 35.4 48.1 20.3 2.9 750 0.9
170
Open circuit Excitation IV curves at 24V V-line
0
20
40
60
100 200 300 400 500 600 700 800
Speed (rpm)
Vol
tage
s (V
) A
C &
DC
-10-50510
Exc
iting
cu
rren
t (A
)
Vline (V) Vdc (V) Vrotor (V) Iexc (A)
Load test maintaining 24V on the lines at 600rpm
Iexc (A) Iline (A) Idc (A) Vline (V) Vdc (V) Vrotor (V) Flux Torque
0.05 0.57 0.6 60.9 80.5 -0.159 6.37 1.80.05 0.57 0.6 60.9 80.5 -0.159 6.37 1.8
-5.04 0.72 1 24.4 30.9 20.2 2.77 1.4-4.6 1.47 2 24.3 30 18.67 3.03 1.7
-4.21 2.2 3 24 29.2 17.17 3.23 2.5-3.71 2.94 4 24.6 30.1 15.3 3.53 3.2-3.28 3.77 5 24.1 29.2 13.61 3.75 3.7-2.67 4.51 6 25 30.3 11.15 4.07 4.2-2.38 5.05 7 24.5 28.6 9.41 4.14 5-1.53 5.96 8 24.5 28.8 7.02 4.39 5.4-1.12 6.48 9 24.3 29.5 4.71 4.75 6.6-0.08 7.69 10 24.1 29.1 0.518 5.28 7.2
Excitation IV curves at 24V V-line 600rpm
0
10
20
30
40
50
60
70
80
0 2 4 6 8 10 12
DC Load current (A)
Vol
tage
s (V
) AC
& D
C
-6-4-2024681012
Cur
rent
s (A
)
Vline (V)
Vdc (V)
Vrotor (V)
Iline (A)
Idc (A)
Iexc (A)
171
Load fixed at Idc = 5.8A & I line 4A and also speed 750rpm V ex Vdc Vline Iex bbd Iex Torque
5.32 64.7 51.3 -1.05 -1.05 4.98.74 52.9 42.3 -1.99 -2 4.8
12.33 40.4 32.7 -3.01 -3.02 4.415.85 22.2 18.6 -3.96 -3.98 319.8 1.89 2.97 -4.97 -5 1.54.59 88.7 69.3 1 1.01 7.98.22 101.4 78.7 1.99 2 8.911.9 112.4 87.4 2.99 3 9.5
15.66 124.2 95.9 3.97 4 10.419.47 136.8 105.2 4.97 5 10.6
Load test at 5.8A and 750rpm
0255075
100125150
-24 -18 -12 -6 0 6 12 18 24
Exciting voltage (V)
Vol
tage
s (V
) A
C &
DC
-10
-5
0
5
10
15
Exc
iting
C
urre
nt (A
)
Vdc Vline Iex Torque
Open circuit test at different speeds without excitation
Speed V DC Back EMF V AC Torque Flux
149 22.1 0.506 16.82 0.7 0.28 200 29.3 0.536 22.37 0.7 0.28 250 36.8 0.569 28.02 0.8 0.28 264 38.5 0.577 29.38 0.8 0.28 300 43.8 0.606 33.33 0.8 0.28 350 51.1 0.651 38.82 0.9 0.28 374 54.7 0.671 40.5 0.9 0.28 400 58.6 0.687 43.5 0.9 0.28 450 66.1 0.734 49 0.9 0.28 500 73.6 0.776 54.6 0.9 0.28 550 80.6 0.815 59.8 0.9 0.28 598 87.4 0.856 64.9 0.9 0.28
172
650 95.2 0.904 70.6 0.99 0.28 700 103.1 0.953 76.4 0.99 0.28 750 110.5 1 82 1 0.28
Speed VS open circuit Voltages
0
20
40
60
80
100
120
100 300 500 700 900Speed (rpm)
Sta
tor v
olta
ges
(V)
V
AC
& V
DC
0
0.2
0.4
0.6
0.8
1
1.2
Bac
k em
f (V
)
V DC V AC Back EMF
Open circuit test maintaining 42Vline for 48V battery bank
Speed Iexc Vrotor Vline Vdc Torque flux
149 5.53 23.1 25.4 34.8 0.8 5.79 200 5.44 22.6 33.9 46.1 0.9 7.81 250 5.33 22.4 42.1 57.1 0.9 9.31 264 4.51 19.04 42.2 57.3 0.9 9.08 300 2.93 12.71 42.4 57.7 0.91 8.42 350 1.14 5.72 42.4 57.7 0.9 7.69 374 0.42 2.66 42.1 57.1 0.9 7.28 400 -0.18 2.01 42.1 56.9 0.86 7.02 450 -1.16 5.72 42 56.8 0.85 6.56 500 -1.95 8.76 42 56.8 0.81 6.10 550 -2.58 11.12 42.1 57 0.8 5.77 600 -3.07 12.95 42.1 57 0.85 5.46 650 -3.5 14.59 42.2 57.2 0.78 5.13 700 -3.92 16.11 42.1 57.1 0.8 4.84 750 -4.23 17.28 42.1 57 0.81 4.64 800 -4.51 18.42 42.2 57.2 0.75 4.47 850 -4.78 19.47 42.1 57.2 0.8 4.21 900 -5.02 20.5 42 57 0.81 4.05 950 -5.2 21.2 42.2 57.4 0.8 3.88
1000 -5.4 21.9 42 57 0.8 3.68 1050 -5.4 21.8 44.6 60.7 0.8 3.73 1100 -5.4 21.7 46.7 63.6 0.8 3.79 1150 -5.4 21.7 48.9 66.5 0.7 3.76 1200 -5.4 21.7 51 69.5 0.75 3.80
173
Open circuit test maintaining 42V Vline for 48V battery bank
01020304050607080
0 200 400 600 800 1000 1200 1400
Speed (rpm)
Indu
ced
volta
ges
(Vro
tor,
Vlin
e &
Vdc
)
-8
-3
2
7
Iexc
(A),
flux(
mT)
&
torq
ue(N
m)
Vrotor
Vline
Vdc
Torque
flux
Iexc
Open circuit test at different speeds without excitation Speed Back emf Vdc Vline Flux Torque
149 0.196 21.5 15.85 2.90 0.7200 0.254 28.8 21.5 3.83 0.78250 0.31 35.7 26.4 4.87 0.85264 0.324 37.8 28.1 5.06 0.9300 0.367 43.1 31.9 5.53 0.85350 0.423 50.6 37.2 6.10 0.9374 0.451 54.1 39.8 6.42 0.8400 0.481 57.9 42.8 6.66 0.9450 0.533 64.9 47.9 7.04 0.9500 0.59 72.3 53.3 7.37 0.9550 0.647 79.7 58.7 7.81 0.9598 0.699 86.7 63.8 8.05 0.99650 0.762 95.1 70 8.33 0.91700 0.814 102.1 75.1 8.54 1750 0.867 109.1 80.2 8.67 1
174
Generator open-circuit voltage magnitudes as per test at 750rpm
over a range of tests Test NO: V-DC V-Line % Drop
1 117.5 88 0.002 115 86.14 2.113 113.65 85.14 3.254 110 82.44 6.325 109.1 81.77 7.086 108.4 81.25 7.67
O/C votage characteristics over a range of tests (750rpm)
70
80
90
100
110
120
0 2 4 6 8
Test NO:
Vol
tage
(AC
& D
C) i
n V
olts
V-DC V-Line
Synchronous line current error chart Line current error comparison chart Ir-Iy Iy-Ib Ib-Ir
0.01 0.05 0.06 0.2 0.21 0.01
0.16 0.17 0.01 0.18 0.12 0.06 0.11 0.04 0.07 0.08 0.05 0.03 0.07 0.03 0.04 0.01 0.03 0.04 0.01 0.02 0.01 0.01 0 0.01 0.02 0.02 0.04
Maximum percentage error is 3.3%
175
Iexc V12 V23 V31 Vline Vrotor Ia Ib Ic IR IY IB Flux (mT) P (kW) S (kVA) Pf T (Nm) 5 86 87 87 87.7 20.3 7 7 7 7.21 7.21 7.28 10.75 -1 1 0.98 15.7
4.5 86 86 88 86.9 19 7 7 7 6.95 6.94 7.05 10.35 -1 1 0.98 15.54 84 84 86 86.3 16.55 7 6 7 6.92 6.91 6.98 10.03 -1 1 0.99 15.453 84 83 85 85.4 12.65 7 6 7 6.73 6.71 6.73 9.45 0 1 1 15.32 83 82 84 83.6 8.53 7 6 7 6.83 6.78 6.89 8.79 -1 1 -0.98 151 80 81 82 82.3 4.54 7 7 7 7.16 7.08 7.23 8.19 -1 1 -0.93 14.90 79 80 81 80.9 0.26 8 8 8 7.81 7.71 7.98 7.57 -1 1 -0.87 14.8
-1 77 77 78 79.2 4.7 8 8 8 8.51 8.32 8.56 6.66 -1 1 -0.78 14.7-2 76 77 78 76.9 8.71 9 9 9 9.76 9.54 9.76 5.45 -1 1 -0.67 14.5
-5 74 74 75 75.8 20.5 8 8 8 8.15 8.16 8.21 4.19 0 1 -0.49 8.1-4 76 76 78 78.1 16.7 6 6 6 6.26 6.06 6.27 4.69 0 1 -0.57 8-3 78 77 78 79.2 12.68 6 6 6 5.84 5.68 5.85 5.32 0 1 -0.66 8-2 78 78 79 79.9 8.51 6 5 5 5.45 5.27 5.39 5.95 0 1 -0.76 8.2-1 79 79 81 81.2 4.35 5 4 4 4.62 4.51 4.55 6.47 0 1 -0.84 8.10 81 80 82 82 0.43 4 4 4 4.23 4.15 4.2 6.82 0 1 -0.92 8.11 81 82 83 83.2 4.57 4 4 4 4.01 3.94 3.97 7.26 0 1 -0.99 8.12 83 83 84 84.2 8.52 4 4 4 3.94 3.93 3.9 7.7 0 1 1 8.13 84 85 85 85.7 12.51 4 4 4 4.14 4.13 4.15 8.15 0 1 0.95 8.14 85 85 86 86.7 16.31 4 4 4 4.39 4.38 4.38 8.49 0 1 0.84 8.25 86 87 87 87.8 20.7 5 5 5 4.92 4.94 4.96 9.04 0 1 0.73 8
Synchronous load test
176
Appendix C
Tests boards-data
Line voltage harmonics
Harmonic magnitude as a % of the fundamental amplitude2 4 6 8 10 12 14 16 18 20 22 24 26 28 30 32 34 36 38 40 42 44 46 48 50
0.0%
0.4%
0.8%
1.2%
1.7%
2.1%
2.5%
2.9%
3.3%
3.7%
4.2%
Line voltage harmonic chart at 598rpm.
Harmonic magnitude as a % of the fundamental amplitude2 4 6 8 10 12 14 16 18 20 22 24 26 28 30 32 34 36 38 40 42 44 46 48 50
0.00%
0.08%
0.16%
0.24%
0.31%
0.39%
0.47%
0.55%
0.63%
0.71%
0.78%
Line current harmonic chart at 598rpm.
177
Load current influence on ripple voltage at zero
exciting current
T1 >1 >1 >1 >
1) Ch 1: 10 Volt 1 ms RPL75000 Ripple waveform at 0A load current and at 0A exciting current when speed is 750rpm.
T1 >1 >1 >1 >
1) Ch 1: 10 Volt 1 ms RPL75010 Ripple waveform at 1A load current and at 0A exciting current when speed is 750rpm.
T1 >1 >1 >1 >
1) Ch 1: 10 Volt 1 ms RPL75020 Ripple waveform at 2A load current and at 0A exciting current when speed is 750rpm.
T1 >1 >1 >1 >
1) Ch 1: 10 Volt 1 ms RPL75030 Ripple waveform at 3A load current and at 0A exciting current when speed is 750rpm.
178
T1 >1 >1 >1 >
1) Ch 1: 10 Volt 1 ms RPL75040 Ripple waveform at 4A load current and at 0A exciting current when speed is 750rpm.
T1 >1 >1 >1 >
1) Ch 1: 10 Volt 1 ms RPL75050 Ripple waveform at 5A load current and at 0A exciting current when speed is 750rpm.
T1 >1 >1 >1 >
1) Ch 1: 10 Volt 1 ms RPL75090 Ripple waveform at 9A load current and at 0A exciting current when speed is 750rpm. From waveform RPL75000 up to waveform RPL75090, it is clear that the ripple voltage decreases with the increasing load current. Exciting current influence on ripple voltage at 2A load current when the generator speed is maintained at 750 rpm
T1 >1 >1 >1 >
1) Ch 1: 10 Volt 2.5 ms RPL75021 Ripple waveform at 2A load current and at 1A exciting current when speed is 750rpm.
179
T1 >1 >1 >1 >
1) Ch 1: 10 Volt 2.5 ms RPL75022 Ripple waveform at 2A load current and at 2A exciting current when speed is 750rpm.
T1 >1 >1 >1 >
1) Ch 1: 10 Volt 2.5 ms RPL75023 Ripple waveform at 2A load current and at 3A exciting current when speed is 750rpm.
T1 >1 >1 >1 >
1) Ch 1: 10 Volt 2.5 ms RPL75024 Ripple waveform at 2A load current and at 4A exciting current when speed is 750rpm.
T1 >1 >1 >1 >
1) Ch 1: 10 Volt 2.5 ms RPL75025 Ripple waveform at 2A load current and at 5A exciting current when speed is 750rpm. With reference to waveform RPL75021 up to waveform RPL75025, it is clear that the ripple voltage decreases with the increasing exciting current.
T
T
1 >1 >1 >1 >
2 >2 >2 >2 >
1) Ch 1: 50 Volt 5 ms 2) Ch 2: 10 mVolt 5 ms
Generated voltage waveform (CH1) and current waveform (CH2) with unity star load.
T
2 >2 >2 >2 >
2) Ch 2: 1 Volt 5 ms
WL2750NX Line current 2A with 6-pulse bridge
180
T
2 >2 >2 >2 >
2) Ch 2: 1 Volt 5 ms
WL5750NX Line current 5A with 6-pulse bridge
T
T
1 >1 >1 >1 >
2 >2 >2 >2 >
1) Ch 1: 2 Volt 10 ms 2) Ch 2: 2 Volt 10 ms
Generator speed pulses waveforms Rotor excitation control waveforms
T
T
1 >1 >1 >1 >
2 >2 >2 >2 >
1) Ch 1: 5 Volt 250 us 2) Ch 2: 2 Volt 250 us
Ramp and PWM waveforms
TT
TT
1 >1 >1 >1 >
2 >2 >2 >2 >
1) Ch 1: 5 Volt 2.5 us 2) Ch 2: 5 Volt 2.5 us
DC/DC converter no load drive signals
T
T
1 >1 >1 >1 >
2 >2 >2 >2 >1) Ch 1: 5 Volt 2.5 us 2) Ch 2: 5 Volt 2.5 us
DC/DC converter on load drive signals
T
T
1 >1 >1 >1 >
2 >2 >2 >2 >
1) Ch 1: 20 Volt 2.5 us 2) Ch 2: 20 Volt 2.5 us
DC/DC converter transformer primary waveforms
181
T
T
1 >1 >1 >1 >
2 >2 >2 >2 >1) Ch 1: 5 Volt 2.5 us 2) Ch 2: 5 Volt 2.5 us
DC/DC converter’s miss matched MOSFETs drive signals
T
T
1 >1 >1 >1 >
2 >2 >2 >2 >
1) Ch 1: 20 Volt 2.5 us 2) Ch 2: 20 Volt 2.5 us
Miss matched MOSFETs effect on DC/DC transformer primary waveforms
T
T
1 >1 >1 >1 >
2 >2 >2 >2 >1) Ch 1: 5 Volt 2.5 us 2) Ch 2: 5 Volt 2.5 us
Escalated MOSFETs miss match effect on the converter drive signals.
T
T
1 >1 >1 >1 >
2 >2 >2 >2 >
1) Ch 1: 20 Volt 2.5 us 2) Ch 2: 20 Volt 2.5 us
Escalated miss matched MOSFETs effect on DC/DC transformer primary waveform
T1 >1 >1 >1 >
1) Ch 1: 20 Volt 2.5 us DC/DC transformer secondary waveform for matched MOSFETs Primary load Secondary load 20V 2A 39.2V 0.83A 20V 2.21A 39.2V 1A 20V 11.93A 38.81 5A
T
1 >1 >1 >1 >
1) Ch 1: 20 Volt 2.5 us Converter 40V DC voltage waveform
182
Rotor back emf waveforms without and with freewheeling diode and zerner diode
T
1 >1 >1 >1 >
1) Ch 1: 10 Volt 2 ms
RNX75020 Open circuit back emf at 750rpm, 2A load current and 0A exciting current
T
1 >1 >1 >1 >
1) Ch 1: 10 Volt 2 ms
RNX75030 Open circuit back emf at 750rpm, 3A load current and 0A exciting current
T
1 >1 >1 >1 >
1) Ch 1: 10 Volt 2 ms
RNX75040 Open circuit back emf at 750rpm, 4A load current and 0A exciting current
TT1 >1 >1 >1 >
1) Ch 1: 10 Volt 2.5 ms RWX75010 The waveform RWX75010 is of the induced back EMF when a freewheeling diode is connected across the rotor terminals. The negative pulses of the EMF are clipped off with a freewheeling diode. This is because every pulse forward biases the diode. During the clipping period, the diode current flows in to the rotor. It flows for as long as the excitation source voltage is less than the back EMF. The sharp positive pulses reverse bias both the freewheeling diode and the blocking diode. Although the positive pulses are blocked by the two diodes, their magnitudes still need to be limited to magnitudes below freewheeling diode avalanche voltage Vdav. Thus, zener-clipping diode is introduced by connecting a zener diode across the free wheeling diode. The zener voltage Vz magnitude has to be greater than that of the excitation voltage Vex to avoid zener continous conduction as it increases losses on board. That is, Vex < Vz < Vdav (diode avalanche voltage).
183
TT1 >1 >1 >1 >
1) Ch 1: 10 Volt 2.5 ms RWX75020 Back emf at 750rpm, 2A load current, 0A exciting current and with freewheeling diode.
TT1 >1 >1 >1 >
1) Ch 1: 10 Volt 2.5 ms RWX75030 Back emf at 750rpm, 2A load current, 0A exciting current and with freewheeling diode.
TT1 >1 >1 >1 >
1) Ch 1: 10 Volt 2.5 ms RWX75040 Back emf at 750rpm, 4A load current, 0A exciting current and with freewheeling diode.
T
1 >1 >1 >1 >
1) Ch 1: 5 Volt 1 ms ONZ75020 No exciting current waveform with freewheeling diode and no zerner diode at 750rpm, 2A load current
T1 >1 >1 >1 >
1) Ch 1: 5 Volt 1 ms OWZ75020 Back emf at 750rpm, 2A load current, 0A exciting current and with freewheeling diode and zerner diode
T1 >1 >1 >1 >
1) Ch 1: 5 Volt 1 ms ONZ75040 No exciting current waveform with freewheeling diode and no zerner diode at 750rpm, 4A load current
184
T1 >1 >1 >1 >
1) Ch 1: 5 Volt 1 ms OWZ75040 Back emf at 750rpm, 4A load current, 0A exciting current and with freewheeling diode and zerner diode
185
Switched exciting voltage and current waveforms
T1 >1 >1 >1 >
1) Ch 1: 20 Volt 2.5 ms LWX75051 Exciting voltage at 750rpm, 5A load current and 1A exciting current
T
T1 >1 >1 >1 >
2 >2 >2 >2 >
1) Ch 1: 100 mVolt 250 us 2) Ch 2: 5 Volt 250 us
FGPWM Exciting current ripple at 30% duty cycle
186
Appendix D
Schematics test results
187
188
189
190
191
192
193
Figure D.1 ULN2803 representative schematic diagram
Table D.1
Ferrite geometries offer a wide selection in shapes and sizes. When choosing a core for power applications, parameters shown in Table D.1 should be evaluated.
Figure D.2 shows the reduction in flux levels for MAGNETICS “P” ferrite material necessary to maintain constant 100mW/cm³ core losses at various frequencies, with a maximum temperature rise of 25°C.
194
Figure D.2
195
196
Flexible PVC Insulated Stranded Copper Panel Wire.
Conductor Cross Section Current Rating
0.2 mm² 0.5 Amperes
0.5 mm² 3 Amperes
0.75 mm² 6 Amperes
1.0 mm² 10 Amperes
1.5 mm² 16 Amperes
2.5 mm² 25 Amperes
4.0 mm² 32 Amperes
6.0 mm² 41 Amperes
10.0 mm² 55 Amperes
Manufactured to SABS 1574 1992 Last Updated on 10th of February, 2000
197
V-BE V- CE
0 15.030.001 15.030.05 15.030.1 15.03
0.15 15.030.2 15.01
0.25 15.010.3 15.01
0.35 15.010.4 15.01
0.45 150.5 14.99
0.55 14.970.6 14.79
0.65 13.530.7 4.41
0.75 0.090.8 0.06
0.85 0.060.9 0.06
0.95 0.061 0.06
1.05 0.061.1 0.06
1K 1K
2K
+15V
VV
2N3904 Switching voltage test
0.01
0.1
1
10
100
0.01 0.11 0.21 0.31 0.41 0.51 0.61 0.71 0.81 0.91
V BE (V)
V C
E (V
)
198
Opto Isolator 4N25 (IV)F Curves Test I fwd V fwd Step mA V
12 0 0.82 12 0.02 0.91
11.98 0.1 0.97 11.38 0.3 1.02 11.24 0.5 1.05 11.09 0.7 1.06 10.94 0.9 1.07 10.76 1.1 1.08 10.6 1.3 1.085 10.4 1.5 1.09
10.19 1.7 1.096 10 1.9 1.1
9.81 2.1 1.104 9.61 2.3 1.108 9.39 2.5 1.111 9.18 2.7 1.114 8.95 2.9 1.117 8.7 3.1 1.119
8.45 3.3 1.122 8.3 3.5 1.124
8.08 3.7 1.126 7.81 3.9 1.128 7.62 4.1 1.131 7.41 4.3 1.132 7.18 4.5 1.134 6.92 4.7 1.136 6.76 4.9 1.138 6.5 5.1 1.139
6.31 5.3 1.141 6.06 5.5 1.142 5.86 5.7 1.144 5.66 5.9 1.145 5.44 6.1 1.146 5.26 6.3 1 5.06 6.5 1.149 4.84 6.7 1.15 4.66 6.9 1.152 4.5 7.1 1.153 4.4 7.3 1.154
4.21 7.5 1.155 3.85 7.7 1.156 3.74 7.9 1.158
3.52 8.1 1.159 3.44 8.3 1.16 3.28 8.5 1.161 3.15 8.7 1.162 2.97 8.9 1.163 2.88 9.1 1.164
4N25 I fwd Vc Curve
02468
101214
0 1 2 3 4 5 6 7 8 9 10
I fwd
Vc
(V)
I fwd mA
4N25 IV fwd Curve
0.85
0.9
0.95
1
1.05
1.1
1.15
1.2
0 1 2 3 4 5 6 7 8 9 10
I fwd
V fw
d
V fwd V
199
I ac Vo
0.085 1.60.1 1.650.5 1.67
1 1.681.5 1.72
2 1.772.5 1.82
3 1.883.5 1.94
4 1.994.5 2
5 2
AD736 IV Curves
1.5
1.6
1.7
1.8
1.9
2
2.1
0 2 4 6
Iac in (mA)
Vdc
out
(V)
V o @ 10Hz
I-line sen-voltage (mV)
0.01 11.30.51 141.03 17.51.51 21.32.02 25.42.5 28.8
3.04 33.83.54 37.64.07 41.64.51 44.75.01 48.85.44 51.65.96 55.66.51 59.67.04 63.77.42 66.58.01 71.18.59 75.5
Line current sensor characteristic curve
01020304050607080
0 1 2 3 4 5 6 7 8 9 10
Line current (A)
Vol
tage
(mV
)
sen-voltage (mV)
200
Vgs (v) Vds (v)
0 29.81 1 29.8 3 28.94
3.1 27.86 3.2 22.87 3.3 10.8
3.31 10.41 3.32 7.9 3.34 4.64 3.35 3.69 3.36 1.007 3.38 0.231 3.42 0.0537 3.43 0.0476 3.45 0.0354 3.47 0.028 3.48 0.025 3.5 0.021
3.54 0.014 3.58 0.0114 3.6 0.0099
3.68 0.0062 3.69 0.006 3.82 0.0037 3.9 0.0029
3.96 0.0024 4.04 0.002 4.14 0.0018 4.2 0.0016 4.4 0.0014 4.5 0.0013 4.7 0.0012 4.8 0.0011 5.5 0.001 6.3 0.0009
7 0.0009 8 0.0009 9 0.0008
10 0.0008 11 0.0008 12 0.0008 13 0.0008 14 0.0008 15 0.0008
IRF9640 Vgs vs Vds
0.0001
0.001
0.01
0.1
1
10
100
0 1 2 3 4 5 6 7 8 9 10 11 12
Vgs (v)
Vds
(v)
Vds
Vgs
Vin Vo
201
Opto Isolator 4N25 VI Curve I fwd Vc V rc IC MA V V mA
0.7 9.15 0.15 0.381 9.07 0.23 0.59
1.3 8.95 0.35 0.901.6 8.81 0.49 1.261.9 8.67 0.63 1.622.2 8.49 0.81 2.082.5 8.35 0.95 2.442.8 8.21 1.09 2.793.1 8.06 1.24 3.183.4 7.91 1.39 3.563.7 7.75 1.55 3.97
4 7.54 1.76 4.514.3 7.38 1.92 4.924.6 7.24 2.06 5.284.9 7.08 2.22 5.695.2 6.91 2.39 6.135.5 6.74 2.56 6.565.8 6.62 2.68 6.876.1 6.45 2.85 7.316.4 6.3 3 7.696.7 6.14 3.16 8.10
7 6.01 3.29 8.447.3 5.85 3.45 8.857.6 5.72 3.58 9.187.9 5.56 3.74 9.598.2 5.46 3.84 9.858.5 5.32 3.98 10.218.8 5.16 4.14 10.629.1 5.07 4.23 10.859.4 4.94 4.36 11.189.7 4.86 4.44 11.3810 4.77 4.53 11.62
10.3 4.63 4.67 11.9710.6 4.56 4.74 12.1510.9 4.51 4.79 12.2811.2 4.45 4.85 12.4411.5 4.4 4.9 12.5611.8 4.36 4.94 12.6712.1 4.31 4.99 12.7912.4 4.27 5.03 12.9012.7 4.23 5.07 13.00
13 4.2 5.1 13.0813.3 4.15 5.15 13.2113.6 4.12 5.18 13.2813.9 4.08 5.22 13.38
14.2 4.05 5.25 13.4614.5 4.01 5.29 13.5614.8 3.97 5.33 13.6715.1 3.94 5.36 13.7415.4 3.91 5.39 13.8215.7 3.88 5.42 13.90
16 3.85 5.45 13.97
Opto IV curves
0
2
4
6
8
10
12
14
16
0 5 10 15 20
I fwd (mA)
V (V
)
Vc I fwd Ic
+
_
0V
15V
V
+ 4.5V
- 4.5V
A
V
1K
10K
10K
1K
202
Freq Voltage Hz V
5 310 315 320 325 330 335 2.940 2.745 2.650 2.555 2.360 2.265 2.0570 1.975 1.780 1.685 1.590 1.395 1.25
100 1.2105 1.1110 1.05120 0.9130 0.8140 0.7150 0.6160 0.5170 0.45180 0.42190 0.4200 0.35
Low Pass Filter VF Curves
0
0.5
1
1.5
2
2.5
3
3.5
0 30 60 90 120 150 180 210Frequency (Hz)
Vol
tage
(V)
Voltage
203
BB 100 hall effect current sensor `+/- 7.5V supplied V-out I-rot
0.0362 0.18 0.0405 0.2 0.0585 0.3 0.0782 0.4 0.0956 0.5 0.1873 1 0.2845 1.5 0.3732 2 0.4646 2.5 0.551 3 0.639 3.5 0.742 4 0.816 4.5 0.923 5
BB-100 current test
00.10.20.30.40.50.60.70.80.9
1
0 1 2 3 4 5 6
Load current (A)
V-o
ut (V
)
BB-100 in circuit test V-out I-load
0.544 0.180.545 0.20.57 0.4
0.596 0.60.619 0.80.646 10.67 1.2
0.696 1.40.723 1.60.74 1.74
0.779 2.020.809 2.250.839 2.450.886 2.780.951 3.210.979 3.421.036 3.811.105 4.311.15 4.56
1.196 4.861.206 5
BB-100 circuit test
0.5
0.6
0.7
0.8
0.9
1
1.1
1.2
1.3
0 2 4 6
Load current (A)
V-o
ut (V
)
204
Appendix E
Project Test boards and schematics
Analog signals test board
I/O interface test board
205
Power test board
Power converter test board
206
Analog signals test board schematic diagram
1 2 3 4 5 6 7 8
A
B
C
D
87654321
D
C
B
A
Title
Number RevisionSize
A3
Date: 17-Nov-2003 Sheet of File: D:\Proschm&pcb\resaerch_A.ddb Drawn By:
CTL5
GN
D1
OUT 3
RST4
THR6
TRIG2 DISC 7
U1
R1
R2
Vin1
GN
D2
Vout 3RG1
Vin1 GN
D2
Vout 3
RG2
C1
C2 C3
C4
+B
-B -10
GND
VCC
-B
GND
R3
R4
R5
VR1
Q1
C5
C6
VCC
GND
GND
VCC
VR2
R6
3
21
411
U2A
5
67
U2B
10
98
U2C
12
1314
U2D
VCC
-10R7
R8
U5
Q2
GNDR9
VR3
VCC
GND
R22
VR4
VR5
R10
R11
R12
R13 R14
R15
R16
R17
R18
R19
R20
R21 3
21
411
U3A
5
67
U3B
10
98
U3C
12
1314
U4D
3
21
411
U4A
5
67
U4B
GND
GNDGND
-10
VCC
VCC
-10
LK
LK
-B
+15V
VgP
R25
R26
R27
R28
R29
R30
R31
R32 R33
VR6
3
21
411
U6A
5
67
U6B
10
98
U6C
12
1314
U6D
C10
C9
R34
R35
R36
R37R38
VR7
COM
COM +4.5 COM
+4.5
-4.5
+BL
+RL
U7
+5V
-B
PLC
R23
R24
C7
C8
+4.5
-4.5
COM
3
26
15
74 U8
3
26
15
74 U10
3
26
15
74 U12
C11
C12
C13
C14
C15
C16
COM
COM
COM
+4.5
+4.5
+4.5
+4.5
+4.5
+4.5
-4.5
-4.5
-4.5
-4.5
-4.5
-4.5
1234 5
678U9
1234 5
678U11
1234 5
678U13
+RL
-RL
+YL
-YL
+BL
-BL
R39R40
R41
R42
R43
R44
VR8 VR9
3
21
411
U14A5
67
U14B
10
98
U14C
C17
C18+4.5
-4.5COMCOMU15
COM
R45
R46 R47
R48
R49
3
21
411
U16A
5
67
U16B
10
98
U16C
12
1314
U16D
-B
+15V
VR10+15V
-B
IR
-B
+PWM
-PWM
R50R51
R52
R53
R54
R55
VR11 VR12
3
21
411
U17A5
67
U17B
10
98
U17C
C19
C20
+4.5
-4.5COMCOMU18
COM
R56
R57 R58
R59
VR13+15V
-B
IY
R61R62
R63
R64
R65
R66
VR14 VR15
3
21
411
U19A5
67
U19B
10
98
U19C
C21
C22
+4.5
-4.5COMCOMU20
COM
R67
R68 R69
R70
VR16+15V
-B
IB
R60
R71
-B
-B
U21-R
R73
R72 R74
R75
VR17+15V
-B
VRt
R76 -B
VRti
+Vcc 1-Vcc 2GND 3
GND 4Vo 5HE
OR1
OR2
OR3
3
26
15
74 OU1
3
26
15
74
OU21234 5
678OU3
-R
+9VVRti
-R
+9V
-B
+15V
VRt
3
26
1 5
74
U22R77
R78
VR18
C23
IRt
+15V
+15V-B
-B
-B
-B
123456
CON1
123456
CON2
123456
CON4
+B
-B
+15V+5V
+9V-R
+4.5-4.5
+RL-RL+YL-YL+BL-BL
VgP
S1S2S3S4+PWM-PWM
1 23 45 67 89 10
HD1
1 23 45 67 89 10
HD2
IRt VRtIR IYIB -B
PLC
S1S2S3S4-B
VRti
123456
CON3
Control Analog Signals Board
C.T. 'Moleli
-R
1 2
J2
R8a
D1
D2
D3
DZ2
DZ3
1
2
J1
12
OJ
1 2
OJ1
DZ1
S1a+BS_+B
PWM Generator
Line current sensors
Generator speed sensor
External component
-10V
VoVo
207
I/O interface test board schematic
1 2 3 4 5 6 7 8
A
B
C
D
87654321
D
C
B
A
Title
Number RevisionSize
A3
Date: 25-Oct-2003 Sheet of File: D:\Proschm&pcb\Research.ddb Drawn By:
2
31
411
U1A
5
67
411
U1B
9
108
411
U1C
1413
12
411
U1D
2
31
411
U2A
5
67
411
U2B
9
108
411
U2C
1413
12
411
U2D
2
31
411
U3A
5
67
411
U3B
9
108
411
U3C
1413
12
411
U3D
R1
R2
R3
R4
R5
R6
R7
R8
R9
R10
R11
R12
R13
R14
R15
R16
R17
R18
VR1
VR2
VR3
-B
+15V
-Vg
+Vg
-IRg
+IRg
-VB
+VB
a1
a3
a4
+5Vd
+5Vd
+5Vd
+5Vd +5Vd
+5Vd +5Vd
+5Vd
-Bd
-Bd
-Bd -Bd
-Bd -Bd
-Bd
-Bd
-B
-B
+15V
+15V
A
C
D
E
F G
H
BIRt VRt
IR IY
IB
1
2
3
4
5
Ain1 2
3 4
5
Ain
Ain
Ain
1234
CON1
1234
CON2
1234
CON3
+15V-B+5Vd-Bd
-Vg+Vg-IRg+IRg
-VB+VB
1 23 45 67 89 10
HD1
1 23 45 67 89 10
HD2
IRt VRtIR IYIB
-BPLC
a1a IRta3 a4VRt IRIY IB
S1S2S3S4S5S6S7S8
O1O2O3O4O5O6O7O8
2
36
74
8
DIS U4
2
36
74
8
DIS U5
2
36
74
8DIS U6
2
36
74
8DIS
U72
36
74
8DIS
U8
2
36
74
8DIS
U92
36
74
8DIS
U10
2
36
74
8DIS
U11
a1a
1 23 45 67 89 1011 1213 1415 1617 1819 20
HD6
A1
B2
C3
G16
G2A4
G2B5
Y0 15
Y1 14
Y2 13
Y3 12
Y4 11
Y5 10
Y6 9
Y7 7
U12
CLR1
LOAD9
ENT10
ENP7
CLK2
RCO 15
A3
QA 14
B4
QB 13
C5
QC 12
D6
QD 11
U13
CLR1
LOAD9
ENT10
ENP7
CLK2
RCO 15
A3
QA 14
B4
QB 13
C5
QC 12
D6
QD 11
U14
-B
-B
-B
IN 11
IN 22
IN 33
IN 44
IN 55
IN 66
IN 77
IN 88
DIODE CLAMP10 OUT 8 11OUT 7 12OUT 6 13OUT 5 14OUT 4 15OUT 3 16OUT 2 17OUT 1 18U18
+5Vd
O0O1O2O3O4O5O6O7
ABCDEFGH
S6S7S8
-Bd
R19
R20
R21
R22
R23
R24
R25
R26
R27
R28
R29
R30
R31
R32
R33
R34
-Bd
+5Vd
+5Vd
-Bd
-Bd
Qc0Qc1Qc2Qc3
TC
PLC
S5
S5
TCQc4Qc5Qc6Qc7
-Bd
-Bd
+5Vd
-Bd
C1
C2
C3-Bd
-Bd
S1 S2S3 S4S5 S6S7 S8
O1 O2O3 O4O5 O6O7 O8
VCCGND
1 23 45 67 89 10
HD5S1S2S3S4
Control I/O Interface Board
C.T. 'Moleli
Qc0Qc1Qc2Qc3Qc4Qc5Qc6Qc7
1 23 45 67 89 1011 1213 1415 16
HD3
1 23 45 67 89 1011 1213 1415 16
HD4
1 2
J1
1 2
J2
a1a
D1
D2
D3
D4
D5
D6
D7
D8
When single channel read analog mode is used jumber J2 has to be disconnected and J1 be connected.On the contrary, J2 has to be connected and J1 disconnected when multi channel (8 channel)read analog mode is used.
208
Power test board schematic diagram
1 2 3 4 5 6 7 8
A
B
C
D
87654321
D
C
B
A
Title
Number RevisionSize
A3
Date: 9-Nov-2003 Sheet of File: D:\Proschm&pcb\researchb.ddb Drawn By:
R7
PT1
PT2
R8
R9
Q1
R11
R12
R13
R14
PT3
Tp1a Tp2 Tp3
C4C7
C5C6
+15V
-B
NC1
CL2
CS3
Inv4
Ninv5
Vref6
-V7 NC 8Vz 9Vout 10Vc 11+V 12FC 13NC 14U2
R15
R16
R17
R18
R19 R20
R21
VR1
C8Tp5 Tp4Q3Tp3Tp6
C9 C10
R23
R24
R25
R26
R27
R28
R29
R30
R31
R32
R33
PT4
PT5
PT6
+B
+15V
Q5
Q6
+5V
Tp7 Tp8
-Vg +Vg
VgP
+VB -VB
R34
R35
R36
R37
Q8
Q9
R38
U4 U5
Rly1
+15V
-B
S4
+5V
-BS2
+5V
+5V
+15V
-B
R39
R40
R41
R42
R43
R44
R45 R46
VR2 1234 5
678U7
C11 C13
C15
C17
C19
C12C14
C16
C18
1 3
2
V V
GNDIN OUT
Reg1
1 3
2
V V
GNDIN OUT
Reg2
1 3
2
V V
GNDIN OUT
Reg3
1 3
2
V V
GNDIN OUT
Reg4
Q16
Q10
Q11
Q12
Q13
Q14
Q15
R47
U6
S1 +5V
+12V
+15V
+12V
+5V
Tp9 Tp10
+B
+15V
+12V
-B
-B
-B
C20C22
C21
R48
R49
R50
R59
R60
PT7
Q19
Q22
Q23
Q21
R51
R52R53
R54
C26
C25
U9R61
+PWM-PWM
+9V
-R
-R
S3-B
Rly2
+15V+5V
R55
R56
R57
R58
R62
R63
R64
C23
C24
VR3
3
21
411
U8A
5
67
U8B
10
98
U8C
12
1314
U8D
+9V
-R
VRti
LK
LK
+R Tp11 Tp12
+B
123456
CON1
123456
CON2
123456
CON7
1234
CON6
1234
CON5
Tp1
Tp2Tp3Tp4Tp5Tp6
Tp7Tp8
Tp9Tp10
Tp11Tp12
+B
+15V
+5V
-B
+R-R
+Vg-Vg
+9V
+PWM-PWM
Rly1Rly2
S1
S2S3S4
VgP
+VB
-VB
Control Power Board
C.T. 'Moleli
1
2
J1
VRti
1234
CON3
1234
CON4
+12V
VRx
3
21
411
U1A
5
67
U1B
10
98
U1C
12
1314
U1D
Vref
Vref X
X
R11a
R28a
R32a
Rx1
Rx2
Dx
12
3
U3A
DZ1
DZ2
DZ3
DZ4
D8
D9
D10
D11
D16
D17
D18
Q18
Q17
Q2 Q7
Q20
Q4
D7
R22
R10
Tp9
T10
dQ18
dQ17
-B
F1
S1
F2
FQ17
FQ18
Tp1
Tp1a
Tp8
+Load
All dotted line enclosed componets are external to the control power board
R22a
209
Power converter test board schematic
1 2 3 4
A
B
C
D
4321
D
C
B
A Title
Number RevisionSize
A4
Date: 25-Oct-2003 Sheet of File: D:\Proschm&pcb\converters.ddb Drawn By:
12
RL
12
YL
12
BL
12
Rly1
12
D7 Con
12
Rot12
Rtpol
12
Rly2
123
Trans
12
Hv1
12
Hv2
R1
R2
R3
R4
R5
R6
Rrly1
Rrly2
D1
D2
D3
D4
D5
D6
D12 D13
D14 D15
D7
Dbd
D16a
L1
L2
HVc1
HVc2C_Trans
Rly2
C1
C2
C3
Tp1
-B
+RL-RL
+YL-YL
+BL
-BL
+15V
Rly1
Q4E
+B
dQ17
dQ18
+B
dQ20
-R
-R
+R
+15vRly2
Rly1 is formed with 3_relays:
Rly1a Rly1b Rly1c
Rly2 is formed with 2_relays:
Rly2a Rly2b
All coils connected in parallel
All coils connected in parallel
Converter Board
C.T. 'Moleli
Rly1
A balanced star load is biult with R4, R5 and R6 as indicated by the schematic.
210
Generator test workstation
Test boards arrangement