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22 High Frequency Electronics

High Frequency Design

RF POWER AMPLIFIERS

R

F and microwavepower amplifiersand transmitters

are used in a wide varietyof applications including wireless communication,

jamming, imaging, radar,and RF heating. Thisarticle provides an intro-duction and historical

background for the subject, and begins thetechnical discussion with material on signals,linearity, efficiency, and RF-power devices. Atthe end, there is a convenient summary of theacronyms usedthis will be provided with allfour installments. Author affiliations and con-tact information are also provided at the endof each part.

1. INTRODUCTIONThe generation of significant power at RF

and microwave frequencies is required notonly in wireless communications, but also inapplications such as jamming, imaging, RFheating, and miniature DC/DC converters.Each application has its own unique require-ments for frequency, bandwidth, load, power,efficiency, linearity, and cost. RF power can begenerated by a wide variety of techniquesusing a wide variety of devices. The basictechniques for RF power amplification viaclasses A, B, C, D, E, and F are reviewed andillustrated by examples from HF through Kaband. Power amplifiers can be combined intotransmitters in a similarly wide variety of architectures, including linear, Kahn, enve-

lope tracking, outphasing, and Doherty.Linearity can be improved through techniquessuch as feedback, feedforward, and predistor-

tion. Also discussed are some recent develop-ments that may find use in the near future.

A power amplifier (PA) is a circuit for con- verting DC input power into a significantamount of RF/microwave output power. Inmost cases, a PA is not just a small-signalamplifier driven into saturation. There existsa great variety of different power amplifiers,and most employ techniques beyond simplelinear amplification.

A transmitter contains one or more poweramplifiers, as well as ancillary circuits such assignal generators, frequency converters, mod-ulators, signal processors, linearizers, andpower supplies. The classic architectureemploys progressively larger PAs to boost alow-level signal to the desired output power.However, a wide variety of different architec-tures in essence disassemble and thenreassemble the signal to permit amplificationwith higher efficiency and linearity.

Modern applications are highly varied.Frequencies from VLF through millimeterwave are used for communication, navigation,and broadcasting. Output powers vary from 10mW in short-range unlicensed wireless sys-tems to 1 MW in long-range broadcast trans-mitters. Almost every conceivable type of mod-ulation is being used in one system or anoth-er. PAs and transmitters also find use in sys-tems such as radar, RF heating, plasmas, laserdrivers, magnetic-resonance imaging, andminiature DC/DC converters.

With this issue, we begin afour-part series of articles

that offer a comprehensiveoverview of power amplifiertechnologies. Part 1 coversthe key topics of amplifier

linearity, efficiency andavailable RF power devices

RF and Microwave Power

Amplifier and TransmitterTechnologies Part 1

By Frederick H. Raab, Peter Asbeck, Steve Cripps, Peter B. Kenington,Zoya B. Popovic, Nick Pothecary, John F. Sevic and Nathan O. Sokal

This series of articles is an expanded version of the paper, Power Amplifiers and Transmitters for RF andMicrowave by the same authors, which appeared in the the 50th anniversary issue of the IEEE Transactions on

Microwave Theory and Techniques , March 2002. 2002 IEEE. Reprinted with permission.

From May 2003 High Frequency ElectronicsCopyright 2003 Summit Technical Media, LLC


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RF POWER AMPLIFIERS

No single technique for poweramplification nor any single trans-mitter architecture is best for allapplications. Many of the basic tech-niques that are now coming into use

were devised decades ago, but haveonly recently been made practicalbecause of advances in RF-powerdevices and supporting circuitry suchas digital signal processing (DSP).

2. HISTORICAL DEVELOPMENTThe development of RF power

amplifiers and transmitters can bedivided into four eras:

Spark, Arc, and AlternatorIn the early days of wireless com-

munication (from 1895 to the mid1920s), RF power was generated byspark, arc, and alternator techniques.The original RF-power device, thespark gap, charges a capacitor to ahigh voltage, usually from the ACmains. A discharge (spark) throughthe gap then rings the capacitor, tun-ing inductor, and antenna, causing radiation of a damped sinusoid.Spark-gap transmitters were rela-tively inexpensive and capable of generating 500 W to 5 kW from LF toMF [1].

The arc transmitter, largelyattributed to Poulsen, was a contem-porary of the spark transmitter. Thearc exhibits a negative-resistancecharacteristic which allows it to oper-ate as a CW oscillator (with somefuzziness). The arc is actually extin-guished and reignited once per RFcycle, aided by a magnetic field andhydrogen ions from alcohol drippedinto the arc chamber. Arc transmit-ters were capable of generating asmuch as 1 MW at LF [2].

The alternator is basically an ACgenerator with a large number of poles. Early RF alternators by Teslaand Fessenden were capable of oper-ation at LF, and a technique devel-oped by Alexanderson extended theoperation to LF [3]. The frequencywas controlled by adjusting the rota-tion speed and up to 200 kW could be

generated by a single alternator. Onesuch transmitter (SAQ) remainsoperable at Grimeton, Sweden.

Vacuum Tubes

With the advent of the DeForestaudion in 1907, the thermoionic vac-uum tube offered a means of elec-tronically generating and controlling RF signals. Tubes such as the RCA UV-204 (1920) allowed the transmis-sion of pure CW signals and facilitat-ed the transition to higher frequen-cies of operation.

Younger readers may find it convenient to think of a vacuum tube asa glass-encapsulated high-voltageFET with heater. Many of the con-

cepts for modern electronics, includ-ing class-A, -B, and -C power ampli-fiers, originated early in the vacuum-tube era. PAs of this era were charac-terized by operation from high volt-ages into high-impedance loads andby tuned output networks. The basiccircuits remained relatively un-changed throughout most of the era.

Vacuum tube transmitters weredominant from the late 1920sthrough the mid 1970s. They remainin use today in some high powerapplications, where they offer a rela-tively inexpensive and rugged meansof generating 10 kW or more of RFpower.

Discrete TransistorsDiscrete solid state RF-power

devices began to appear at the end of the 1960s with the introduction of sil-icon bipolar transistors such as the2N6093 (75 W HF SSB) by RCA.Power MOSFETs for HF and VHFappeared in 1974 with the VMP-4 bySiliconix. GaAs MESFETs introducedin the late 1970s offered solid statepower at the lower microwave fre-quencies.

The introduction of solid-stateRF-power devices brought the use of lower voltages, higher currents, andrelatively low load resistances.Ferrite-loaded transmission linetransformers enabled HF and VHF

PAs to operate over two decades of bandwidth without tuning. Becausesolid-state devices are temperature-sensitive, bias stabilization circuitswere developed for linear PAs. It also

became possible to implement a vari-ety of feedback and control tech-niques through the variety of op-amps and ICs.

Solid-state RF-power deviceswere offered in packaged or chipform. A single package might includea number of small devices. Power out-puts as high as 600 W were availablefrom a single packaged push-pulldevice (MRF157). The designer basi-cally selected the packaged devicethat best fit the requirements. How

the transistors were made wasregarded as a bit of sorcery thatoccurred in the semiconductor housesand was not a great concern to theordinary circuit designer.

Custom/Integrated TransistorsThe late 1980s and 1990s saw a

proliferation variety of new solid-state devices including HEMT,pHEMT, HFET, and HBT, using a

variety of new materials such as InP,SiC, and GaN, and offering amplifica-tion at frequencies to 100 GHz ormore. Many such devices can be oper-ated only from relatively low voltages.However, many current applicationsneed only relatively low power. Thecombination of digital signal process-ing and microprocessor control allowswidespread use of complicated feed-back and predistortion techniques toimprove efficiency and linearity.

Many of the newer RF-powerdevices are available only on a made-to-order basis. Basically, the designerselects a semiconductor process andthen specifies the size (e.g., gateperiphery). This facilitates tailoring the device to a specific power level, aswell as incorporating it into an RFICor MMIC.

3. LINEARITYThe need for linearity is one of the

principal drivers in the design of


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RF POWER AMPLIFIERS

modern power amplifiers. Linearamplification is required when thesignal contains both amplitude andphase modulation. It can be accom-plished either by a chain of linearPAs or a combination of nonlinearPAs. Nonlinearities distort the signalbeing amplified, resulting in splatterinto adjacent channels and errors indetection.

Signals such as CW, FM, classicalFSK, and GMSK (used in GSM) haveconstant envelopes (amplitudes) andtherefore do not require linear ampli-fication. Full-carrier amplitude mod-ulation is best produced by high levelamplitude modulation of the final RFPA. Classic signals that require lin-ear amplification include single side-band (SSB) and vestigal-sideband(NTSC) television. Modern signalsthat require linear amplificationinclude shaped-pulse data modula-tion and multiple carriers.

Shaped Data PulsesClassic FSK and PSK use abruptfrequency or phase transitions, orequivalently rectangular data pulses.The resultant RF signals have con-stant amplitude and can therefore beamplified by nonlinear PAs with goodefficiency. However, the resultantsinc-function spectrum spreads sig-nal energy over a fairly wide band-width. This was satisfactory for rela-

tively low data rates and a relativelyuncrowded spectrum.

Modern digital signals such asQPSK or QAM are typically generat-ed by modulating both I and Q sub-carriers. The requirements for bothhigh data rates and efficient utiliza-tion of the increasingly crowded spec-trum necessitates the use of shapeddata pulses. The most widely usedmethod is based upon a raised-cosinechannel spectrum, which has zerointersymbol interference during detection and can be made arbitrari-ly close to rectangular [4]. A raised-cosine channel spectrum is achievedby using a square-root raised-cosine(SRRC) filter in both the transmitterand receiver. The resultant SRRCdata pulses (Figure 1) are shapedsomewhat like sinc functions whichare truncated after several cycles. Atany given time, several different datapulses contribute to the I and Q mod-ulation waveforms. The resultantmodulated carrier (Figure 2) hassimultaneous amplitude and phasemodulation with a peak-to-averageratio of 3 to 6 dB.

Multiple Carriers and OFDM Applications such as cellular base

stations, satellite repeaters, andmulti-beam active-phased-arraytransmitters require the simultane-ous amplification of multiple signals.

Depending on the application, thesignals can have different ampli-tudes, different modulations, andirregular frequency spacing.

In a number of applicationsincluding HF modems, digital audiobroadcasting, and high-definitiontelevision, it is more convenient touse a large number of carriers withlow data rates than a single carrierwith a high data rate. The motiva-tions include simplification of themodulation/demodulation hardware,equalization, and dealing with multi-path propagation. Such OrthogonalFrequency Division Multiplex(OFDM) techniques [5] employ carri-ers with the same amplitude andmodulation, separated in frequencyso that modulation products from onecarrier are zero at the frequencies of the other carriers.

Regardless of the characteristicsof the individual carriers, the resul-tant composite signal (Figure 2) hassimultaneous amplitude and phasemodulation. The peak-to-averageratio is typically in the range of 8 to13 dB.

NonlinearityNonlinearities cause imperfect

reproduction of the amplified signal,resulting in distortion and splatter.

Amplitude nonlinearity causes theinstantaneous output amplitude or

Figure 1 SRRC data pulses. Figure 2 RF waveforms for SRRC and multicarrier signals.


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RF POWER AMPLIFIERS

envelope to differ in shape from thecorresponding input. Such nonlinear-ities are the variable gain or satura-tion in a transistor or amplifier.

Amplitude-to-phase conversion is aphase shift associated with the signalamplitude and causes the introduc-tion of unwanted phase modulationinto the output signal. Amplitude-to-phase conversion is often associatedwith voltage-dependent capacitancesin the transistors. While imperfectfrequency response also distorts asignal, it is a linear process andtherefore does not generate out-of-band signals.

Amplitude nonlinearity andamplitude-to-phase conversion aredescribed by transfer functions that

act upon the instantaneous signal voltage or envelope. However, memo-ry effects can also occur in high-power PAs because of thermal effectsand charge storage. Thermal effectsare somewhat more noticeable in III-

V semiconductors because of lowerthermal conductivity, while charge-storage effects are more prevalent inoverdriven BJT PAs.

Measurement of LinearityLinearity is characterized, mea-

sured, and specified by various tech-niques depending upon the specificsignal and application. The linearityof RF PAs is typically characterizedby C/I, NPR, ACPR, and EVM(defined below).

The traditionalmeasure of lineari-ty is the carrier-to-in termodula t ion(C/I) ratio. The PA

is driven with twoor more carriers(tones) of equala m p l i t u d e s .N o n l i n e a r i t i e scause the produc-tion of intermodu-lation products atfrequencies corre-sponding to sumsand differences of multiples of thecarrier frequencies

[6]. The amplitude of the third-orderor maximum intermodulation distor-tion (IMD) product is compared tothat of the carriers to obtain the C/I.

A typical linear PA has a C/I of 30 dBor better.

Noise-Power Ratio (NPR) is a tra-ditional method of measuring the lin-earity of PAs for broadband andnoise-like signals. The PA is drivenwith Gaussian noise with a notch inone segment of its spectrum.Nonlinearities cause power to appearin the notch. NPR is the ratio of thenotch power to the total signal power.

Adjacent Channel Power Ratio(ACPR) characterizes how nonlinear-ity affects adjacent channels and iswidely used with modern shaped-pulse digital signals such as NADCand CDMA. Basically, ACPR is theratio of the power in a specified bandoutside the signal bandwidth to therms power in the signal (Figure 3).In some cases, the actual power spec-trum S ( f ) is weighted by the frequen-cy response H ( f ) of the pulse-shaping filter; i.e. (eq. 1)

ACPR

H f S f df

H f S f df lower

f f BW

f f BW

f

f c o

c o

L

U =

( ) ( )

( ) ( )

+

2

2

2

2

/

/

Figure 3 ACPR offsets and bandwidths. Figure 4 Error vector.

Frequency Offset from Carrier (MHz)

EDGE ACPR OffsetsI

I N T E N

D E D V

E C T O

R

E R R O R V E C T O R

A C T U

A L V E

C T O

R

Q

N o r m a l i z e d M a g n i t u d e ( d B )

STANDARD Offset 1 Offset 2 BW Integration EVM(kHz) Filter (peak/rms)

NADC [ 13 ] 30 kHz 60 kHz 32.8 kHz RRC 25%/12%26 dBc 45 dBc =0.35

PHS [ 14 ] 600 kHz 900 kHz 37.5 kHz RRC 25%/12%50 dBc 55 dBc =0.50

EDGE [ 15 ] 400 kHz 600 kHz 30 kHz None 22%/7.0%58 dBc 66 dBc

TETRA [ 16 ] 25 kHz 50 kHz 25 kHz RRC 30%/10%60 dBc 70 dBc =0.35

IS-95 CDMA [17] 885 kHz 1980 kHz 30 kHz None N/A 45 dBc 55 dBcW-CDMA (3G-PP) 5.00 MHz 10.0 MHz 4.68 MHz RRC 25%/N/A [18 ] 33 dB 43 dB =0.22

Table 1 ACPR and EVM requirements of various systems.


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May 2003 29

where f c is the center frequency, B isthe bandwidth, f o is the offset, and f Land f U are the band edges. Theweighting, frequency offsets, andrequired ACPRs vary with applica-tion as shown in Table 1. ACPR canbe specified for either upper or lowersideband. In many cases, two differ-ent ACPRs for two different frequen-cy offsets are specified. ACPR2, basedupon the outer band, is sometimescalled Alternate Channel PowerRatio.

Error Vector Magnitude (EVM) isa convenient measure of how nonlin-earity interferes with the detectionprocess. EVM is defined (Figure 4) asthe distance between the desired andactual signal vectors, normalized to afraction of the signal amplitude.Often, both peak and rms errors arespecified (Table 1).

4. EFFICIENCYEfficiency, like linearity, is a criti-

cal factor in PA design. Three defini-tions of efficiency are commonly used.Drain efficiency is defined as theratio of RF output power to DC inputpower:

= P out / P in (2)

Power-added efficiency (PAE)incorporates the RF drive power bysubtracting it from the output power;

i.e. ( P out P DR )/ P in . PAE gives a rea-sonable indication of PA performancewhen gain is high; however, it canbecome negative for low gains. Anoverall efficiency such as Pout /( P in +

P DR ) is useable in all situations. Thisdefinition can be varied to includedriver DC input power, the powerconsumed by supporting circuits, andanything else of interest.

Average EfficiencyThe instantaneous efficiency is

the efficiency at one specific outputlevel. For most PAs, the instanta-neous efficiency is highest at thepeak output power (PEP) anddecreases as output decreases.Signals with time-varying ampli-tudes (amplitude modulation) there-fore produce time-varying efficien-cies. A useful measure of performanceis then the average efficiency, whichis defined [7] as the ratio of the aver-age output power to the average DC-input power:

AVG = P outAVG / P inAVG (3)

This concept can be used with anyof the three definitions of efficiency.

The probability-density function(PDF) of the envelope gives the rela-tive amount of time an envelopespends at various amplitudes (Figure5). Also used is the cumulative distri-

bution function (CDF), which givesthe probability that the envelopedoes not exceed a specified ampli-tude. CW, FM, and GSM signals haveconstant envelopes and are thereforealways at peak output. SRRC datamodulation produces PDFs that areconcentrated primarily in the upperhalf of the voltage range and havepeak-to-average ratios on the order of 3 to 6 dB. Multiple carriers [8] pro-duce random-phasor sums much likerandom noise and therefore haveRayleigh-distributed envelopes; i.e.,

p ( E ) = 2E exp(V 2 ) (4)

Peak-to-average ratio is typicallybetween 6 and 13 dB.

The average input and outputpowers are found by integrating theproduct of their variation with ampli-tude and the PDF of the envelope.Two cases are of special interest.When the DC input current is con-stant (class-A bias), the DC inputpower is also constant. The averageefficiency is then PEP / . If the DCinput current (hence power) is pro-portional to the envelope (as in class-B), the average efficiency is (4/ )1/2

PEP , for a Rayleigh-distributed sig-nal. Thus for a multicarrier signalwith a 10 dB peak-to-average ratio,ideal class-A and B PAs with PEPefficiencies of 50 and 78.5 percent,

Figure 5 Envelope PDFs. Figure 6 Power-output PDFs.


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RF POWER AMPLIFIERS

respectively, have average efficienciesof only 5 and 28 percent, respectively.

Back-OffThe need to conserve battery

power and to avoid interference toother users operating on the samefrequency necessitates the transmis-sion of signals whose peak ampli-tudes well below the peak outputpower of the transmitter. Since peakpower is needed only in the worst-case links, the back-off is typicallyin the range of 10 to 20 dB.

For a single-carrier mobile trans-mitter, back-off rather than envelopePDF is dominant in determining theaverage power consumption and

average efficiency. The PDF of thetransmitting power (Figure 4)depends not only upon the distance,but also upon factors such as attenu-ation by buildings, multipath, andorientation of the mobile antenna [8],[9], [10]. To facilitate prediction of thepower consumption, the envelope andback-off PDFs can be combined [11].

5. RF POWER TRANSISTORS A wide variety of active devices is

currently available for use in RF-power amplifiers, and RF-powertransistors are available in packaged,die, and grown-to-order forms.Packaged devices are used at fre-quencies up to X band, and are domi-nant for high power and at VHF andlower frequencies. A given packagecan contain one or more die connect-ed in parallel and can also includeinternal matching for a specific fre-quency of operation. Dice (chips) canbe wire-bonded directly into a circuitto minimize the effects of the packageand are used up to 20 GHz. InMMICs, the RF-power device isgrown to order, allowing its size andother characteristics to be optimizedfor the particular application. Thisform of construction is essential forupper-microwave and millimeter-wave frequencies to minimize theeffects of strays and interconnects.

Virtually all RF power transistors

are npn or n-channel types becausethe greater mobility of electrons (vs.holes) results in better operation athigher frequencies.

Bipolar Junction Transistor (BJT)The Si BJT is the original solid-state RF power device, originating inthe 1960s. Since the BJT is a verticaldevice, obtaining a high collector-breakdown voltage is relatively sim-ple and the power density is veryhigh. Si BJTs typically operate from28 V supplies and remain in use atfrequencies up to 5 GHz, especially inhigh-power (1 kW) pulsed applica-tions such as radar. While Si RFpower devices have higher gain at

high frequencies, their fundamentalproperties are basically those of ordi-nary bipolar transistors. The positivetemperature coefficient of BJTs tendsto allow current hogging, hot-spot-ting, and thermal runaway, necessi-tating carefully regulated base bias.Since RF power BJTs are generallycomposed of multiple, small BJTs(emitter sites), emitter ballasting (resistance) is generally employed toforce even division of the currentwithin a given package.

Metal-Oxide-Silicon Field-EffectTransistor (MOSFET)

MOSFETs are constructed withinsulated gates. Topologies with both

vertical and later current flow areused in RF applications, and most areproduced by a double-diffusion pro-cess. Because the insulated gate con-ducts no DC current, MOSFETs are

very easily biased.The negative temperature coeffi-

cient of a MOSFET causes its draincurrent to decrease with tempera-ture. This prevents thermal runawayand allows multiple MOSFETs to beconnected in parallel without ballast-ing. The absence of base-charge stor-age time allows fast switching andalso eliminates a mechanism for sub-harmonic oscillation. An overdriven(saturated) MOSFET can conductdrain current in either direction,

which is very useful in switching-mode operation with reactive loads.

Vertical RF power MOSFETs areuseable through VHF and UHF.Gemini-packaged devices can deliver

up to 1 kW at HF and 100s of wattsat VHF. VMOS devices typically oper-ate from 12, 28, or 50-V supplies,although some devices are capable of operation from 100 V or more.

Laterally Diffused MOS (LDMOS)LDMOS is especially useful at

UHF and lower microwave frequen-cies because direct grounding of itssource eliminates bond-wire induc-tance that produces negative feed-back and reduces gain at high fre-

quencies. This also eliminates theneed for the BeO insulating layercommonly used in other RF-powerMOSFETs.

LDMOS devices typically operatefrom 28-V supplies and are currentlyavailable with power outputs of 120W at 2 GHz. They are relatively lowin cost compared to other devices forthis frequency range and are current-ly the device of choice for use in high-power transmitters at 900 MHz and 2GHz.

Junction FET (JFET)JFETs for power applications are

often called Static InductionTransistors (SITs). Impressive powerand efficiency have been obtainedfrom RF JFETs based upon Si, SiGe,and SiC at frequencies through UHF.However, the JFET has never becomeas popular as other RF-power FETs.

GaAs MEtal Semiconductor FET

(GaAs MESFET)GaAs MESFETs are JFETs basedupon GaAs and a Schottky gate junc-tion. They have higher mobility thando Si devices and are therefore capa-ble of operating efficiently at higherfrequencies. GaAs MESFETs arewidely used for the production of microwave power, with capabilities of up 200 W at 2 GHz and 40 W at 20GHz in packaged devices. These


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RF POWER AMPLIFIERS

devices have relatively low break-down voltages compared toMOSFETs or JFETs and are typical-ly operated from supply voltages(drain biases) of 5 to 10 V. Most

MESFETs are depletion-modedevices and require a negative gatebias, although some enhance-modedevices that operate with a positivebias have been developed. Linearityis often poor due to input capacitance

variation with voltage; the outputcapacitance is also often stronglybias- and frequency-dependent.

Heterojunction FET (HFET) / High-Electron-Mobility Transistor (HEMT)

HFETs and HEMTs improve upon

the MESFET geometry by separating the Schottky and channel functions.

Added to the basic MESFET struc-ture is a heterojunction consisting of an n-doped AlGaAs Schottky layer,an undoped AlGaAs spacer, and anundoped GaAs channel. The disconti-nuity in the band gaps of AlGaAs andGaAs causes a thin layer of electrons(two-dimensional electron gas or 2-DEG) to form below the gate at theinterface of the AlGaAs and GaAslayers. Separation of the donors fromthe mobile electrons reduces colli-sions in the channel, improving themobility, and hence high-frequencyresponse, by a factor of about two.

AlGaAs has crystal-lattice proper-ties similar to those of GaAs, and thismakes it possible to produce a poten-tial difference without lattice stress.The GaAs buffer contributes to a rel-atively high breakdown voltage.Their fabrication employs advancedepitaxial technologies (MolecularBeam Epitaxy or Metal OrganicChemical Vapor Deposition) whichtends to increase their cost.

The GaAs HEMT is known in theliterature by a wide variety of differ-ent names, including MODFET(Modulation-Doped FET), TEGFET(Two-dimensional Electron-GasFET), and SDFET (Selectively DopedFET). It is also commonly called anHFET (Heterostructure FET),

although technically an HFET hasa doped channel that provides thecarriers (instead of the heterojunc-tion). The acronyms HFET andHJFET (HeteroJunction FET)

appear to be used interchangeably.GaAs HEMTs/HFETs with f T ashigh as 158 GHz are reported. PAsbased upon these HEMTs exhibit 15-W outputs at 12 GHz with a power-added efficiency (PAE) of 50 percent.Outputs of 100 W are available at Sband from packaged devices.

Pseudomorphic HEMTThe pseudomorphic HEMT

(pHEMT) further improves upon thebasic HEMT by employing an

InGaAs channel. The increasedmobility of In with respect to GaAsincreases the bandgap discontinuityand therefore the number of carriersin the two-dimensional electron gas.The lattice mismatch between theGaInAs channel and GaAs substrateis also increased, however, and thislimits the In content to about 22 per-cent.

The efficiency of PAs using pHEMTs does not begin to drop untilabout 45 GHz and pHEMTs are use-able to frequencies as high as 80GHz. Power outputs vary from 40 Wat L band to 100 mW at V band.While pHEMTs are normally grownto order, a packaged device pHEMThas recently become available.

InP HEMTThe InP HEMT places an

AlInAs/GaInAs heterojunction on anInP substrate. The lattices are moreclosely matched, which allows an Incontent of up to about 53 percent.This results in increased mobility,which in turn results in increasedelectron velocity, increased conduc-tion-band discontinuity, increasedtwo-dimensional electron gas, andhigher transconductance. The ther-mal resistance is 40 percent lowerthan that of a comparable devicebuilt on a GaAs substrate.

The InP HEMT has higher gain

and efficiency than the GaAspHEMT, with the PA efficiency begin-ning to drop at 60 GHz. However, ithas a lower breakdown voltage (typi-cally 7 V) and must therefore be oper-

ated from a relatively low drain-volt-age supply (e.g., 2 V). This results inlower output per device and possiblyloss in the combiners required toachieve a specified output power.Nonetheless, the InP HEMT general-ly has a factor-of-two efficiencyadvantage over the pHEMT andGaAs HEMT.

InP HEMTs have been fabricatedwith fmax as high as 600 GHz (0.1m gate length), and amplificationhas been demonstrated at frequen-

cies as high as 190 GHz. The efficien-cy does not begin to drop until about60 GHz. Power levels range from 100to 500 mW per chip.

Metamorphic HEMT (mHEMT)The mHEMT allows channels

with high-In content to be built onGaAs substrates. The higher electronmobility and higher peak saturation

velocity result in higher gain than ispossible in a pHEMT. mHEMTs aregenerally limited to low-power appli-cations by their relatively low break-down voltage (


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RF POWER AMPLIFIERS

FET) vertical so that surface imper-fections have less effect upon perfor-mance. The use of a semi-insulating substrate and the higher electronmobility result in reduced parasitics.

The DC curves are somewhat similarto those of a conventional BJT, butoften contain a saturation resistanceas well as saturation voltage.Currently available AlGaAs/GaAsHBTs are capable of producing sever-al watts and are widely used in wire-less handsets, GaAs HBTs are alsowidely used in MMIC circuits at fre-quencies up to X band and can oper-ate in PAs at frequencies as high as20 GHz.

SiGe HBTThe use of SiGe rather than Si in

the base of the HBT both increasesthe maximum operating frequencyand decreases the base resistance.However, they are generally less effi-cient than GaAs HBTs and can havelower breakdown voltages. Oneexperimental SiGe HBT is capable of delivering over 200 W at L band.

InP HBTThe use of InP in an HBT further

boosts mobility and therefore thehigh frequency response. In addition,InP HBTs have lower turn-on andknee voltages, resulting in highergain and efficiency. InP HBTs for RF-power applications incorporate twoheterojunctions (AlInAs/GaInAs andGaInAs/InP). The InP in the collectorincreases the breakdown voltage,allowing higher output power. Todate, outputs of about 0.5 W at 20GHz have been demonstrated, but itis anticipated that operation to 50 or60 GHz will be possible.

SiC MESFETThe wide band gap of SiC results

in both high mobility and high break-down voltage. An SiC MESFET cantherefore have a frequency responsecomparable to that of a GaAs MES-FET, but breakdown voltages doublethat of Si LDMOS. This results in

power densities of 10 W/mm, which isten times that of a GaAs MESFET.The high thermal conductivity of theSiC substrate is particularly usefulin high-power applications. The high-

er operating voltage and associatedhigher load impedance greatly sim-plify output networks and powercombining. SiC MESFETs typicallyoperate from a 48-V supply. Deviceswith outputs of 10 W are currentlyavailable, and outputs of 60 W ormore have been demonstrated exper-imentally. The cost of SiC devices isat presently about ten times that of Si LDMOS.

GaN HEMTGaN offers the same high break-

down voltage of SiC, but even highermobility. Its thermal conductivity is,however, lower, hence GaN devicesmust be built substrate such as SiCor diamond. While the GaN HEMToffers the promise of a high-power,high-voltage device operating at fre-quencies of 10 GHz or more, it is stillin an experimental state. Power out-puts of 8 W at 10 GHz with 30 per-cent efficiency have been demonstrat-ed.

Monolithic Microwave IntegratedCircuit (MMIC)

MMICs integrate RF powerdevices and matching/decoupling ele-ments such as on-chip inductors,capacitors, resistors, and transmis-sion lines. The proximity of these ele-ments to the RF-power devices isessential for input, output, and inter-stage matching at microwave andmillimeter-wave frequencies.

References1. W. J. Bryon, Arcs and sparks,

Part 1, Communications Quarterly , vol. 4, no. 2, pp. 27-43, Spring 1994.

2. W. J. Bryon, The arc method of producing continuous waves,Communications Quarterly , vol. 8, no.3, pp. 47-65, Summer 1998.

3. K. M. MacIlvain and W. H.Freedman, Radio Library, Vol. III:

Radio Transmitters and CarrierCurrents , Scranton PA: InternationalTextbook Company, 1928.

4. J. B. Groe and L. E. Larson,CDMA Mobile Radio Design,

Norwood, MA: Artech House, 2000.5. R. van Nee and R. Prasad,OFDM for Wireless MultimediaCommunications , Norwood, MA:

Artech House, 2000.6. H. L. Krauss, C. W. Bostian, and

F. H. Raab, Solid State Radio Engineering , New York: Wiley, 1980.

7. F. H. Raab, Average efficiencyof power amplifiers, Proc. RF Technology Expo '86 , Anaheim, CA,pp. 474-486, Jan. 30-Feb. 1, 1986.

8. N. Pothecary, Feedforward lin-

ear power amplifiers, in Workshop Notes WFB , Intl. Microwave Symp.,Boston, MA, June 16, 2001.

9. J. F. Sevic, Statistical charac-terization of RF power amplifier effi-ciency for wireless communicationsystems, Proc. Wireless Commun.Conf. , Boulder, CO, pp. 1-4, Aug. 1997.

10. G. Hanington, P.-F. Chen, P. M. Asbeck, and L. E. Larson, High-effi-ciency power amplifier using dynam-ic power-supply voltage for CDMA applications, IEEE Trans.

Microwave Theory Tech. , vol.47, no. 8,pp. 1471-1476, Aug. 1999.

11. I . Kipnis, Refining CDMA mobile-phone power control,

Microwaves & RF , vol. 39, no. 6, pp.71-76, June 2000.

12. J. Staudinger, Applying switched gain stage concepts toimprove efficiency and linearity formobile CDMA power amplification,

Microwave Journal , vol. 43, no. 9, pp.152-162, Sept. 2000.

Table 1 References13. Mobile station base station

interoperability standard for dual-mode cellular system, ANSI-136Standard, Telecommun. Industries

Assoc., 2000.14. Digital cellular communica-

tion systems, RCR STD-27, Researchand Development Center for RadioSystems (RCR), April 1991.


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RF POWER AMPLIFIERS

15. Digital cellular telecommunications system(phase 2+), radio transmission and reception, GSM 5.05Standard, v. 8.4.1, European Telecommun. StandardsInst., 1999.

16. Terrestrial trunked radio (TETRA) voice+data airinterface, TETRA Draf t Standard, EuropeanTelecommun. Standards Inst., 1999.

17. Mobile station base station interoperabilitystandard for dual-mode wideband spread-spectrum cellu-lar system, TIA/EIA IS-95 Interim Standard,Telecommun. Industries Assoc., July 1993.

18. UE radio transmission and reception (FDD), TS25.101, v. 3.4.1, Third Generation Partnership Project,Technical Specification Group, 1999.

Series Notes1. The remaining three parts of this series will appear

in successive issues of High Frequency Electronics (July,September and November 2003 issues).

2. To maintain continuity, all figures, tables, equationsand references will be numbered sequentially throughoutthe entire series.

3. Like all articles in High Frequency Electronics , thisseries will be archived and available for downloading (forpersonal use by individuals only) online at the magazinewebsite: www.highfrequencyelectronics.com

Author InformationThe authors of this series of articles are: Frederick H.

Raab (lead author), Green Mountain Radio Research, e-mail: [email protected]; Peter Asbeck, University of California at San Diego; Steve Cripps, Hywave

Associates; Peter B. Kenington, Andrew Corporation;Zoya B. Popovic, University of Colorado; Nick Pothecary,Consultant; John F. Sevic, California EasternLaboratories; and Nathan O. Sokal, Design Automation.Readers desiring more information should contact thelead author.

AC Alternating Current ACPR Adjacent-Channel Power RatioBJT Bipolar-Junction TransistorC/I Carrier-to-IntermodulationCDF Cumulative Distribution FunctionCDMA Code-Division Multiple AccessCW Continuous WaveDC Direct CurrentDSP Digital Signal Processing EVM Error-Vector MagnitudeFET Field-Effect TransistorFSK Frequency-Shift Keying GMSK Gaussian Minimum Shift Keying GSM Global System for Mobile communicationHBT Heterojunction bipolar transistor

HEMT High Electron-Mobility TransistorHFET Heterojunction FET (also HJFET)IC Integrated CircuitJFET Junction Field-Effect TransistorLDMOS Laterally Diffused MOS (FET)

MESFET MEtal Semiconductor FETmHEMT Metamorphic HEMTMMIC Microwave Monolithic Integrated CircuitMOSFET Metal-Oxide-Silicon Field-Effect TransistorNADC North American Digital CellularNPR Noise-Power RatioNTSC National Television Standards CommitteeOFDM Orthogonal Frequency-Division MultiplexPA Power AmplifierPAE Power-Added EfficiencyPDF Probability-Density FunctionPEP Peak-Envelope PowerpHEMT Pseudomorphic HEMTPSK Phase-Shift Keying QAM Quadrature Amplitude ModulationQPSK Quadrature Phase Shift Keying RF Radio FrequencySRRC Square-Root Raised CosineSSB Single SideBand

Acronyms Used in Part 1


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RF POWER AMPLIFIERS



By Frederick H. Raab, Peter Asbeck, Steve Cripps, Peter B. Kenington,Zoya B. Popovich, Nick Pothecary, John F. Sevic and Nathan O. Sokal

P

art 1 of this seriesintroduced basicconcepts, discussed

the characteristics of sig-nals to be amplified, andgave background infor-mation on RF powerdevices. Part 2 reviewsthe basic techniques, rat-ings, and implementation

methods for power amplifiers operating at HFthrough microwave frequencies.

6a. BASIC TECHNIQUES FORRF POWER AMPLIFICATION

RF power amplifiers are commonly desig-nated as classes A, B, C, D, E, and F [19]. Allbut class A employ various nonlinear, switch-ing, and wave-shaping techniques. Classes of operation differ not in only the method of operation and efficiency, but also in theirpower-output capability. The power-outputcapability (transistor utilization factor) isdefined as output power per transistor nor-malized for peak drain voltage and current of 1 V and 1 A, respectively. The basic topologies(Figures 7, 8 and 9) are single-ended, trans-former-coupled, and complementary. Thedrain voltage and current waveforms of select-ed ideal PAs are shown in Figure 10.

Class AIn class A, the quiescent current is large

enough that the transistor remains at alltimes in the active region and acts as a cur-rent source, controlled by the drive.

Consequently, the drain voltage and currentwaveforms are (ideally) both sinusoidal. Thepower output of an ideal class-A PA is

P o = V om2 / 2R (5)

where output voltage V om on load R cannotexceed supply voltage V DD . The DC-powerinput is constant and the efficiency of an idealPA is 50 percent at PEP. Consequently, theinstantaneous efficiency is proportional to thepower output and the average efficiency isinversely proportional to the peak-to-averageratio (e.g., 5 percent for x = 10 dB). The uti-lization factor is 1/8.

For amplification of amplitude-modulatedsignals, the quiescent current can be varied inproportion to the instantaneous signal enve-lope. While the efficiency at PEP isunchanged, the efficiency for lower ampli-

Our multi-part series onpower amplifier tech-

nologies and applicationscontinues with a review of

amplifier configurations,classes of operation,

device characterizationand example applications

This series of articles is an expanded version of the paper, Power Amplifiers and Transmitters for RF andMicrowave by the same authors, which appeared in the the 50th anniversary issue of the IEEE Transactions on

Microwave Theory and Techniques , March 2002. 2002 IEEE. Reprinted with permission.

Figure 7 A single-ended power amplifier.

From May 2003 High Frequency ElectronicsCopyright 2003 Summit Technical Media, LLC


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RF POWER AMPLIFIERS

tudes is considerably improved. In anFET PA, the implementationrequires little more than variation of the gate-bias voltage.

The amplification process in class A is inherently linear, hence increas-ing the quiescent current or decreas-ing the signal level monotonicallydecreases IMD and harmonic levels.Since both positive and negativeexcursions of the drive affect thedrain current, it has the highest gainof any PA. The absence of harmonicsin the amplification process allowsclass A to be used at frequencies closeto the maximum capability (fmax) of the transistor. However, the efficiencyis low. Class-A PAs are therefore typ-ically used in applications requiring low power, high linearity, high gain,broadband operation, or high-fre-quency operation.

The efficiency of real class-A PAsis degraded by the on-state resistance

or saturation voltage of the transis-tor. It is also degraded by the pres-ence of load reactance, which inessence requires the PA to generatemore output voltage or current todeliver the same power to the load.

Class BThe gate bias in a class-B PA is

set at the threshold of conduction sothat (ideally) the quiescent drain cur-rent is zero. As a result, the transis-tor is active half of the time and thedrain current is a half sinusoid.Since the amplitude of the drain cur-rent is proportional to drive ampli-tude and the shape of the drain-cur-rent waveform is fixed, class-B pro-

vides linear amplification.The power output of a class-B PA

is controlled by the drive level and varies as given by eq. (5). The DC-input current is, however, proportion-al to the drain current which is in

turn proportional to the RF-outputcurrent. Consequently, the instanta-neous efficiency of a class-B PA

varies with the output voltage andfor an ideal PA reaches /4 (78.5 per-

cent) at PEP. For low-level signals,class B is significantly more efficientthan class A, and its average efficien-cy can be several times that of class A at high peak-to-average ratios (e.g.,28 vs. 5 percent for = 10 dB). Theutilization factor is the same 0.125 of class A.

In practice, the quiescent currentis on the order of 10 percent of thepeak drain current and adjusted tominimize crossover distortion causedby transistor nonlinearities at low

outputs. Class B is generally used ina push-pull configuration so that thetwo drain-currents add together toproduce a sine-wave output. At HFand VHF, the transformer-coupledpush-pull topology (Figure 8) is gen-erally used to allow broadband oper-ation with minimum filtering. Theuse of the complementary topology(Figure 9) has generally been limitedto audio, LF, and MF applications bythe lack of suitable p-channel tran-sistors. However, this topology isattractive for IC implementation andhas recently been investigated forlow-power applications at frequen-cies to 1 GHz [20].

Class CIn the classical (true) class-C PA,

the gate is biased below threshold sothat the transistor is active for lessthan half of the RF cycle (Figure 10).Linearity is lost, but efficiency isincreased. The efficiency can beincreased arbitrarily toward 100 per-cent by decreasing the conductionangle toward zero. Unfortunately,this causes the output power (utiliza-tion factor) to decrease toward zeroand the drive power to increasetoward infinity. A typical compromiseis a conduction angle of 150 and anideal efficiency of 85 percent.

The output filter of a true class-CPA is a parallel-tuned type that

Figure 8 Transformer-coupledpush-pull PA.

Figure 9 Complementary PA. Figure 10 Wavefrorms for ideal PAs.


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RF POWER AMPLIFIERS

bypasses the harmonic componentsof the drain current to ground with-out generating harmonic voltages.When driven into saturation, effi-ciency is stabilized and the output

voltage locked to supply voltage,allowing linear high-level amplitudemodulation.

Classical class C is widely used inhigh-power vacuum-tube transmit-ters. It is, however, little used insolid-state PAs because it requireslow drain resistances, making imple-mentation of parallel-tuned outputfilters difficult. With BJTs, it is alsodifficult to set up bias and drive toproduce a true class-C collector-cur-rent waveform. The use of a series-

tuned output filter results in amixed-mode class-C operation that ismore like mistuned class E than trueclass C.

Class DClass-D PAs use two or more tran-

sistors as switches to generate asquare drain-voltage waveform. A series-tuned output filter passes onlythe fundamental-frequency compo-nent to the load, resulting in poweroutputs of (8/ 2)V DD

2 /R and(2/ 2)V DD 2 /R for the transformer-cou-pled and complementary configura-tions, respectively. Current is drawnonly through the transistor that ison, resulting in a 100-percent effi-ciency for an ideal PA.The utilizationfactor (1/2 = 0.159) is the highest of any PA (27 percent higher than thatof class A or B). A unique aspect of class D (with infinitely fast switch-ing) is that efficiency is not degradedby the presence of reactance in theload.

Practical class-D PAs suffer fromlosses due to saturation, switching speed, and drain capacitance. Finiteswitching speed causes the transis-tors to be in their active regions whileconducting current. Drain capaci-tances must be charged and dis-charged once per RF cycle. The asso-ciated power loss is proportional to

V DD3 /2 [21] and increases directly

with frequency.Class-D PAs with power outputs

of 100 W to 1 kW are readily imple-mented at HF, but are seldom usedabove lower VHF because of losses

associated with the drain capaci-tance. Recently, however, experimen-tal class-D PAs have been tested withfrequencies of operation as high as 1GHz [22].

Class EClass E employs a single transis-

tor operated as a switch. The drain- voltage waveform is the result of thesum of the DC and RF currentscharging the drain-shunt capaci-tance. In optimum class E, the drain

voltage drops to zero and has zeroslope just as the transistor turns on.The result is an ideal efficiency of 100percent, elimination of the lossesassociated with charging the draincapacitance in class D, reduction of switching losses, and good toleranceof component variation.

Optimum class-E operationrequires a drain shunt susceptance0.1836/R and a drain series reac-tance 1.15R and delivers a power out-put of 0.577V DD

2 /R for an ideal PA [23]. The utilization factor is 0.098.

Variations in load impedance andshunt susceptance cause the PA todeviate from optimum operation [24,25], but the degradations in perfor-mance are generally no worse thanthose for class A and B.

The capability for efficient opera-tion in the presence of significantdrain capacitance makes class E use-ful in a number of applications. Oneexample is high-efficiency HF PAswith power levels to 1 kW based uponlow-cost MOSFETs intended forswitching rather than RF use [26].

Another example is the switching-mode operation at frequencies ashigh as K band [27]. The class-DE PA [28] similarly uses dead-spacebetween the times when its two tran-sistors are on to allow the load net-work to charge/discharge the draincapacitances.

Class FClass F boosts both efficiency and

output by using harmonic resonatorsin the output network to shape thedrain waveforms. The voltage wave-

form includes one or more odd har-monics and approximates a squarewave, while the current includes evenharmonics and approximates a half sine wave. Alternately (inverse classF), the voltage can approximate ahalf sine wave and the current asquare wave. As the number of har-monics increases, the efficiency of anideal PA increases from the 50 per-cent (class A) toward unity (class D)and the utilization factor increasesfrom 1/8 (class A) toward 1/2 (class

D) [29].The required harmonics can in

principle be produced by current-source operation of the transistor.However, in practice the transistor isdriven into saturation during part of the RF cycle and the harmonics areproduced by a self-regulating mecha-nism similar to that of saturating class C. Use of a harmonic voltagerequires creating a high impedance(3 to 10 times the load impedance) atthe drain, while use of a harmoniccurrent requires a low impedance(1/3 to 1/10 of the load impedance).While class F requires a more com-plex output filter than other PAs, theimpedances must be correct at only afew specific frequencies. Lumped-ele-ment traps are used at lower fre-quencies and transmission lines areused at microwave frequencies.Typically, a shorting stub is placed aquarter or half-wavelength awayfrom the drain. Since the stubs fordifferent harmonics interact and theopen or short must be created at avirtual drain ahead of the draincapacitance and bond-wire induc-tance, implementation of suitablenetworks is a bit of an art.Nonetheless, class-F PAs are success-fully implemented from MF throughKa band.

A variety of modes of operation in-between class C, E, and F are possi-


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RF POWER AMPLIFIERS

ble. The maximum achievable effi-ciency [30] depends upon the numberof harmonics, (0.5, 0.707, 0.8165,0.8656, 0.9045 for 1 through 5 har-monics, respectively). The utilizationfactor depends upon the harmonicimpedances and is highest for idealclass-F operation.

6b. LOAD-PULLCHARACTERIZATION

RF-power transistors are charac-terized by breakdown voltages andsaturated drain currents. The combi-nation of the resultant maximumdrain voltage and maximum draincurrent dictates a range of loadimpedances into which useful powercan be delivered, as well as animpedance for delivery of the maxi-mum power. The load impedance formaximum power results in drain

voltage and current excursions fromnear zero to nearly the maximumrated values.

The load impedances correspond-ing to delivery of a given amount of RF power with a specified maximumdrain voltage lie along parallel-resis-

tance lines on the Smith chart. Theimpedances for a specified maximumcurrent analogously follow a series-resistance line. For an ideal PA, theresultant constant-power contour isfootball-shaped as shown in Figure11.

In a real PA, the ideal drain isembedded behind the drain capaci-tance and bond-wire/package induc-tance. Transformation of the idealdrain impedance through these ele-ments causes the constant-powercontours to become rotated and dis-torted [31]. With the addition of sec-ond-order effects, the contoursbecome elliptical. A set of power con-tours for a given PA somewhatresembles a set of contours for a con-

jugate match. However, a true conju-gate match produces circular con-tours. With a power amplifier, theprocess is more correctly viewed asloading to produce a desired poweroutput. As shown in the example of Figure 12, the power and efficiencycontours are not necessarily aligned,nor do maximum power and maxi-mum efficiency necessarily occur forthe same load impedance. Sets of such load-pull contours are widelyused to facilitate design trade-offs.

Load-pull analyses are generallyiterative in nature, as changing one

parameter may produce a new set of contours. A variety of differentparameters can be plotted during aload-pull analysis, including not onlypower and efficiency, but also distor-

tion and stability. Harmonicimpedances as well as driveimpedances are also sometimes var-ied.

A load-pull system consists essen-tially of a test fixture, provided withbiasing capabilities, and a pair of low-loss, accurately resettable tuners,usually of precision mechanical con-struction. A load-pull characteriza-tion procedure consists essentially of measuring the power of a device, to agiven specification (e.g., the 1-dB

compression point) as a function of impedance. Data are measured at alarge number of impedances andplotted on a Smith chart. Such plotsare, of course, critically dependent onthe accurate calibration of the tuners,both in terms of impedance and loss-es. Such calibration is, in turn, highlydependent on the repeatability of thetuners.

Precision mechanical tuners, withmicrometer-style adjusters, were thetraditional apparatus for load-pullanalysis. More recently, a new gener-ation of electronic tuners hasemerged that tune through the use

varactors or transmission linesswitched by pin diodes. Such elec-tronic tuners [32] have the advantageof almost perfect repeatability andhigh tuning speed, but have muchhigher losses and require highly com-plex calibration routines. Mechanicaltuners are more difficult to controlusing a computer, and move veryslowly from one impedance setting toanother.

In an active load-pull system, asecond power source, synchronized infrequency and phase with the deviceinput excitation, is coupled into theoutput of the device. By controlling the amplitude and phase of theinjected signal, a wide range of impedances can be simulated at theoutput of the test device [33]. Such a

Figure 11 Contant power contoursand transformation.

Figure 12 Example load-pull con-tours for a 0.5-W, 836 MHz PA.(Courtesy Focus Microwaves anddBm Engineering)


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RF POWER AMPLIFIERS

system eliminates the expensivetuners, but creates a substantial cali-bration challenge of its own. The wideavailability of turn-key load-pull sys-tems has generally reduced the appli-

cation of active load-pull to situationswhere mechanical or electronic tun-ing becomes impractical (e.g., mil-limeter-wave frequencies).

6c. STABILITYThe stability of a small-signal RF

amplifier is ensured by deriving a setof S-parameters from using mea-sured data or a linear model, andthen establishing the value of the k-factor stability parameter. If the k-factor is greater than unity, at the

frequency and bias level in question,then expressions for matching impedances at input and output canbe evaluated to give a perfect conju-gate match for the device. Amplifierdesign in this context is mainly amatter of designing matching net-works which present the prescribedimpedances over the necessary speci-fied bandwidth. If the k factor is lessthan unity, negative feedback or lossymatching must be employed in orderto maintain an unconditionally stabledesign.

A third case is relevant to PA design at higher microwave frequen-cies. There are cases where a devicehas a very high k-factor value, but

very low gain in conjugate matchedcondition. The physical cause of thiscan be traced to a device which hasgain roll-off due to carrier-mobilityeffects, rather than parasitics. Insuch cases, introduction of some posi-tive feedback reduces the k-factorand increases the gain in conjugatelymatched conditions, while maintain-ing unconditional stability. This tech-nique was much used in the early eraof vacuum-tube electronics, especiallyin IF amplifiers.

6d. MICROWAVE IMPLEMENTATION At microwave frequencies, lumped

elements (capacitors, inductors)become unsuitable as tuning compo-

nents and are used primarily aschokes and by-passes. Matching, tun-ing, and filtering at microwave fre-quencies are therefore accomplishedwith distributed (transmission-line)

networks. Proper operation of poweramplifiers at microwave frequenciesis achieved by providing the requireddrain-load impedance at the funda-mental and a number of harmonicfrequencies.

Class FClass-F operation is specified in

terms of harmonic impedances, so itis relatively easy to see how trans-mission-line networks are used.Methods for using transmission lines

in conjunction with lumped-elementtuned circuits appear in the originalpaper by Tyler [34]. In modernmicrowave implementation, however,it is generally necessary to use trans-mission lines exclusively. In addition,the required impedances must beproduced at a virtual ideal drain thatis separated from the output networkby drain capacitance, bond-wire/leadinductance.

Typically, a transmission linebetween the drain and the load pro-

vides the fundamental-frequencydrain impedance of the desired value.

A stub that is a quarter wavelengthat the harmonic of interest and openat one end provides a short circuit atthe opposite end. The stub is placedalong the main transmission line ateither a quarter or a half wavelengthfrom the drain to create either anopen or a short circuit at the drain[35]. The supply voltage is fed to thedrain through a half-wavelength linebypassed on the power-supply end oralternately by a lumped-elementchoke. When multiple stubs are used,the stub for the highest controlledharmonic is placed nearest the drain.Stubs for lower harmonics are placedprogressively further away and theirlengths and impedances are adjustedto allow for interactions. Typically,open means three to ten times thefundamental-frequency impedance,

and shorted means no more 1/10 to1/3 of the fundamental-frequencyimpedance [FR17].

A wide variety of class-F PAs havebeen implemented at UHF and

microwave frequencies [36-41].Generally, only one or two harmonicimpedances are controlled. In the X-band PA from [42], for example, theoutput circuit provides a match at thefundamental and a short circuit atthe second harmonic. The third-har-monic impedance is high, but notexplicitly adjusted to be open. The 3-dB bandwidth of such an output net-work is about 20 percent, and the effi-ciency remains within 10 percent of its maximum value over a bandwidth

of approximately 10 to 15 percent.Dielectric resonators can be used

in lieu of lumped-element traps inclass-F PAs. Power outputs of 40 Whave been obtained at 11 GHz withefficiencies of 77 percent [43].

Class EThe drain-shunt capacitance and

series inductive reactance requiredfor optimum class-E operation resultin a drain impedance of R + j0.725Rat the fundamental frequency, j1.7846R at the second harmonic,and proportionately smaller capaci-tive reactances at higher harmonics.

At microwave frequencies, class-Eoperation is approximated by provid-ing the drain with the fundamental-frequency impedance and preferablyone or more of the harmonicimpedances [44].

An example of a microwaveapproximation of class E that pro-

vides the correct fundamental andsecond-harmonic impedances [44] isshown in Figure 13. Line l2 is a quar-ter-wavelength long at the secondharmonic so that the open circuit atits end is transformed to a short atplane AA'. Line l1 in combinationwith L and C is designed to be also aquarter wavelength to translate theshort at AA' to an open at the tran-sistor drain. The lines l1 to l4 providethe desired impedance at the funda-


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July 2003 31

mental. The implementation using anFLK052 MESFET is shown in Figure14 produces 0.68 W at X band with adrain efficiency of 72 percent and

PAE of 60 percent [42].Methods exist for providing the

proper impedances through thefourth harmonic [45]. However, theharmonic impedances are not critical[30], and many variations are there-fore possible. Since the transistoroften has little or no gain at the high-er harmonic frequencies, thoseimpedances often have little or noeffect upon performance. A single-stub match is often sufficient to pro-

vide the desired impedance at thefundamental while simultaneouslyproviding an adequately highimpedance at the second harmonic,thus eliminating the need for anextra stub and reducing a portion of the losses associated with it. Mostmicrowave class-E amplifiers operatein a suboptimum mode [46].Demonstrated capabilities rangefrom 16 W with 80-percent efficiencyat UHF (LDMOS) to 100 mW with60-percent efficiency at 10 GHz [47],[48], [44], [49], [50], [51]. Optical sam-pling of the waveforms [52] has veri-fied that these PAs do indeed operatein class E.

ComparisonPAs configured for classes A (AB),

E, and F are compared experimental-ly in [50] with the following conclu-sions. Classes AB and F have essen-tially the same saturated output

power, but class F has about 15 per-cent higher efficiency. Class E has thehighest efficiency. Gain compressionoccurs at a lower power level for class

E than for class F. For a given effi-ciency, class F produces more power.For the same maximum outputpower, the third order intermodula-tion products are about 10 dB lowerfor class F than for class E. Lower-power PAs implemented with smallerRF power devices tend to be moreefficient than PAs implemented withlarger devices [42].

Millimeter-Wave PAsSolid-state PAs for millimeter-

wave (mm-W) frequencies (30 to 100GHz) are predominantly monolithic.Most Ka-band PAs are based uponpHEMT devices, while most W-bandPAs are based upon InP HEMTs.Some use is also made of HBTs at thelower mm-W frequencies. Class A isused for maximum gain. Typical per-formance characteristics include 4 Wwith 30-percent PAE at Ka band [53],250 mW with 25-percent PAE at Qband [54], and 200 mW with 10-per-cent PAE at W band [55]. Devices foroperation at mm-W are inherentlysmall, so large power outputs areobtained by combining the outputs of multiple low-power amplifiers in cor-porate or spatial power combiners.

6e. EXAMPLE APPLICATIONSThe following examples illustrate

the wide variety of power amplifiersin use today:

HF/VHF Single SidebandOne of the first applications of

RF-power transistors was linearamplification of HF single-sideband

signals. Many PAs developed byHelge Granberg have been widelyadapted for this purpose [56, 57]. The300-W PA for 2 to 30 MHz uses a pairof Motorola MRF422 Si NPN transis-tors in a push-pull configuration. ThePA operates in class AB push-pullfrom a 28-V supply and achieves acollector efficiency of about 45 per-cent (CW) and a two-tone IMD ratioof about 30 dBc. The 1-kW amplifieris based upon a push-pull pair of MRF154 MOSFETs and operatesfrom a 50-V supply. Over the frequen-cy range of 2 to 50 MHz it achieves adrain efficiency of about 58 percent(CW) with an IMD rating of 30 dBc.

13.56-MHz ISM Power SourcesHigh-power signals at 13.56 MHz

are needed for a wide variety of Industrial, Scientific, and Medical(ISM) applications such as plasmageneration, RF heating, and semicon-ductor processing. A 400-W class-EPA uses an International RectifierIRFP450LC MOSFET (normallyused for low-frequency switching-mode DC power supplies) operatesfrom a 120-V supply and achieves adrain efficiency of 86 percent [58, 26].Industrial 13.56-MHz RF power gen-erators using class-E output stageshave been manufactured since 1992by Dressler Hochfrequenztechnik(Stolberg, Germany) and Advanced

Figure 13 Idealized microwave class-E PA circuit. Figure 14 Example X-band class-E PA.


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RF POWER AMPLIFIERS

Energy Industries (Ft. Collins, CO).They typically use RF-powerMOSFETs with 500- to 900-V break-down voltages made by DirectedEnergy or Advanced PowerTechnology and produce output pow-ers of 500 W to with 3 kW with drainefficiencies of about 90 percent. The

Advanced Energy Industries amplifi-er (Figure 15) uses thick-film-hybridcircuits to reduce size. This allowsplacement inside the clean-roomfacilities of semiconductor-manufac-turing plants, eliminating the needfor long runs of coaxial cable from anRF-power generator installed outsidethe clean-room.

VHF FM Broadcast TransmitterFM-broadcast transmitters (88 to

108 MHz) with power outputs from50 W to 10 kW are manufactured byBroadcast Electronics (Quincy,Illinois). These transmitters use up to32 power-combined PAs based uponMotorola MRF151G MOSFETs. ThePAs operate in class C from a 44-V supply and achieve a drain efficiencyof 80 percent. Typically, about 6 per-cent of the output power is dissipatedin the power combiners, harmonic-suppression filter, and lightning-pro-tection circuit.

MF AM Broadcast TransmittersSince the 1980s, AM broadcast

transmitters (530 to 1710 kHz) havebeen made with class-D and -E RF-output stages. Amplitude modulationis produced by varying the supply

voltage of the RF PA with a high-effi-ciency amplitude modulator.

Transmitters made by Harris(Mason, Ohio) produce peak-envelopeoutput powers of 58, 86, 150, 300, and550 kW (unmodulated carrier powersof 10, 15, 25, 50, and 100 kW). The100-kW transmitter combines theoutput power from 1152 transistors.The output stages can use eitherbipolars or MOSFETs, typically oper-ate in class DE from a 300-V supply,and achieve an efficiency of 98 per-cent. The output section of the Harris3DX50 transmitter is shown inFigure 16.

Transmitters made by BroadcastElectronics (Quincy, IL) use class-ERF-output stages based upon

APT6015LVR MOSFETs operating from 130-V maximum supply volt-ages. They achieve drain efficienciesof about 94 percent with peak-enve-lope output powers from 4.4 to 44 kW.The 44-kW AM-10A transmitter com-bines outputs from 40 individual out-put stages.

900-MHz Cellular-TelephoneHandset

Most 900-MHz CDMA handsetsuse power-amplifier modules from

vendors such as Conexant and RFMicro Devices. These modules typi-cally contain a single GaAs-HBTRFIC that includes a single-endedclass-AB PA. Recently developed PA modules also include a silicon controlIC that provides the base-bias refer-ence voltage and can be commandedto adjust the output-transistor basebias to optimize efficiency whilemaintaining acceptably low amplifierdistortion. over the full ranges of temperature and output power. A typ-ical module (Figure 17) produces 28dBm (631 mW) at full output with aPAE of 35 to 50 percent.

Cellular-Telephone Base

Station TransmitterThe Spectrian MCPA 3060 cellu-lar base-station transmitter for 1840-1870 MHz CDMA systems providesup to 60-W output while transmitting a signal that may include as many 9modulated carriers. IMD is mini-mized by linearizing a class-AB mainamplifier with both adaptive predis-tortion and adaptive feed-forwardcancellation. The adaptive control

Figure 15 3-kW high efficiency PA for 13.56 ISM-band operation.

(Courtesy Advanced Energy)

Figure 16 Output section of a 50-kW AM broadcast transmitter.

(Courtesy Harris)


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RF POWER AMPLIFIERS

system adjusts operation as needed

to compensate for changes due totemperature, time, and output power.The required adjustments arederived from continuous measure-ments of the system response to aspread-spectrum pilot test signal.The amplifier consumes a maximumof 810 W from a 27-V supply.

S-Band Hybrid Power Module A thick-film-hybrid power-ampli-

fier module made by UltraRF (nowCree Microwave) for 1805 to 1880MHz DCS and 1930-1960 MHz PCSis shown in Figure 18. It uses four140-mm LDMOS FETs operating from a 26-V drain supply. The indi-

vidual PAs have 11-dB power gainand are quadrature-combined to pro-duce a 100-W PEP output. The aver-age output power is 40 W for EDGEand 7 W for CDMA, with an ACPR of 57 dBc for EDGE and 45 dBc forCDMA. The construction is basedupon 0.02-in. thick film with silvermetalization.

GaAs MMIC Power Amplifier A MMIC PA for use from 8 to 14

GHz is shown in Figure 19. Thisamplifier is fabricated with GaAsHBTs and intended for used inphased-array radar. It produces a 3-W output with a PAE of approxi-mately 40 percent [59].

References19. H. L. Krauss, C. W. Bostian, and

F. H. Raab, Solid State Radio Engineering, New York: Wiley, 1980.

20. R. Gupta and D. J. Allstot,Fullymonolithic CMOS RF power amplifiers:Recent advances, IEEE Communi-cations Mag. , vol. 37, no. 4, pp. 94-98,

April 1999.21. F. H. Raab and D. J. Rupp, HF

power amplifier operates in both classB and class D, Proc. RF Expo West 93 ,San Jose, CA, pp. 114-124, March 17-19,1993.

22. P. Asbeck, J. Mink,T. Itoh, and G.Haddad,Device and circuit approaches

for next-generation wireless communi-cations, Microwave J. , vol. 42, no. 2, pp.22-42, Feb. 1999.

23. N. O. Sokal and A. D. Sokal,Class Ea new class of high efficiency

tuned single-ended switching poweramplifiers, IEEE J. Solid-StateCircuits , vol. SC-10, no. 3, pp. 168-176,June 1975.

24. F. H. Raab, Effects of circuit variations on the class E tuned poweramplifier, IEEE J. Solid State Circuits ,

vol. SC-13, no. 2, pp. 239-247, April1978.

25. F. H. Raab, Effects of VSWRupon the class-E RF-power amplifier,

Proc. RF Expo East 88 , Philadelphia,PA, pp. 299-309, Oct. 25-27, 1988.

26. J. F. Davis and D. B. Rutledge, A low-cost class-E power amplifier withsine-wave drive, Int. Microwave Symp.

Digest , vol. 2, pp. 1113-1116, Baltimore,MD, June 7-11, 1998.

27. T. B. Mader and Z. B. Popovic,The transmission-line high-efficiencyclass-E amplifier, IEEE Microwaveand Guided Wave Letters , vol. 5, no. 9,pp. 290-292, Sept. 1995.

28. D. C. Hamill, Class DE invert-ers and rectifiers for DC-DC conver-sion, PESC96 Record, vol. 1, pp. 854-860, June 1996.

29. F. H. Raab, Maximum efficiencyand output of class-F power amplifiers,

IEEE Trans. Microwave Theory Tech. , vol. 47, no. 6, pp. 1162-1166, June 2001.

30. F. H. Raab, Class-E, -C, and -Fpower amplifiers based upon a finitenumber of harmonics, IEEE Trans.

Microwave Theory Tech. , vol. 47, no. 8,pp. 1462-1468, Aug. 2001.

31. S. C. Cripps, RF Power Amplif iers for Wireless Communi-cation, Norwood, MA: Artech, 1999.

32. A load pull system with har-monic tuning, Microwave J. , pp. 128-132, March 1986.

33. B. Hughes, A. Ferrero, and A.Cognata, Accurate on-wafer power andharmonic measurements of microwaveamplifiers and devices, IEEE Int.

Microwave Symp. Digest , Albuquerque,NM, pp. 1019-1022, June 1-5, 1992.

34. V. J.Tyler, A new high-efficiency

Figure 17 Internal view of a dual-band (GSM/DCS) PA module forcellular telephone handsets.(Courtesy RF Micro Devices)

Figure 18 Thick-film hybrid S-bandPA module. (Courtesy UltraRF)

Figure 19 MMIC PA for X- and K-bands.

Acronyms Used in Part 2BJT Bipolar Junction

TransistorDSP Digital Signal

ProcessorIC Integrated CircuitIMD Intermodulation

DistortionMOSFET Metal Oxide Silicon

FET


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RF POWER AMPLIFIERS

high power amplifier, The Marconi Review , vol. 21, no. 130, pp. 96-109, Fall1958.

35. A. V. Grebennikov, Circuitdesign technique for high-efficiency

class-F amplifiers, Int. Microwave Symp. Digest , vol. 2, pp. 771-774,Boston, MA, June 13-15, 2000.

36. P. Colantonio, F. Giannini, G.Leuzzi, and E. Limiti, On the class-Fpower-amplifier design, RF and Micro-wave Computer-Aided Engin-eering ,

vol. 32, no. 2, pp. 129-149, March 1999.37. A. N. Rudiakova and V. G.

Krizhanovski, Driving waveforms forclass-F power amplifiers, Int.

Microwave Symp. Digest , vol. 1,pp. 473-476, Boston, MA, June 13-15, 2000.

38. A. Inoue, T. Heima, A. Ohta, R.Hattori, and Y. Mitsui, Analysis of class-F and inverse class-F amplifiers,

Int. Microwave Symp. Digest , vol. 2, pp.775-778, Boston, MA, June 13-15, 2000.

39. F. van Rijs et al., Influence of output impedance on power added effi-ciency of Si-bipolar power transistors,

Int. Microwave Symp. Digest , vol. 3, pp.1945-1948, Boston, MA, June 13-15,2000.

40. F. Huin, C. Duvanaud, V. Serru,F. Robin,and E.Leclerc,A single supply

very high power and efficiency integrat-ed PHEMT amplifier for GSM applica-tions, Proc. 2000 RFIC Symp. , Boston,MA, CD-ROM, June 11-13, 2000.

41. B. Ingruber et al.,Rectangularly driven class-A harmon-ic-control amplifier, IEEE Trans.

Microwave Theory Tech. , pt. 1, vol. 46,no. 11, pp. 1667-1672, Nov. 1998.

42. E. W. Bryerton, M. D. Weiss, andZ. Popovic, Efficiency of chip-level ver-sus external power combining, IEEETrans. Microwave Theory Tech. , vol. 47,no. 8, pp. 1482-1485, Aug. 1999.

43 S. Toyoda, Push-pull poweramplifiers in the X band, Int.

Microwave Symp. Digest , vol. 3, pp.1433-1436, Denver, CO, June 8-13, 1997.

44. T. B. Mader and Z. Popovic, Thetransmission-line high-efficiency class-E amplifier, IEEE Micro-wave andGuided Wave Lett. , vol.5, no. 9, pp. 290-292, Sept. 1995.

45. A. J. Wilkinson and J. K. A.Everard, Transmission line load net-work topology for class E amplifiers,

IEEE Trans. Microwave Theory Tech. , vol. 47, no. 6, pp. 1202-1210, June 2001.

46. F. H. Raab, Suboptimum opera-tion of class-E power amplifiers, Proc. RF Technology Expo. , Santa Clara, CA,pp. 85-98, Feb. 1989.

47. S. Li, UHF and X-band class-Eamplifiers, Ph.D. Thesis, CaliforniaInstitute of Technology, Pasadena, 1999.

48. F. J. Ortega-Gonzalez, J. L.Jimenez-Martin, A. Asensio-Lopez, G.Torregrosa-Penalva, High-efficiencyload-pull harmonic controled class-Epower amplifier, IEEE MicrowaveGuided Wave Lett. , vol. 8, no. 10, pp.

348-350, Oct. 1998.49. E. Bryerton, High-efficiency

switched-mode microwave circuits,Ph.D. dissertation, Univ. of Colorado,Boulder, June 1999.

50. T. B. Mader, E. W. Bryerton, M.Markovic, M. Forman, and Z. Popovic,Switched-mode high-efficiency micro-wave power amplifiers in a free-spacepower-combiner array, IEEE Trans.

Microwave Theory Tech. , vol. 46, no. 10,pt. I, pp. 1391-1398, Oct. 1998.

51. M. D. Weiss and Z. Popovic, A 10GHz high-efficiency active antenna,

Int. Microwave Symp. Digest , vol. 2, pp.663-666, Anaheim, CA, June 14-17,1999.

52. M. Weiss, M. Crites, E. Bryerton,J. Whitacker, and Z. Popovic, "Timedomain optical sampling of nonlinearmicrowave amplifiers and multipliers,

IEEE Trans. Microwave Theory Tech. , vol. 47, no.12, pp. 2599-2604, Dec. 1999.

53. J. J. Komiak, W. Kong, P. C.Chao, and K. Nichols, Fully monolithic4 watt high efficiency Ka-band poweramplifier, Int. Microwave Symp.

Digest , vol. 3, pp. 947-950, Anaheim,CA, June 14-17, 1999.

54. S.-W. Chen et al., A 60-GHzhigh-efficiency monolithic power ampli-fier using 0.1-m pHEMTs, IEEE

Microwave and Guided Wave Lett. , vol.5, pp. 201-203, June 1995.

55. D. L. Ingram et al., Compact W-band solid-state MMIC high power

sources, Int. Microwave Symp. Digest , vol. 2, pp. 955-958, Boston, MA, June13-15, 2000.

56. H. Granberg, Get 300 wattsPEP linear across 2 to 30 MHz from

this push-pull amplifier, BulletinEB27A, Motorola SemiconductorProducts, Phoenix, Feb. 1980.

57. H. Granberg, A compact 1-kW2-50 MHz solid-state linear amplifier,QEX , no. 101, pp. 3-8, July 1990. Also

AR347, Motorola SemiconductorProducts, Feb. Oct. 1990.

58. N. O. Sokal, Class-E RF poweramplifiers ... , QEX , No. 204, pp. 9-20,Jan./Feb. 2001.

59. M. Salib, A. Gupta, A. Ezis, M.Lee, and M. Murphy, A robust 3W high

efficiency 8-14 GHz GaAs/AlGaAs het-erojunction bipolar transistor poweramplifier, Int. Microwave Symp.

Digest , vol. 2, pp. 581-584, Baltimore,MD, June 7-11, 1998.

Author InformationThe authors of this series of arti-

cles are: Frederick H. Raab (leadauthor), Green Mountain RadioResearch, e-mail: [email protected];Peter Asbeck, University of California at San Diego; SteveCripps, Hywave Associates; Peter B.Kenington, Andrew Corporation;Zoya B. Popovic, University of Colorado; Nick Pothecary,Consultant; John F. Sevic, CaliforniaEastern Laboratories; and Nathan O.Sokal, Design Automation. Readersdesiring more information shouldcontact the lead author.

Notes1. In Part 1 of this series (May

2003 issue), the references containedin Table 1 were not numbered cor-rectly. The archived version has beencorrected and may be downloadedfrom: www.highfrequencyelectronics.com click on Archives, selectMay 2003 Vol. 2 No. 3 then clickon the article title.

2. This series has been extendedto five parts, to be published in succe-sive issues through January 2004.


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RF POWER AMPLIFIERS



By Frederick H. Raab, Peter Asbeck, Steve Cripps, Peter B. Kenington,Zoya B. Popovich, Nick Pothecary, John F. Sevic and Nathan O. Sokal

T

he building blocksused in transmit-ters are not only

power amplifiers, but a variety of other circuitelements including oscil-lators, mixers, low-levelamplifiers, filters, match-ing networks, combiners,and circula tors. The

arrangement of building blocks is known asthe architecture of a transmitter. The classictransmitter architecture is based upon linearPAs and power combiners. More recently,transmitters are being based upon a variety of different architectures including stagebypassing, Kahn, envelope tracking, outphas-ing, and Doherty. Many of these are actuallyfairly old techniques that have been recentlymade practical by the capabilities of DSP.

7a. LINEAR ARCHITECTUREThe conventional architecture for a linear

microwave transmitter consists of a basebandor IF modulator, an up-converter, and a power-amplifier chain (Figure 20). The amplifierchain consists of cascaded gain stages withpower gains in the range of 6 to 20 dB. If thetransmitter must produce an amplitude-mod-ulated or multi-carrier signal, each stage musthave adequate linearity. This generallyrequires class-A amplifiers with substantialpower back-off for all of the driver stages. Thefinal amplifier (output stage) is always themost costly in terms of device size and currentconsumption, hence it is desirable to operatethe output stage in class B. In applicationsrequiring very high linearity, it is necessary touse class A in spite of the lower efficiency.

The outputs of a driver stage must bematched to the input of the following stagemuch as the final amplifier is matched to theload. The matching tolerance for maintaining power level can be significantly lower thanthat for gain [60], hence the 1-dB load-pullcontours are more tightly packed for powerthan for gain.

To obtain even modest bandwidths (e.g.,above 5 percent), the use of quadrature bal-anced stages is advisable (Figure 21). Themain benefit of the quadrature balanced con-figuration is that reflections from the transis-tors are cancelled by the action of the inputand output couplers. An individual device cantherefore be deliberately mismatched (e.g., toachieve a power match on the output), yet thequadrature-combined system appears to bewell-matched. This configuration also acts asan effective power combiner, so that a givenpower rating can be achieved using a pair of devices having half of the required power per-formance. For moderate-bandwidth designs,the lower-power stages are typically designedusing a simple single-ended cascade, which insome cases is available as an RFIC. Designswith bandwidths approaching an octave or

Transmitter architectures isthe subject of this install-

ment of our continuingseries on power amplifiers,

with an emphasis ondesigns that can meet

todays linearity and highefficiency requirements

Figure 20 A conventional transmitter.

RF/

BasebandExciter

Mixer

LO RX

3-stage PA

From September 2003 High Frequency ElectronicsCopyright 2003 Summit Technical Media, LLC


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RF POWER AMPLIFIERS

more require the use of quadrature-balanced stages throughout theentire chain.

Simple linear-amplifier chains of

this kind have high linearity but onlymodest efficiency. Single-carrierapplications usually operate the finalamplifier to about the 1-dB compres-sion point on amplitude modulationpeaks. A thus-designed chain inwhich only the output stage exhibitscompression can still deliver an

ACPR in the range of about 25 dBcwith 50-percent efficiency at PEP.

Two practical problems are fre-quently encountered in the design of linear PA chains: stability and lowgain. Linear, class-A chains are actu-ally more susceptible to oscillationdue to their high gain, and single-path chains are especially prone tounstable behavior. Instability can besubdivided into the two distinct cate-gories: Low-frequency oscillation andin-band instability. In-band instabili-ty is avoided by designing the indi-

vidual gain stages to meet the crite-ria for unconditional stability; i.e.,the Rollet k factor [61] must begreater than unity for both in-bandand out-of-band frequencies. Meeting this criterion usually requires sacri-ficing some gain through the use of absorptive elements. Alternatively,the use of quadrature balancedstages provides much greater isola-tion between individual stages, andthe broadband response of thequadrature couplers can eliminatethe need to design the transistor

stage itself with k>1. This is anotherreason for using quadrature coupledstages in the output of the chain.

Large RF power devices typically

have very high transconductance, andthis can produce low-frequency insta-bility unless great care is taken toterminate both the input and outputat low frequencies with impedancesfor unconditional stability. Because of large separation from the RF band,this is usually a simple matter requir-ing a few resistors and capacitors.

At X band and higher, the powergain of devices in the 10 W and abovecategory can drop well below 10 dB.To maintain linearity, it may be nec-essary to use a similarly size deviceas a driver. Such an architectureclearly has a major negative impactupon the cost and efficiency of thewhole chain. In the more extremecases, it may be advantageous to con-sider a multi-way power combiner,where 4, 8, or an even greater num-ber of smaller devices are combined.Such an approach also has otheradvantages, such as soft failure, bet-ter thermal management, and phaselinearity. However, it typically con-sumes more board space.

7b. POWER COMBINERSThe need frequently arises to

combine the outputs of several individual PAs to achieve the desiredtransmitter output. Whether to use anumber of smaller PAs vs. a singlelarger PA is one of the most basicdecisions in selection of an architec-

ture [60]. Even when larger devicesare available, smaller devices oftenoffer higher gain, a lower matching Qfactor (wider bandwidth), betterphase linearity, and lower cost. Heat

dissipation is more readily accom-plished with a number of smalldevices, and a soft-failure modebecomes possible. On the other hand,the increase in parts count, assemblytime, and physical size are significantdisadvantages to the use of multiple,smaller devices.

Direct connection of multiple PAsis generally impractical as the PAsinteract, allowing changes in outputfrom one to cause the load impedanceseen by the other to vary. A constant

load impedance, hence isolation of one PA from the other, is provided bya hybrid combiner. A hybrid combinercauses the difference between thetwo PA outputs to be routed to anddissipated in a balancing or dumpresistor. In the event that one PA fails, the other continues to operatenormally, with the transmitter out-put reduced to one fourth of nominal.

The most common power combin-er is the quadrature-hybrid combiner.

A 90 phase shift is introduced atinput of one PA and also at the out-put of the other. The benefits of quadrature combining include con-stant input impedance in spite of

variations of input impedances of theindividual PAs, cancellation of oddharmonics, and cancellation of back-ward-IMD (IMD resulting from a sig-nal entering the output port). Inaddition, the effect of load impedanceupon the system output is greatlyreduced (e.g., to 1.2 dB for a 3:1SWR). Maintenance of a nearly con-stant output occurs because the loadimpedance presented to one PA decreases when that presented to theother PA increases. As a result, how-ever, device ratings increase and effi-ciency decreases roughly in propor-tion to the SWR [65]. Becausequadrature combiners are inherentlytwo-terminal devices, they are usedin a corporate combining architecture

Figure 21 Amplifier stages with quadrature combiners.

0

90


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RF POWER AMPLIFIERS

(Figure 21). Unfortunately, the physical construction of such couplers poses some problems in a PC-board envi-

ronment. The very tight coupling between the two quar-ter-wave transmission lines requires either very fine gapsor a three-dimensional structure. This problem is circum-

vented by the use of a miniature co-axial cable having apair of precisely twisted wires to from the coupling sec-tion or ready-made, low-cost surface mount 3-dB couplers.

The Wilkinson or in-phase power combiner [62] isoften more easily fabricated than a quadrature combiner.In the two-input form (as in each section in Figure 22),the outputs from two quarter-wavelength lines summedinto load R 0 produce an apparent load impedance of 2 R 0,which is transformed through the lines into at the loadimpedances R PA seen by the individual PAs. The differ-ence between the two PA outputs is dissipated in a resis-tor connected across the two inputs. Proper choice of thebalancing resistor (2 R PA ) produces a hybrid combinerwith good isolation between the two PAs. The Wilkinsonconcept can be extended to include more than two inputs[63].

Greater bandwidth can be obtained by increasing thenumber of transforming sections in each signal path. A single-section combiner can have a useful bandwidth of about 20 percent, whereas a two-section version can havea bandwidth close to an octave. In practice, escalating cir-cuit losses generally preclude the use of more than twosections.

All power-combining techniques all suffer from circuitlosses as well as mismatch losses. The losses in a simpletwo-way combiner are typically about 0.5 dB or 10 per-cent. For a four-way corporate structure, the intercon-nects typically result in higher losses. Simple openmicrostrip lines are too lossy for use in combining struc-tures. One technique that offers a good compromiseamong cost, produceability, and performance, uses sus-pended stripline. The conductors are etched onto double-sided PC board, interconnected by vias, and then sus-

pended in a machined cavity. Structures of this kind allowhigh-power 8-way combiners with octave bandwidths and

of 0.5 dB. A wide variety of other approaches to power-combin-

ing circuits are possible [62, 64]. Microwave power canalso be combined during rad

hfd rfpas raab

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