generalized filter topology using grounded components and single novel active element

Circuits Syst Signal ProcessDOI 10.1007/s00034-014-9807-4

SHORT PAPER

Generalized Filter Topology Using GroundedComponents and Single Novel Active Element

Parveen Beg · Sudhanshu Maheshwari

Received: 28 January 2014 / Revised: 18 April 2014 / Accepted: 19 April 2014© Springer Science+Business Media New York 2014

Abstract This paper introduces a new active element combining the useful features ofdifferential voltage, dual-X and first generation current conveyors. The new proposedactive element is further utilized to introduce a new generalized filter topology employ-ing grounded components only. The proposed single active element-based topologybenefits from first-order and second-order filter realization by appropriate impedancespecialization. The circuit topology with single current input provides two output cur-rents and voltages in each case. A thorough study of proposed active element alongwith extensive simulations is carried out to validate the filter topology. A detailednon-ideal study is also given. To further support the usefulness of filter topology,higher-order filters are also realized. The new active element and the new filter struc-ture provide advancement to the existing knowledge; with the scope of active elementbeing further exploited for analog signal processing applications in general. The pro-posed differential voltage dual-X first generation current conveyor (DV-DXCCI) andits filtering applications are simulated using TSMC 0.25 µm technology.

Keywords Current mode · Active filters · Current conveyors · Analog signalprocessing

1 Introduction

Current conveyors have become the most common building blocks for design-ing high performance analog filters. The first current conveyors namely, the CCI

P. Beg (B) · S. MaheshwariDepartment of Electronics Engineering, Z. H. College of Engineering and Technology,Aligarh Muslim University, Aligarh 202002, Indiae-mail: [email protected]

S. Maheshwarie-mail: [email protected]

Circuits Syst Signal Process

and the CCII, were proposed in [25,28], respectively. Later, several other vari-ants such as the DOCCII, DCCDVCC, DDCC, DVCC, DXCCII, DVCCTA, CBTA,FDCCII, CCCCTA, ICCII, CFTA with buffered output etc. were also proposed[1,5,8,9,14,19,23,24,26,27,29,31,32,34]. Most of these blocks are a result of com-bination of the known blocks like CCII, CCCII, differential pairs, transconductanceamplifiers, current followers, voltage followers, etc. Despite being combination ofknown blocks, these developed active elements not only aim at greater design flexibil-ity for analog signal processing, but also be referred to, and counted as single activeelement in each case. A detailed classification, review along with model descriptionsof active elements was published in the literature, with prospective continuation of thenew additions, based on concurrence of known simpler building blocks [7]. Quite a fewblocks have already been added to the literature, ever since the appearance of Ref. [7],proving the assertion made therein. These building blocks have played a significantrole in realizing a number of current mode (CM) filters with a number of distinctivefeatures. With a view to being adaptable towards low cost portable systems, the useof single active elements for such filtering functions has been a topic of continuousresearch [2,3,6,8,22–24,26,27,30,34]. Out of these, grounded components-based fil-ters [8,23,24,26,27] are of special interest as these facilitate easy integration and para-sitic reduction. Filters proposed in [8,24] provide simultaneous responses while thosein [23,26,27] offer high output impedance features. Single active element-based filtersrealized with passive components in both grounded and floating form do not allowsimultaneous outputs [2,22] and lack in high impedance [2,6,22,34], but do have lowcomponent count in some cases [6,34]. Designs presented in [2,3,8,24,30,34] requireonly a single input in contrast to those presented in [6,12,22,23,26,27] which needmultiple inputs and is a design overhead.

The filter proposed in this paper is based on single input, hence simpler in configura-tion than multiple input circuits. The active element used is the newly introduced DV-DXCCI, which is a versatile building block since it combines the features of DVCC,DXCCII, and CCI. The CM filter realized using the proposed DV-DXCCI is capableof realizing first- or second-order filter responses. The proposed filter employs onlya single DV-DXCCI and four grounded passive components, offering several advan-tages such as: (i) use of a single active element, (ii) need of only a single input, (iii)grounded passive components, (iv) two simultaneous responses, (v) output at highimpedance terminals, (vi) ability to obtain first- or second-order responses by properimpedance specialization, (vii) simultaneous CM and transresistance mode outputs insecond-order filter. Table 1 summarizes the superiority of the proposed design oversimilar existing designs. The active element, filter circuit, practical considerationsrelated to the new active element and proposed circuit are given along with appli-cation in higher-order filters in the subsequent sections. PSPICE results are furtherincluded for verification purpose.

2 Circuit Description

DV-DXCCI is a new CM active building block. Its symbol is shown in Fig. 1. Itmay be noted that the new DV-DXCCI combines the advantages of the dual-X CCII


Tabl

e1

Com

para

tive

stud

yof

prop

osed

circ

uitw

ithot

her

sing

leac

tive

elem

ent-

base

dfil

ter

Ref

no.

Act

ive

elem

ent

No.

ofpa

ssiv

eco

mpo

nent

sG

roun

ded

Com

pone

nts

Num

ber

ofin

puts

No.

ofSi

mul

tane

ous

outp

uts

Hig

hou

tput

impe

danc

eIm

peda

nce

spec

ializ

atio

nfo

rot

her

func

tions

Supp

lyvo

ltage

(V)

Hig

hest

oper

atin

gfr

eque

ncy

[2]

CC

I9

No

11

No

––

<10

0kH

z

[3]

OTA

5N

o1

1Y

es–

±12

1.0

MH

z

[6]

ZC

-CD

TA3

No

32

No

–±1

.25

10M

Hz

[8]

DV

CC

4Y

es1

3N

o–

±2.5

9.7

MH

z

[22]

CC

II6

No

31

No

–±1

222

5K

Hz

[23]

DV

CC

TA3

Yes

31

Yes

–±1

.25

1.5

MH

z

[24]

CB

TA4

Yes

13

No

–±1

.515

.9M

Hz

[26]

FDC

CII

5Y

es4

1Y

es–

±1.5

1M

Hz

[27]

MO

-CC

CC

TA2

Yes

31

Yes

–±2

.51.

8M

Hz

[30]

ICC

II4

No

11

Yes

–±2

.510

MH

z

[34]

CFT

A3

No

13

No

–±3

159

KH

z

Pro

pose

dD

V-D

XC

CI

4Y

es1

4Y

esF

irst

orde

rL

P,H

P,A

P±1

.25

15.9

MH

z


Fig. 1 Symbol of DV-DXCCI

DV-DXCCIY2

ZpY1

Xp Xn

IZp

Zn

IXp IXn

I1V1

I2V2

IZn

VXp VXn

VZp

VZn

M15 M16 M17 M18 M19 M20

Xp

M1 M2 M3 M7 M8 Xn

CcCc

M13 M11

M12 M14

M4 M5 M6 M9 M10

VDD

VSS

Y2

M21 M22 M23 M24

M25 M26

M27

M28 M29

Y1

VB2

M33

VB1

Zn

Z p

M30 M31

M32M34

Fig. 2 CMOS implementation of DV-DXCCI

[4,18,21], the differential voltage DVCC [10], and the first generation current conveyor[28]. The properties of the DV-DXCCI can be characterized as in Eq. (1) while itsCMOS structure is shown in Fig. 2. The CMOS circuitry of Fig. 2 is designed usingthe well-known DVCC (M21–M30) with unused Z-stages such that the X-terminal(gate of M24) drives the Y-terminal (gate of M2) of the well known dual-X CCII. Thefeedback from Z p to Y2 (gate of M22 to drain of M13) and Zn to Y1 (gate of M21to drain of M14) ensures first generation current conveyor property. Final Z p and Zn

stages of DV-DXCCI are realized at the drain of M33 and M31, respectively.

⎡⎢⎢⎢⎢⎢⎢⎣

I1I2VXpVXnIZpIZn

⎤⎥⎥⎥⎥⎥⎥⎦

=

⎡⎢⎢⎢⎢⎢⎢⎣

0 0 0 10 0 1 01 −1 0 0

−1 1 0 00 0 1 00 0 0 1

⎤⎥⎥⎥⎥⎥⎥⎦

⎡⎢⎢⎣

VY1VY2IXpIXn

⎤⎥⎥⎦ (1)

The CMOS circuit of DV-DXCCI is extensively simulated for various parametersof the circuit, which are tabulated as in Table 2.

The proposed CM single-input multi-output second-/first-order filter is shown inFig. 3. It employs four grounded passive components. By proper impedance special-ization (i.e., appropriate selection of resistors and capacitors), the same structure iscapable of generating second-/first-order responses. Second-order responses can beobtained when two capacitors and two resistors are used while first-order responses areobtained by selecting three resistors and a single capacitor as shown in Fig. 4a and b,


Table 2 Various simulatedparameters for DV-DXCCI

Parameter Value

Parasitic resistance at Y1 RY1= 32.4 K�

Parasitic capacitance at Y1 CY1= 9.5 fF

Parasitic resistance at Y2 RY2= 32.4 K�

Parasitic capacitance at Y2 CY2= 16 fF

Parasitic resistance at X p RXP = 29.7 �

Parasitic inductance at X p LXP = 0.75 µH

Parasitic resistance at Xn RXn = 29.3�

Parasitic inductance at Xn LXn = 0.29 µH

Parasitic resistance at Z p RZP= 32.4 K�

Parasitic capacitance at Z p CZP= 14.4 fF

Parasitic resistance at Zn RZn= 32.4 K�

Parasitic capacitance at Zn CZn= 8.04 fF

Power supply voltage ±1.25 V

Voltage transfer gain β1 = 1.01 β2 = 1.02

Current transfer gain αP = 0.999 αn = 0.999

Voltage transfer gainbandwidth

fβP = 415 MHz,fβN = 305 MHz

Current transfer gainbandwidth

fαP = 2.51 GHz,fαN = 2.51 GHz

Fig. 3 Proposed generalstructure for second-/first-orderfilter

DV-DXCCI

Y2

ZpY1

XP Xn

Iin

Z1

Z3

I01

Zn I02

Z2

V01

V02Z4

respectively. Further, second-order responses are obtained both in CM and transresis-tance mode simultaneously, while first-order responses are obtained in CM only. Thefilter employs grounded passive components which makes the proposed filter ideal forIC implementation.

Using the matrix of Eq. (1) the DV-DXCCI is characterized by:

I1 = IXn, I2 = IXp, VXp =VY1−VY2, VXn =−VY1+VY2, IZp = IXp, IZn = IXn.

(2)From the circuit in Fig. 3, the current and transresistance mode transfer functions byusing Eq. (2) can be obtained as:

Current transfer functions:

Io1

Iin= Z1 Z3

Z1 Z2 + Z2 Z3 + Z3 Z4, (3)


DV-DXCCI

Y2

ZpY1

XP Xn

Iin

R1

I01

Zn I02

R2

V01

V02C2

C1

DV-DXCCI

Y2

ZpY1

XP Xn

Iin

C1

I01

Zn I02

C2

V01

V02R2

R1

(a) (b)

(c) (d)

DV-DXCCI

Y2

ZpY1

XP Xn

Iin

R1

I01

Zn I02

R2

V01

V02R3

C

DV-DXCCI

Y2

ZpY1

XP Xn

Iin

R1

I01

Zn I02

CV01

V02

R2

R2

Fig. 4 Proposed cascadable filters: a, b second order c, d first order

Io2

Iin= Z1 Z2

Z1 Z2 + Z2 Z3 + Z3 Z4. (4)

Transresistance transfer functions:

Vo1

Iin= Z1 Z2 Z3

Z1 Z2 + Z2 Z3 + Z3 Z4, (5)

Vo2

Iin= Z1 Z3 Z4

Z1 Z2 + Z2 Z3 + Z3 Z4. (6)

The function of the circuit shown in Fig. 3 can be explained with the help of Tables 3and 4.

Table 3 depicts CM and transresistance mode transfer functions. In Case 1, low-pass (LP), and inverting high-pass (HP) responses are obtained simultaneously athigh impedance terminals. In addition to CM outputs, transresistance mode LP, andband-pass (BP) outputs are also obtained simultaneously without the need of addi-tional components. Functions obtained in Case 2 are BP and inverting LP, while intransresistance mode inverting LP and BP are obtained.

Table 4 shows that the same structure provides first-order transfer functions byappropriate impedance specialization. CM LP and inverting HP responses are simul-taneously obtained at high impedance terminals in Case 3. In Case 4, HP and invertingLP responses are obtained simultaneously. By connecting HP and LP responses, the


Tabl

e3

Impe

danc

esp

ecia

lizat

ion

for

seco

nd-o

rder

filte

rs

Filte

ror

der

Impe

danc

esp

ecia

lizat

ion

Cur

rent

tran

sfer

Func

tions

Tra

nsre

sist

ance

func

tions

Seco

ndor

der

Cas

e1:

Fig.

4aZ

1=

R1,

Z2

=R

2,

Z3

=1/

sC1,

Z4

=1/

sC2

I BP

I in

=I 0

1I i

n=

s/C

1R

2D

1(s

)

I HP

I in

=I 0

2I i

n=

−s2

D1(s

)

VB

PI i

n=

V01 I in

=s/

C1

D1(s

)

VL

PI i

n=

V02 I in

=1/

C1

C2

R2

D1(s

)

Cas

e2:

Fig.

4bZ

1=

1/sC

1,

Z2

=1/

sC2,

Z3

=R

1,

Z4

=R

2

I BP

I in

=I 0

1I i

n=

s/C

1R

2D

2(s

)

I LP

I in

=I 0

2I i

n=

−1/

C1

C2

R1

R2

D2(s

)

VL

PI i

n=

V01 I in

=−

1/C

1C

2R

2D

2(s

)

VB

PI i

n=

V02 I in

=−

s/C

1D

2(s

)

D1(s

)=

s2+

sC

1R

1+

1C

1C

2R

1R

2,

D2(s

)=

s2+

sC

2R

2+

1C

1C

2R

1R

2


Tabl

e4

Impe

danc

esp

ecia

lizat

ion

for

first

-ord

erfil

ters

Filte

ror

der

Impe

danc

esp

ecia

lizat

ion

Cur

rent

tran

sfer

func

tions

Firs

tord

erC

ase

3:Fi

g.4c

Z1

=R

1,

Z2

=R

2,

Z3

=1/

sC,

Z4

=R

3

I LP

I in

=I 0

1I i

n=

1/C

RD

3(s

)

I HP

I in

=I 0

2I i

n=

−s

D3(s

),

I AP

I in

=(s

−1/C

R)

D3(s

)

Taki

ngR

1=

2R

2an

dR

3=

R2

=R

Cas

e4:

Fig.

4dZ

1=

R1,

Z2

=1/

sC,

Z3

=R

2,

Z4

=R

3

I HP

I in

=I 0

1I i

n=

s2

D3(s

),

I LP

I in

=I 0

2I i

n=

−1

2CR

D3(s

),

I AP

I IN

=(s

−1/C

R)

2D

3(s

)

Taki

ngR

1=

R2

=R

and

R3

=2

R2

=R

D3(s

)=

s+

1 CR


inverting all-pass (AP) and non-inverting AP responses can also be obtained withoutadditional components in Case 3 and Case 4, respectively.

The pole frequency (ω0) and pole-Q of the second-order filter can be obtained fromTable 3 as:

ω0 =√

1

C1C2 R1 R2, Q =

√C1 R1

C2 R2. (7)

From Eq. (7) it is clear that the sensitivity of the filter parameters is analyzed andfound to be within half in magnitude for pole-frequency and pole-Q.

Selecting R1 = R2 = R and C1 = C2 = C , the pole frequency and pole-Q of thesecond-order filter can be expressed as:

ω0 = 1

C R, Q = 1. (8)

3 Practical Considerations

3.1 Non-ideal Analysis

This section deals with the non-ideal analysis of the circuit proposed in Fig. 3. Thematrix equation defining a non-ideal DX-DVCCI may be given as:

⎡⎢⎢⎢⎢⎢⎢⎣

I1I2VXpVXnIZpIZn

⎤⎥⎥⎥⎥⎥⎥⎦

=

⎡⎢⎢⎢⎢⎢⎢⎣

0 0 0 αp

0 0 αn 0βp −βn 0 0−βp βn 0 00 0 γp 00 0 0 γn

⎤⎥⎥⎥⎥⎥⎥⎦

⎡⎢⎢⎣

VY1VY2IXpIXn

⎤⎥⎥⎦ , (9)

where αp(αn) is the current transfer gain from the X to Y1(Y2), βp(βn) is the voltagetransfer gain from the Y1(Y2) to X p(Xn) and γp(γn) are the current transfer gain fromthe X to Z p(Zn). Using Eq. (9) the non-ideal second- order and first-order transferfunctions can be expressed as shown in Tables 5 and 6, respectively.

For second-order filter the non-ideal pole frequency and pole-Qn can be expressedas:

Case1 : ω0n =√

αpβ2

αnβ1C1C2 R1 R2, Qn =

√αnαpβ1β2C1 R1

C2 R2, (10)

Case2 : ω0n =√

αnβ1

αpβ2C1C2 R1 R2, Qn =


C2 R2(11)

From Eqs. (10) and (11), the active and passive sensitivities of filter performance areanalyzed and found to be within 0.5 in magnitude.


Tabl

e5

Non

-ide

altr

ansf

erfu

nctio

nsof

the

seco

nd-o

rder

filte

r

Filte

ror

der

Impe

danc

esp

ecia

lizat

ion

Cur

rent

tran

sfer

func

tions

Tra

nsre

sist

ance

func

tions

Seco

ndor

der

Cas

e1:

Figu

re4a

Z1

=R

1,

Z2

=R

2,

Z3

=1/

sC1,

Z4

=1/

sC2

I 01

I in

=s

γp

αn

C1

R2

Dn1

(s)

,I 0

2I i

n=

−s2

γn

αn

Dn1

(s)

V01 I in

=s

1α

nC

1D

n1(s

),

V02 I in

=α

pα

nC

1C

2R

2D

n1(s

)

Cas

e2:

Figu

re4b

Z1

=1/

sC1,Z

2=

1/sC

2,

Z3

=R

1,

Z4

=R

2

I 01

I in

=s

γpβ

1α

pβ

2C

1R

2D

n2(s

),

I 02

I in

=

−γ

nβ

1α

nβ

2C

1C

2R

1R

2D

n2(s

)

V01 I in

=

−β

1α

pβ

2C

1C

2R

2D

n2(s

)

V02 I in

=−

s(β

1β

2C

1

)

Dn2

(s)

Dn1

(s)=

s2+

sα

nβ

1C

1R

1+

αpβ

2α

nβ

1C

1C

2R

1R

2,

Dn2

(s)=

s2+

sα

pβ

2C

2R

2+

αnβ

1α

pβ

2C

1C

2R

1R

2


Table 6 Non-ideal transfer functions of the first-order filter

Filter order Impedancespecialization

Current transfer functions

First order Case 3:Fig. 4c

Z1 = R1, Z2 =R2, Z3 =1/sC, Z4 = R3TakingR1 = 2R2 andR3 = R2 = R

I01Iin

=γp

αnC RDn3(s) ,

I02Iin

= − s γnαn

Dn3(s) ,IAPIIN

=

−[s− γp

γnC R

]

Dn3(s)

Case 4:Fig. 4d

Z1 = R1, Z2 =1/sC, Z3 =R2, Z4 = R3TakingR1 = R2 = Rand R3 =2R2 = 2R

I01Iin

= s(

γpβ1αpβ2

)

2Dn4(s) ,I02Iin

=

−γnβ1αpβ2

2C RDn4(s) ,

IAPIIN

=γpβ1

2αpβ2

[s− γn

γpC R

]

Dn4(s)

Dn3(s) = s + R+αpβ2 R2αnβ1C R2 , Dn4(s) = s + R+αnβ1 R

2αpβ2C R2

For integrated circuit realizations αp = αn = α and β1 = β2 = β, then Eqs. (10)and (11) reduce to:

Case1 : ω0n =√

1

C1C2 R1 R2≈ ω0, Qn =


C2 R2≈ αβQ, (12)

Case2 : ω0n =√

αnβ1

αpβ2C1C2 R1 R2≈ ω0, Qn =


C2 R2≈ αβQ. (13)

Equations (12) and (13) show that non-ideal pole frequency approaches to ideal valuewhen αp = αn = α and β1 = β2 = β.

The non-ideal pole frequency and pole-Qn for the first-order filter can be analyzedfrom Table 6 as:

Case1 : ω0n = (1 + αpβ2)

2αnβ1C R, (14)

Case2 : ω0n = (1 + αnβ1)

2αpβ2C R(15)

Equations (14) and (15) are analyzed for sensitivities which are found to be unity inmagnitude, thus exhibiting good sensitivity performance.

As far as the circuit topology is concerned, it is to be noted that resistive X-terminations are desirable from the viewpoint of absorbing the intrinsic X-terminalresistance, while capacitive X-terminations do pose high frequency limitations. More-over, the capacitive terminations at Y and Z terminals are also desirable from the par-asitic capacitance absorption at those terminals [11]. Many recent works have often


presented general parasitic considerations in current conveyor-based circuits [16,17].Thus, the next study is on the effect of parasitic present at X-terminals in the proposedcircuits of Fig. 4, in which a capacitor is connected at one of the X-terminals whilethe other X-terminal has an external resistor. If the effect of finite X-terminal parasiticresistance (Rx1, Rx2 for Xn , X p, respectively) is taken into account, the second ordercurrent transfer functions for Fig.4a and b are modified as:

For Case 1, referring to Fig. 4a the modified band-pass (Eq. 16) and high-pass(Eq. 17) transfer functions become:

IBP

Iin= I01

Iin= s R1

C1(R1+RX1)(R2+RX2)+s2 R1 RX1

(R1+RX1)(R2+RX2)

s2 + s[

1C1(R1+RX1)

+ RX1C2(R1+RX1)(R2+RX2)

]+ 1

C1C2(R1+RX1)(R2+RX2)

, (16)

IHP

Iin= I02

Iin=− s2 R1

(R1+RX1)

s2+s[

1C1(R1+RX1)

+ RX1C2(R1+RX1)(R2+RX2)

]+ 1


. (17)

Similarly for Case 2, referring to Fig. 4b the modified BP (Eq. 18) and low-pass(Eq. 19) transfer functions become:

IBP

Iin= I01

Iin=

sC1(R2+RX2)

s2+s[

1C2(R2+RX2)

+ RX2C1(R1+RX1)(R2+RX2)

]+ 1


(18)

ILP

Iin= I02

Iin=−

1C1C2(R1+RX1)(R2+RX2)

+s RX2C1(R1+RX1)(R2+RX2)

s2+s[

1C2(R2+RX2)

+ RX2C1(R1+RX1)(R2+RX2)

]+ 1


(19)

Equation (16) shows that additional zero (of order 2, at origin) is introduced inthe BP current transfer function with a high-pass nature; whose gain is very small,since Rx values are quite small (measured as ∼30 Ohms) in comparison to externalresistor. The high-pass current transfer function in Eq. 17), is not altered except forsome deviations in filter parameters namely, pole-frequency and pole-Q, an effect alsopresent in BP case.

Equation (18) shows that the BP function remains free from any additional zerowhile Eq. (19) shows that the low-pass transfer function is also disturbed by thepresence of an additional zero, with the BP nature, whose gain is again quite small, withthe aforementioned reasoning. However, the additional zero’s appearance (whereverapplicable out of four above eqns.) does cause for high frequency errors. This is dueto the fact that at high frequencies, the gain term tends to increase, proportional to thepower of s, with s replaced by j2π f , f being the frequency of operation.

3.2 Applications as 4th and 6th Order Filters

Some of the recent works reemphasize the realization of higher-order CM filter [13,35].The proposed second-order filter is employed to realize 4th and 6th order filter using


Table 7 Transistors Aspectratios used in simulation

MOS Transistors Aspect ratio

M1, M2, M3, M4, M5, M15, M16,M17, M18, M19, M20, M32, M34

2/0.25

M3, M6, M7, M8, M9, M10 4/0.25

M11, M12, M13, M14, M31, M33 16/0.25

M21, M22, M23, M24 1/0.25

M25, M26, M27 5/0.25

M28, M29, M30 3/0.25

DV-DXCCI

Y2

Y1

XP Xn

Iin

C1

ZnILP

C2

R2

R1

DV-DXCCI

Y2

Y1

XP Xn

C3

Zn

C4

R4

R3

DV-DXCCI

Y2

Y1

XP Xn

C5

Zn

C6

R6

R5

Fig. 5 6th order Butterworth LP filter

VY1- VY2

-100mV 0 200mV

-200mV

0

200mV VXn VXp

-200mV 100mV

Am

plitu

de

Fig. 6 DC transfer characteristic showing VXn(VXp) with respect to VY1–VY2

cascade approach. Table 7 summarizes the design of so obtained filters as shown inFig. 5.

4 Design and Verification

To verify the proposed theory, the new active device DV-DXCCI and the filters aresimulated using PSPICE with TSMC 0.25 µm technology. The MOS transistor aspectratios for the CMOS DV-DXCCI are given in Table 7. The supply voltage is takento be ±1.25V. The dc transfer characteristic of DV-DXCCI is shown in Fig. 6 whichshows that the linear differential input range is ±200 mV. The circuit was designedfor a pole frequency ( f0) = ωo/2π = 15.9 MHz. The passive components were chosenas R1 = R2 = R =1 K� and C1 = C2 = C = 10 pF. Figure 7 shows that thesecond-order BP, LP and HP responses are obtained at pole-Q = 1. The BP response


Frequency1.0MHz 10MHz 100MHz

-80

-40

0

20

Gai

n (d

B)

Fig. 7 Frequency response of second-order BP, LP, and HP at Q = 1

100KHz 1.0MHz 10MHz 100MHz-150

-100

-50

Gai

n (d

B)

Q =10

Frequency

Fig. 8 Frequency response of second-order BP at Q = 10

at Q = 10 which is obtained by selecting in Fig. 4b R1 = R2 = R = 1 K�, C1 =1 pF and C2 = 100 pF is shown in Fig. 8. Next, the first-order filter responses HP andLP are used to obtain the AP response which is also designed for a pole frequency of15.9 MHz by selecting R1 = 2R2 = 2R3 = 2 K� and C = 10 pF. The time domainresponse of AP is also shown in Fig. 9 at 15.5 MHz with THD of 0.8 %. The Fourierspectrum of the AP output signal for the applied signal frequency of 15.5 MHz isgiven in Fig. 10. The proposed circuit is next used to implement 4th and 6th orderButterworth responses. The design of these higher-order filters are given in Table 8.Figure 11 shows 4th and 6th order LP responses for pole frequency of 1 MHz. It showsthe stopband attenuation of 80 dB/decade and 120 dB/decade for the 4th and 6th orderLP function, respectively.


Time100ns 150ns 200ns 250ns 300ns 350ns 400ns

-10uA

0

10uA

-10uA

0

10uAA

mpl

itude

Am

plitu

de

Fig. 9 Time domain input and output of first-order AP filter at 15.5 MHz

Frequency

0Hz 25MHz 50MHz 75MHz

-175

-150

-125

Out

put C

urre

nt (

dB)

-100

Fig. 10 Fourier spectrum of Fig. 9 output showing a minimum of −35 dB suppression of harmonics

Table 8 Passive componentvalues for 4th and 6th orderButterworth filter responses

Filter order Passive components values Pole-Q values

4th order R1 = R2 = R3 = R4 = R =1 K� C1 = 121 pF, C2 =208 pF, C3 = 121 pF, C4 = 294pF

Q1 = 1.3071andQ2 = 0.5411

6th order R1 = R2 = R3 = R4 = R5 =R6 = R =1 K� C1 = 82 pF, C2 =307 pF, C3 = 225 pF, C4 = 112pF, C5 = 307 pF, C6 = 82 pF

Q1 = 1.93, Q2= 0.707 andQ3 = 0.516


Gai

n (d

B)

Frequency10KHz 30KHz 100KHz 300KHz 1.0MHz 3.0MHz 10MHz 30MHz 100MHz

-150

-100

-50

4th order

6th order

-0

Fig. 11 Frequency response of 4th and 6th order Butterworth LP filter

5 Conclusion

A generalized circuit for realizing first and second-order filters with grounded com-ponents is introduced in this paper. The circuit is built around a new proposed activeelement namely the differential voltage dual-X first generation current conveyor. Thesingle current input circuit realizes two output current functions and two voltagefunctions thereby providing current transfer functions and transresistance functions,respectively. Exhaustive non-ideal study of the proposed filter is included. The appli-cation of the circuit in realizing higher-order filters is also demonstrated. As far asthe applications of current conveyors are concerned, these active elements continueto find recent attention in the literature [15,20,33]. The new proposed DV-DXCCIis also expected to find an increasing number of linear and non-linear analog signalprocessing applications.

Acknowledgments Thanks are due to anonymous reviewers for useful comments and editors for recom-mending the paper.

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http://dx.doi.org/10.1002/cta.1880

generalized filter topology using grounded components and single novel active element

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