Experimental HF circular array withdirection finding and null steering

capabilitiesN. Karavassilis. Ph.D., Prof. D.E.N. Davies. D.Sc. F.I.E.E.. F.Eng., F.R.S.. and

C.G. Guy. B.Sc. C.Eng., M.I.E.E.

Indexing terms: Antennas (Arrays), Antennas(Adaptive)

Abstract: The paper describes an HF-circular-array receiving and direction finding system which also offers acapability for null steering. The experimental array consists of four wideband active loop antennas which feed a4 x 4 Butler matrix. The action of the Butler matrix is to analyse the signals received by the array in terms ofphase-mode components. Direction finding can then be carried out by measuring the phase difference betweenphase-mode outputs of the matrix, and either beam or null formation can be achieved by combining such phasemodes with appropriate phase and amplitude weightings. The experimental system uses a microcomputer basedadaptive technique for the bearing measurement. The advantage of this technique is that it employs a singleunmodified HF receiver and it also offers a capability for null steering by means of software control of thereceiving system.

1 Introduction

HF receiving systems employ wide bandwidth antennas(2-30 MHz) and often need 360° azimuth coverage plussome antenna directivity capable of improving the ratio ofwanted signal to the high level of background interference.There may also be requirements for a direction findingfeature or a null steering capability (where the suppressionof a single high-level interfering signal is more valuablethan general reduction of background). This paperdescribes an HF receiving array based on circulargeometry offering both direction finding and null steeringfeatures and incorporating microcomputer based arraypattern control plus a standard HF receiver. This micro-computer is used to control the array performance and todisplay data such as the mean and standard deviation ofsignal bearing.

The concept employs active loop antenna elements,which can also be transportable units. The experimentalarray employs only four elements in a circular array con-figuration, but the concept is directly applicable to largerarrays consisting of 8, 16 or 32 elements for increased per-formance. .The work is based upon the excitation of circu-lar arrays in terms of phase modes. This approach tocircular arrays has been studied at University CollegeLondon for several years, but here it is combined withwideband active loop antenna elements plus a simple formof single loop adaptive receiver which measures theazimuth angle of arrival while making use of a single stan-dard HF receiver. The project has been a collaborativeexercise between the Electronic and Electrical EngineeringDepartment at UCL and C & S Antennas.

2 Circular arrays

Circular arrays appear to represent an obvious choice forthose antenna applications requiring 360° coverage wherethere is a need to steer beams and nulls, or to performdirection finding. In practice, their use has been somewhatrestricted, as electronic rotation of a circular array excita-tion function, in general, requires variable control of both

Paper 4443H (Ell), first received 25th March and in revised form 7th November1985

The authors are with the Department of Electronic and Electrical Engineering, Uni-versity College London, Torrington Place, London WC1E 7JE, United Kingdom

amplitude and phase at each antenna element. This is farmore complex than the linear array case, which requiresphase control alone coupled with fixed amplitude tapers.However, by making use of the symmetrical properties ofcircular arrays it is possible to excite the arrays in terms ofa set of excitation functions called phase modes. Theseoffer complete flexibility for the design of directional pat-terns, plus the ability to rotate beams and nulls or topreform direction finding by the use of phase control (orphase measurement) alone. It is convenient to give a briefdescription of the properties of phase modes here beforeproceeding to describe the HF direction finding array, butreaders are referred to Reference 1 for more details.

Consider a circular array in which the interelementspacing is sufficiently small so that the array excitationF(6) can be considered as a continuous function, given by

F(6)= jNO

where

(1)

The excitation terms of eqn. 1 represent N complete cyclesof phase variation round the array. For the case N = 1, theamplitude excitation of the array is constant and the phasevaries linearly with angle through one complete cycle asthe azimuth direction varies through 360°. For otherinteger values of N, the excitation will comprise N com-plete cycles of phase change round the circumference.Clearly such excitations are radically symmetrical, and theresultant far-field patterns must be similarly symmetrical.It can be shown that this far-field pattern is given by

JNO (2)

Where r is the array radius. We see that this far-fieldpattern has exactly the same form as the array excitation,although the amplitude of this function depends upon thearray radius and (if we consider angles outside the plane ofthe circle) upon elevation angle. These excitation terms arecalled phase modes. Negative phase modes correspond tophase variations in the reverse direction with azimuth, andthe zero-order phase mode corresponds to constant ampli-tude with zero phase change, resulting in an omnidirec-tional far-field pattern with no phase variation. Such

IEE PROCEEDINGS, Vol. 133, Pt. H, No. 2, APRIL 1986 147

phase-mode terms may be regarded as the exponentialFourier terms of a far-field directional pattern, and theconcept can therefore be used as a far-field-pattern synthe-sis technique. It has also been shown that, in order for theabove properties to apply, the interelement spacing of thearray should be X/2 or less at the highest operating fre-quency of the array, otherwise higher-order mode distor-tion terms will appear [1] in the shape of the pattern.

To make use of such phase modes it is necessary todevise a network which feeds the circular array. Thisphase-mode forming network would have N output portsconnected to the circular array and N input ports, each ofwhich so excite the array to produce the N separatemodes. For N elements there are only N independentphase modes that can be formed without ambiguity. Thesephase modes and the corresponding far-field patterns rep-resent, mathematically, an orthogonal set. The Butlermatrix network (often used for forming multiple beams forlinear arrays) may be used to form the phase modes forcircular arrays [2].

four elementarray

Butlermatrix

0 2 - 1phase-mode outputs

180#hybrid

90 hybrid

Fig. 1 A four element array feeding a Butler matrix

148

= 180*

= -90*

3 Direction finding and null steering with circulararrays

We will consider the example of a four-element circulararray fed from a 4 x 4 Butler matrix network, as shown inFig. 1. It should be noted that, although the conventionalway of depicting a four-port Butler matrix is to use four180° hybrids plus a 90° phase shifter, we have used three180° hybrids and one 90° hybrid. The effect is identical,but ours has the advantage that the phase shift incorpo-rated within the 90° hybrid will have a wide bandwidthproperty. The four mode outputs correspond to the0 + 1 — 1 and ± 2 modes. This second-order mode turnsout to have identical excitation for the plus or minus direc-tions, as both would correspond to a 180° phase shiftbetween adjacent elements.

If such an array is used on reception, the output of thezero-order mode will give an amplitude and phase which isindependent of the azimuth direction of any far-fieldsource, neglecting the mode distortion terms which will bediscussed later. The output from the + 1 mode would havea constant amplitude with a phase shift (relative to the far-field source or relative to the zero-order mode) which waslinearly dependent upon azimuthal bearing. Thus, if wemeasure the difference in phase in the zero- and first-orderphase mode for any received signal, it will give the bearing.This is the basis of the direction finding systems.

This direct relationship between phase shift and bearingapplies to any pair of phase modes whose order differ byone. We could therefore use the zero and — 1 or the — 1and — 2 modes. If we used the +1 and — 1 modes, themeasured phase shift would double, leading to an increaseof bearing accuracy but also to a bearing ambiguity.Clearly, the output information from the four modes maybe used in a variety of ways to obtain the best bearinginformation.

Ideally, the relation between the measured phase shift <Pand the signal bearing 6 should be a straight line throughthe origin of unity slope (or of slope N, where the differ-ence in mode order is N). Analysis has shown that theactual shape of this graph is modified (by the mode distor-tion terms) giving small cyclic errors of bearing measure-ment. These errors are a function of interelement spacing,mode order and the azimuthal shape of the element direc-tional pattern. In the case of four-element circular arraysemploying omnidirectional elements, these errors can beconsidered to be equivalent to the well known octantalbearing errors of a four-element Adcock directionfinder[3]. The effect of such errors may be minimised byemploying circular arrays with adequately close elementspacing. However, as the errors are completely predictable,an alternative approach is to use a microcomputer to storecorrection curves and compensate for such errors. This isthe approach adopted in the experimental system.

We can also use the arrangements of Fig. 1 to steerdirectional nulls. If we combine the output from twoadjacent-order modes (modes whose order differ by one),then the resulting pattern will always contain one directionin which the phases of the two modes are in opposition.This gives rise to one null, and the resulting pattern will bea cardioid. A variation in phase of $ applied to eithermode before combination will thus produce a rotation ofthe direction of the null by <j) degrees in azimuth. Thisprocess of null steering with circular arrays has beendescribed in a number of previous experimental systems[4]. Use may be made of more than two phase modes inorder to reduce null width or to generate more than oneindependently steerable null [5]. The maximum number of

IEE PROCEEDINGS, Vol. 133, Pt. H, No. 2, APRIL 1986

independently steerable nulls is (N — 1) for N phase modes(and N elements) as expected.

From this discussion, it is easy to see that a receivingsystem with access to such phase modes can use them fordirection finding or for null steering, or for both simulta-neously. This opens the way to a number of extremelydesirable features, including the capability of measuringthe bearing of a high-level signal, employing a null toremove its effect and then measuring the bearing of asecond lower-level signal in the presence of the nulled outhigh-level interference.

4 Experimental receiver

The objective of this study was to produce an experimentalfour-element HF receiving system capable of operatingover part of the HF band (2-30 MHz) with computercontrol. In order to compute the expected direction findingperformance of the system, the experimental measured pat-terns of the active loop antennas used in the arrays wereused to compute the expected performance of bearingagainst phase shift (or indicated bearing). These were alsocomputed as DF error against indicated bearing, (errorcorrection curves) for a range of frequencies (3, 6, 15 and30 MHz).

Fig. 2a indicates the predicted value of indicatedbearing against true bearing for the performance of such adirection finding system at 6 MHz using the +1 and — 1modes. Fig. 2b shows the same information presented as aDF error against indicated bearing response. Figs. 2c and2d represent the same information for 15 MHz. The fre-quencies relate to the measurement of the respective polar

0 90true bearing, degree

abearing, degree

b

90

£ 0

i °-2

0 90true bearing,degree bearing , degree

d

90

Fig. 2 DF curves for + 1, —I phase mode comparison, for a fourelement array of active loop antennas

a and b Radius of array = 3.20 m, frequency = 6 MHz, interelement spacing inwavelengths = 0.10c and d Radius of array = 3.20 m, frequency 15 MHz, interelement spacing inwavelengths = 0.25

responses for the loop elements. It will be observed thatthe DF error curves display a noise-like component. Thisarises from quantisation and interpolation effects withinthe computation, which occurred because the basic direc-tional response of the HF loop was only known at a verylimited number of angles.

Fig. 3A shows a corresponding DF error response at6 MHz, where the error curve displays an approximatelysinusoidal form with a peak error of about 8°. By

assuming a theoretical sinusoidal model for this error func-tion, as shown in Fig. 3B, the predicted peak error can be

10r

bearing, degree

Fig. 3A DF error against indicated bearing for a four-element array andfor — 1, +1 phase mode comparison. Frequency 6 MHz

& 8

enT3

ot 0itcn

-890

bearing .degree

Fig. 3B First-order theoretical sinusoidal approximation to the DFerror curve of Fig. 3 A

true bearing,degree90

Fig. 3C Indicated against true bearing before the application of anysinusoidal correction

+ 1 , -1 phase mode comparison. Four element array. Frequency 6 MHz

cn 90•8

= 0

true bearing.degree90

Fig. 3D Indicated against true bearing after the application of sinus-oidal corrections+ 1 , -1 phase mode comparison. Four element array. Frequency 6 MHz

reduced to less than 3°, and a second-order sinewave cor-rection can reduce this to 1.6°. In practice, the potentialvalue of such correction terms will depend upon the appli-cation, and it is important to realise that such terms willhave to be separately stored for a variety of frequenciesover the band. Fig. 3C and 3D show the change in theresponse of indicated against true bearing for the basic DFperfc rmance and with first-order correction.

Cne of the purposes of the experimental receiver was toevaluate such bearing error correction schemes, but it wasfirst necessary to devise a satisfactory way of making thephase measurement corresponding to the bearing determi-nation. The simplest approach would appear to involveincorporating identical HF receivers after the two phasemodes (whose relative phase shift is to be determined) and

IEE PROCEEDINGS, Vol. 133, Pt. H, No. 2, APRIL 1986 149

to make the phase measurement from the IF outputs. Thedifficulty with this approach is that it requires two nearidentical HF receivers running from a common local oscil-lator, plus adequate confidence in the relative phase track-ing of the two separate receivers. Systems based upon fastswitching the receiver between two of the phase modes arealso possible, although this can have the disadvantage ofincreasing signal bandwidths and complicating the abilityto use the receivers for communication reception whilemaking bearing measurements.

Fig. 4 represents a form of adaptive processing used to

phaseperturber

controlphaseshifter

perturbingoscillator

PSD

B'

bandpass,filtertunedat L

detected

amplitude modulationat in

Fig. 4A Basic outline of the adaptive processing scheme incorporated inthe experimental phase comparator unit

signals not in phase

Fig. 4B Phasor diagrams of the signals input to the phase comparatorunit and their resultant

make the bearing measurement with a single receiver. Theoutput from one of the phase modes is passed via an elec-tronically controlled phase shifter and a phase perturbingdevice which imposes a periodic phase deviation on thisphase-mode signal (at a rate of 70 Hz). The output fromthis phase perturber is combined with the output from theother phase-mode port and the resultant signal thusobtained is fed to a single unmodified HF receiver. Inpractice, we have chosen a deviation of ± 90°, but this isnot critical. As a result of the above phase modulation, thecombined signal from the two modes will contain envelopemodulation at the same 70 Hz frequency. If this envelope

modulation is detected and the phase of the demodulatedsinewave compared with the original 70 Hz, then this com-parison can be used to produce a servoerror signal used tochange the quiescent phase shift point of the phase modu-lator to bring the two modes into phase coincidence. Thisalso points the maximum of the cardioid directionalpattern in the direction of the incoming signal, a 3 dBincrease of directional gain.

Various forms of feedback systems are possible, but theuse of a microcomputer to control the system suggestedquantised techniques. The above concept merely describesa way of driving the control phase shifter to point themaximum amplitude of the cardioid directional patterntowards the incoming signal. However, if the control phaseshifter is calibrated (electronically) in phase shift, then thevalue so chosen represents the uncorrected value of thesignal bearing. We therefore see that this simple adaptivesystem has the dual advantage of making a bearing mea-surement based upon a single conventional form of HFreceiver and, in addition, achieves a small improvement inSNR (or directivity) by pointing the main beam of thearray towards the incoming signal.

The computer can also collect and process DF statisticswhile the receiver is in operation, or can be instructed tosteer a null in one particular direction while listening in adifferent direction. The optimum way of achieving thislatter function requires control of amplitude and phase ofone phase mode, and, although this has not been incorpo-rated in the initial experimental model, work is in progressto incorporate this capability. The choice of a symmetricalpair of phase modes (e.g. +1) will result in reduced levelsof bearing error terms, in addition to the factor of two inphase sensitivity. For this reason, the experimental receiverhas access to the —1,0 and + 1 modes for processing twoat a time.

5 Details of experimental system

The requirement for the HF DF system described here,was to provide DF measurements from 0° to 360° in stepsof 1° and maintain its performance for the whole of the HFband. The requirement for wide bandwidth suggested theuse of 'In phase' and 'quadrature' channels for phaseshifter design. Correct bipolar amplitude control of therelative weighting of these two components before they arecombined gives any desired phase shift.

In this particular design, a wideband 90° hybrid wasused to split the signal into I and Q channels. Control ofthe relative weighting of these two channels is provided bythe use of two (type PAS-1) electronic attenuators. Varia-tion in the current injected into the control input of theseattenuators results in control of the amplitude of the mainRF signal through them, whereas reversal of the directionof this current results in a 180° phase shift. The correctcurrents into the control inputs of the attenuators wereprovided by a combination of an EPROM and a D-A con-verter for each attenuator, as shown in Fig. 5. The first 360locations of the EPROMs were loaded with appropriatedigital words, each of which corresponded to a particularattenuation setting of the corresponding electronic attenu-ator. The digital words were chosen in such a way thatwhen the two EPROMs were commonly addressed by amodulo-360 digital counter, as shown in Fig. 5, the overallphase shift introduced by the phase shifter to the main RFsignal, would be equal in degrees to the counter output.

The device that produces the phase perturbation issimilar in design to the main shifter. This phase perturberis introduced into the other phase mode channel, as shown

150 IEE PROCEEDINGS, Vol. 133, Pt. H, No. 2, APRIL 1986

in Fig. 6. In this way all the phase delays introduced by thehybrids balance out between the two channels. For simpli-city in the current prototype system, the phase perturber

RF input

JH-6-4

PA5-1

PA5-1

digitalwordfrom outputof digitalcounter

D/A EPROM

X D/A EPROM

"output

Fig. 5 Outline of experimental I-Q phase shifter

Omodephaseperturbator

-1 mode

phaseshifter

into the receiver

Fig. 6 Outline of experimental phase perturbator

produces a ±90° square-wave phase modulation. This isdone by completely attenuating the I channel and feedingthe Q channel attenuator with a control signal thatremains constant in amplitude but alternately reverses itssign at the required frequency f0, as shown in Fig. 6. Whenno phase perturbance is required (as in the case of a null orbeam steering procedure), the control input to the Qchannel attenuator is connected to a constant positivesignal.

An unmodified RACAL RA17 receiver is used for theresultant signal. The envelope detection is performedwithin the receiver itself, which provides a basebandoutput. The fundamental component of the amplitudemodulation produced by the introduction of the phaseperturbance is filtered out by a 10 Hz wide fourth-orderChebyshev active filter. The output of this filter is theneither in phase, or antiphase, with the square-wave signalthat drives the phase perturber depending on which of thetwo phase modes is phase advanced relative to the other.A digital phase sensitive detector (PSD) is used to checkthe phase of the detected signal relative to the perturbingsignal. The output of the digital PSD is a binary bit whichindicates whether one of the phase modes has to be phaseadvanced or phase delayed. The PSD output is thus usedto control the direction of the counter that drives thephase shifter, as shown in Fig. 7. The microcomputer auto-

•1mode Omode

7--RFswitch

-1 mode (used as phasereference)

perturbingsquare wavesignal at Fo

phaseaerturbator

phaseshifter

digitalcounterU/D

tbaseband

BPF tuned at F

digitalPSD

binaryerrorsignal

Fig. 7 Block diagram of experimental signal processing unit

matically reads the settings of the phase shifter and, byaveraging them, obtains the required phase difference. Anaccurate bearing is obtained by comparing the phases of+ 1 and —1 modes, but with a 180° ambiguity which isresolved by a rough measurement carried out using 0 and— 1 modes.

The experimental system was primarily designed tooperate for speech broadcast signals, which typicallyoccupy the frequency range between 300 Hz to 3 KHz in a3 KHz HF channel. For this reason, a 70 Hz phase modu-lation frequency was chosen, which is below the usableinformation bandwidth. In practice though, it was foundthat HF signals often carried some information at fre-quencies below 300 Hz which could cause the measuredbearing to oscillate around the correct value. Overall, thesystem has a satisfactory performance for AM signals,high-frequency teletype signals, PSK and FSK modulatedsignals, It was also made to handle OOK signals by theintroduction of an RF level detector as shown in Fig. 8.An IF output from the receiver was used for this purposein conjunction with a RMS to DC converter and a voltagecomparator. When the RF signal is in the OFF state, the

IEE PROCEEDINGS, Vol. 133, Pt. H, No. 2, APRIL 1986 151

IF level drops below a specified threshold and the counteris disabled from clocking on. The system in its present

digitalcounter

H=}-Hl'

voltagecomparator

set thresholdlevel

Fig. 8 Outline of the feedback path of the adaptive loop after the inclu-sion of the RF level detector

form will not operate with SSB signals, but plans are inhand to overcome this limitation.

In the present experimental system, the microcomputercan both read and control the setting of the phase shifter.This means that the microcomputer can also set the phaseshifter in such a way that the two phase modes add eitherin phase (in the case of beam formation) or in antiphase (inthe case of null formation) at a particular bearing. For adeep null to be formed, the amplitudes of the two phasemodes should be equalised before they are added in anti-phase. This equalisation is currently limited by the factthat the two phase modes have gains which vary over theHF band and, at the higher end of the frequency range,contain mode-distortion terms resulting in cyclic gainripples in the mode patterns. Despite the lack of such anamplitude equalisation, nulls of up to 22 dB deep havebeen observed and steered in the direction of varioustransmissions. Clearly, if the adaptive receiver had theability to control both amplitude and phase of the phasemode signals, then deep nulls could be steered in additionto a direction finding capability.

6 Operation of experimental system

The operation of the system is outlined in the flowchart ofFig. 9. Upon starting up, the computer switches on the

phase perturbation and waits for the operator to tune thereceiver to the desired station. Then the operator keys thefrequency of the signal into the computer so that the com-puter loads the appropriate table of theoretical ripple cor-rections. In the present system there are correction tablesfor four distinct frequencies namely 3, 6, 15 and 30 MHz.The resolution of such data may clearly be increased ifnecessary.

Subsequently, the computer asks for the number of con-secutive DF readings required, before some simple sta-tistics (such as their mean and standard deviation) arecomputed. One DF reading consists of averaging the set-tings of the control phase shifter over a complete huntcycle of the adaptive loop. Normally, once the loop is inlock, consecutive readings are taken every other huntcycle, which is equivalent to taking readings every second.If a spike of noise or a frequency component of the speechsignal near the frequency of the imposed phase per-turbance appears, this may force the loop out of lock andthe DF reading might take longer.

Before the computer starts taking the specified numberof DF readings, it asks the operator whether it should do acoarse bearing measurement by comparing the phases ofthe — 1 and 0 phase-mode signals to resolve the 180°ambiguity. The computer then calculates the mean and thestandard deviation of the specified number of readings anddisplays the results.

7 Initial experimental results

Initial experimental results were taken by setting up afour-element circular array on the roof of the Engineeringbuilding at University College London. The site is a verybad one for HF DF with a very limited ground plane andwith a lot of reradiators (in the form of pipe, rails, nearbybuildings etc.) near to the array.

The method adopted for calculating the probablebearing of a transmitter was that suggested by W. Ross[6]. Approximately ten instantaneous readings are takenfor a period of about ten seconds. The mean of these read-ings constitutes a 'snap reading'. Then ten such snap read-ings are taken over a period of not more than ten minutesand not less than five minutes. The mean of these snapreadings constitutes an 'observation'. The calculation of asnap reading averages out the random errors due to noiseand due to the sudden appearance of frequency com-ponents in the modulated signal which interfere with thedetected error signal of the adaptive loop. In a traditionalaural null type direction finder, this is equivalent to tryingto find the centre of a flat null by carrying out about ten'swings' around the apparent centre of the null. The calcu-lation of an 'observation' (over a period of time) averages

Table 1: Tables of DF results obtained from direction finding exercises carried out upon realHF broadcast transmissions

Number

123456789

Frequency

kHz

15084153951506015635152201524521265

95356165

Mean ofoneobservation

n113188122147116194

29108

96

True locationand bearing oftransmitter

(°)

TehranSpainJedahAthensAnkaraDW SinesVOA PhilippinesSwissGermany

94190106119103208

5712596

DF error

(°)

19- 2162813

-14-28-17

0

Approximatedistance oftransmitter

km

600018006000300030001800

1500012001200

V,

n3.02.05.43.65.03.51.84.33.0

v?

(°)3.73.73.75.24.35.01.42.44.6

n1.21.21.21.61.41.60.50.81.5

n1.41.01.41.21.21.01.60.90.9

152 IEE PROCEEDINGS, Vol. 133, Pt. H, No. 2, APRIL 1986

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out the errors due to fast varying effects of the ionosphere.Averaging out errors due to slow varying effects of the ion-osphere would require measurements taken over manyhours and probably days.

Although many HF transmissions were detected andtheir bearings measured, only a few of them could be iden-tified. Table 1 shows results for direction finding carriedout on some identified international broadcasting stations.It can be seen that most of them are in the 15 MHz band.Some statistical figures are also shown in the same tablefor each of the transmissions. Hence VL is the average stan-dard deviation over ten instantaneous readings (that con-stitute a snap reading), V2 is the standard deviation of thesnap readings (that constitute an observation) calculatedaround their mean; whereas K4 is the expected standarddeviation of the observations taken over a long time, dueto slowly varying effects of the ionosphere. VA is obtainedby using the curves of Fig. 10 and the approximate dis-tances of the transmitters.

210

180

S.150

£ 120o

"2 90

60

30

/ x

A 6

4xA

/ ideal line

5x

0 30 60 90 120 150 180 210 2A0true bearing,degree

Fig. 10 Plot of measured against true bearing of transmitters<g> 15 MHz bandx 6 MHz band

this receiving system with active loop antennas representsan effective combination and can also offer a transportablecapability. Apart from the obvious advantage of 360° DFcoverage, a circular array has the notable advantage ofminimising bearing errors due to changes in the elevationof the received signal.

Although the lack of a means to control the amplitudesof the signals from the phase modes did not allow verydeep nulls to be formed, the experimental system did showthat both DF and null steering can be incorporated in afinal version of the system without modification to thesingle HF receiver.

The incorporation of a microcomputer to drive thesystem to apply on-line corrections to the DF resultsmeans that a small number of antenna elements could beemployed in the array. The use of a microcomputer alsoadded a substantial amount of flexibility to the system, asdifferent features and applications can be added by simplymodifying the software. The limited null steering capabilityof the experimental system is an example of this flexibility,as it was added at a later stage of this project. The abilityto analyse and display DF studies on the microcomputerVDU is a further attractive feature.

The experimental system operates well with most typesof received signals, but modifications are required for oper-ation with SSB modulation. Future development of thesystem will include this feature, in addition to amplitudecontrol of the phase modes for a better null steering capa-bility.

Studies are also in progress on the advantages of usingmore than two phase modes. This feature can offer thecapability of increased antenna gain, plus the potential toremove a high-level interfering signal by null steering whiledirection finding on the residue weaker signal. The abilityto change the mode of operation by software control of asmall microcomputer will continue to be a major advan-tage as the system becomes more complex.

9 Acknowledgments

The authors wish to express their thanks to C & SAntennas Ltd., for providing the financial support and totheir technical director, Mr. B. Collins, for many helpfuldiscussions and advice.

Fig. 10 shows graph of indicated against true bearingfor the observed transmissions. The discrepancy of theexperimental points from the theoretically expectedstraight line of slope 45° is mainly due to site errors thatcould not be corrected.

10 References

8 Conclusions

The principal conclusion to be drawn from this work isthat the use of phase-mode analysis of signals from a circu-lar array represents a very effective form of HF directionfinding. The particular receiver configuration developedfor this system had the additional advantage that it madeuse of a single HF receiver in unmodified form. The use of

1 RUDGE, A.W., MILEN, K., OLVER, A.D., and KNIGHT, P.: 'Hand-book of antenna design. Vol. 2' (Peter Peregrinus, 1983) Chap. 12

2 DAVIES, D.E.N.: 'Transformation between the phasing techniquesrequired for linear and circular arrays', Proc. IEE, 1965, 112, (11), pp.2041-2045

3 GUY, J.R.F., and DAVIES, D.E.N.: 'Studies of the Adcock directionfinder in terms of phase mode excitations around circular arrays',Radio & Electron. Eng. 1983, 53, (1), pp. 33-38

4 RIZK, M, and DAVIES, D.E.N.: 'A broadband experimental nullsteering antenna for mobile communications', ibid., 1978, 48, pp.509-517

5 DAVIES, D.E.N., and RIZK, M.: 'Electronic steering of multiple nullsfor circular arrays', Electron, Lett, 1977, 13, (22), pp. 669-670

6 ROSS, W.: 'The estimation of the probable accuracy of high frequencydirection finding bearings', J. IEE, '1947, 94, Pt. Ill A, pp. 722-726

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