e shaped antenna design

RESOLUTION Reconfigurable Systems for Mobile Local Communication and Positioning

RESOLUTION Project

IST - 026851

Design of antennas and antenna arrays [ D12 ]

Document Information

Title D12 – Design of antennas and antenna arrays

Workpackage WP3 – Antennas and Propagation

Responsible Warsaw University of Technology (WUT)

Due Date Project Month 23 (December 2007)

Type Report

Status Version 1.0

Security Public

Authors Paweł Bajurko, WUT

Yevhen Yashchyshyn, WUT

Rafał Szumny, WUT

Krzysztof Kurek, WUT

Project URL http://www.ife.ee.ethz.ch/RESOLUTION/

D12 – Design of antennas and antenna arrays 2

RESOLUTION 2007-12-31

Summary This report presents the results of design of antennas and antenna arrays for High-

Precision-Localization-System. Several structures, meeting particular requirements of the

system, have been designed, realized and measured. Depending on the system

configuration different types of antennas have been proposed for the base stations and

mobile terminals in the system. Antennas realized as planar structures have been preferred

due to their advantages: easy fabrication and integration with Tx/Rx modules, low profile

and lightweight.

Also analysis of dependence of distance determination error in localization system based on

time of arrival measurements, on beam width of antenna radiation pattern has been

presented. Using narrow beam antennas some multipath components with small delays,

causing errors of calculation of distance between antennas, can be eliminated and accuracy

of the system can be increased. This improvement depends on detection method

implemented in the system. When leading edge detection, resistant to multipath

propagation, is used influence of antenna radiation characteristics is smaller.

Table of contents

SUMMARY ...............................................................................................................2

TABLE OF CONTENTS ..............................................................................................2

1 INTRODUCTION.................................................................................................3

2 PROJECT AND REALIZATION OF SINGLE ANTENNAS .........................................7

2.1 OPTIMISATION PROCEDURE APPLIED TO DESIGNING PROCESS .......................................7

2.2 PROJECT OF E-SHAPED PATCH ANTENNA ...................................................................8

2.3 PROJECT OF ELLIPTICALLY POLARIZED CIRCULAR PATCH ANTENNA................................10

2.4 PROJECT OF MONOPOLE ANTENNA .........................................................................16

2.5 PROJECT OF PLANAR ANTENNA WITH CIRCULAR SLOT.................................................19

2.6 PROJECT OF BOW-TIE ANTENNA............................................................................23

2.7 PROJECT OF INVERTED-F ANTENNA .......................................................................26

3 PROJECT AND REALIZATION OF ANTENNA ARRAYS.........................................29

3.1 4-ELEMENT SWITCHED ARRAY ..............................................................................29

3.2 SEMICIRCULAR ARRAY OF 4 E-SHAPED PATCH ANTENNAS ...........................................30

3.3 LINEAR ARRAY OF 4 E-SHAPED PATCH ANTENNAS.....................................................32

3.4 SEMICIRCULAR ARRAY OF 8 CIRCULAR PLANAR ANTENNAS WITH ELLIPTICAL POLARIZATION

33

3.5 2×2 ARRAY OF MONOPOLE ANTENNA .....................................................................36

3.6 2×2 ARRAY OF PLANAR ANTENNAS WITH CIRCULAR SLOT...........................................38

3.7 TWO INVERTED-F ANTENNAS ...............................................................................42

4 ANTENNAS FEEDING INTERFACES ...................................................................45

5 ANALYSIS OF ANTENNA BEAM WIDTH INFLUENCE ON DISTANCE ERROR

MEASUREMENT .....................................................................................................46

6 CONCLUSION...................................................................................................53

7 REFERENCES....................................................................................................54



1 Introduction Antenna is an important element of each radio transmission system. The use of proper

antennas, with appropriate radiation pattern, can improve the system performance, allowing

to eliminate some unwanted interfering signals and increase quality of the signal reception.

Different characteristics of antenna can be necessary for the transmitter and the receiver

side, depending on of wireless system configuration. Topology of the considered indoor

localization system is presented in Fig. 1.1. System consists of stationary base stations (BS)

and mobile stations (STA). The BSs serve as absolute spatial references for the mobile

STAs. System architecture of proposed High Precision Localization System (HPLS) is

described in details in [1]. Two modes of the system operation are considered in the

RESOLUTION:

• bi-directional down- and up-link transmission between BS and STA, with active reflector

in STA. Each BS sequentially realizes communication with chosen STA, and then position

of the STA is calculated in the central station of the system. Such procedure is repeated

for each STA in area of system coverage.

• unidirectional down-link transmission from BS to STA (concept similar to GPS). All STAs

receive signals transmitted sequentially by the BSs and then each of STA calculates its

position.

Fig. 1.1 The High-Precision-Localization-System (HPLS) topology [1].

Two types of the mobile terminals are foreseen to be used in the system, depending on

application:

• larger terminals mounted on vehicles for Automated Guided Vehicle (AGV) applications

• small terminals integrated with Personal Digital Assistant (PDA) or mobile handsets for

interactive guiding applications

Additionally localization terminal can be integrated with WLAN terminal to assure localization

and communication applications in one system.

Depending on system configuration and side of antenna placement different types of

antenna are preferred. The BS antenna shall to assure coverage of considered area

(typically single room in the building to assure line-of-sight LOS conditions for transmission

to STA), so for BSs placed on walls sector antennas can be used. For the STA the antenna

with omnidirectional radiation pattern in azimuth plane should be used to allow realize



transmission with each BS independently on orientation of STA. Additional improvement of

the system parameters can be obtained by using antenna arrays with adaptive steered

narrow main beam. Such array used in the BS or in STA can increase received power and

can eliminate some multipath components arrived to the receiver from directions different

from direction of direct LOS one. As it was shown in D3 [2] multipath components with very

small delays and strong amplitude in relation to LOS one can cause serious degradation of

obtained accuracy of indoor localization system based on measurements of time of arrival

(TOA). It is possible to eliminate some of such components by use of antenna arrays with

relatively narrow beam. Such arrays can be used in BSs if bidirectional transmission

between BS and STA is realized, or in STA, if AGV applications are considered. But in last

case due to dimension and complexity limitations simple arrays with possibility of beam

steering in all directions are preferred.

In first version of the system concept it was assumed that the same antenna would be used

for localization system working in 5.725-5.875 GHz band and for communications

applications using WLAN in frequency band 5.15-5.35 GHz or 5.47-5.725 GHz, so wideband

antenna was necessary. The Fig. 1.2 presents frequency range exploited in the system and

required for designed antennas. Relative bandwidth for the range 5.150-5.875 GHz equals

13.15%.

Fig. 1.2 Frequency range exploited in RESOLUTION project for localization and

communications systems.

During the realization of the project the concept of the system has been changed. To make

the proposed solution more universal and allowing to use other communication systems i.e.

WLANs working in 2.4 GHZ band, the use of two separate antennas for communications and

localization has been decided, during the project meeting in Athens in December 2006 [3].

In this case for localization system narrowband antennas, working in frequency range

5.725-5.875 GHz (Fig. 1.3), will be used. Only antennas for HPLS have been considered in

this case.

Fig. 1.3 Frequency range exploited in RESOLUTION project for localization system only.

Antennas designed for the HPLS during the RESOLUTION project are presented in Fig. 1.4

and antenna arrays are presented in Fig. 1.5. All of the antennas and arrays are described

in further part of the document.

Planar antennas - microstrip patch antenna, planar dipole antenna, circular slot antenna -

are expected to satisfy requirements due to its conformability to mounting in system and

other eligible features – easy fabrication, low profile and light weight.

5GHz 6GHz

5.725GHz

frequency

5.875GHz

5GHz 6GHz

5.150GHz 5.350GHz 5.470GHz

frequency

5.875GHz



a) b) c)

d) e) f)

Fig. 1.4 Single element antennas designed for Work Package 3 of RESOLUTION project.

a) E-shaped wideband patch antenna, b) Elliptically polarized circular patch antenna, c)

Monopole antenna, d) Planar antenna with circular slot, e) Bow-tie antenna, f) Inverted-F

antenna.

a) b) c)

d) e) f)

g)

Fig. 1.5 Antenna arrays designed for Work Package 3 of RESOLUTION project. a) 4-element

switched array of E-shaped patch antennas, b) Semicircular array of 4 E-shaped patch

antennas, c) Linear array of 4 E-shaped patch antennas, d) Semicircular array of 8 circular

planar antennas with elliptical polarization, e) 2×2 array of monopole antenna, f) 2×2 array

of planar antennas with circular slot, g) Two inverted-F antennas.

For the base station (BS) antenna in the case of system with bidirectional transmission

between BS and STA, semicircular array of 4 or 8 antennas are considered to be the

optimum solution. That antenna allows adaptive beam forming and following the STA with

the beam. As a single element in the array E-shaped patch antenna for wideband

applications or circular patch antenna for narrowband applications have been chosen.



In the case of system concept using only down-link transmission from BS to STA, wide-

beam or sector antenna seems to be optimum solution for BSs, because STA tracking is

unnecessary. Also elliptical polarization has been taken into account for transmission

improvement – odd-reflected rays are suppressed in that case. Thus elliptically polarized

antenna has been designed and semicircular array of such antennas has been proposed.

For mobile stations (STA) mounted on vehicles or large objects different types of antennas

have been considered as optimum solutions:

• single monopole antenna for wideband solution

• single planar antenna with circular slot for narrow band solution

• 4-element switched array of E-shaped wideband patch antennas

• 2×2 array of monopole or planar antenna with circular slot

In case of transmission with elliptically polarized waves, arrays of elliptically polarized patch

antennas will be used.

However if the mobile station is integrated in mobile handset (PDA), small, omnidirectional

antenna is needed. For that purpose small and very flat bow-tie antenna and inverted-F

antenna have been designed. The frequency band of these antennas is adequate for the

narrow bandwidth of the localization system.

All of proposed antennas will work covered. Covering materials and mounting boxes may

have influence on antenna parameters. Thus final project of antenna needs to be verified

with external elements taken into account.



2 Project and realization of single antennas

2.1 Optimisation procedure applied to designing process

Simulations of designed antennas were performed using two electromagnetic simulators –

QuickWave 3D [4], based on FDTD (finite-difference time-domain) method, and FEKO [5]

based on MoM (method of moments).

In order to obtain antenna structure providing parameters accordant to the system

requirements, an automatic procedure finding optimal geometrical dimensions is necessary.

The optimisation procedure built in Matlab computation environment has been applied. The

optimisation procedure changes structure parameters, runs repeatedly simulation in the EM

simulator and analyses simulation results in order to find results best-matched to the

assumed requirements.

Optimisation process is presented in details in Fig. 2.1. Optimisation script generates vector

containing values of dimensions of structure, then runs structure editor, that rebuilds

structure model using the vector of values, then simulator is running and after termination

of simulation results are stored into file. Optimisation script reads the file and computes

desirable parameters as values of goal function, i.e. reflection coefficient in specified

frequency band as well as radiation pattern in specified plane, or other parameter, or any of

them combined. The computed parameters are taken back to the optimisation function as a

current iteration result, afterwards the function makes decision about further search using a

sequential quadratic programming (SQP) method.

Fig. 2.1 The optimisation procedure applied for E-shaped patch antenna design.

Optimisation takes significant amount of time. Each iteration typically takes half an hour up

to a few hours on a single-processor workstation. Single optimisation procedure needs

about 20 up to several hundred iterations to find a solution, depending on structure

complexity, number of optimised variables and starting point of optimisation. Typically

optimisation procedure takes one or two days, but if it fails to find acceptable solution the

procedure needs to be executed again.

By using two simulation environments based on different methods for designing purposes,

self-verification route is able to be introduced, just before physical realization and

measurement verification will be executed.

Other verification method, feasible over designing stage, is to re-simulate the structure

using the same computation environment but applying higher mesh density in order to get

more accurately structure mapping and field represent.

|S11| < –10.94dB



2.2 Project of E-shaped patch antenna

E-shaped patch antenna with operating frequency 5.150 – 5.875 GHz (with bandgap 5.350

– 5.470 GHz) was designed and performed. After literature survey, an E-shaped patch

antenna has been chosen to design due to its simple construction, conformability to

mounting in system and expected characteristics satisfying the project requirements. Patch

of the investigated antenna is located on the dielectric substrate, and the substrate is

located on metallic ground surface. Fig. 2.2 and Fig. 2.3 show structure of the antenna. Tab.

2.1 contains information about physical dimensions and other parameters of designed

structure. The antenna has been realized and measured recently. Fig. 2.4 shows performed

structure.

Fig. 2.2 Structure of E-shaped patch antenna, top view and side view.

a) b)

Fig. 2.3 Coordinates (a) and symbols designation (b) for E-shaped patch antenna.

Parameter Value

laminate RO3003

substrate thickness 1.524 mm

substrate permittivity 3.00

substrate conductance 0.0014 S/m

metal thickness 35 µm

metal conductance 5.847e7 S/m

antenna feed RG-405, semi-rigid

X

Y

φ

θ

Z

dp

lb la

width

pa wa w2 wb pb

gmx gmx

gmy

gmy



gmx 7.00 mm

gmy 10.00 mm

width 14.05 mm

pa = pb 13.40 mm

wa = wb 1.55 mm

w2 4.25 mm

la = lb 8.55 mm

dp 2.25 mm

Tab. 2.1 Parameters of the structure shown in the Fig. 2.2.

Fig. 2.4 Performed structure of E-shaped patch antenna.

Following figures show measurements results. Fig. 2.5 presents reflection coefficient of the

designed antenna. There is good agreement between results of measurement and

simulation. Frequency bands coincide in both cases as well as resonance frequencies.

However there is –8.17dB reflection provided in the considered frequency band in

comparison with –10dB obtained in the simulation.

Fig. 2.5 Reflection coefficient of the antenna shown in the Fig. 2.2.

Fig. 2.6 and Fig. 2.7 show radiation pattern of the antenna. Good agreement between

simulation and measurements can be also observed.



a)

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0

Measured

Simulated

b)

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0

Measured

Simulated

Fig. 2.6 XZ-plane cross-section of radiation pattern of antenna shown in the Fig. 2.2 at 5.5

GHz. a) Co-polarization, b) cross-polarization.

a)

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0

Measured

Simulated

b)

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0

Measured

Simulated

Fig. 2.7 YZ-plane cross-section of radiation pattern of antenna shown in the Fig. 2.2 at 5.5

GHz. a) Co-polarization, b) cross-polarization.

The antenna is optimised to work in system based on first concept with wide frequency

band. The antenna array based on E-shaped patch is described in further part of document.

2.3 Project of elliptically polarized circular patch antenna

The design of elliptically polarized antenna was realized. The assumptions of designed

antenna are listed below:

• frequency band: 5.725 – 5.875 GHz (bandwidth equals 150MHz)

• polarization: circularly polarized antenna

• small electrical dimensions in order to allow antenna array design.

Propagation features of the electromagnetic wave of elliptical polarization give some profits

for the system, as it was shown in [6]. Using circularly polarized antennas for the

|E|/|Emax| |E|/|Emax|




transmitter and the receiver some multipath components received after odd number of

reflections can be attenuated.

The antenna presented in the Fig. 2.9 and Fig. 2.8 contains circular patch with oblique slot

as a diversity element. The antenna is fed directly from semi-rigid coax cable. Optimisation

procedure has been using four variables in order to get appropriate radiation characteristics:

patch diameter, slot length, rotation of slot and feed location. The goal function was defined

as a combination of reflection coefficient value in the frequency band and circularity of

polarization for the straight direction (direction of axis Z). Tab. 2.2 contains parameters and

dimensions of designed antenna.

Fig. 2.8 Circular patch with oblique slot, 3-D view.

Fig. 2.9 Circular patch with oblique slot, top view and side view.

Parameter Value

laminate RO3003





metal conductance 5.847e +007 S/m


substrate dimensions 32 mm × 32 mm

patch diameter 15.45 mm

slot dimensions 6.85 mm × 0.7 mm

rotation of slot 50.2° (from X-axis)

feed point location

(distance from the

3.075 mm


Y

Z X

Z

X Y



Fig. 2.10 presents ellipses of polarization at three frequencies: at limits and at centre of

frequency band. Shapes of ellipses are almost circular, however their shape changes with

frequency. The quotient of ellipse axes equals 0.76, 0.71 and 0.44 for three considered

frequencies respectively. Obtained values permit to achieve benefits of polarization effects

(i.e. suppression of certain multipath components).

-2

-1,5

-1

-0,5

0

0,5

1

1,5

2

-2 -1,5 -1 -0,5 0 0,5 1 1,5 2

Etheta

Ep

hi

f = 5.725 GHz

f = 5.8 GHz

f = 5.875 GHz

Axes of ellipse

Fig. 2.10 Ellipses of polarization for Z-axis radiation for the antenna shown in the Fig. 2.9.

The antenna has been manufactured, Fig. 2.11 presents matrix of antennas on the sheet of

laminate. The antenna is fed directly from semi-rigid coax cable. Fig. 2.12 shows performed

structure.

Fig. 2.11 Matrix of designed structures prepared for manufacturing.



Fig. 2.12 Performed structure of circular patch with oblique slot.

Fig. 2.13 shows reflection coefficient of the designed antenna. There are some differences

between simulation and measurement (i.e. resonance frequencies slightly drifted away), but

demanded reflection coefficient inside considered frequency band is still kept. The frequency

band at –10dB reflection coefficient level is much wider then it was assumed, however there

are also other parameters (i.e. polarization parameters), which changes with frequency and

needs to be maintained in the assumed frequency band. It will be discussed hereunder.

5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7-25

-20

-15

-10

-5

0

frequency [GHz]

|S11| [dB]

Measured

Simulated

Demanded


Fig. 2.14 and Fig. 2.15 present XZ and YZ cross-section of radiation pattern of the antenna

respectively. It is necessary to know phase pattern in order to determine polarization of the

antenna. Thus phase patterns have been measured as well as amplitude pattern.

Measured amplitudes of field components are not equal, as expected according to the

simulation results. It means that oblique slot used as a diversity element does not work

exactly like simulation predicted. Probably antenna could be performed imprecisely.



a)

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0

Measured

Simulated

b)

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0

Measured

Simulated

c)

-180 -150 -120 -90 -60 -30 0 30 60 90 120 150 180-450

-400

-350

-300

-250

-200

-150

-100

Theta [deg]

Phase [deg]

Measured

Simulated

d)

-180 -150 -120 -90 -60 -30 0 30 60 90 120 150 180-600

-500

-400

-300

-200

-100

0

100

200

300

400

Theta [deg]

Phase [deg]

Measured

Simulated

Fig. 2.14 XZ-plane cross-section of radiation pattern of antenna shown in the Fig. 2.9 at

5.8 GHz. Amplitude of E-component orthogonal (a) / parallel (b) to the cross-section plane.

Phase of E-component orthogonal (c) / parallel (d) to the cross-section plane.

a)

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0

Measured

Simulated

b)

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0

Measured

Simulated





c)

-180 -150 -120 -90 -60 -30 0 30 60 90 120 150 180-900

-800

-700

-600

-500

-400

-300

-200

-100

0

Theta [deg]

Phase [deg]

Measured

Simulated

d)

-180 -150 -120 -90 -60 -30 0 30 60 90 120 150 180-500

-400

-300

-200

-100

0

100

Theta [deg]

Phase [deg]

Measured

Simulated

Fig. 2.15 YZ-plane cross-section of radiation pattern of antenna shown in the Fig. 2.9 at

5.8 GHz. Amplitude of E-component parallel (a) / orthogonal (b) to the cross-section plane.

Phase of E-component parallel (c) / orthogonal (d) to the cross-section plane.

Measured amplitude and phase patterns were used in order to determine quality of

polarization. It was assumed that quality of polarization is represented by circular ratio

(CR), defined as ratio of minor axis to major axis of ellipse of polarization (please refer to

the Fig. 2.10). CR falls into the range from 0 to 1. In case of linear polarization CR equals 0,

in case of circular polarization CR equals 1. All values between 0 and 1 relate to the elliptical

polarization.

Fig. 2.16 shows circular ratio versus frequency for the wave propagating in the main

direction (along Z axis). Measured CR reaches maximum value for the centre frequency of

the antenna (5.8 GHz), and exceeds 0,56 in the considered frequency band. It is very good

result, especially taking into account some differences between simulation and

measurements of reflection coefficient and radiation pattern.

5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 70

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1

frequency [GHz]

CR

Measurement

Simulation

Frequency band

Fig. 2.16 Circular ratio vs. frequency for the main direction.

Designed antenna radiates in wide range of directions and thus it is not enough to

determine CR value for main direction. Fig. 2.17 presents measured CR values versus



direction for two main planes (XZ and YZ). Presented results are absolutely satisfying – high

CR values are held on in very wide range of angle values in both planes.

-180 -150 -120 -90 -60 -30 0 30 60 90 120 150 1800

0.1

0.2

0.3

0.4

0.5

0.6

0.7

0.8

0.9

1

Theta [deg]

CR

XZ plane

YZ plane

Fig. 2.17 Circular ratio vs. theta at 5.8 GHz for XZ and YZ-plane cross-section.

Concluding, obtained characteristics of circular antenna with elliptical polarization meet

established assumptions for situation when antenna will be used only in localization system.

In spite of some differences between simulation and measurements, it is characterized by

low reflection coefficient, CR value exceeding 56% within frequency band and polarization

purity held on wide range of directions.

2.4 Project of monopole antenna

According to the decisions made during the RFIC meeting (Munich, 5-6th of July, 2007),

project of the single monopole antenna with centre operating frequency

5.8 GHz and bandwidth 150 MHz has been realized. Then 2×2 array on the basis

of the designed structure has been modelled. The reflection coefficient and radiation

patterns of both single and quadruple antenna have been simulated as well as mutual

transmission for the quadruple antenna has been calculated.

Structure of the antenna consists of ground plane and vertical monopole feed directly from

SMA junction. The structure is presented in the Fig. 2.18 and Fig. 2.19.

Ground dimensions are chosen to be somewhat smaller than the length of electromagnetic

wave at resonant frequency 5.8 GHz. The wavelength is 51.7 mm, and the length of ground

square side is 40 mm. The chosen dimension allows combining multiple structures in order

to get antenna array, and is sufficient to perform a role of ground plane.

The length of monopole is chosen to ensure resonant frequency equal to 5.8 GHz. Tab. 2.3

contains information about physical dimensions of designed structure.



Fig. 2.18 Structure of single monopole antenna, top view and side view (in scale 1:1).

Fig. 2.19 Structure of single monopole antenna – 3-D view.

Parameter Value

monopole length 11.7 mm (0.226 λ)

monopole diameter 1.28 mm (0.025 λ)

ground plane 40 mm × 40 mm × 0.5 mm


The antenna has been modelled and calculated in electromagnetic simulator QuickWave 3D.

Fig. 2.20 presents reflection coefficient of the antenna. Distinctive feature of the simulated

structure is frequency band much higher than required.

Y

Z X

Z

X Y



4 4.2 4.4 4.6 4.8 5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7 7.2 7.4 7.6 7.8 8-25

-20

-15

-10

-5

0

frequency [GHz]

[dB]

|S11|

|S11| demanded


Fig. 2.21 and Fig. 2.22 show radiation pattern of the antenna. Obtained results are typical

for the monopole on the ground antenna. The radiation pattern is omnidirectional. Side- and

back-radiation result from finite ground plane dimensions and is desirable in the considered

application. XZ cross-section of radiation pattern is almost the same as at the plane of

phi=45°.

0.5

1

1.5

2

30

210

60

240

90

270

120

300

150

330

180 0

ϕ

|E|/|EI|

|Eθ|, θ = 90°

|Eθ|, θ = 75°

|Eθ|, θ = 60°

|Eθ|, θ = 45°

|Eϕ|, θ = 90°

|Eϕ|, θ = 75°

|Eϕ|, θ = 60°

|Eϕ|, θ = 45°

Fig. 2.21 Radiation pattern of antenna shown in the Fig. 2.18 at the surface of constant

value of theta at 5.8 GHz. The spherical system coordinates are presented next to the

figure.



a)

0.5 1 1.5

2

30

210

60

240

90 270

120

300

150

330

180

0

θ

|E|/|EI|

|Eθ| @ f = 5.725 GHz

|Eθ| @ f = 5.8 GHz

|Eθ| @ f = 5.875 GHz

|Eϕ

| @ f = 5.725 GHz

|Eϕ

| @ f = 5.8 GHz

|Eϕ

| @ f = 5.875 GHz b)

0.5 1 1.5

2

30

210

60

240

90 270

120

300

150

330

180

0

θ

|E|/|EI|

|Eθ| @ f = 5.725 GHz

|Eθ| @ f = 5.8 GHz

|Eθ| @ f = 5.875 GHz

|Eϕ

| @ f = 5.725 GHz

|Eϕ

| @ f = 5.8 GHz

|Eϕ

| @ f = 5.875 GHz

c)

0.5 1 1.5

2

30

210

60

240

90 270

120

300

150

330

180

0

θ

|E|/|EI|

|Eθ| @ f = 5.725 GHz

|Eθ| @ f = 5.8 GHz

|Eθ| @ f = 5.875 GHz

|Eϕ

| @ f = 5.725 GHz

|Eϕ

| @ f = 5.8 GHz

|Eϕ

| @ f = 5.875 GHz d)

0.5 1 1.5

2

30

210

60

240

90 270

120

300

150

330

180

0

θ

|E|/|EI|

|Eθ| @ f = 5.725 GHz

|Eθ| @ f = 5.8 GHz

|Eθ| @ f = 5.875 GHz

|Eϕ

| @ f = 5.725 GHz

|Eϕ

| @ f = 5.8 GHz

|Eϕ

| @ f = 5.875 GHz

Fig. 2.22 Cross-sections of radiation pattern of antenna shown in the Fig. 2.18 a) at the

plane of phi=0°, b) at the plane of phi=45°, c) at the plane of phi=90°, d) at the plane of

phi=135°.

2.5 Project of planar antenna with circular slot

Structure of monopole antenna is not convenient to perform and practical application

because of its high profile. That was a reason why alternative planar antenna structure is

proposed. The antenna has low profile with similar transversal dimensions. Radiation

pattern of the planar antenna and monopole antenna are almost identical. Because of that

these structures can be considered to be electromagnetically complementary. However

frequency band of the planar antenna is much narrow.

The complementary planar structure consists of circular patch fed in the centre and metallic

surface separated by a slot from the patch. There is ground plane at the bottom side of

substrate and the outer metallic surface lying on the top is short-circuited to the ground

plane by means of a number of metallic vias. Structure of considered antenna is presented

in the Fig. 2.23, and Tab. 2.4 contains parameters and dimensions of designed antenna.

The antenna structure has been optimised in order to ensure frequency band of the antenna

in the range 5.725 – 5.875 GHz (bandwidth equals 150MHz).



a)

b)

Fig. 2.23 Structure of single planar antenna with circular slot (in scale 1:1): a) top view; b)

side view; c) 3-D view

Parameter Value

laminate RO3003







circular patch diameter 31.9 mm

slot size 1.0 mm

length of side of vias

rectangle

40.0 mm × 40.0 mm

vias radius 0.25 mm



Fig. 2.24 shows reflection coefficient of the designed antenna. The frequency band is

narrower than the frequency band of monopole antenna, but still satisfying requirements.

Radiation pattern in the frequency band is almost the same.

Y

Z X Z

X Y

c)



4 4.2 4.4 4.6 4.8 5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7 7.2 7.4 7.6 7.8 8-25

-20

-15

-10

-5

0

frequency [GHz]

[dB]

|S11|

|S11| demanded


The antenna has been performed and measured, Fig. 2.25 presents one of performed

antennas.

Fig. 2.25 Performed structure of single planar antenna with circular slot.

Fig. 2.26 presents results of measured reflection coefficient for two performed antennas. For

both antennas the resonant frequency is shifted from 5.8 GHz to ca. 6.0 GHz.

In order to find the reason of this disagreement further simulation has been executed. The

structure has been simulated again in the QuickWave 3D environment but using higher

mesh density (200 cells per wavelength in the dielectric instead of 100 cells per wavelength

in the dielectric). Also there was performed simulation of the same structure in FEKO

electromagnetic simulator. All of the simulation results are presented in Fig. 2.27.

Simulation results presented in Fig. 2.27Fig. 1.1 allow drawing conclusion that mesh density

used in simulations has been not enough for that purpose, nevertheless density 100 cells

per wavelength is commonly sufficient in the most of simulated electromagnetic problems.



4 4.2 4.4 4.6 4.8 5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7 7.2 7.4 7.6 7.8 8-30

-25

-20

-15

-10

-5

0

frequency [GHz]

[dB]

|S11| measured (piece 1)


|S11| simulated

|S11| demanded

Fig. 2.26 Measured reflection coefficient of the antenna shown in the Fig. 2.23 and Fig.

2.25.

4 4.2 4.4 4.6 4.8 5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7 7.2 7.4 7.6 7.8 8-30

-25

-20

-15

-10

-5

0

frequency [GHz]

[dB]



|S11| simulated 100 cells / λd

|S11| simulated 200 cells / λd

|S11| simulated FEKO

|S11| demanded

Fig. 2.27 Measured and simulated reflection coefficient the antenna shown in the Fig. 2.23

and Fig. 2.25. Figure presents measured reflection coefficient of two pieces of performed

antennas as well as simulation results obtained on the QuickWave 3D software at 100 and

200 cells per wave length inside dielectric and simulation results obtained on the FEKO

electromagnetic simulator.

Fig. 2.28 shows measured and simulated XZ-plane cross-section of radiation pattern at 5.8

GHz. Results are roughly similar. Differences in the back radiation results from an antenna

under test mounting system.

Parameters of the antenna are optimal in order to applying the structure in mobile stations

in the system. In particular, upraised radiation pattern is very adequate for system with

highly located base stations. The radiation pattern causes reduction of rays reflected from

the floor.



0.2 0.4 0.6 0.8 1

30

210

60

240

90 270

120

300

150

330

180

0

θ

|E|/|Emax

|

|Eθ| measured

|Eϕ

| measured

|Eθ| simulated

|Eϕ

| simulated

Fig. 2.28 Measured and simulated XZ-plane cross-section of radiation pattern of the

antenna shown in the Fig. 2.23 and Fig. 2.25 at 5.8 GHz.

2.6 Project of bow-tie antenna

A single bow-tie antenna with centre operating frequency 5.8 GHz and bandwidth 150 MHz

has been designed. The antenna is assigned for mobile handset, thus it needs to be flat,

slight and able to easy-integrate with the system.

Antenna presented in Fig. 2.29 and Fig. 2.30 consists of two parts: grounded (with feeding

microstrip line of impedance equal 50Ω, quarter-wave impedance transformer, power

divider and balloon circuit) and ungrounded (with dipole).

Fig. 2.29 Structure of bow-tie antenna, top view.

Y

Z X



Fig. 2.30 Structure of bow-tie antenna, 3-D view.

Substrate of the antenna is high frequency laminate with reinforced woven glass in to

ensure greater mechanical strength for very thin layer (0.254 mm). Dimensions of the

structure have been chosen first accordingly to the centre operating frequency 5.8 GHz, and

then have been optimised in order to minimize reflection coefficient within the operating

frequency band. Fig. 2.31 shows symbols designations for structure dimensions, and Tab.

2.5 contains parameters of designed antenna.

Fig. 2.31 Symbols designations for bow-tie antenna.

Parameter Value

laminate RO3203





w1

w2

w3

sy

l1 l2

l3

l4

l5 d

w

rb rc

d

sx

dl

w1




sx × sy 28 mm × 32 mm

l1 20.5 mm

l2 8.0 mm

l3 1.5 mm

l4 1.4 mm

l5 5.7 mm

d 0.8 mm

dl 10.5 mm

dw 3.7 mm

rb 3.7 mm

rc 1.0 mm

w1 0.6 mm

w2 0.7 mm

w3 1.0 mm


Fig. 2.32 shows simulated reflection coefficient of the designed antenna. Fig. 2.33 and Fig.

2.34 present simulated radiation pattern of the antenna. The antenna will be performed in

the nearest future in order to confirm simulation results.

4 4.2 4.4 4.6 4.8 5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7 7.2 7.4 7.6 7.8 8-30

-25

-20

-15

-10

-5

0

frequency [GHz]

[dB]

|S11| simulated

|S11| demanded




0.5

1

1.5

2

30

210

60

240

90

270

120

300

150

330

180 0

ϕ

|E|/|EI|

|Eϕ

| @ f = 5.725 GHz

|Eϕ

| @ f = 5.8 GHz

|Eϕ

| @ f = 5.875 GHz

|Eθ| @ f = 5.725 GHz

|Eθ| @ f = 5.8 GHz

|Eθ| @ f = 5.875 GHz

Fig. 2.33 XY-plane cross-section of radiation pattern of antenna shown in the Fig. 2.29. The

spherical system coordinates are presented next to the figure.

0.5 1 1.5

2

30

210

60

240

90 270

120

300

150

330

180

0

θ

|E|/|EI|

|Eθ| @ f = 5.725 GHz

|Eθ| @ f = 5.8 GHz

|Eθ| @ f = 5.875 GHz

|Eϕ

| @ f = 5.725 GHz

|Eϕ

| @ f = 5.8 GHz

|Eϕ

| @ f = 5.875 GHz

0.5 1 1.5

2

30

210

60

240

90 270

120

300

150

330

180

0

θ

|E|/|EI|

|Eθ| @ f = 5.725 GHz

|Eθ| @ f = 5.8 GHz

|Eθ| @ f = 5.875 GHz

|Eϕ

| @ f = 5.725 GHz

|Eϕ

| @ f = 5.8 GHz

|Eϕ

| @ f = 5.875 GHz

Fig. 2.34 Radiation pattern of antenna shown in the Fig. 2.29. a) YZ-plane cross-section of

radiation pattern, b) XZ-plane cross-section of radiation pattern.

2.7 Project of inverted-F antenna

A single Inverted-F Antenna (IFA) with centre operating frequency 5.8 GHz and bandwidth

1.62 GHz has been designed. It has been optimised for mobile handset usage. It is slim,

small and easy to integrate with different types of feeding. Influence of handset PCB has

been also considered in design process.

Antenna consists of three layers. Lower one is 1.5 mm high layer of FR4 substrate, which

simulates handset PCB. Upper one is 0.3 mm layer of Rogers RO3203 substrate that is used

as substrate for the antenna. They are separated by the ground plane layer. In simulations

structure has been feed by 50 Ω microstrip line made on FR4 dielectric layer and feed pin



Y

X

φ

Θ

Z

that goes through both dielectric substrates and ground plane. Fig. 2.35 shows antenna

geometry.

Fig. 2.35 IFA antenna geometry and coordinate system used in radiation pattern plots.

Structure has been optimised in order to have 5.8 GHz centre frequency and later to

achieve wide bandwidth. Area below antenna is not grounded. FR4 substrate and ground

plane are 10 mm shorter than RO3203 substrate. Antenna utilizes two pins. One of them is

used to feed the antenna (left one in Fig. 1) and another to ground the antenna to the

ground plane. In Fig. 2.36 structure dimensions are presented. In Tab. 2.6 detailed

parameters of used substrate can be found.

6.7

100 100

0.3

80

32.6

10

1.5

Fig. 2.36 IFA antenna: dimensions of the antenna and dimensions of structure where

antenna is placed

Parameter Value

laminate RO3203







Tab. 2.6 Parameters of used substrate.

Fig. 2.37 shows simulated antenna input reflection coefficient, and Fig. 2.38 shows

simulated antenna radiation patterns.

Fig. 2.37 Input reflection coefficient of designed IFA antenna. With grey background

demanded bandwidth is marked.

a)

|E|/|EI|

0°15°

30°

45°

60°

75°

90°

105°

120°

135°

150°

165°±180°

-165°

-150°

-135°

-120°

-105°

-90°

-75°

-60°

-45°

-30°

-15°

0.5

1

1.5

|EΘ| @ f = 5.725 GHz

|EΘ| @ f = 5.800 GHz

|EΘ| @ f = 5.875 GHz

|Eφ| @ f = 5.725 GHz

|Eφ| @ f = 5.800 GHz

|Eφ| @ f = 5.875 GHz

Θ

b)

|E|/|EI|

0°15°

30°

45°

60°

75°

90°

105°

120°

135°

150°

165°±180°

-165°

-150°

-135°

-120°

-105°

-90°

-75°

-60°

-45°

-30°

-15°

0.4

0.8

1.2

1.6

|EΘ| @ f = 5.725 GHz

|EΘ| @ f = 5.800 GHz

|EΘ| @ f = 5.875 GHz

|Eφ| @ f = 5.725 GHz

|Eφ| @ f = 5.800 GHz

|Eφ| @ f = 5.875 GHz

Θ

c)

|E|/|EI|

0°

15°

30°

45°

60°

75°90°

105°

120°

135°

150°

165°

±180°

-165°

-150°

-135°

-120°

-105°-90°

-75°

-60°

-45°

-30°

-15°

0.40.81.2

1.6

|EΘ| @ f = 5.725 GHz

|EΘ| @ f = 5.800 GHz

|EΘ| @ f = 5.875 GHz

|Eφ| @ f = 5.725 GHz

|Eφ| @ f = 5.800 GHz

|Eφ| @ f = 5.875 GHz

φ

Fig. 2.38 Radiation pattern of antenna shown in Fig. 2.35. a) YZ-plane cross-section. b)

XZ-plane cross-section. c) XY-plane cross-section.



3 Project and realization of antenna arrays

3.1 4-element switched array

Structure of proposed and realized switched array is presented in Fig. 3.1. As single

elements of the array wideband E-shaped patch antennas were used. Antennas are placed

on sides of metal construction in shape of pyramid. Switching between antennas allows

receiving signals from all directions in azimuth plane, and in the same time some multipath

components are eliminated due to radiation pattern of single antenna in the array.

Measured reflection coefficients and radiation patterns of antenna array are presented in

Fig. 3.2 and Fig. 3.3.

Fig. 3.1 Structure of 4-element switched array for mobile reflector in HPLS: a) electrical

structure of the array; b) mechanical structure of the array; c) dimensions of pyramid side.

Fig. 3.2 Measured reflection coefficients of each antenna in the array.



Fig. 3.3 Radiation patterns of 4-element switched pyramid array: a) in azimuth plane; b) in

elevation plane.

3.2 Semicircular array of 4 E-shaped patch antennas

Structure of proposed and realized (without beam forming network) adaptive array is

presented in Fig. 3.4. This is basic part of proposed 16-element circular array with switching

and adaptive beam forming using 4 neighbouring elements. Wideband E-shaped patch

antennas are placed on part of metal cylinder.

Measured radiation patterns of single antenna in the antenna array are presented in Fig. 3.5

and Fig. 3.6 presents example of resultant pattern of the array.

Fig. 3.4 Structure of 4-element adaptive array for BS in HPLS: a) electrical structure of the

array; b) mechanical structure of the array; c) dimensions of the array.



Fig. 3.5 Measured radiation pattern of single antennas in the array: a) antenna 1; b)

antenna 2; c) antenna 3; d) antenna 4.

Fig. 3.6 Radiation pattern of 4-element adaptive array for equal amplitude and phase

combining.



3.3 Linear array of 4 E-shaped patch antennas

The linear array configuration has been built out of four E-shaped antennas arranged with

equal distances (Fig. 3.7). Distances between antennas may vary. Fig. 3.8 presents

reflection coefficient measured independently of each antenna (outside the array).

Fig. 3.7 Linear antenna array built out of four E-shaped patches.

4 4.2 4.4 4.6 4.8 5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7-25-24

-22

-20

-18

-16

-14

-12

-10

-8

-6

-4

-2

0

frequency [GHz]

|S11| [dB]

4 4.2 4.4 4.6 4.8 5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7-25-24

-22

-20

-18

-16

-14

-12

-10

-8

-6

-4

-2

0

frequency [GHz]

|S11| [dB]

Antenna 1 Antenna 2

1 2 3 4



4 4.2 4.4 4.6 4.8 5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7-25-24

-22

-20

-18

-16

-14

-12

-10

-8

-6

-4

-2

0

frequency [GHz]

|S11| [dB]

4 4.2 4.4 4.6 4.8 5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7-25-24

-22

-20

-18

-16

-14

-12

-10

-8

-6

-4

-2

0

frequency [GHz]

|S11| [dB]

Antenna 3 Antenna 4

Fig. 3.8 Reflection coefficient of particular E-shaped antennas.

3.4 Semicircular array of 8 circular planar antennas with elliptical polarization

The semicircular array configuration has been built out of eight antennas with circular patch

and oblique slot. The antenna array is presented in the Fig. 3.9. Fig. 3.10 presents reflection

coefficient measured independently of each antenna (outside the array).

Fig. 3.9 Semicircular antenna array built out of eight circular patches with oblique slots.

Analogically to the linear array measurements describer in the previous chapter, reflection

coefficient as well as radiation pattern of four of eight antennas were measured. Central

antennas of the array were chosen for measurements. Measurements results and synthesize

radiation patterns are presented in Fig. 3.10 - Fig. 3.14.



5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7-25

-20

-15

-10

-5

0

frequency [GHz]

|S11| [dB]

5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7-25

-20

-15

-10

-5

0

frequency [GHz]

|S11| [dB]

Antenna 3 Antenna 4

5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7-25

-20

-15

-10

-5

0

frequency [GHz]

|S11| [dB]

5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7-25

-20

-15

-10

-5

0

frequency [GHz]

|S11| [dB]

Antenna 5 Antenna 6

Fig. 3.10 Reflection coefficient of particular circular antennas.

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0

E-component orthogonal to the plane E-component parallel to the plane

Fig. 3.11 Measured XZ plane cross-section of radiation pattern of the 3rd antenna of

semicircular array.



0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0


Fig. 3.12 Measured XZ plane cross-section of radiation pattern of the 4th antenna of

semicircular array.

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0



semicircular array.

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0

0.2

0.4

0.6

0.8

1

30

210

60

240

90

270

120

300

150

330

180 0



semicircular array.



3.5 2×2 array of monopole antenna

Fig. 3.15 and Fig. 3.16 show structure of quadruple monopole antenna. The antenna is built

of four previously designed single monopole antenna combined together. Tab. 3.1 contains

information about physical dimensions of designed structure.

Fig. 3.15 Structure of quadruple monopole antenna, top view and side view (in scale 1:1).

Fig. 3.16 Structure of quadruple monopole antenna – 3-D view.

Parameter Value

monopole length 11.7 mm (0.226 λ)

monopole diameter 1.28 mm (0.025 λ)

monopole distance 40.0 mm (0.77 λ)

ground plane 80 mm × 80 mm × 0.5 mm


Y

Z X

Z

X Y

1 2

3 4



Fig. 3.17 shows simulated reflection coefficient and mutual transmission between integral

radiators. Reflection coefficient is similar to the one for single monopole antenna and mutual

transmission is less than –16 dB in the considered frequency band.

4 4.2 4.4 4.6 4.8 5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7 7.2 7.4 7.6 7.8 8-25

-20

-15

-10

-5

0

frequency [GHz]

[dB]

|S11|

|S21| = |S41|

|S31|

|S11| demanded

Fig. 3.17 Reflection coefficient and mutual transmission of the antenna shown in the Fig.

3.15.

Fig. 3.18 and Fig. 3.19 show radiation pattern of the first monopole of the antenna. The

result of greater ground plane area and monopoles situated out of the ground centre is that

radiation pattern is not as regular as in case of single monopole antenna. However the

antenna still can be considered as an omnidirectional.

0.5

1

1.5

2

30

210

60

240

90

270

120

300

150

330

180 0

ϕ

|E|/|EI|

|Eθ|, θ = 90°

|Eθ|, θ = 75°

|Eθ|, θ = 60°

|Eθ|, θ = 45°

|Eϕ|, θ = 90°

|Eϕ|, θ = 75°

|Eϕ|, θ = 60°

|Eϕ|, θ = 45°

Fig. 3.18 Radiation pattern of 1st radiator of antenna shown in the Fig. 3.15 at the surface

of constant value of theta at 5.8 GHz. The spherical system coordinates are presented next

to the figure.



a)

0.5 1 1.5

2

30

210

60

240

90 270

120

300

150

330

180

0

θ

|E|/|EI|

|Eθ| @ f = 5.725 GHz

|Eθ| @ f = 5.8 GHz

|Eθ| @ f = 5.875 GHz

|Eϕ

| @ f = 5.725 GHz

|Eϕ

| @ f = 5.8 GHz

|Eϕ

| @ f = 5.875 GHz b)

0.5 1 1.5

2

30

210

60

240

90 270

120

300

150

330

180

0

θ

|E|/|EI|

|Eθ| @ f = 5.725 GHz

|Eθ| @ f = 5.8 GHz

|Eθ| @ f = 5.875 GHz

|Eϕ

| @ f = 5.725 GHz

|Eϕ

| @ f = 5.8 GHz

|Eϕ

| @ f = 5.875 GHz

c)

0.5 1 1.5

2

30

210

60

240

90 270

120

300

150

330

180

0

θ

|E|/|EI|

|Eθ| @ f = 5.725 GHz

|Eθ| @ f = 5.8 GHz

|Eθ| @ f = 5.875 GHz

|Eϕ

| @ f = 5.725 GHz

|Eϕ

| @ f = 5.8 GHz

|Eϕ

| @ f = 5.875 GHz d)

0.5 1 1.5

2

30

210

60

240

90 270

120

300

150

330

180

0

θ

|E|/|EI|

|Eθ| @ f = 5.725 GHz

|Eθ| @ f = 5.8 GHz

|Eθ| @ f = 5.875 GHz

|Eϕ

| @ f = 5.725 GHz

|Eϕ

| @ f = 5.8 GHz

|Eϕ

| @ f = 5.875 GHz

Fig. 3.19 Cross-sections of radiation pattern of 1st radiator of antenna shown in the Fig.

3.15 a) at the plane of phi=0°, b) at the plane of phi=45°, c) at the plane of phi=90°, d) at

the plane of phi=135°.

3.6 2×2 array of planar antennas with circular slot

The complementary planar antenna array structure is built of four single planar antennas

with circular slot combined into quadruple antenna structure. Fig. 3.20 shows the antenna

structure. Tab. 3.2 contains information about physical dimensions and other parameters of

the structure.



Fig. 3.20 Structure of quadruple monopole antenna, top view and side view (in scale 1:1).

Parameter Value

laminate RO3003







circular patch diameter 31.9 mm

slot size 1.0 mm

length of side of vias

rectangle

40.0 mm × 40.0 mm

vias radius 0.25 mm



The quadruple planar antenna with circular slot has been performed and measured. Fig.

3.21 presents the performed antenna.

Y

Z X

Z

X Y

1 2

3 4



Fig. 3.21 Performed structure of quadruple planar antenna with circular slot.

Fig. 3.22 presents measured reflection coefficient for all of four radiators. The same effect

as for single planar antenna can be observed – resonant frequency is shifted from 5.8 GHz

to ca. 6.0 GHz.

4 4.2 4.4 4.6 4.8 5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7 7.2 7.4 7.6 7.8 8-30

-25

-20

-15

-10

-5

0

frequency [GHz]

[dB]

|S11| measured

|S22| measured

|S33| measured

|S44| measured

|Sii| simulated

|Sii| demanded

Fig. 3.22 Measured reflection coefficient of the antenna shown in the Fig. 3.20 and Fig.

3.21.

Fig. 3.23 and Fig. 3.24 show mutual transmission between radiators. In all cases its value is

less than –16 dB in the considered frequency band.



4 4.2 4.4 4.6 4.8 5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7 7.2 7.4 7.6 7.8 8-30

-25

-20

-15

-10

-5

0

frequency [GHz]

[dB]

|S21| measured

|S32| measured

|S41| measured

|S43| measured

|Sii| simulated

Fig. 3.23 Mutual transmission for the neighbouring patches of the antenna shown in the Fig.

3.20 and Fig. 3.21.

4 4.2 4.4 4.6 4.8 5 5.2 5.4 5.6 5.8 6 6.2 6.4 6.6 6.8 7 7.2 7.4 7.6 7.8 8-30

-25

-20

-15

-10

-5

0

frequency [GHz]

[dB]

|S31| measured

|S42| measured

|Sii| simulated

Fig. 3.24 Mutual transmission for the opposite patches of the antenna shown in the Fig. 3.20

and Fig. 3.21.

Fig. 3.25 shows measured and simulated XZ-plane cross-section of radiation pattern at 5.8

GHz. Results are roughly similar. Differences in the back radiation results from an antenna

under test mounting system, multiple narrow back lobes occurring in the measurement

results are caused by interferences with mounting tripod situated behind the antenna.



a)

0.2 0.4 0.6 0.8 1

30

210

60

240

90 270

120

300

150

330

180

0

|E|/|Emax

|

|Eθ| measured

|Eϕ

| measured

|Eθ| simulated

|Eϕ

| simulated

b)

0.2 0.4 0.6 0.8 1

30

210

60

240

90 270

120

300

150

330

180

0

|E|/|Emax

|

|Eθ| measured

|Eϕ

| measured

|Eθ| simulated

|Eϕ

| simulated

c)

0.2 0.4 0.6 0.8 1

30

210

60

240

90 270

120

300

150

330

180

0

|E|/|Emax

|

|Eθ| measured

|Eϕ

| measured

|Eθ| simulated

|Eϕ

| simulated

Fig. 3.25 Measured and simulated cross-sections of radiation pattern of 1st radiator of

antenna shown in the Fig. 3.20 and Fig. 3.21 at 5.8 GHz. a) at the plane of phi=0°, b) at

the plane of phi=45°, d) at the plane of phi=135°.

Measurement results show that proposed structure can meet imposed requirements,

although design needs to be improved in order to avoid resonant frequency shift.

3.7 Two inverted-F antennas

A two Inverted-F Antennas (IFAs) system with centre operating frequency 5.8 GHz and

bandwidth 1.67 GHz has been designed. The project is based on single inverted F antenna

for mobile handset usage. It is slim, small and easy to integrate with different types of

feeding. Influence of handset PCB has been also considered in design process. Two antenna

elements (separated by 0.85λ distance) allow introducing diversity into system.

Antenna consists of three layers. Lower one is 1.5 mm high layer of FR4 substrate, which

simulates handset PCB. Upper one is 0.3 mm layer of Rogers RO3203 substrate that is used

as substrate for the antennas. They are separated by the ground plane layer. In simulations



Y

X

φ

Θ

Z

structure has been feed by two 50 Ω microstrip lines made on FR4 dielectric layer and feed

pins that go through both dielectric substrates and ground plane. Fig. 3.26 shows antenna

geometry.

Fig. 3.26 Two IFA antenna system geometry and coordinate system used in radiation

pattern plots.

Structure has been optimised in order to have 5.8 GHz centre frequency and later to

achieve wide bandwidth. Area below antenna is not grounded. FR4 substrate and ground

plane are 10 mm shorter than RO3203 substrate. Each antenna utilizes two pins. One of

them is used to feed the antenna (inner one in Fig. 3.27) and another to ground the

antenna to the ground plane. In Fig. Fig. 3.27 structure dimensions are presented. In Tab.

3.3 detailed parameters of used substrate can be found.

80

62.6

6.7

100100

0.3

10

17.4

6.7

1.5

Fig. 3.27 Two IFA antennas: dimensions of antenna and structure

Parameter Value

laminate RO3203





Tab. 3.3 Parameters of used substrate.



Fig. 3.28 shows simulated antenna system S-parameters. Using S-parameters results

envelope correlation has been computed using equation (3-1) [7]. Envelope correlation plot

is shown in Fig. 3.29.

( )( )

2* *

11 21 12 22

2 2 2 2

11 21 22 121 1

e

S S S S

S S S S

ρ+

=− − − −

(3-1)

Fig. 3.28 S-parameters of the designed two IFA antenna systems. With grey background


Fig. 3.29 Envelope correlation of the designed antenna system. With grey background




4 Antennas feeding interfaces E-shaped patch antenna, elliptically polarized circular patch antenna and planar antenna

with circular slot are fed by semirigid coaxial cable. Cable parameters are given in the Tab.

4.1. For the measurement purpose the cable was terminated by SMA junction, as shown in

Fig. 4.1.

Parameter Value

Type of cable RG-405

Impedance 50 ohm

Conductor diameter 0.020 in.

Insulation diameter 0.062 in.

Overall nominal

diameter 0.085 in.

Tab. 4.1 Parameters of feeding cable.

Fig. 4.1 Antenna feeding.

Monopole antenna is designed to be fed directly from SMA junction, however that kind of

antenna has not been performed during investigation.

Bow-tie antenna is fed by unsymmetrical microstrip transmission line with impedance equal

to 50 ohm. Line parameters are given in the Tab. 4.2.

Parameter Value

Type of laminate RO3203

Substrate thickness 0.254 mm

Substrate relative

permittivity

3.00

Line width 0.6 mm

Tab. 4.2 Parameters of microstrip transmission line.



5 Analysis of antenna beam width influence on distance error measurement

Multipath propagation can cause degradation of accuracy of distance determination in

localization system based on time of arrival (TOA) method. Due to existence in the received

signal multipath components with small delays and large amplitudes, in relation to line-of –

sight LOS component (closely spaced paths - CSP), distance between transmitting and

receiving antennas is calculated with error. This error depends on the system bandwidth:

the larger bandwidth, the smaller error.

Additional decreasing of the distance error can be obtained by:

• use of appropriate method of detection of time of TOA, resistant to CSP

• elimination of some multipath components with small delays using antennas with narrow

beam of radiation pattern – steered antenna arrays.

Using the narrow beam antenna multipath components received from directions different

from direction of LOS one can be attenuated (Fig. 5.1) and accuracy of distance calculation

can be increased. Degree of improvement depends on spatial distribution of received

multipath components and width of antenna radiation patterns. Report D3 [2] presents

analysis of the indoor mutlipath propagation channel for scenarios typical for planed HPLS.

Special attention has been paid to determine existence of small delay multipath components

for different configurations of propagation environment (factory and office halls). It has

been shown that significant part of such components arrived from directions different from

LOS direction, so such components can be eliminated by narrow beam antenna.

Fig. 5.1 Elimination of some multipath components using narrow beam antenna in the

receiver

This analysis bases on results of D13 and presents influence of beam width of antenna

radiation patterns on error of calculated distance between antennas. Special software (DEES

– Distance Error Estimator) has been implemented in Matlab to calculate distance between

antennas using the channel impulse responses (CIR) obtained from simulations or

measurements. Different detection methods of TOA of LOS component have been

considered during the analysis (Fig. 5.2):

• detection of maximum of the entire signal – TOA-MD, component of the channel impulse

response with largest amplitude is considered as LOS signal, when there are multipath

components with amplitude larger than amplitude of LOS one , errors occurs

• detection of first peak of the signal – TOA-FP, first pick above the assumed threshold in

the channel impulse response is detected as LOS



• leading edge detection – TOA-LE, first signal above assumed threshold is detected and

TOA of LOS signal is calculated considering time correction (adequate for direct clear

LOS signal delay with the same threshold)

Fig. 5.2 TOA detection techniques (a): TOA-MD (τMD), TOA-FP(τFP), TOA-LE (τLE). Description

of techniques is in text. On (b) method of ld value calculation for TOA-LE method with LEP

threshold level for signal of bandwidth B.

TOA-LE detection method is resistant to existence of strong multipath signals with small

delays, but it needs to calculate proper approximation of correction time parameter ld that

depends on signal shape (modulation, Tx and Rx filters bandwidths, which are responsible

for side lobes levels, and main lobe width). Such detection method is implemented in

designed HPLS.

Analysis of influence of antenna radiation patterns on accuracy of localization bases on

results of ray-tracing propagation simulator RPS. Following steps are performed during this

analysis (Fig. 5.3):

• simulations of the propagation channel in RPS propagation software for omnidirectional

antennas and chosen location of Tx antenna and Rx antennas uniformly distributed in all

the room

• influence of antenna characteristics is calculated in Matlab, using simple models of

antenna radiation patterns based on sin(x)/x function and the channel impulse

responses (CIR’s) obtained from RPS.

• influence of the system bandwidth is calculated using Root-Raised-Cosine (RRC) filtering

as band limiting filter - convolution of two RRC filters with CIR gives band limited

representation of the received signal (In simulation filter with 0.5 roll-off have been

used)

• distance measurement error (DME) is calculated using developed DEES software,

considering three different TOA detection techniques

• statistics of obtained DME (mean, median, standard deviation) have been calculated for

obtained data in Matlab

0.1

1

Measure

d C

IR

τ τFP τLE τMD

ld

TOA-FP levell (np.:10%)

TOA-LE level (LEP)

TOA-MD level (100%)

a)

τ1

τ

b)

ld

1

LEP

1/B

|h(τ)| |h(τ)|



Fig. 5.3 CIR generation and filtering process using DEES program

In the first step average channel impulse responses for considered propagation environment

and for different types of the receiver antennas has been calculated, basing on results form

RPS. In calculations for each point of the receiver it was assumed that the antenna was

always pointed towards the transmitter (model of scanning antenna array). In the

transmitter that was placed near a wall, sector antenna, allowing to coverage all the room,

was used. Results are presented in Fig. 5.4 and Fig. 5.5. It is seen that when antenna with

narrower beam is used, multipath components with small delays (having largest impact on

accuracy of distance determination) are attenuated. The use of antenna with beam width in

a range 60-90o can already significantly decrease amplitude of small delay components in

the channel impulse response.

Fig. 5.4 Average channel impulse responses for LOS cases: a) Tx in the corner, b) Tx in

quarter length of shorter wall.

a) b)

|h(

ττ ττ )|

|h(

ττ ττ)|

|h(

ττ ττ)|

|h(

ττ ττ )|

a) b)

CIR files 2xRRC

filtration

CIR after

filtration

DME

estimation

R-T simulation data

τ τ τ τ



Fig. 5.5 Average channel impulse responses for: a) LOS case only, TX in half length of

shorter wall, b) NLOS case only, Tx in half length of shorter wall

Analysis of dependence of distance error DME on different detection methods, different the

system bandwidths and different antenna beam widths and polarisation has been done. Two

configurations of the propagation environment have been considered:

• empty room

• room with cubic objects, placed uniformly inside the room

Visualization of DME as function of the system bandwidth and polarisation for empty room

are presented in Fig. 5.6 and Fig. 5.7. It was assumed that omnidirectional antennas were

used for the transmitter and the receiver. Better results are obtained when circular

polarisation is used. In this case single reflected multipath waves are additionally attenuated

in the receiving antenna (after changing direction of polarization rotation).

When there are objects inside the room for some locations of Rx antenna direct visibility

between Tx and Rx antennas are obstructed, what corresponds to additional attenuation of

LOS signal and increasing of DME (Fig. 5.8). But error is smaller when TOA-LE detection

method is implemented.

a) B=100 MHz

b) B= 10 MHz

Fig. 5.6 2D visualisation of distance measurement error DME for empty room situation (Tx

placed in the corner, both antennas with linear polarisation): a) system bandwidth - 100

MHz; b) system bandwidth – 10 MHz



a) B=100 MHz

b) B= 10 MHz

Fig. 5.7 2D visualisation of distance measurement error DME for empty room situation (Tx

placed in the corner, both antennas with circular polarisation): a) system bandwidth - 100

MHz; b) system bandwidth – 10 MHz

a) TOA-MD

b) TOA-LE



Fig. 5.8 2D visualisation of distance measurement error DME for room with objects inside

(Tx placed in the corner, both antennas with linear polarisation, 100 MHz system

bandwidth): a) TOA-MD detection; b) TOA-LE detection

Fig. 5.9 and Fig. 5.10 present dependence of distance error DME, calculated in DEES

software, on the receiver antenna beam width for 100 MHz and 10 MHz system bandwidths

respectively. Four values of the antenna beam width: omni (360o), 60o, 38o, 19 o Different

detection methods are considered. The largest errors are for maximum pick TOA-MD

detection and the smallest for leading edge TOA-LE that is used in RESOLUTION. The use of

antenna with narrower main beam allows to decrease DME, but changes are smaller when

TOA-LE detection is implemented and the system bandwidth is larger. Over 50 % accuracy

improvement in distance measurement can be achieved, for 100 MHz bandwidth, using 20o

beam width antenna instead the omnidirectional one, when leading edge TOA-LE is

implemented.

Fig. 5.9 Mean value and standard deviation of distance measurement error DME as function

of the mobile Rx antenna beam width for different detection methods and 100 MHz

bandwidth



Fig. 5.10 Mean value and standard deviation of distance measurement error DME as

function of the mobile Rx antenna beam width for different detection methods and 10 MHz

bandwidth

It follows from simulation, that much smaller DME error can be obtained for the mobile

terminal antenna with 60° beam width. In each analysed case (different Tx locations)

narrowing antenna beam width causing decreasing of DME error. Additionally using TOA-LE

and TOA-FP methods results are few times better than for traditional TOA-MD method. In

RESOLUTION project edge detection (TOA-LE) is used for the distance calculation. This

method is resistant for multipath components, but needs to have proper ld parameter (Fig.

4.3) calculation. Additionally signal shaping filters should have low level side lobes, which

can cause deterioration of precision, and cause even measurement distance values smaller

than real distance.

Results concerning discussed techniques are also partially presented in [8], [9], [10], [11]

by participants of this part of the project.



6 Conclusion Different types of antennas and antenna arrays for High-Precision-Localization-System have

been designed, realized and measured. Proposed antenna arrays meet the system

requirements and seem to be suitable for practical application.

Considering the use of one wideband antenna for localization and communication in 5.5 GHz

systems E-shaped patch antenna could be used as single antenna or element of antenna

array in the base station BS and in mobile terminal STA mounted on vehicle (AGV

applications). In handheld mobile terminal in this case inverted F antenna could be used.

When communications and localization parts of the system are separated (i.e.

communication is realized using WLAN working in 2.4 GHz band) narrowband antennas can

be used in localization system. In this case standard patch antennas are preferable. The use

of circular polarization can be considered in this case. The use of the transmitting and

receiving antennas with circular polarization can attenuate some multipath components with

odd number of reflections and small delays (i.e. single reflections from ceiling, floor, walls,

objects in a room), and can increase accuracy of localization system. Additionally circularly

polarized antenna in the BS can decrease polarization mismatch in handheld terminals with

linearly polarized antennas.

To achieve possibility of mobile terminal of receiving signals form all directions the use of

omnidirectional antenna has been proposed. Quarter wavelength monopole antenna could

be used in this case for wideband transmission, or in narrowband case planar

complementary solution – circular slot antenna (that has the same characteristics as

monopole antenna, but it is narrowband solution) could be used. The use of such antenna in

the mobile terminal has additional advantage – attenuation of waves reflected from floor,

due to tilt of radiation pattern in elevation plane.

Antenna arrays can be used in the system to improve its characteristics. Array with

electronically steered narrow beam can increase power of the received signal (increasing

coverage range of the system) and attenuate some strong multipath components with small

delays, causing errors of position determination in localization system. The use of antenna

arrays depends on HPLS configuration. If localization is realized using active reflector in the

mobile terminal STA, antenna arrays could be used in the BS (to received signal

retransmitted by STA) and in STA (to receive signal from chosen BS and retransmit it in the

same direction). In case of STA a simple switching array seems to be a good solution. If

localization is realized basing on GPS concept with unidirectional down-link transmission

from the BS to the mobile terminal STA, sector antennas could be used in the BS and

antenna arrays could be used in the STA to receive signal from each BS. To obtain

possibility of receiving signals from all directions four-element array with planar circular slot

antennas has been proposed.

Following antennas and antenna arrays have been designed, realized and investigated:

• for the BS

1. E-shaped wideband patch antenna

2. circular patch antenna with circular polarization

3. quarter-circular array of 4 elements

4. semicircular array of 8 elements

5. linear array of 4 elements (for testing purposes)

• for the mobile terminal

1. quarter wavelength monopole antenna (only designed)



2. circular slot antenna

3. 2x2 array of circular slot antennas (for AGV applications)

4. switched array with pyramid shape of 4 patch antennas (for AGV applications)

5. planar dipole for handheld terminal (only designed, will be realized soon)

6. inverted F antenna for handheld terminal (only designed, will be realized soon)

Analysis of dependence of error of distance measurement on characteristics of antennas has

been done, basing on results of propagation analysis presented in D3 Report [2]. The use of

leading edge TOA-LE detection technique allows to minimize errors caused by multipath

propagation. Additionally increasing of accuracy of localization system can be achieved by

using narrow beam antennas (steered antenna arrays) in the mobile terminal or the base

station, depending on system configuration. Performed investigations show that the use of

narrow beam antennas can improve accuracy of the system based on measurements of time

of arrival (TOA), independent on used method to detect direct LOS component delay. But

when TOA-LE detection is implemented in localization system influence of antenna radiation

patterns is smaller in comparison to classical maximum pick TOA-MD detection. Distance

measurement error DME decreases about 20-30% when antenna with beam width in a

range 60-100o is used in the mobile terminal in HPLS. It is good compromise between

dimensions and complication of used antenna array and increasing of positioning accuracy.

7 References [1] RESOLUTION, Half-Year Report, Q2

[2 RESOLUTION, D3 Deliverable - Characterization and modelling of radio channel

[3] Minutes of the Meeting in Athens

[4] http://www.qwed.com.pl/

[5] http://www.feko.info/

[6] RESOLUTION, Technical Interim Report, Q5

[7] Constantine A. Balanis, Antenna Theory. Analysis and Design, Wiley, 1997

[8] R. Szumny, K. Kurek, S. Kozlowski, J. Modelski, “Measurements and analysis of the

propagation channel for different indoor environments”, IEEE EUROCON 2007,

Warsaw, Poland, 2007

[9] S. Kozlowski, K. Kurek, R. Szumny, J. Modelski, “Computer Simulations of The

Propagation Channel for Various Indoor Environments”, IEEE EUROCON 2007,

Warsaw, Poland, 2007

[10] R. Szumny, K. Kurek, J. Modelski, „Attenuation of Multipath Components Using

Directional Antennas and Circular Polarization for Indoor Wireless Positioning

Systems”, European Microwave Conference EuMC 2007, Munich, Germany, 2007

[11] R. Szumny, K.Kurek, J. Modelski, Antenna Diversity Impact to Indoor Wireless TOA-

based Positioning Systems Accuracy, accepted to IEEE Radio and Wireless

Symposium, Orlando, USA, 2008

e shaped antenna design

Documents