Microelectronics Journal 42 (2011) 790–797

Contents lists available at ScienceDirect

Microelectronics Journal

0026-26

doi:10.1

n Corr

E-m

journal homepage: www.elsevier.com/locate/mejo

Design of a wideband multi-standard antenna switch for wirelesscommunication devices

Vlad Marian a,n, Jacques Verdier b, Bruno Allard a, Christian Vollaire a

a Amp�ere, UMR CNRS 5005, Universite de Lyon, Ecole Centrale de Lyon, 36 Av. Guy de Collongues, F-69134 Ecully, Franceb Institut de Nanotechnologies de Lyon (INL), UMR CNRS 5270, Universite de Lyon, INSA Lyon, F-69621 Villeurbanne, France

a r t i c l e i n f o

Article history:

Received 18 November 2010

Received in revised form

7 January 2011

Accepted 17 January 2011Available online 22 February 2011

Keywords:

MMIC

Antenna switch

Wideband

Mobile telecommunications

pHEMT transistors

GaAs

92/$ - see front matter & 2011 Elsevier Ltd. A

016/j.mejo.2011.01.005

esponding author. Tel.: + 33 472186117; mo

ail address: vlad.marian@ec-lyon.fr (V. Maria

a b s t r a c t

A wideband Low Power Single Pole 6-Throw (SP6T) antenna switch has been designed for GSM/DCS/

802.11b mobile standards using a newly improved architecture and fabricated using a pseudomorphic

depletion mode 0.18 mm HEMT GaAs process. The switch exhibits less than 1 dB insertion loss and isolation

performances from up to 53 dB at 0.8 GHz down to 42 dB at 2.5 GHz. The circuit DC power consumption is

less than 500 mW in full power transmission condition and makes it suitable for use in mobile terminals

like mobile phones or PDAs. The paper presents simulation results validated by experimental measure-

ments on an IC prototype.

& 2011 Elsevier Ltd. All rights reserved.

1. Introduction

Wireless devices are embedding more and more functions,which renders communication issues increasingly complex.Today mobile phones include many other features like 802.11band Bluetooth connectivity or even a GPS transceiver. Thisincreases the RF system complexity while in the mean time it iscombined with the need for lower fabrication cost and more andmore compact devices.

This trend pushes for the development of integrated RF front-endsthat share the same antenna for transmitting (Tx) or receiving (Rx)signals at different frequencies. The role of an antenna switch is toimplement an interface between the antenna and the Tx/Rx ports ofthe different standards that need to use it.

Traditionally PIN diodes have been used to switch signals inmobile communication systems, because they offer a good linearityand high power handling capabilities [1,2], but have the majordisadvantage of high DC power consumption. This is mainly due tothe fact that in order to obtain a good isolation or a low insertionloss, an important polarization current is required. In the case of aseries connected diode, the insertion loss (IL) is given by [3]

IL¼ 20log 1þRS

2Z0

� �ð1Þ

ll rights reserved.

bile: +33 632782155.

n).

where RS is the series resistance of the PIN diode and Z0 is thesystem characteristic impedance. The resistance of the intrinsicregion, which is an important portion of the forward bias resis-tance, can be changed from high to low by the application of aforward bias current [4]:

RI �L2

i

2mIdc

1

tinterfaceþvperim

P

A

� �ð2Þ

where Li is the thickness of the intrinsic region, m is the averageelectron end hole mobility, Idc is the forward bias current, tinterface isthe P+I interface minority carrier lifetime, vperim is the effectivehole surface recombination velocity and P/A is the PIN diodeperiphery-to-area ratio. Thus for a given diode geometry, one hasto lower the series resistance by increasing the forward biascurrent to obtain a low insertion loss.

In addition a complex circuitry is needed to ensure the suitableDC polarization of these devices and an external driver is neededin order to control their switching speed.

RF CMOS transistors have been introduced as antenna switches [5].A RF CMOS technology offers the advantage of integration on thesame die of the RF front-end standard components like charge pumpsor decoders. These components can work using a single positivevoltage source (like a battery) and are easily controlled using standarddigital signals [6].

The standard bulk CMOS technology seems to be the bestcandidate for realizing RF transceivers for use in short-rangewireless communications with not so strict requirements in noise

V. Marian et al. / Microelectronics Journal 42 (2011) 790–797 791

and power, such as Bluetooth and WLANs, at least from the point ofview of low-cost and high-level integration with baseband LSIs [7].The finite resistivity of the silicon substrate generates important lossat high frequency due to degradations in quality factors of on-chippassive elements such as spiral inductors and capacitors. Thisshortcoming has been overcome using Silicon-on-Insulator (SOI)technology. SOI switches offer satisfying isolation performances(larger than 50 dB) and a good insertion loss of less than 1 dB inthe 0.8–2.5 GHz range [8,9]. Unfortunately these switches handle alimited amount of power, around 12 dBm, considerably less thanwhat is currently obtained using HEMT processes.

Several demonstrations of antenna switches using HEMTtechnologies have been presented in literature usually usingGaAs, AlGaN/GaN or InP substrates. Low losses at high frequenciesprovided by the GaAs substrate have made it possible to obtainaround 1 dB of insertion loss and isolation performances between30 and 40 dB in a SP2T switch [10] and in a SP6T switch [11],respectively. It has also been shown [12] that the combination ofseries-shunt mounted HEMT transistors and resonant circuitscould result in an improved isolation in the frequency rangerelated to the resonant circuit.

MEMS have also been considered as a possible solution forhigh performances and low-cost RF switches. These devicesare operated by electrostatic forces: they therefore draw noquiescent current other than a very small leakage current [13].Low loss dielectrics and high conductivity metals used forthe fabrication of these devices give them a very low loss.MEMS-based RF switches suffer from limited lifetime (in theorder of 1–100 million cycles) and high operating voltage require-ments (in the range of 30–50 V) [14]. These devices remaininferior to their semiconductor competitors in terms of switchingspeed, making them unsuitable for switching GSM signals [15].

The paper presents a new SP6T structure demonstrationcapable of improved isolation performances between 42 and53 dB in the 0.8–2.5 GHz frequency range and an insertion losskept less than 1 dB. These state-of-the-art figures represent agood trade-off between isolation and insertion loss covering theentire frequency range used in mobile communication devices. Itis a first work to our knowledge that presents a full multi-standard switch, including both a 2.45 GHz branch together withGSM and DCS branches traditionally embedded.

In the first section, the paper summarizes the main charac-teristics of the GaAs technology used for the fabrication ofthe prototype integrated circuit. The second section presentsthe general structure of the circuit and the awaited improve-ments with respect to series-shunt structures. The last section

Fig. 1. Small signal equivalent circuit mo

describes the prototype integrated circuit and comparesmeasurements with simulation results. A good agreement isobtained.

2. Technology overview

A prototype SP6T circuit was fabricated using the ED02AHprocess provided by OMMIC, a supplier of epitaxy, foundryservices and MMICs based on advanced III–V processes. ED02AHis a depletion/accumulation pseudomorphic HEMT process with0.18 mm gate length [16]. The transistors have a transitionfrequency fT of 60 GHz. This process was developed for bothdigital and analog applications in the millimeter wave andmicrowave range. The ED02AH process is also adapted for high-speed digital circuits for optical connections.

The transistors generally exhibit 2, 4 or 6 gate fingers and eachgate finger can be of 15–100 mm long. Total transistor gatelengths range between 30 and 600 mm. The equivalent circuit ofa transistor for small signal simulation is presented in Fig. 1.

In this equivalent circuit model, Cg, Cd, Cs, Lg, Ld and Ls are theexternal parasitic devices to the P-HEMT, Rg, Rs and Rd arethe P-HEMT access resistances, Cgs, Cgd, Cds, Rds, Rgs, gm and tdare the classical GaAs P-HEMT equivalent circuit elements whilegate-drain conductance (Ggd), gate-source conductance (Ggs) andgate-drain series resistance (Rgd) are additional elements useful fora better accuracy in some particular operating conditions.

For most applications, the on-state impedance is primarilyresistive while the off-state impedance is mainly capacitive [17].

Only ‘‘normally off’’ transistors are used in this design. A‘‘normally off’’ transistor is characterized by a threshold voltageof Vt¼ +0.225 V. The typical value of the supply voltage is +3.3 V.A decision was made to use standard CMOS logic signals tocontrol the state of the switch: low logic level under 0.1 V (andless than 10 mA leakage current) and high logic level above 2.5 V(and less than 10 mA leakage current). Transistor models havebeen verified and used for simulation-based design of the SP6Tcircuit.

3. Circuit design and optimization

Most of the transistor-based antenna switches are built using aclassic series-shunt topology, as shown in Fig. 2. They consist ofmain switches (T1 and T2) in series with the signal path. Theyblock the signal when in off-state. The incoming signal then leaks

del of a ED02AH GaAs P-HEMT [6].

Fig. 2. Circuit schematic of a SPDT Tx/Rx switch based on the series-shunt

topology.

Fig. 3. Influence of series transistor gate width on insertion loss for frequencies

from 0.8 to 2.5 GHz.

Fig. 4. Influence of series transistor gate width on isolation in 0.8–2.5 GHz range.

V. Marian et al. / Microelectronics Journal 42 (2011) 790–797792

through a derivation transistor to the ground using secondarytransistors (T3 and T4) in on-state. Transistors operate by pairs inan exclusive manner except if the antenna must be isolated.

As shown in Fig. 2, it is possible to connect 2 branches usingthe same topology to the same antenna to realize the transmis-sion and reception paths. The two control voltages V1 and V2 areused to control both the series transistor of one branch as well asthe shunt transistor of the other branch.

Gate resistances (R1–R4) are used to control the transistor gatecurrent transients. The gate voltages’ swings remain low when ahigh RF power signal is transmitted. Capacitors C1 and C2 serve asDC isolation devices to ground.

3.1. Classical structure performances

Simulations have been performed using Advanced DesignSystem (ADS) 2008 by Agilent Technologies, in order to evaluatethe performances of a switch based on the series-shunt standardtopology using OMMIC GaAs process.

The influence of the gate width of each of the series and shunttransistors on insertion loss and isolation has been evaluated inorder to settle a trade-off. Fig. 3 shows the influence of the gatewidth of a 6-gate-finger series transistor on its insertion loss.The smaller the gate width, the larger the insertion loss andthe lower the transconductance. There is then a trade-off to besettled between isolation, the insertion loss and the maximumoperating power.

A 100 mm gate width per gate finger has been chosen torespect power capabilities. This is true for the entire frequencyrange from 800 MHz to 2.5 GHz, because frequency has practi-cally no influence on the insertion loss when the series transistoris in on-state and the gate-voltage is steady. This is mainly due tothe fact that the HEMT-transistor channel behaves much as asimple resistor (Rds) of small value (around 4 O) with negligiblecapacitive parasitic elements. The insertion loss in the seriestransistor can thus be written as

ILseries ¼ 20log 1þRds

2Z0

� �ð3Þ

At the same time

Rds ¼ Rds0=Wgate ð4Þ

where Wgate is the gate width of the transistor and Rds0 is thetechnology-specific resistance (measured in O mm). A large gatewidth allows to minimize the insertion loss. The maximum gatewidth per gate finger allowed by the ED02AH process is 100 mm.

The isolation level drops as the gate width increases, as shownin Fig. 4. In addition frequency has an important influence onisolation. This can be easily explained when analyzing theexpression for the isolation provided by the series transistor

during its off-state. The transistor mainly behaves as a capacitor(Cds) whose value is Wgate dependent:

Cds ¼ Cds0WgateþCdseNf ð5Þ

where Nf is the number of gate fingers, Cds0 is a technology-specific capacitance (in F/mm) and Cdse is the supplementaryparasitic capacitance introduced by connection paths between thegate fingers with respect to the first-level metal. The isolation ofthe series transistor can be expressed as

ISseries ¼ 10log 1þ1

2oCdsZ0

� �2 !

ð6Þ

In this case the isolation is the best for low values of thegate width.

The choice was made to prioritize the handling power and thelow insertion loss over a high isolation, because isolation can beimproved through the use of the shunt transistor and the inser-tion loss has the tendency to become more important as thenumber of branches connected to the same antenna increases.

V. Marian et al. / Microelectronics Journal 42 (2011) 790–797 793

The total gate width of the series transistor is then

Wtotal ¼Nf Windividual gate ð7Þ

This amounts to a total equivalent gate length of 600 mm foreach series transistor, the maximum allowed by this process.

Simulation has shown that the presence of a shunt transistordoes not significantly deteriorate the insertion loss of the switch(Fig. 5).

The insertion loss due to the shunt transistor off-state is givenby (8). In this mode, the shunt transistor behaves like a very smallcapacitor; hence it slightly deteriorates the adaptation:

ILshunt ¼ 10log 1þ1

2oCdsZ0

� �2 !

ð8Þ

The generated additional loss is less than 0.025 dB in fullfrequency range and disregard of the gate width. The isolation ishowever strongly dependent on the shunt transistor gate width:

ISshunt ¼ 20log 1þZ0

2Rds

� �ð9Þ

Fig. 5. Influence of shunt transistor gate width on insertion loss in 0.8–2.5 GHz range.

Fig. 6. Influence of shunt transistor gate width on isolation in 0.8–2.5 GHz range.

A higher isolation is obtained for larger gate width per finger(Fig. 6). This is technology-limited to 100 mm.

The presence of a shunt transistor would improve isolation byas much as 16 dB with hardly any influence on the insertion loss.The overall SPST structure is thus characterized by a totalinsertion loss of 0.48 dB and an isolation between 28 and 38 dBin the desired frequency domain. This is one branch of the multi-standard switch.

3.2. Improved SP6T switch structure

One possible solution when using this single branch in order tobuild a six-branch switch would be to connect six identicalbranches to the common antenna. Table 1 presents the evolutionof switch parameters at 2 GHz as a function of the number ofswitch branches connected to the antenna.

The number of branches connected to the antenna increasesthe insertion loss while the isolation remains fairly constant.Once 6-series-shunt structures are connected to the antenna asin Fig. 1, 1.1 dB insertion loss and 36 dB isolation are simulated ata frequency of 2 GHz.

The isolation was considered to be unsatisfactory compared toprevious designs. The structure is therefore modified in order toimprove the isolation according to Fig. 7a and b, containing apartial transistor-level representation on circuit layout.

Compared to the classical approach that would connect all thesix branches directly to the antenna, the proposed switch topol-ogy adds two extra switches (K1 and K2) between the antenna andthe Rx and Tx branches, respectively. Transistors K1 and K2 have600 mm gate length to preserve power handling capabilities. Thistopology is equivalent to a two-branch structure, because onlytwo branches are directly connected to the antenna at the sametime. This results in an increase of 13 dB in isolation that nowreaches 49 dB at 2 GHz. This is to our knowledge the highestisolation value obtained for a SP6T transistor-based switch in the0.8–2.5 GHz frequency range.

The presence of an additional transistor channel on the signalpath would have a tendency to increase the insertion loss, but theinsertion loss remains around 1.1 dB because only two branchesare now directly connected to the antenna as in the case with theclassical structure.

It was also considered important to evaluate the influenceof these additional switches (K1 and K2) on power handling capabil-ities and the distortion. The simulated 1 dB compression point of theclassical switch structure has been evaluated to be 23 dBm. Thepresence of switches K1 and K2 in the modified structure lowers the1 dB compression point to around 22 dBm. Their influence isnegligible due to their large gate width. The influence of theseswitches on harmonics is shown in Fig. 8. An input signal of 10 dBmat 2 GHz is injected into a transmission branch of the switch. Thespectrum presented is at the antenna level.

The presence of the additional switches K1 and K2 has very littleinfluence on the fundamental signal. However most of the harmo-nics are attenuated, sometimes by as much as 10 dB like in the caseof the second harmonics. Only 7th and 8th harmonics are slightlyincreased, but their value is very low (less than �60 dBm).

Simulated data shows that it is possible to lower the insertionloss value by inserting small series inductors at each extremity ofthe switch ports in order to improve the 50 O adaptation of thesignal paths. The needed inductors range from 1 nH to 2 nH andtheir presence lowers the insertion loss to around 0.9 dB through-out the 0.8–2.5 GHz frequency range.

Simulations have also predicted switching times as low as10 ns. This is excellent compared to other demonstrations espe-cially those using a CMOS process like in [5], where the switching

Table 1Performance comparison at 2 GHz between a classic SP6T switch structure and the proposed architecture.

Classical approach Improved structure

Number of branches Insertion loss (dB) Isolation (dB) Insertion loss (dB) Isolation (dB)

1 0.48 30 0.9 38

2 0.57 36 0.94 43

3 0.65 36 0.95 46

4 0.8 36 1 46

5 0.9 36 1.15 49

6 1.1 36 1.15 49

Fig. 7. Modified SP6T switch topology (a); partial representation of circuit

layout (b).

Fig. 8. Comparison of harmonic behavior between new and classical structures.

V. Marian et al. / Microelectronics Journal 42 (2011) 790–797794

time is around 300 ns and is comparable with other GaAsprototypes.

3.3. Limiting the DC power consumption

In order to limit the DC power consumption of the circuit, alltransistors have been polarized in the triode region with VDS

values close to zero. The DC polarization voltages of the transistordrain and source were chosen with the purpose of maximizingthe RF voltage swing without accidentally changing the state of atransistor.

A DC offset of 1.65 V was selected and applied through 10 kOon-chip resistors connected to the RF signal path.

The transistor gates are controlled using CMOS-standard logiclevels. High value resistors are used to control the transistor gateleakage current. The overall power consumption of the circuit isevaluated to be less than 500 mW.

4. Experimental results and discussion

4.1. Circuit layout and fabrication

The circuit layout was realized on a 100 mm GaAs substrate,which is characterized by a resistivity exceeding 107 O cm. Thetotal die area is approximately 1.5�2 mm2.

As seen in Section 3, DC capacitors are needed to connect theshunt transistors to the ground. The OMMIC technology offers twotypes of on-chip capacitors. The first type consists of a layer of850 nm of SiO2 between two metal layers and it features a typicalcapacity of 49 pF/mm2. The second type consists of a 150 nmlayer of SiN between two metal layers. It features a capacity of400 pF/mm2. Simulations have shown that in order to limit theinfluence of the DC capacitors on circuit performances, their valuesshould be in excess of 50 pF. Such a capacitor would occupy asemiconductor area of 350�350 mm2, resulting in an importantincrease in the circuit size. It was therefore decided to report the DCcapacitors on the IC test board.

Simulations have also shown that the presence of serial inductorsat the RF input and output ports allows an improvement of theinsertion loss by as much as 0.2 dB. As bonding wires are character-ized by a typical inductance of roughly 1 nH/mm for a 25 mmdiameter gold wire, the choice was made to use appropriate lengthsof bonding wire to add the necessary inductance values. The lengths

Fig. 9. Photograph of the fabricated GaAs integrated circuit (top). Die is

3 mm�2 mm (switches are located on the left half of the IC). Test board (bottom)

carries the external passive components.

Fig. 10. Simulated and measured insertion losses of the fabricated GaAs antenna

switch at full RF power.

V. Marian et al. / Microelectronics Journal 42 (2011) 790–797 795

of the bonding wires connected to the RF ports are between 1 and2 mm. They have been optimized after primary measurements ofthe adaptation quality.

Fig. 9 shows a photograph of the fabricated integrated circuitand the test board. The test board is fabricated on a standard1.6 mm FR-4 substrate. It includes the surface-mounted DCcapacitors and gate resistors that would have otherwise cost alarge die area penalty.

Fig. 11. Simulated and measured isolation of the fabricated circuit.

4.2. Test results

Measurements are performed with input signals from a radiofrequency signal generator, thus allowing to control the injectedpower. The logic signals came from a set of batteries in order tolimit insertion noise. The signal coming out of the switch isobserved using a spectrum analyzer.

All external cables, connectors as well as the printed circuitboard metal interconnections have been previously characterizedand the following measurement results take into account all thelosses that these elements generate.

Fig. 10 represents the evolution of the insertion loss valuecompared to the values predicted by simulation for frequenciesranging from 800 MHz to 2.5 GHz. The input signal is insertedinto a transmission port and measured on the antenna port, whilethe reception part of the switch as well as the other transmissionports are configured in off-state. The same analysis is repeated byapplying a signal into the antenna port and directing it towards areception port, while all the other branches are in their off-state.The results are in very good agreement, confirming the processlow variability or the negligible effect of the process variability oninsertion loss and isolation.

The measured values of the insertion loss are very close tothe results obtained during simulation with less than 0.1 dB

difference. Any of the fabricated switch exhibits alone an inser-tion loss value between 1 and 1.1 dB throughout the frequencyrange. This confirms the fact that carefully dimensioned bondingwires allow to improve the 50 O adaptation of the switch portsand to decrease the overall insertion loss.

The isolation is measured with the reception port in off-statewhile a signal is inserted in the transmission port in on-statetowards the antenna. The experimental and simulation results arepresented in Fig. 11.

Again a good agreement is observed between simulation andmeasurement results. The difference between the two is less than1.5 dB at most. The measured isolation ranges from up to 53 dB at800 MHz down to 42 dB at 2.5 GHz.

The total DC power consumption amounts to 480 mW, whichcan be considered as a low value when compared to the severalmW or even several tens of mW consumed by a standard LNAdesigned to work in this frequency range [18].

Measurements have been performed in order to evaluate thepower handling capabilities of the newly designed antenna switch(Fig. 12). The 1 dB compression point (P1 dB) of the circuit is over

Fig. 12. Measured P1 dB of the SP6T antenna switch at 2.45 GHz.

Fig. 13. Measured IP3 of the SP6T antenna switch at 2.45 GHz.

Table 2Summary of fabricated GaAs switch performances.

Symbol Parameter Min. Typ. Max.

Vcc Supply voltage (V) 3.0 3.8 4.0

Vlow Low level gate control voltage (V) 0.0

Vhigh High level gate control voltage (V) 3.3

Pdc DC power consumption (mW) 0.48

f Frequency range (MHz) 800.0 2500.0

IL Insertion loss (dB) 1.0 1.1

IS Isolation (dB) 42.0 53.0

P1 dB 1 dB compression point (dBm) 21.0

IP3 3rd order harmonics (dBm) interception point 47.0

Ts Switching time (ns) 10.0

Table 3Comparison of performances of this work with previous demonstrations.

Reference Technology Type Band

(GHz)

IL

(dB)

Isolation

(dB)

P1 dB

(dBm)

[5] CMOS SPDT 2.4 0.8 42 16

[6] CMOS SPDT 0–5 1.4 30 12

[7] CMOS SPDT 2.4 1.5 24 11

[8] CMOS-SOI SPDT 2.5–5 0.7 50 12

[9] CMOS-SOI SP9T 0.9–2.1 1 29–45 –

[10] HEMT GaN SPDT 0.9–2.1 1 41–46 30

[11] HEMT GaAs SP6T 0.9–1.9 1 40 35

[12] HEMT GaAs SPST 3–4 2 63 32

[15] MEMS SP9T 0–2.1 0.5 49–67 33

[19] PIN Diode SPST 2–18 0.69 28 16

[20] PIN Diode SP6T 0.9–2.1 1.2 30–36 32

[21] PIN Diode SPST 2–38 1.4 30 –

[4] PIN Diode SPST 2–16 1 19–42 –

This work HEMT GaAs SP6T 0.8–2.5 1 42–53 21

V. Marian et al. / Microelectronics Journal 42 (2011) 790–797796

21 dBm of input power and the simulation was predicting 22 dBm.The 3rd order harmonics interception point (IP3) is estimated to bearound 47 dBm, as shown in Fig. 13.

These power handling figures make the circuit suitable formost of the communication standards used by mobile handsets.

Due to the lack of necessary equipment, it is not possible tomeasure the switching time of the designed switches. However itseems reasonable to think that the experimental switching time isroughly around 10 ns as predicted by simulation. This would becoherent with other GaAs switch demonstration for high-speedlogic applications using the same technology (Table 2).

Table 3 compares the performances of proposed switch withother previous designs. The proposed circuit introduces a novelSP6T topology that allows excellent isolation while maintaining alow insertion loss. Compared to the circuit in [11], the proposedswitch has about the same amount of insertion loss, but a betterisolation over a wider bandwidth. The power handling capabilityis lower but limited by the process. It is so far sufficient for mobilecommunication standards used by modern handsets. This work isto our knowledge the only realization to include an 802.11bbranch together with the usual GSM/DCS branches in a SP6Ttopology. Moreover this device consumes very low DC power, butwe were unable to make a full comparison of this parameter withother realizations because this parameter is seldom mentioned inpublications.

5. Conclusion

The design and optimization of an integrated SP6T antennaswitch have been presented in a pseudomorphic depletion mode0.18 mm HEMT GaAs commercial process. The classical series-shuntconfiguration is modified into an original structure in order toimprove the isolation while preserving the insertion loss at full RFpower. A multi-standard and wideband switch is obtained. Theexperiments on a fabricated integrated circuit confirm the simula-tion results. The proposed switch features excellent performancesfor a 6-branch structure in terms of insertion loss, isolation, powerhandling capability and switching time. The DC power consumptionis small and the circuit is easily controlled by the standard CMOSlogic signals. These characteristics make it suitable for use in circuitsof battery-operated communication handsets where autonomy is acrucial factor.

Acknowledgments

The authors wish to thank OMMIC for the fabrication of thecircuits, as well as to Mr. Y. Gamberini for his contribution to thisproject.

V. Marian et al. / Microelectronics Journal 42 (2011) 790–797 797

References

[1] Chin-Leong Lim (Avago Technologies), Design of a PIN diode switch for high –linearity applications, RF Designline, November 2008.

[2] D. Gotch, A review of technological advances in solid-state switches, Micro-wave Journal 50 (11) (2007) 24.

[3] C. Straelhi, J.V. Bouvet, D. Goral, PIN and varactor diodes, in: B.L. Smith,M.-H. Carpentier (Eds.), The Microwave Engineering Handbook, vol. 1,Chapman & Hall, London, , 1993.

[4] P. Sun, P. Upadhyaya, D. Jeong, D. Heo, G.S.La Rue, A novel SiGe PIN diodeSPST switch for broadband TIR module, Microwave Wireless ComponentsLetters 17 (5) (2006) 352–354.

[5] C. Hove, J. Lysholm, M. Bohl, S. Lindsfors, 0.35 mm CMOS T/R switch for2.45 GHz short range wireless applications, Analog Integrated Circuits andSignal Processing 38 (2004) 35–42.

[6] P. Crippa, S. Orcioni, F. Ricciardi, C. Turchetti, A DC–5 GHz NMOSFET SPDT T/Rswitch in 0.25-mm SiGe BiCMOS technology, Applied Surface Science 224(2004) 434–438.

[7] K. Yamamoto, T. Heima, A. Furukawa, M. Ono, Y. Hashizume, H. Komurasaki,S. Maeda, H. Sato, N. Kato, A 2.4 GHz-band 1.8-V operation single-chip Si-CMOS T/R-MMIC front-end with a low insertion loss switch, IEEE Journal ofSolid-State Circuits 36 (8).

[8] C. Tinella, J.M. Fournier, D. Belot, V. Knopik, A 0.7 dB insertion loss CMOS –SOI antenna switch with more than 50 dB isolation over the 2.5 to 5 GHzband, European Solid-State Circuits Conference, 2002.

[9] A. Tombak, C. Iversen, J.-B. Pierres, D. Kerr, M. Carroll, P. Mason, E. Spears,T. Gillenwater, Cellular antenna switches for multimode applications basedon a silicon-on-insulator technology, in: Proceedings of the 2010 IEEE RadioFrequency Integrated Circuits Symposium.

[10] V. Kaper, R. Thomson, T. Prunty, J.R. Shealy, Monolithic AlGaN/GaN HEMT SPDTswitch, in: Proceedings of the 12th GAAS Symposium, Amsterdam, 2004.

[11] D. Gotch, T. Goh, R. Jackson, State-of-the-Art Low Loss High, Isolation SP6Tswitch for handset applications, European Conference on Wireless Technol-ogy, 2004, Amsterdam.

[12] D.P. Chang, Y.S. Noh, I.B. Yom, Design of High Performance HEMT Switch forS-band MSM of Satellite Transponder, IEEE Vehicular Technology Conference,2008.

[13] C. Goldsmith, J. Kleber, B. Pillans, RF MEMS: benefits & challenges of anevolving RF switch technology, IEEE GaAs Digest, 2001.

[14] R.E. Mihailovich, M. Kim, J.B. Hacker, E.A. Sovero, J. Studer, J.A. Higgins,J.F. DeNatale, MEM Relay for Reconfigurable RF Circuits, IEEE Microwave andWireless Components Letters 12 (2) (2001) 53–55.

[15] S. Lee, J.-M. Kim, Y.-K. Kim, Y. Kwon, A single-pole nine-throw antenna switchusing radio-frequency microelectromechanical systems technology forbroadband multi-mode and multi-band front-ends, Journal of Micromecha-nics and Microengineering 18 (1).

[16] ED02AH Design Manual, OMMIC, 24 October 2008.[17] M. Kameche, N.V. Drozdovski, GaAs-, InP- and GaN HEMT-based microwave

control devices: what is best and why, Microwave Journal 48 (5) (2005)164–178.

[18] Datasheet of the DCS 1800 MHz STB7002 LNA from ST Microelectronics:/http://www.datasheetcatalog.org/datasheet/stmicroelectronics/7673.pdfS.

[19] P. Sun, P. Liu, P. Upadhyaya, D. Jeong, D. Heo, E. Mina, Silicon-based PIN RFswitches for improved linearity, Microwave Symposium Digest (MTT), 2010IEEE MTT-S International.

[20] T. Yamashita, K. Fukamachi, S. Kemmochi, Low distortion and compact RFswitch circuitry with the combination of PIN-diode and GaAs-FET for GSMquartet-band cellular phone in: Proceedings of the 34th European MicrowaveConference, Amsterdam, 2004.

[21] J.G. Yang, H. Eom, S. Choi, K. Yang, 2–38 GHz broadband compact InGaAsPIN switches using a 3-D MMIC technology, in: Proceedings of theInternational Conference on Indium Phosphide and Related Materials,Matsue, Japan, 2007.