selected to be Zo � 36.04 � and Ze � 69.37 �, in such a way toconstruct 10 dB coupling.

The ADS simulated single-ended S-parameter matrix [Sstd] is

�Sstd�

� �1.71E � 5�0 0.316�0 0.949� � 90 5.69E � 6�90

0.316�0 1.71E � 5�0 5.69E � 6�90 0.949� � 900.949� � 90 5.69E � 6�90 1.71E � 5�0 0.316�0

5.69E � 6�90 0.949� � 90 0.316�0 1.71E � 5�0�

(38)

As seen above, the network essentially serves as a 10-dB direc-tional coupler. After ignoring the trivial elements, the correspond-ing mixed-mode S-parameter matrix [Smm] using (5) reads asfollows:

�Smm�

� �0.316�180 0.949� � 90 0 0

0.949� � 90 0.316�180 0 00 0 0.316�180 0.949�00 0 0.949�0 0.316�180

�(39)

Obviously, the directional coupler analyzed is a reciprocal andloss-less network. [Smm] in (39) is symmetric, which verifies thatthe MMS matrix of reciprocal network is symmetric. In addition,because

�Smm��Smm�� � �1 0 0 00 1 0 00 0 1 00 0 0 1

� � �I�, (40)

thus the unitary property of the MMSP is verified.Next, by shifting the reference plane for one of the differential

ports outward from the original location by the electronic length of90 degree and following the procedure described in (34), we obtain[SS

mm], the mixed-mode S parameter matrix of network under thenew reference planes:

�SSmm� � �

0 1.0�180 0 01.0�180 0 0 0

0 0 0 1.0�1800 0 1.0�180 0

� (41)

Meanwhile, we construct a new ADS schematic window, which isthe same as the one shown in Figure 5, except that we increase theelectric length of the lines from 90 degrees to 180 degrees. Thedirectly simulated single-ended S-parameters through the transfor-mation lead to the mixed-mode S-parameter matrix [SD

mm] underthe shifted references that reads

�SDmm� � �

0 1.0�180 0 01.0�180 0 0 0

0 0 0 1.0�1800 0 1.0�180 0

� (42)

Again the identical form of (41) and (42) validates the property ofthe shift of reference planes for the coupled transmission lines [theprocedure (34)] deduced previously.

5. CONCLUSIONS

In this paper, we have studied the properties of the mixed-modeS-parameter. It is found that

1. If a differential microwave network is reciprocal, its re-sulted mixed-mode S-parameter matrix has symmetricproperty.

2. If a differential network is lossless, both the single-endedand the mixed-mode S-parameter matrixes are unitary.

3. With the transformation matrix [P�], the MMSP of a dif-ferential network under the shifted reference planes can bedirectly obtained from those in the original referenceplanes for uncoupled network connection lines.

4. With the defined procedure, the MMSP under the shiftedreference planes can be obtained for coupled connectiontransmission lines.

ACKNOWLEDGMENT

The authors acknowledge the donation of the powerful softwareAdvanced Design System (ADS) from Agilent to the University ofSouth Carolina and the Visiting Scholar Program from NanjingUniversity of Aeronautics and Astronautics.

REFERENCES

1. D.E. Bockelman and W.R. Eisenstadt, Combined differential-mode andcommon-mode scattering parameters: Theory and simulation, IEEETrans Microwave Theory Tech 43 (1995), 1530–1539.

2. W.R. Eisenstadt, B. Stengel, and B.M. Thompson, Microwave differ-ential circuit design using mixed-mode s-parameters, Artech House,Boston, MA, 2006.

3. A. Ferrero and M. Pirola, Generalized mixed-mode S-parameters, IEEETrans Microwave Theory Tech 54 (2006), 458–463.

4. H. Shi, W. T. Beyene, J. Feng, B. Chia, and X. Yuan, Properties ofmixed-mode parameters of cascaded balanced networks and their ap-plications in modeling of differential interconnects, IEEE Trans Micro-wave Theory Tech 54 (2006), 360–372.

5. H. Erkens and H. Heuermann, Mixed-mode chain scattering parameters:theory and verification, IEEE Trans Microwave Theory Tech 55 (2007),1704–1708.

6. G. Gonzalex, Microwave transistor amplifiers: Analysis and design, 2nded., Prentice Hall, Upper Saddle River, NJ, 1997.

7. D.M. Pozar, Microwave engineering, Addison-Wesley, Reading, MA,1990, pp. 220–230.

8. D. E. Bockelma, The theory, measurement and application of scatteringparameters of propagation, Ph.D. dissertation, Dept. Elect. ComputerEng., University Florida, Gainesville, 1997.

© 2008 Wiley Periodicals, Inc.

DESIGN OF A NEW DUAL-BANDMICROSTRIP BANDPASS FILTER WITHMULTIPLE TRANSMISSION ZEROS

Lin Li and Zheng-Fan LiDepartment of Electrical and Electronic Engineering, ShanghaiJiaotong University, 800 Dongchuan Road, Shanghai, China 200240;Corresponding author: lilin_door@hotmail.com

Received 21 March 2008

ABSTRACT: A novel dual-band microstrip bandpass filter with multi-ple transmission zeros has been proposed in this article. Dual-band res-onators composed of one shunt short-circuited stub and one shunt open-circuited stub are connected by dual-band admittance inverters (J-

2874 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 50, No. 11, November 2008 DOI 10.1002/mop

inverters) to create dual-passband responses. The three transmissionzeros brought by the resonator plus the transmission zero resulted fromthe inverter will produce sharp attenuations near the passbands, andlarge isolations between the passbands. Design formulas of this type offilters are also provided in detail. To verify the presented concept, afilter with a dual-band response has been designed and fabricated withmicrostrip technology. The measured results are in good agreement withthe full-wave simulation results. © 2008 Wiley Periodicals, Inc.Microwave Opt Technol Lett 50: 2874–2877, 2008; Published online inWiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.23861

Key words: dual-band resonator; dual-band filter; dual-band J-invert-er; transmission zero

1. INTRODUCTION

The rapid progress in mobile and wireless communication tech-nology in recent years has increased the need of integrating morethan one communication standard (or mode) into a single system[1]. Under this trend, compact, high-performance radio frequency(RF), and microwave components or devices capable of operatingat more than one frequency band thus play an important role in avariety of modern communication systems.

As one of the most important components of multi-band wire-less systems, dual-band and dual-band bandpass filters are gainingwide attentions in recent years [2–7]. To meet the demand, muchresearch has been carried out and a number of topologies havebeen studied and developed to realize high-quality filters withdual-passband or multipassband responses. In [2], a dual-bandbandpass filter (BPF) was achieved by a cascade connection of aBPF and a bandstop filter, with the drawback of a large circuit size.Two individual filters are combined to provide two specific singlepassbands in [3]. However, extra impedance-matching networksmust be used to design the input and output structure in this kindof filters. Furthermore, dual-band stepped-impedance resonators[4–7] are also employed to achieve dual-passband responses.However, all the filters presented in [4–7] fail to get good transi-tions from the passbands to stopbands owing to the lack of enoughtransmission zeros beside the passbands. Dual-band filters capableof generating multiple transmission zeros still need to be pursued.

In this article, we propose a new microstrip bandpass filter thatcan be operated at two different bands, respectively. Two-band

resonators composed of one shunt short-circuited stub and oneshunt open-circuited stub are proposed in this article, which fea-tures in multiple transmission zeros, and thereby, sharp attenua-tions near the passbands and large isolations between the pass-bands can be provided. The methodology to realize this kind offilter is offered using an exemplary case. Using this method, athree-pole dual-band BPF operating at 1.9 and 4.34 GHz is de-signed and measured. The simulated frequency response of thefilter is confirmed well by the measured result.

2. ANALYSIS OF THE PROPOSED DUAL-BAND FILTER

2.1. The Proposed Dual-Band ResonatorFigure 1 shows the topology of the proposed dual-band resonator.Each resonator consists of one shunt short-circuited stub and oneshunt open-circuited stub, the characteristic impedances and elec-trical lengths of the stubs are also depicted in Figure 1. The inputimpedance of the proposed resonator is

Yin � jYo1tan2�fl1��e

c� jYo2cot

2�fl2��e

c(1)

The overall susceptance of dual-band resonator is expressed by

bin � Yo1tan2�fl1��e

c� Yo2cot

2�fl2��e

c(2)

where c is the speed of light in free space, and �e denotes theeffective dielectric constant of the substrate.

Figure 1 The topology of the proposed resonator

Figure 2 The input admittance versus frequency for the exemplary case

Figure 3 Equationuivalent circuit of the proposed dual-band filter

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 50, No. 11, November 2008 2875

Then, the dual-band resonator resonates when

Yin� foi� � 0 (3)

for i � 1, 2, 3, . . . n.where foi are the desired resonant frequencies.

Obviously, there are many solutions for the above equation,however only the lowest two nonzero ones in frequencies will bestudied to compose the dual-band resonator.

And the susceptance slope parameter �i at frequency foi isderived as follows:

�i �foi

2

�bin

�f�f�foi �

foi

2�Yo1

2�l1��e

csec2

2�foil1��e

c

� Yo2

2�l2��e

cCSC 2

2�foil2��e

c � (4)

If the two short-circuited lines’ dimensions are determined before-hand, the two resonant frequencies and the susceptance slopeparameters at the two frequencies can be calculated easily on agiven substrate. Figure 2 display the input admittance for theresonator with the dimensions arranged below: W1 � 1.5 mm, l1 �14.9 mm, W2 � 4.5 mm, l2 � 14.5 mm.

Two resonant frequencies positioned at 1.9 and 4.34 GHz canbe observed in the graph. Significantly, four zeros also can be seenat 0, 3.39, and 5.85 GHz, respectively. The zeros are brought bythe two stubs. One zero is generated once any of the short-circuitedstub’s electrical length is equal to k� (k � 0,1, 2,. . . n) or theopen-circuited stub’s electrical length is equal to 0.5k� (k � 1,2,. . . n). It can be got easily using the follow two simultaneousequations.

fzli �kc

2l1��e

(5)

fz2i �kc

4l2��e

(6)

The lowest three solutions are the required zeros, which can bedenoted as fz1, fz2, and fz3, respectively.

2.2. The Proposed Dual-Band FilterFigure 3 shows an equivalent circuit of the dual-band bandpassfilter using the proposed resonator. In the graph, the dual-bandadmittance inverter (J-inverters) is inserted into the two adjacentresonators. And the admittances of the inverter are determined bythe following well-known formulas:

J01 � �Y0FBW�1

g0g1, Jn,n�1 � �Y0FBW�n

gngn�1, Ji, j � FBW��i�j

gigj

(7)

where FBW is the fractional bandwidth of the BPF, gi (i � 0,1,2,. . . , n) is the element value of the prototype lowpass filter, �i

is the susceptance slope parameter for the ith resonator.The method to get dual-band J inverter has been provided in

[8]. The equivalent circuit of the dual-band inverter has beendepicted in Figure 4, the characteristic impedances of the trans-mission line and open-circuited stub, i.e. Zc and Zd, with theelectrical length of 90° at the average frequency of two passbands,i.e., fm � (f1 � f2)/2, can then be obtained [8]

Zc �1

Jcos��

2��, Zd �

1

Jsin��

2��tan��

2��, � �

f2 � f1

f2 � f1(8)

Figure 4 Equationuivalent circuit of the dual-band inverter

Figure 5 The layout of proposed filter

2876 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 50, No. 11, November 2008 DOI 10.1002/mop

Significantly, the open-circuited stub in the dual-band inverter willalso introduce a transmission zero at fm, which can further improvethe selectivity. Thus, there exist four transmission zeros in theresponse of the proposed filter.

3. FABRICATED Dual-BAND BPF AND MEASUREDRESULTS

To validate the method, an exemplary case is designed, simulated,and measured. The substrate used in this design has a relativedielectric constant of 2.65, a thickness of 1 mm, and a loss tangentof 0.004. The filter’s layout is demonstrated in Figure 5, whosedimensions are W1 � 1.5 mm, l1 � 15.1 mm, W2 � 4.5 mm, l2 �14.8 mm, Wd01 � 0.89 mm , Wd23 � 0.89 mm, Wd12 � 0.49 mm,ld01 � 17.04 mm, ld12 � 17.3 mm, ld23 � 17.04 mm, Wd0 � 0.19mm, Wd4 � 0.19 mm, wc01 � 1.31 mm, wc12 � 0.48 mm, wc23 �0.48 mm, wc34 � 1.31 mm, lc01 � 16.64 mm, lc34 � 16.64 mm,lc12 � 17.12 mm, lc23 � 17.12 mm, ld0 � 17.42 mm, ld4 � 17.42mm, we � 2 mm, le � 1 mm, re � 0.6 mm. And the simulated andmeasured results are illustrated in Figure 6. The measurementresults agree quite well with those obtained from simulation usingHFSS. The measured 3-dB frequency ranges (fractional band-widths) for the three passbands centered at 1.9, and 4.38 GHz, andare found to be 1640–2115 MHz (25%), and 4248–4524 MHz(6.3%), respectively. Four transmission zeros positioned at 0,3.181, 3.655, and 5.551 GHz can be observed as expected, whichhelp to produce sharp attenuations near the passbands, and largeisolations between the passbands. And the minimum insertionlosses measured for these three passbands in the same sequence are0.48 and 1.44 dB, respectively.

4. CONCLUSIONS

A novel microstrip bandpass filter capable of providing dual-bandresponses with multiple transmission zeros has been proposed inthis article. The BPF consists of dual-band resonators connected bydual-band admittance inverters. Each resonator is constructed bycombining one shunt short-circuited stub and one shunt open-circuited stub. The newly proposed resonator possesses the advan-tage of multiple transmission zeros to upgrade the filter’s perfor-mance. On the basis of the proposed concept, a third-order filterwith two bands centered at 1.9 and 4.38 GHz is designed, fabri-

cated, and measured. The close agreement between simulated andmeasured results validates well the proposed approach.

REFERENCES

1. M.I. Lai and S.K. Jeng, Compact microstrip dual-band bandpass filtersdesign using genetic-algorithm technique, IEEE Trans Microwave The-ory Tech 54 (2006), 160–168.

2. L.-C. Tsai and C.-W. Huse, Dual-band bandpass filters using equal-length coupled-serial-shunted lines and Z-transform techniques, IEEETrans Microwave Theory Tech 52 (2004), 1111–1117.

3. H. Miyake, S. Kitazawa, T. Ishizaki, T. Yamada, and Y. Nagatom, Aminiaturized monolithic dual band filter using ceramic lamination tech-nique for dual mode portable telephones, IEEE MTT-S Int MicrowaveSymp Dig 2 (1997), 789–792.

4. T.-H. Huang, H.-J. Chen, C.-S. Chang, L.-S. Chen, Y.-H. Wang, andM.-P. Houng, A novel compact ring dual-mode filter with adjustablesecond-passband for dual-band applications, IEEE Microwave WirelessCompon Lett 16 (2006), 360–362.

5. S. Sun and L. Zhu, Compact dual-band microstrip bandpass filterwithout external feeds,” IEEE Microwave Wireless Compon Lett 15(2005), 644–646.

6. S.-F. Chang, Y.-H. Jeng, and J.-L. Chen, Dual-band step-impedancebandpass filter for multimode wireless LANs, Electron Lett 40 (2004),38–39.

7. J. Wang, Y.-X. Guo, B.-Z. Wang, L.C. Ong, and S. Xiao, High-selectivity dual-band stepped-impedance bandpass filter, Electron Lett42 (2006), 538–540.

8. H.Y. Anita Yim and K.K. Michael Cheng Novel dual-band planarresonator and admittance inverter for filter design and applications,IEEE MTT-S Int Microwave Symp (2005), Long Beach, CA, pp.2187–2190.

© 2008 Wiley Periodicals, Inc.

A NOVEL LOW-COST MULTISENSOR-TAG FOR RFID APPLICATIONS INHEALTHCARE

Luca Catarinucci, Mauro Cappelli, Riccardo Colella, andLuciano TarriconeDepartment of Innovation Engineering, University of Salento, ViaMonteroni, 73100 Lecce, Italy; Corresponding author:luca.catarinucci@unile.it

Received 28 February 2008

ABSTRACT: Healthcare represents one of the most promising sectorsfor future RFID applications. RFID, indeed, candidates as a naturalsolution to a large amount of biomedical problems, from care giving totelemedicine. For a large scale introduction into this market, the inte-gration of RFID systems with a sensor network could be the turnpoint.We focus here on the implementation of a multisensor RFID system ableto send information about the physiological status of a patient by simplyexploiting its intrinsic characteristics. Without changing the digital de-sign of the chip (hence its basic cost), it is possible to transmit extra-information on sensor parameter variations. Results proposed with re-spect to the wireless remote monitoring of the patient temperaturedemonstrate the quality of the proposed technology, and the possibilityto be extended to many other biomedical sensors. © 2008 Wiley Period-icals, Inc. Microwave Opt Technol Lett 50: 2877–2880, 2008; Publishedonline in Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.23837

Key words: RFID; wireless systems; biomedical engineering; sensors

Figure 6 Simulated and measured responses of the dual-band microstripBPF

DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 50, No. 11, November 2008 2877