design, and control of a sic isolated bidirectional power

Design, and Control of a SiC Isolated Bidirectional Power Converter for V2L Applications to both DC

and AC Load

Xiaorui Wanga, Yunting Liub, Wei Qiana, Ameer Janabia, Bingsen Wanga, Xi Luc, Ke Zouc, Chingchi Chenc, Fang Z. Pengd

a Electrical and Computer Engineering, Michigan State University, East Lansing, Michigan [email protected]

b CURENT, Department of EECS, The University of Tennessee, Knoxville, Tennessee c Research and Innovation Center, Ford Motor Company, Dearborn, Michigan

d Center for Advanced Power Systems, Florida State University, Tallahassee, Florida

Abstract—EV can help export its unused energy to the external load (such as home appliance) when the vehicle is stationary. Hence a bidirectional power converter with such function is of need. Due to the safety concern, a galvanic isolation transformer is required. This paper presents an isolated universal power converter which can perform vehicle to load (V2L) feature for both AC load and DC load in one single unit. Design details such as SiC MOSFET gate drive and high frequency isolation transformer are disclosed in this paper. Control algorithm and analysis for both operation modes are also performed. Experimental results are shown to validate the hardware design and control algorithm development.

Keywords—SiC MOSFET, Vehicle to Load (V2L), On-board Charger, Dual Active Bridge, Bidirectional DC/DC, Electric Vehicle, High frequency Transformer, Nanocrystalline Core

I. INTRODUCTION

Electric vehicle (EV) is gaining more attention from customers due to its ample compelling features such as environment friendly technology, lower cost per mile, and competitive driving performance. What is more, EV can extend more functionalities besides of being only the transportation tool. Nowadays, EV with a few tens of kWh battery is the perfect candidate for smart and green home system. Hence, there is a great need to equip EV with certain power converters to make it qualified and capable for this important role to serve as the power hub interacting between grid and the EV. Bidirectional on-board charger would be the perfect power conversion unit for this job. There exists extensive research on on-board-charger design [1~3], which only focuses on system design and control for charging. However, most of the designs are fixed output, which increase more component cost whenever additional output or feature are needed. A unique circuit topology with isolation transformer was proposed in [4] to perform various essential functions such as grid to vehicle on board charging (G2V), vehicle to grid connection (V2G), and vehicle to load (V2L). In this paper, more focus is put on the V2L operation modes with both AC and DC load. Section II introduces this circuit, and section III covers the gate drive design, transformer design, and the whole prototype. Section IV shows the control analysis for the AC load along with simulation and experimental results, and Section V discusses the control for

DC load with experimental results. Section VI ends with some conclusion and future work.

II. PROPOSED CIRCUIT

The circuit shown in Fig.1 is consisting of two sections. This first part would be Q1~Q8 (SiC MOSFET), and it can be referred as load side inverter. These switches are interacting with the grid/load as the PWM rectifier or the grid connected inverter. The fourth leg is added to provide the option of neutral connection for safety concern. T1, T2, Tx, Ty are the AC side terminals. The maximum switching frequency is set to be 100 kHz. The second part would be P1~ P4, isolation transformer, and S1~S4, which are known as the dual active bridge (DAB) [1,2]. The DAB is the main workhorse to transfer energy directionally from battery to the other DC side. Since sinusoidal charging is still not fully accepted in automotive industry when operating in single phase mode, large capacitance value of 3400 uF C1 is placed to absorb the 2ω power.

1Q 3Q

2Q 4Q

1P

1C

3P

2P 4P

1S 3S

2S 4S

2C

5Q

6Q

1L

3L

2LsL

1T

2T

xT

9T

0T

1fC 2fC 3fC

3C4C

yT4L

Load Side inverter Dual Active Bridge

7Q

8Q

5L

Fig. 1. Power Converter Building Block (PCBB)

Dual active bridge is utilizing the isolation transformer’s leakage inductance as the energy transfer component. The turns ratio n of the transformer is selected as “1” [4]. The reason can be summarized to cover wide operation range and maintain equal current stress for bidirectional application scenarios. T9 and T0 are the connection points to the battery. Some other circuit parameter can be found in Table I.

III. HARDWARE PROTOTYPE DESIGN

In this section, focus will be shifted to the gate drive design and the high frequency isolation transformer design. Gate Drive: To ensure the robust gate drive performance and eliminate any possible influence from EMI and exploit potential from the SiC MOSFET [7], isolated gate drive structure is adopted. Fig. 2 [5] demonstrates the three basic elements in the gate

978-1-7281-3761-2/19/$31.00 ©2019 IEEE 143

Authorized licensed use limited to: Michigan State University. Downloaded on December 04,2020 at 18:52:33 UTC from IEEE Xplore. Restrictions apply.

drive design, isolated gate drive IC, isolated power supply, and external buffer amplifier.

Table I. Circuit Parameter

The IC selection criteria is based on the isolation technology in the IC; protection features such as Desat, active miller clamping, and UVLO; sourcing and sinking current capability; propagation delay and common mode noise immunity performance. The final choice is the IC from Analog Device [6] which meets the design requirements in terms of speed and protection. More details such as Desat tuning is not covered in this paper and can be found in [8]. The gate drive power supply selection is based upon the MOSFET characteristics such as gate charge.

_ lim

1(V V )Q

20.5*24 *570 *200 1.37

gdsw dd ss g sw itP f

V nC kHz W

(1)

2* 2.7gdPower gdswP P W (2)

Hence, a 3W power supply with dual output is used. A Zener diode is placed between the negative and ground to achieve negative bias voltage for turning off the switch [8]. Double pulse test is performed at the lower switch of the half bridge module. Fig. 3 shows the double pulse test setup, and an air-core inductor is used to reduce the influence of the core saturation.

In

Flt

Desat

+V

Out

-V

Vs1 Vs2

PWM

+Vdd

-Vss

+Vdd

-Vss

Gate Drive IC

Isolated Power Supply

External Buffer

Amplifier

Rg

Fig. 2: Gate Drive Structure

Ch1, Ch3, Ch4 in Fig.4 and Fig. 5 are drain to source votlage Vds, inductor current and gate to source votlage Vgs. The turn on time is around 120ns and the turn off time is close to 70ns.

Special attention needs to be paid on the turn-off Vgs ringing, since false turn on can be triggered with these spikes.

Fig. 3: Double pulse test

Fig. 4: Turn ON Transients

Fig. 5: Turn OFF Transients

Transformer Design:

The transformer design procedure proposed in this paper is targeted at finding a viable and feasible starting point for the isolation transformer design. There are plenty of rooms left for optimization which is not covered in this paper. Most of the converter design starts off with the simulation validation. And the transformer design can embark on with the simulation data using non-ideal transformer (known as T-shape) model shown in Fig. 6 in the process. The Llp , Lls and Lmag are the primary/secondary side leakage inductance and the magnetizing inductance. Since the turns ratio is 1:1, the leakage inductance would be equal on both sides. The goal of the transformer is to provide: 1) galvanic isolation; 2) accurate voltage step up/down ratio; 3) energy transfer using its own

L1, L2, L3, L4, L5 10uH

Ls 22uH

Cf1, Cf2, Cf3 50uF

fsw 100kHz ~ 200kHz

C1, C2 3400uF

C3 140uF

C4 100uF

144


leakage inductance. And the most important rule behind these functions is that the core can not saturate. Hence, in the simulation, voltage waveform across the magnetizing inductance is being monitored, and it can help predict the flux waveform upon the core with different combination of core shapes and winding layout. For example, smaller core area and less number of turns will quickly lead to high flux density in the core. And the core will saturate if the flux being applied is larger than its own limit. The whole design philosophy is summarized in Fig. 7.

Fig. 6 Transformer T-shape model

Get the voltage waveform on the magnetizing inductance

Calculate conductor area

Put T-shape model into simulation

Is Bmax < Bsat

Get the current information

Find flux linkage by integration

Vdt

Pick one core and get its core area

Find another coreLarger Area;Higher Bsat

Yes

Find the correct conductor based on

current rating (Litz wire gauge)

Is conductor area < core window area

Yes

Calculate number of turns 1

n

BA

B VdtnA

Change number of turns

Final Design

Fig. 7 Transformer Design Flowchart

Since the transformer proposed is designed to meet such requirements shown in Table II, the Nanocrystalline core is selected. The advantage of such core have been discovered in previous research work [9] and the comparison between other material can be found in Table III.

Design Specs Value

Voltage rating 700V

Current rating 130A

VA rating 25kVA

Leakage inductance 22uH

Operating frequency 100kHz

Table II. Design Specs for Transformer

The reasons behind the selection are the higher saturation flux density against ferrite core and relatively low core loss compared to silicon steel material under high frequency over 10kHz. The shinning part of using wide band gap device is the lower loss operating under high frequency, and the associated passive component should not incur more loss under this frequency. Higher flux saturation density limit means higher power density under same power level since it requires less area or space to process the flux.

Material Saturation Limit

Core Loss (mW/cm3)

@ 3kHz

Permeability

Silicon Steel 1.48T 147 High

Ferrite 0.4T 7.4 Medium

Nano-crystalline 1.23T 4.4 High

Table III. Core Material Comparison

Based upon the availability of the cores, F3CC0125 from Hitachi [10] is chosen along with its bobbin AMCC0125. And the number of turns on each side is chosen following the procedure in Fig.7. The design paramteres are summarized in Table IV and Fig.8 shows the core flux wavefrom of the selected core and winding pattern. The final assembly of the transformer is shown in Fig. 9.

Core Area 545 mm2

Maximum Flux 0.5T

Winding type Litz wire: 6 x 3 x 45 AWG-38 braid

No of Turns 10 (Each turn is insulated by Kapton tape)

Table IV. Transformer Assembly Parameter

To verfiy the insulation/isolation of the transformer, insulation test between primary and secondary is performed by a Fluke insulation tester shown in Fig. 10. To find out the leakage inductance of the transformer, short circuit test is performed by shorting the secondary side while giving a pulse on the primary side. From the test result shown in Fig 11, the slope of the Ch3 along with the voltage change of Ch1 can indicate the leakage inductance by (3):

lpL lsL

magL

145


diV L

dt (3)

The leakge inductance is around 22uH. Open circuit test is conducted as well to verify the turns ratio between two sides is indeed unity and magnetizing inductance is 480uH which is 20 times larger than the leakage inductance.

Fig. 8 Transformer voltage and flux waveform

Fig. 9 Transformer Prototype

Fig. 10 Transformer Insualtion Test

Fig. 11 Transformer Short Circuit Test (Left) and Open Circuit Test (Right)

The total converter prototype is shown in Fig. 12 which includes the power stage, transformer, filters and digital control platform.

Fig. 12 Proposed Converter Prototype

IV. CONTROL OF V2L – AC LOAD

Vehicle to AC load would be a very common case when the EV is stationary at home [21] when the utility power is not available due to grid fault. In this section, control structure for V2L AC load is elaborated. Simulation and experimental results are presented to validate the control structure proposed.

The converter consists of two stages 1) DAB and 2) load side inverter, and these two sections are decoupled from each other by the proposed control which will allow separate control. DAB is responsible for transferring power from the battery and stabilizing the primary DC bus. Closed loop control on the primary DC bus voltage is needed. Since a 3400uF cap is placed which means a large time constant, a relatively slow control loop whose bandwidth is around 1kHz is high enough to regulate the voltage. Meanwhile, load side inverter is providing the AC voltage to the AC load. In this paper, the most classical sine PWM is adopted. However, there exists other carrier based PWM which can perform better from the aspects of switching loss reduction and common mode voltage reduction which are not covered in this literature.

• DAB voltage closed loop: it works as inner loop whose bandwidth needs to higher than the inverter voltage loop. The primary DC bus voltage reference is not an easy decision. For example, it needs to be close enough to the battery voltage on the secondary side to minimize the reactive power and current stress inside the DAB to reduce the conduction loss. However, in this paper, the dc bus voltage is not optimally selected considering this factor. The DC bus voltage reference can also be modified by the load voltage reference if the DC bus is not high enough to provide the desired load line-line voltage when under highest modulation index. This idea is also summarized in Fig. 13 as red “if” loop and green “if” loop. The main goal for this loop is to dynamically adjust the phase shift between primary bridge and secondary bridge to realize the power flow hence regulating the primary DC bus voltage in a closed loop manner [11~16].

• Inverter voltage loop: As mentioned above, the inverter loop may generate a feedforward value to change the DC voltage reference depending on the load voltage requirement. The main goal for this loop is to dynamically change the modulation index to regulate the output voltage and control it as a “stiff” voltage source.

146


Experiments are carried out to verify the V2L control algorithm and system parameters are summarized in Table V. A step change of primary side voltage reference is commanded. Fig 14 shows the response time of primary side voltage is around 15ms. And the load side voltage is changing accordingly. The dynamic adjustment of phase shift is the reason of the primary side DC link is being regulated. And these adjustments can be identified in Fig 15 and 16.

Hardware Setup Value Switching frequency 100kHz

Load resistance per phase 13 Ω

Filtering inductance 10uH

Filtering capacitance 50uF

DC Source voltage 120V

Inverter modulation index 1.0 Primary voltage reference

step up change 80V --- > 140VTable V. Experiment setup for V2L AC load

Fig.14 Inverter voltage output and current during step up

(Ch1, Ch2 output voltage; Ch3 primary side DC bus voltage; Ch4 output current)

Fig. 15 DAB voltage output and current during step up

(Ch1 DAB primary side voltage; Ch2 primary side DC link voltage, Ch3 DAB secondary side voltage; Ch4 DAB leakage inductance current)

Fig. 16 Details of transition of DAB

When the converter operates under V2L modes, it needs to provide “stiff” AC voltage no matter what load is connected. Assuming the primary bus voltage is constant by regulating the DAB, simulation of dynamic load change including linear and non-linear load is carried out to help validate the inverter side control algorithm as shown in Fig 17. The simulation result in Fig. 19 shows that even the current is highly distorted,

1Q 3Q

2Q 4Q

1P

1C

3P

2P 4P

1:11S 3S

2S 4S

2C

5Q

6Q

1L

3L2L sL1T

2T

xT

9T

0T

1fC 2fC 3fC

5L

*1dcV D

D

*sV

sV* 3

22dc

s

Vif V MI

* 3

22dc

s

Vif V MI

Fig 13 V2L- AC load Control Structure

147


the output voltage seen by the load is still sinusoidal which the same case as the grid voltage output is.

Fig. 17 Simulation of V2L-AC Load

Simulation Parameter Value Switching frequency 100kHz

Load resistance 13 Ω


Filtering capacitance 50uF

Inverter modulation index 1.0

DC source voltage 400V

Load step change @0.05s

Load Change Add more linear load/ Add rectifier bridge

Table VI. V2L AC Load Simulation Parameter

Another reason behind the stiffness is the filtering inductance is small, which means the internal impedance of the voltage source is small. If the load impedance is not comparable to internal impedance, the voltage source inverter can be viewed as ideal voltage source. To further verify the idea, filtering inductance is changed to 10mH and with load resistance is 0.1 ohm. Pronounced voltage drop (@ 0.02s) can be observed in Fig. 20, since the load and the internal impedance is working as a voltage divider. It is also important to point that low filtering inductance can help to improve the system stability with LC filter and reduce the need for resistive damping. That is another advantage of adopting high frequency switching which is enabled by using SiC MOSFET.

Fig. 18 Output voltage and current under linear load change

(Top – Output voltage, Bottom – Output current )

Fig. 19 Output voltage and current under non-linear load change


Fig. 20 Output voltage and current with large filter


V. CONTROL OF V2L - DC LOAD

Besides being able to provide the AC load, this converter is capable of another feature which is powering DC load such as 48V DC motor. In this section, control structure for the proposed converter is examined along with experiment results shown. To output DC, the load side switches need to work as interleaved buck converter mode [17~21], while the DAB is being controlled the same way as in AC load. The control structure is shown in Fig 21. The hardware setup is summarized in Table VII.

Hardware Setup Value Switching frequency 100kHz

Phase shift among each leg 120°

Load resistance 10 Ω


Filtering capacitance 150uF(3*50uF)

DC source voltage 200V

Output voltage reference 48V

Duty ratio of each leg 48/200 = 0.24

Table VII. Experimental setup for V2L – DC load

148


Fig. 22 Phase shifted PWM signal to each leg

Fig. 23 Ch1- DC load voltage; Ch2- Single leg current before capacitor;

Ch3- Secondary DC bus voltage; Ch4 – Final load current

Fig. 24 Ch1,2,3 Each leg current before capacitor; Ch4 – Merged leg current before capacitor

Fig. 22 shows the 120° phase shifted PWM signal of each leg. The experiment results in Fig. 23 verify the high step down buck function by the converter from 200V to 48V.

VI. CONCLUSION AND FUTURE WORK

This paper presents the gate drive design, transformer design of an isolated bidirectional power converter which can provide both AC and DC power in one single unit. 25kW hardware prototype is built to validate the design and control algorithm. However, there are some future work which can be explored to optimize the system. For example, efforts can be put to find the best the modulation methods of the load side inverter under AC load in terms of minimal DC link voltage ripple. Another thread is to locate the optimized primary DC bus voltage to reduce both the switching loss of the inverter and DAB under different operation modes.

REFERENCES

[1] M. Yilmaz and P. T. Krein, "Review of Battery Charger Topologies,

Charging Power Levels, and Infrastructure for Plug-In Electric and Hybrid Vehicles," in IEEE Transactions on Power Electronics, vol. 28, no. 5, pp. 2151-2169, May 2013.

[2] Y. Du, X. Zhou, S. Bai, S. Lukic and A. Huang, "Review of non-isolated bi-directional DC-DC converters for plug-in hybrid electric vehicle charge station application at municipal parking decks," 2010 Twenty-Fifth Annual IEEE Applied Power Electronics Conference and Exposition (APEC), Palm Springs, CA, 2010, pp. 1145-1151.

[3] B. Whitaker et al., "A High-Density, High-Efficiency, Isolated On-Board Vehicle Battery Charger Utilizing Silicon Carbide Power Devices," in IEEE Transactions on Power Electronics, vol. 29, no. 5, pp. 2606-2617, May 2014.

[4] X. Wang et al., "A 25kW SiC Universal Power Converter Building Block for G2V, V2G, and V2L Applications," 2018 IEEE International Power Electronics and Application Conference and Exposition (PEAC), Shenzhen, 2018, pp. 1-6.

[5] David Levett, Tim Frank, and Dave Divins “Infineon APEC 2018 Professional Education Seminar Gate Driver Design for IGBT and SiC MOSFET Based Power Device and Modules”

[6] ADum4135 Gate drive IC https://www.analog.com/en/products/adum4135.html

1Q 3Q

2Q 4Q

1P

1C

3P

2P 4P

1:11S 3S

2S 4S

2C

5Q

6Q

1L

3L2L sL1T

2T

xT

9T

0T

1fC 2fC 3fC

5L

*1dcV D

D

4T

*loadV

D

1

3 sf

Fig. 21 V2L DC load Control Structure

149


[7] Rohm SiC MOSFET Power module https://www.rohm.com/products/sic-power-devices/sic-power-module/bsm120d12p2c005-product

[8] Xiaorui Wang, “Design, Analysis, and Control of A SiC Bidirectional G2V, V2L and V2G Universal Power Converter in Next Generation Electric Vehicle”, Ph.D dissertation, Michigan State University, East Lansing, 2019

[9] Y. Du, S. Baek, S. Bhattacharya and A. Q. Huang, "High-voltage high-frequency transformer design for a 7.2kV to 120V/240V 20kVA solid state transformer," IECON 2010 - 36th Annual Conference on IEEE Industrial Electronics Society, Glendale, AZ, 2010, pp. 493-498

[10] Hitachi Core: https://www.hitachimetals.com/materials-products/amorphous-nanocrystalline/powerlite-c-cores/documents/AMCC-F3CC-series-Cut-Cores_hl-fm27-c.pdf

[11] Hui Li, Fang Zheng Peng and J. S. Lawler, "A natural ZVS medium-power bidirectional DC-DC converter with minimum number of devices," in IEEE Transactions on Industry Applications, vol. 39, no. 2, pp. 525-535, March-April 2003.

[12] S. S. Shah and S. Bhattacharya, "Large and small signal modeling of dual active bridge converter using improved first harmonic approximation," 2017 IEEE Applied Power Electronics Conference and Exposition (APEC), Tampa, FL, 2017, pp. 1175-1182.

[13] Hua Bai, Chunting Mi, Chongwu Wang and S. Gargies, "The dynamic model and hybrid phase-shift control of a dual-active-bridge converter," 2008 34th Annual Conference of IEEE Industrial Electronics, Orlando, FL, 2008, pp. 2840-2845.

[14] J. A. Mueller and J. W. Kimball, "An Improved Generalized Average Model of DC–DC Dual Active Bridge Converters," in IEEE Transactions on Power Electronics, vol. 33, no. 11, pp. 9975-9988, Nov. 2018.

[15] D. Segaran, B. P. McGrath and D. G. Holmes, "Adaptive dynamic control of a bi-directional DC-DC converter," 2010 IEEE Energy Conversion Congress and Exposition, Atlanta, GA, 2010, pp. 1442-1449.

[16] D. Segaran, D. G. Holmes and B. P. McGrath, "Enhanced Load Step Response for a Bidirectional DC–DC Converter," in IEEE Transactions on Power Electronics, vol. 28, no. 1, pp. 371-379, Jan. 2013.

[17] I. Lee, S. Cho and G. Moon, "Interleaved Buck Converter Having Low Switching Losses and Improved Step-Down Conversion Ratio," in IEEE Transactions on Power Electronics, vol. 27, no. 8, pp. 3664-3675, Aug. 2012.

[18] J. Abu-Qahouq, Hong Mao and I. Batarseh, "Multiphase voltage-mode hysteretic controlled DC-DC converter with novel current sharing," in IEEE Transactions on Power Electronics, vol. 19, no. 6, pp. 1397-1407, Nov. 2004.

[19] J. Zhang, J. S. Lai, R. Y. Kim and W. Yu, "High-Power Density Design of a Soft-Switching High-Power Bidirectional dc–dc Converter," in IEEE Transactions on Power Electronics, vol. 22, no. 4, pp. 1145-1153, July 2007.

[20] O. Garcia, P. Zumel, A. de Castro and A. Cobos, "Automotive DC-DC bidirectional converter made with many interleaved buck stages," in IEEE Transactions on Power Electronics, vol. 21, no. 3, pp. 578-586, May 2006

[21] S. Yang, Q. Lei, F. Z. Peng and Z. Qian, "A Robust Control Scheme for Grid-Connected Voltage-Source Inverters," in IEEE Transactions on Industrial Electronics, vol. 58, no. 1, pp. 202-212, Jan. 2011.

[22] Y. Fu, Y. Huang, H. Bai, X. Lu, K. Zou and C. Chen, "A High-efficiency SiC Three-Phase Four-Wire inverter with Virtual Resistor Control Strategy Running at V2H mode," 2018 IEEE 6th Workshop on Wide Bandgap Power Devices and Applications (WiPDA), Atlanta, GA, 2018, pp. 174-179.

150


design, and control of a sic isolated bidirectional power

Documents