compact k/ka-band downlink antenna array elements for next

Laura Van Messem

Next-Generation Avionic Satellite CommunicationsCompact K/Ka-Band Downlink Antenna Array Elements for

Academic year 2017-2018Faculty of Engineering and ArchitectureChair: Prof. dr. ir. Bart DhoedtDepartment of Information Technology

Master of Science in Electrical EngineeringMaster's dissertation submitted in order to obtain the academic degree of

Counsellors: Ir. Kamil Yavuz Kapusuz, Dr. ir. Sam Lemey, Ir. Thomas DeckmynSupervisors: Prof. dr. ir. Hendrik Rogier, Prof. dr. ir. Dries Vande Ginste

Preface

First and foremost, I would like to express my sincerest gratitude to Prof. dr. ir. HendrikRogier and Prof. dr. ir. Dries Vande Ginste, for allowing me to perform research regardingthis master’s dissertation at the Electromagnetics Group of the Department of InformationTechnology. Moreover, I would like to thank them for their vast experience and knowledge,which have guided me towards well substantiated decisions, and also for their endless passionregarding their field of expertise.

Special thanks go to Ir. Kamil Yavuz Kapusuz, who led me towards rewarding results withhis extensive guidance and profound feedback. He was always available, not only for theoreticalsupport, but also as a personal coach. Subsequently, I would like to thank Ir. Thomas Deckmyn,for teaching me the fundamental basics of good research and stating well-founded conclusions,and also for the support during the entire year. My sincere gratitude is shown to Dr. ir. SamLemey, who offered aid in any way he could, always taking time to explain his point of view.

Distinct credit goes to Ing. Quinten Van den Brande, Ir. Olivier Caytan, Prof. dr. ir. PieterRombouts, Ir. Martijn Huynen and Ir. Niels Lambrecht for the occasional advice, aid and peptalks. Furthermore, I have to thank Dries Bosman, Igor Lima de Paula, Stijn Cuyvers, StijnPoelman, Lars De Brabander, Pieter Decleer, Stefan Wouters, Joran Claeys and Ozan Catal forparticipating in my brainstorm sessions and for giving meticulous feedback on enquiries.

Ultimately, I would like to give special thanks to my parents, who believed in me every step ofthe way. Last, but definitely not least, special thanks and appreciation goes to Sam Buysse, forbeing my mental and emotional backbone throughout the years, providing me with indispensableencouragement.

You are all one of a kind. Expressing my gratitude through words will never suffice, since theywill always fall short.

Laura Van Messem, May 2018

Admission to Loan

The author gives permission to make this master’s dissertation available for consultation andto copy parts of this master’s dissertation for personal use. In the case of any other use, thelimitations of the copyright have to be respected, in particular with regard to the obligation tostate expressly the source when quoting results from this master’s dissertation.

Laura Van Messem, May 2018

Compact K/Ka-Band Downlink AntennaArray Elements for Next-Generation

Avionic Satellite Communicationsby

Laura VAN MESSEM

Master’s Dissertation submitted to obtain the academic degree ofMaster of Science in Electrical Engineering

Academic 2017–2018

Promotors: Prof. dr. ir. Hendrik ROGIER, Prof. dr. ir. Dries VANDE GINSTESupervisors: Ir. Kamil YAVUZ KAPUSUZ, Dr. ir. Sam LEMEY, Ir. Thomas DECKMYN

Faculty of Engineering and ArchitectureGhent University

Department of Information TechnologyChairman: Prof. dr. ir. Bart DHOEDT

Summary

A novel, circular, cavity backed air-filled substrate integrated waveguide (afsiw) antenna element,covering the civil K/Ka-band downlink [17.7GHz, 20.2GHz], is designed to have a small footprint,low profile and to be lightweight, cost-effective and robust. The antenna is suited for phasedarray applications that make use of electric beam steering. The proposed stacked pcb antennatopology achieves a realized gain of 4.8 dBi and a 3 dB beamwidth of 97° at the centre frequencyof 18.95GHz with dimensions of 12mm x 12mm x 3.88mm. The radiation efficiency is 96.8%and the total efficiency is 94.8%, while the axial ratio (ar) stays below 3 dB for elevation anglesθ between [−44°, 44°] at the centre frequency.

Keywords

Circular cavity-backed afsiw; K/Ka-band downlink; satellite communication; cost-effective;small footprint; low profile; stacked pcb.

1

Compact K/Ka-Band Downlink Antenna ArrayElements for Next-Generation Avionic Satellite

CommunicationsLaura Van Messem

Promotors: prof. dr. ir. H. Rogier and prof. dr. ir. D. Vande GinsteSupervisors: ir. K. Yavuz Kapusuz, ir. T. Deckmyn and dr. ir. S. Lemey

Abstract—A novel, circular, cavity backed air-filled substrateintegrated waveguide (AFSIW) antenna element, covering the civilK/Ka-band downlink [17.7 GHz, 20.2 GHz], is designed to havea small footprint, low profile and to be lightweight, cost-effectiveand robust. The antenna is suited for phased array applicationsthat make use of electric beam steering. The proposed stackedPCB antenna topology achieves a realized gain of 4.8 dBi anda 3 dB beamwidth of 97 at the centre frequency of 18.95 GHzwith dimensions of 12 mm x 12 mm x 3.88 mm. The radiationefficiency is 96.8% and the total efficiency is 94.8%, while theaxial ratio (AR) stays below 3 dB for elevation angles θ between[−44, 44] at the centre frequency.

Index Terms—circular cavity-backed AFSIW; K/Ka-banddownlink; satellite communication; cost-effective; small footprint;low profile; stacked PCB.

I. INTRODUCTION

TODAY’S society has an ever-growing demand for dataavailability. Therefore, it is interesting to start investing in

high data-rate connectivity for moving vehicles. Fundamentalin case of satellite communications is the realisation of smallfootprint, low profile, light weight, cost-effective and robustantenna systems [1]. The integration of user-terminals in high-mobility applications, such as aeroplanes, moving vehicles,etc., is more likely when the system meets the aforementionedrequirements.

Higher frequency bands usually have a larger availablebandwidth, facilitating higher data rates. Therefore the civilK/Ka-band downlink [17.7 GHz, 20.2 GHz] is studied for thenext-generation satellite communication systems. However,these benefits have their counterparts. High frequency signalsrequire expensive radio frequency (RF) equipment to be mea-sured and analysed. This also leads to increased complexitywhen processing these signals. Moreover, a higher path lossis inevitable and, therefore, more gain is required for a properoperation. Sufficient gain is usually obtained by using largerantenna arrays, implying larger feeding networks and thuscreating more losses. Furthermore, frequencies higher than 10GHz are also more susceptible to rain fade, for which solutionsare proposed in the literature [2].

Reflect arrays or lens antennas are only used for basestations, since they are bulky and heavy [3], [4], [5], [6],[7]. Moreover, they make use of mechanical beamsteering,which prevents the system from being responsive. Therefore,an investigation is carried out towards planar phased antenna

arrays that make use of electrical beamsteering, suited forfuture user terminals [8].

A Ka-band dual frequency shorted annular patch, fed by amicrostrip line coupled through a slot, was proposed in [9].The dimensions of the prototype with casing are much largerthan λ/2, 40 mm x 40 mm x 11.5 mm, it lacks robustness, thedesign is rather expensive to fabricate and the reserved civilfrequency band is not entirely covered. However, a maximumgain of 7.1 dBi is attained and the paper suggests an interestingarray configuration.

A low profile, small footprint patch antenna for phasedarrays was proposed in [10], targeting the frequency spectrumaround 30 GHz. The maximum gain for a one-by-eight arraywas 8.6 dBi. A rather mediocre maximum gain and a limitedscanning range of 60 was obtained, while requiring a bulkyfeeding structure.

A compact circular polarized cavity-backed stacked-patchantenna with four-port capacitive feeding is illustrated in [11],covering the reserved civil and military K/Ka-band downlink.Subsequently, a compact low profile and circularly polarizedK/Ka-band antenna with a very large bandwidth, covering ap-proximately from 17.6 GHz to 30.4 GHz, is presented in [12].A dual-band five-layer circularly polarized antenna element,that covers the reserved civil and the military frequencies andexhibits a gain of about 8 dBi, is discussed in [13]. All ofthese designs can be used as measures for the performance ofthe proposed solution of this manuscript. However, fabricationcomplexity increases, due to their printed circuit board (PCB)multi-layer structure.

In conclusion, it is possible to state that most of theproposed solutions make trade-offs. Often, not the entire K/Ka-band downlink and/or uplink are covered, or they suffer fromhigh losses, large footprints, large profiles or high costs.

II. ANTENNA DESIGN

A. Design Specifications

Aforementioned requirements should be fulfilled, being thecreation of a small footprint, low profile, light weight, cost-effective and robust antenna that can be integrated in a phasedantenna array for the next-generation satellite communication.Basic specifications can be formulated as a guideline basedon the literature, since no ITU regulations are predeterminedfor the K/Ka-downlink. A height below 5 mm and a footprint

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smaller than 15 mm x 15 mm seem realistic. Moreover, a gainof more than 4 dBi and a 3 dB beamwidth of more than 60 isdesirable. The choice of fabrication materials should be basedon creating a reliable and cost-effective antenna system.

B. Antenna Topology

Using a cylindrical resonator cavity, as illustrated in Figure1, allows easy excitation of circular polarization (CP) modesand it increases the realized gain compared to traditional patchantennas.

wd

whhd

FR-4t

ROt

cut

afsiww afsiwh

cus

spacing mlinw,top mlinl,top

mlinl,bottom

x

z

Figure 1: Side view of the circular cavity-backed AFSIW.

Left-handed CP (LHCP) and right-handed CP (RHCP) canboth be induced, depending on the direction of the sequentialfeeding of the microstrip lines in Figure 2b.

The four microstrip lines on the bottom plane, drawnin Figure 2b, are fed with 90 phase shifts in clockwiseor counter-clockwise sequential feeding, to obtain LHCP orRHCP, respectively. Thereafter, these fields are coupled intothe AFSIW and propagate to the top plane, where they coupleto the four microstrip lines that feed the cylindrical resonatorcavity.

C. Operation Principle

This feeding topology induces the circular transverse mag-netic (TM)11 excited mode. Constructive interference willoccur, since the height of this cavity is designed to be approx-imately λ18.95GHz/4, allowing the fields to radiate efficiently.The wavelength λ18.95GHz is based on the centre frequencyand the effective relative permittivity of the cavity.

Additional grounded vias are added all around the structure,such that the fields are more confined within the waveguide, in-creasing the antenna efficiency. The via diameter is representedby viad, while the spacing between the centres of adjacentvias is called vias and the spacing between a via edge and thecopper edge is viae [14], [15].

D. Fabrication technology

This cavity is constructed with standard PCB technology,using two layers of FR-4 laminates and two PCB layers withRO4350BTM substrate as feeding layers on the top and bottom.Cost is kept to a minimum by using cheap standard PCBsand relatively inexpensive Rogers substrates. The RO4350BTM

laminate exhibits a relative permittivity of εr = 3.66 at theK/Ka-downlink band and shows excellent behavior at highfrequencies. The stacked PCBs are assembled by providingnon-plated through holes (NPTH) for screws.

ROw

ROlhd

mlinl,top mlinw,topwd

margin

spacing

cus

x

y

(a)ROw

ROlhd

mlinl,bottom mlinw,bottom

spacing

x

y

(b)

afsiwα

FR-4w

FR-4l

hd

wdplating afsiww

cus

x

y

(c)

Figure 2: Layout of the cavity-backed AFSIW antenna topologyin (a) top view (b) bottom view. (c) Top view of both drilledand edge plated standard PCBs with FR-4 substrate.

The manufacturer imposes certain fabrication limits, whichare carefully documented in their design guidelines. Conse-quently, straight corners in a PCB can not be manufactured,the milling tool has a certain radius, called milr.

E. Antenna dimensions

Initial antenna dimensions are based on well-known cut-offfrequencies for the desired propagating modes [16]. Cut-offfrequencies for circular excited modes should be calculatedfor the circular resonator cavity, while excited modes for arectangular waveguide can be used for the AFSIW structures.The widths of the microstrip lines are matched to a lineimpedance of 50 Ω for measurement purposes, to ensure goodpower transfer. The optimized dimensions obtained from CSTMWS R© are presented in Table I.

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Parameter Dimensionscut 0.035 mmFR-4t 1.55 mmFR-4w 12 mmFR-4l 12 mmROt 0.25 mmROw 12 mmROl 12 mmmilr 0.5 mmwd 8 mmwh 3.56 mmhd 1.0 mmAFSIWw 0.5 mmAFSIWα 82

Parameter Dimensionsmlinw,top 0.5 mmmlinl,top 3.9 mmmlininset 2.4 mmmlinw,bottom 0.475 mmmlinl,bottom 2.2 mmmargin 0.4 mmspacing 0.1 mmcus 0.2 mmviad 0.25 mmvias 0.5 mmviae 0.15 mm

Table I: Optimized dimensions for the cavity-backed AFSIWantenna topology.

III. SIMULATION RESULTS

Four sequentially fed microstrip lines are excited at the sametime to achieve CP. Therefore, it is beneficial to look at theactive scattering parameters Sa, as defined in Equation (1).

S11,a = S11(0) + S12(90

) + S13(180) + S14(270

)

S22,a = S21(0) + S22(90

) + S23(180) + S24(270

) (1)S33,a = S31(0

) + S32(90) + S33(180

) + S34(270)

S44,a = S41(0) + S42(90

) + S43(180) + S44(270

)

The simulated active S-parameters cover the entire K/Ka-downlink, as can be seen from Figure 3. The symmetry of thedesign ensures that the S-parameters are approximately equalto each other. The simulated radiation efficiency does not dropbelow 96% for the entire desired frequency range, while thetotal efficiency does not drop below 87%.

16 18 20 22−70

−60

−50

−40

−30

−20

−10

017.65 20.42

2.77GHz

Frequency [GHz]

|Sa|[dB]

|S11,a||S22,a||S33,a||S44,a|

Figure 3: Active S-parameters of the circular cavity-backedAFSIW antenna topology.

As a consequence of the symmetry of this design, similarradiation patterns arise for φ = 0 and φ = 90. Therefore,solely the radiation patterns for φ = 0 are plotted in Figure4. An overview of the farfield characteristics is given in TableII. The front-to-back ratio is sufficiently large, but could beimproved by replacing the microstrip lines on the bottomlayer by grounded coplanar waveguide (GCPW) feeding lines.Consequently, these GCPWs should be optimized for a lineimpedance of 50 Ω.

Simulating the axial ratio (AR) reveals a reliable design,since the AR stays below 0.033 dB for the entire downlinkfrequency range when the elevation angle θ = 0. Moreover,

030

60

90

120

150180

−150−120

−90

−60−30

−10−50510

θ

gain

[dBi]

(a)

030

60

90

120

150180

−150−120

−90

−60−30

−10−50510

θ

gain

[dBi]

(b)0

30

60

90

120

150180

−150−120

−90

−60−30

−10−50510

θ

gain

[dBi]

(c)

Figure 4: Radiation patterns for the cylindrical cavity-backedAFSIW topology for φ = 0 at (a) 17.7 GHz (b) 20.2 GHz and(c) 18.95 GHz.

Frequency 17.7 GHz 18.95 GHz 20.2 GHzMain Lobe Magnitude [dBi] 5.05 4.83 4.52Main Lobe Direction [] 0 0 0Angular Width (3 dB) [] 81.0 97.1 106.0Side Lobe Level [dB] -6.7 -7.5 -5.8

Table II: Overview of radiation pattern characteristics for thecircular cavity-backed AFSIW topology.

the AR stays below 3 dB for a range of 88 for the centrefrequency fc = 18.95 GHz, which means that CP can beassumed in this region.

A sensitivity analysis is performed before manufacturingthe antenna element. In general, the thickness of the FR-4substrates and all dimensions related to the microstrip linesare critical parameters. However, they can withstand errors inthe order of standard PCB manufacturing tolerances.

IV. CONCLUSION

In this manuscript, a circular cavity-backed AFSIW antennawas proposed for operation in the civil K/Ka-downlink fre-quency band. The bandwidth of the circular cavity-backedAFSIW topology covers the desired frequency band of op-eration, whilst not collecting undesired frequencies and thusminimizing interference. A broadside realized gain of 5.05 dBiis achieved and a maximum angular 3 dB beamwidth of 106 isrealised. Circular polarization can be presumed within a rangeof 88 for the elevation angle θ, since the axial ratio staysbelow 3 dB in this region. In conclusion, a highly-efficient,small footprint, low profile, lightweight, cost-effective, robustantenna for phased array applications has been developedusing stacked PCB technology.

REFERENCES

[1] J. S. Herd and M. D. Conway, “The Evolution to Modern Phased ArrayArchitectures”, Proceedings of the IEEE, vol. 104, no. 3, pp. 519-529,March 2016.

[2] L. J. Ippolito and L. J. Ippolito Jr, “Satellite communications systemsengineering: atmospheric effects, satellite link design and system perfor-mance”, John Wiley & Sons, 2017.

4

[3] R. Głogowski and J. Zurcher and C. Peixeiro and J. R. Mosig, “Alow-loss planar Ka-band antenna subarray for space applications”, IEEETransactions on antennas and propagation, vol. 61, no. 9, pp. 4549–4557,September 2013.

[4] J. S. Silva and E. B. Lima and J. R. Costa and C. A. Fernandes and J. R.Mosig, “Design and analysis of a Ka-band coaxial-to-quad-ridged circularwaveguide transition”, The 8th European Conference on Antennas andPropagation (EuCAP 2014), pp. 7-9, April 2014.

[5] J. S. Silva and E. B. Lima and J. R. Costa and C. A. Fernandes and J.R. Mosig, “Tx-Rx Lens-Based Satellite-on-the-Move Ka-Band Antenna”,IEEE Antennas and Wireless Propagation Letters, vol. 14, pp. 1408-1411,2015.

[6] J. S. Silva and M. Garcıa-Vigueras and T. Debogovic and J. R. Costa andC. A. Fernandes and J. R. Mosig, “Stereolithography-Based Antennas forSatellite Communications in Ka-Band”, Proceedings of the IEEE, vol.105, no. 4, pp. 665-667, April 2017.

[7] J. S. Silva and M. Garcıa-Vıgueras and T. Debogovic and J. R. Mosig,“3D-printed Ka-band antenna based on stereolithography”, 2017 11thEuropean Conference on Antennas and Propagation (EUCAP), pp. 589-593, March 2017.

[8] Q. Luo and S. Gao, “Smart antennas for satellite communications onthe move”, Antenna Technology: Small Antennas, Innovative Structures,and Applications (iWAT), 2017 International Workshop on, pp. 260-263,March 2017.

[9] E. Arnieri and L. Boccia and G. Amendola, “A Ka-Band Dual-Frequency Radiator for Array Applications”, IEEE Antennas and WirelessPropagation Letters, vol. 8, pp. 894-897, 2009.

[10] T. Lambard and O. Lafond and M. Himdi and H. Jeuland and S. Bolioliand L. Le Coq, “Ka-Band Phased Array Antenna for High-Data-RateSATCOM”, IEEE Antennas and Wireless Propagation Letters, vol. 11,pp. 256-259, 2012.

[11] J. S. Silva and M. Garcıa-Vigueras and M. Esquius-Morote and J. R.Costa and C. A. Fernandes and J. R. Mosig, “Lens-based Ka-band antennasystem using planar feed”, 2015 9th European Conference on Antennasand Propagation (EuCAP), pp. 1-4, May 2015.

[12] P. Gorski and J. S. Silva and J. R. Mosig, “Wideband, low profile andcircularly polarized K/Ka band antenna”, 2015 9th European Conferenceon Antennas and Propagation (EuCAP), pp. 1-3, May 2015.

[13] H. Hasani and J. S. Silva and J. R. Mosig and M. Garcia-Vigueras,“Dual-band 20/30 GHz circularly polarized transmitarray for SOTMapplications”, 2016 10th European Conference on Antennas and Propa-gation (EuCAP), pp. 1-3, April 2016.

[14] J. E. Rayas-Sanchez and V. Gutierrez-Ayala, “A general EM-baseddesign procedure for single-layer substrate integrated waveguide inter-connects with microstrip transitions”, Microwave Symposium Digest,2008 IEEE MTT-S International, pp. 983–986, 2008.

[15] M. Bozzi and A. Georgiadis and K. Wu, “Review of substrate-integrated waveguide circuits and antennas”, IET Microwaves, Antennas& Propagation, vol. 5, no. 8, pp. 909–920, June 2011.

[16] H. D. Young and R. A. Freedman, “University physics with modernphysics”, Pearson, 2015.

CONTENTS I

Contents

List of Figures i

List of Tables iv

List of Abbreviations v

1 Introduction 11.1 Context . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11.2 Frequency Spectrum . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11.3 Goals and Outline . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

2 Literature Review 52.1 Concept . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52.2 State-of-the-Art . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52.3 Proposed Solution . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6

3 Antenna Element with Microstrip Line Feed 73.1 Proposed Solutions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7

3.1.1 Antenna Topology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 73.1.2 Fabrication Technology . . . . . . . . . . . . . . . . . . . . . . . . . . . . 93.1.3 Antenna Dimensions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103.1.4 CST Solver Remarks : Meshing . . . . . . . . . . . . . . . . . . . . . . . . 12

3.2 Simulation Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 133.2.1 Linear Feeding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 143.2.2 Orthogonal Feeding . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 193.2.3 Sensitivity Analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 233.2.4 Loss Mechanisms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 323.2.5 Axial Ratio . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33

CONTENTS II

3.3 Measurement Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 343.4 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38

4 Antenna Element with Substrate Integrated Waveguide Feed 394.1 Proposed Solutions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39

4.1.1 Antenna Topology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 394.1.2 Antenna Dimensions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40

4.2 Simulation Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 424.3 Measurement Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 474.4 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49

5 Final Single Antenna Element 505.1 Proposed Solution . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50

5.1.1 Antenna Topology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 505.1.2 Antenna Dimensions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52

5.2 Simulation Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 535.3 Measurement Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 585.4 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 58

6 Conclusion 596.1 Evaluation of the Final Antenna . . . . . . . . . . . . . . . . . . . . . . . . . . . 596.2 Future Work . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 59

Bibliography I

LIST OF FIGURES i

List of Figures

1.1 Overview of the frequency spectrum. . . . . . . . . . . . . . . . . . . . . . . . . . 21.2 Overview of the K/Ka-band frequency spectrum. . . . . . . . . . . . . . . . . . . 21.3 Smart antenna. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

3.1 Antenna topology with microstrip line feed. . . . . . . . . . . . . . . . . . . . . . 83.2 Example of time domain meshing and staircasing. . . . . . . . . . . . . . . . . . . 133.3 Excited tm11 mode at 18.95GHz for the large cylindrical waveguide. . . . . . . . 143.4 General overview of mode profiles for a circular cavity and their cut-off frequencies

w.r.t the lowest cut-off frequency of the te11 mode [37]. . . . . . . . . . . . . . . 143.5 Active S-parameters for a resonator cavity with diameter wd = 7.8mm and a

linear microstrip line feed. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 153.6 Active S-parameters for a resonator cavity with diameter wd = 14.4mm and a

linear microstrip line feed. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 153.7 Radiation patterns for a resonator cavity with diameter wd = 7.8mm and a linear

microstrip line feed in case of the time domain solver. . . . . . . . . . . . . . . . 163.8 Radiation patterns for a resonator cavity with diameter wd = 7.8mm and a linear

microstrip line feed in case of the frequency domain solver. . . . . . . . . . . . . 173.9 Radiation patterns for a resonator cavity with diameter wd = 14.4mm and a

linear microstrip line feed in case of the time domain solver. . . . . . . . . . . . . 183.10 Radiation patterns for a resonator cavity with diameter wd = 14.4mm and a

linear microstrip line feed in case of the frequency domain solver. . . . . . . . . . 183.11 Active S-parameters for orthogonal microstrip line feeding of the resonator cavity. 213.12 Trade-off between small cavity diameter and optimized S-parameters. . . . . . . 213.13 Radiation patterns for a resonator cavity based on two plated fr-4 substrates

and orthogonal microstrip line feed. . . . . . . . . . . . . . . . . . . . . . . . . . 223.14 Radiation patterns for a resonator cavity based on three plated fr-4 substrates

and orthogonal microstrip line feed. . . . . . . . . . . . . . . . . . . . . . . . . . 22

LIST OF FIGURES ii

3.15 Sensitivity analysis in case of the small resonator cavity. . . . . . . . . . . . . . . 243.16 Sensitivity analysis in case of the large resonator cavity. . . . . . . . . . . . . . . 263.17 Sensitivity analysis in case of the two layered cavity. . . . . . . . . . . . . . . . . 283.18 Sensitivity analysis in case of the three layered cavity. . . . . . . . . . . . . . . . 303.19 Loss mechanisms in case of the two layered resonator cavity with orthogonal feeding. 323.20 Axial ratio for the small two layered resonator cavity with orthogonal feeding in

the case of the time domain solver. . . . . . . . . . . . . . . . . . . . . . . . . . . 333.21 Axial ratio for the small two layered resonator cavity with orthogonal feeding in

the case of the frequency domain solver. . . . . . . . . . . . . . . . . . . . . . . . 333.22 Prototype with a small resonator cavity using microstrip line feeding. . . . . . . 343.23 Measured S-parameters for three prototypes of the small resonator cavity with

linear feeding. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 343.24 Comparison of one of the measured S-parameters with the simulated S-parameters

for the small resonator cavity with linear feeding. . . . . . . . . . . . . . . . . . . 353.25 Measured S-parameters for three prototypes of the large resonator cavity with

linear feeding. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 363.26 Comparison of one of the measured S-parameters with the simulated S-parameters

for the large resonator cavity with linear feeding. . . . . . . . . . . . . . . . . . . 363.27 Measured S-parameters for three prototypes in case of a two layered cavity. . . . 373.28 Measured S-parameters for three prototypes in case of a three layered cavity. . . 37

4.1 Antenna topology with siw feed. . . . . . . . . . . . . . . . . . . . . . . . . . . . 394.1 Antenna topology with siw feed in (b) top view with linear feeding. (cont.) . . . 404.2 Substrate integrated waveguide topology. . . . . . . . . . . . . . . . . . . . . . . 414.3 Active S-parameters for a resonator cavity with diameter wd = 8mm and a linear

siw feeding topology. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 434.4 Active S-parameters for a resonator cavity with diameter wd = 14.3mm and a

linear siw feeding topology. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 434.5 Radiation patterns for a resonator cavity with diameter wd = 8mm and a linear

siw feeding topology in case of the time domain solver. . . . . . . . . . . . . . . 444.6 Radiation patterns for a resonator cavity with diameter wd = 8mm and a linear

siw feeding topology in case of the frequency domain solver. . . . . . . . . . . . . 444.7 Radiation patterns for a resonator cavity with diameter wd = 14.3mm and a

linear siw feeding topology in case of the time domain solver. . . . . . . . . . . . 45

LIST OF FIGURES iii

4.8 Radiation patterns for a resonator cavity with diameter wd = 14.3mm and alinear siw feeding topology in case of the frequency domain solver. . . . . . . . . 46

4.9 Prototype with a small resonator cavity using siw linear feeding. . . . . . . . . . 474.10 Measured S-parameters for three prototypes of the small resonator cavity with a

linear siw feeding topology. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 474.11 Comparison of one of the measured S-parameters with the simulated S-parameters

for the small resonator cavity with a linear siw feeding topology. . . . . . . . . . 484.12 Measured S-parameters for three prototypes of the large resonator cavity with a

linear siw feeding topology. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 484.13 Comparison of one of the measured S-parameters with the simulated S-parameters

for the resonator cavity with a linear siw feeding topology. . . . . . . . . . . . . 49

5.1 Layout of the cavity-backed afsiw antenna topology. . . . . . . . . . . . . . . . . 515.2 Active S-parameters for the cylindrical cavity-backed afsiw topology. . . . . . . 545.3 Radiation patterns for the cylindrical cavity-backed afsiw topology. . . . . . . . 545.4 Sensitivity analysis for the cylindrical cavity-backed afsiw topology. . . . . . . . 555.5 Axial ratio for the small two layered resonator cavity with orthogonal feeding in

the case of the frequency domain solver. . . . . . . . . . . . . . . . . . . . . . . . 575.6 Prototype of the final antenna element. . . . . . . . . . . . . . . . . . . . . . . . 58

LIST OF TABLES iv

List of Tables

3.1 Optimized dimensions for linear microstrip line feeding of the resonator cavity. . 103.2 Roots of the Bessel function Jm for tm modes : Xmn = n-th root of the m-th

Bessel function. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 113.3 Roots of the Bessel function derivative J ′

m for te modes : X ′mn = n-th root of

the m-th Bessel function derivative. . . . . . . . . . . . . . . . . . . . . . . . . . . 113.4 Overview of radiation pattern characteristics for linear microstrip line feeding of

the small resonator cavity. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 173.5 Overview of radiation pattern characteristics for linear microstrip line feeding of

the large resonator cavity. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 193.6 Optimized dimensions for orthogonal microstrip line feeding of the resonator cavity. 203.7 Overview of radiation pattern characteristics for orthogonal microstrip line feeding

of the resonator cavity. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22

4.1 Optimized dimensions for linear siw feeding of the resonator cavity. . . . . . . . 414.2 Overview of radiation pattern characteristics for linear siw feeding of the small

resonator cavity. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 454.3 Overview of radiation pattern characteristics for linear siw feeding of the large

resonator cavity. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46

5.1 Optimized dimensions for final single antenna element. . . . . . . . . . . . . . . . 525.2 Overview of radiation pattern characteristics for the circular cavity-backed afsiw

topology. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54

List of Abbreviations v

List of Abbreviations

afsiw Air-Filled Substrate Integrated Waveguidecp Circular Polarizationcst Computer Simulation Technologydpu Digital Processing Unitfr-4 Flame Retardant 4gcpw Grounded CoPlanar Waveguidegps Global Positioning Systemieee Institute of Electrical and Electronics Engineersitu International Telecommunication Unionlhcp Left-Hand Circular Polarizationlmds Local Multipoint Distribution Systemlna Low Noise Amplifiermws MicroWave Studiopa Power Amplifierpcb Printed Circuit Boardpec Perfect Electric Conductorpth Plated Through Holerf Radio Frequencyrhcp Right-Hand Circular Polarizationrms Root Mean Squaresiw Substrate Integrated Waveguidete Transverse Electrictm Transverse Magnetictrl Thru - Reflect - Line

INTRODUCTION 1

Chapter 1

Introduction

1.1 Context

Today’s society has an ever-growing demand for data availability, going from simple searchengines to full resolution video streaming. The mobility demands for the user terminals haveincreased over the years. Therefore, it is interesting to start investing in connectivity for movingvehicles, such as aeroplanes, trains, buses, ferries, etc., while still maintaining a high data rate.Since offering internet is not a fundamental functionality of a moving vehicle, the implementedstructures should be affordable, reliable and small.

First of all, it should be possible to mount these devices onto the vehicles, while requirementssuch as lightweight, small footprint and low profile are satisfied. In addition, high data rates canbe achieved by shifting towards higher frequencies. Moreover, sufficient gain should be achievedto overcome the increasing path loss that goes along with higher frequency bands of operation.

The use of satellite communication to address the matter is a natural consequence of the user’sdemands for high data rates with low latency. However, only a limited amount of frequencybands are available for satellite communication, as can be seen from Figure 1.1. The mainfrequency bands that can be used for satellite communication are the C-, X-, Ku- and Ka-band.

1.2 Frequency Spectrum

A lot of trade-offs have to be taken into account when choosing the operating frequency. Forexample, going to higher frequencies will allow smaller end-user terminals, but the design willalso be subjected to fabrication limits. Therefore, it is worth looking into the different aspectsbefore deciding on the operating frequency band.

INTRODUCTION 2

Figure 1.1: Overview of the frequency spectrum [18].

Operating at higher frequencies has multiple advantages. Smaller end user terminals in combinationwith increased mobility, higher bandwidths and speeds are all key words in today’s society.

However, these benefits have their counterparts. High frequency signals require expensive radiofrequency (rf) equipment to be measured and analysed. This also leads to increased complexitywhen processing these signals. A higher path loss is inevitable and, therefore, more gain isrequired for a proper operation. Sufficient gain is obtained by using larger antenna arrays,implying larger feeding networks and thus creating more losses. Furthermore, frequencies higherthan 10GHz are also more susceptible to rain fade, for which solutions are proposed in theliterature [29], [49], [51].

CivilReserved(Civil) Military

17.7 19.7 20.2 21.2

downlink

CivilReserved(Civil) Military

27.5 29.5 30.0 31.0

uplink

f[GHz]

Figure 1.2: Overview of the K/Ka-band frequency spectrum.

INTRODUCTION 3

Nonetheless, to enable better servicing of mobile users in the future, the operating frequencyband is chosen to be the K/Ka-band, since the lower parts of the spectrum become too crowded.The International Telecommunication Union (itu) predetermines which parts of the frequencyspectrum for satellite communications are used for civil, military or other purposes, as can beseen from Figure 1.2. This master’s dissertation will focus on the civil K/Ka-band downlink,which stretches from 17.7GHz to 20.2GHz with a centre frequency of 18.95GHz, a bandwidthof 2.5GHz and a fractional bandwidth of 13.19%.

1.3 Goals and Outline

The goal of this master’s dissertation is to develop a K/Ka-band downlink antenna element, thatcan be used as a basis for the next generation avionic satellite communications. The concept issketched in Figure 1.3.

antenna

polarization converter polarization converter polarization converter

pol. 1 pol. 1 pol. 1pol. 2 pol. 2 pol. 2

tracking unit tracking unit tracking unit

beamforming network

down converter unit

controller

dpu

Controller:1- Beam pointing2- Polarization adjustment rhcp lhcp

Figure 1.3: Smart antenna.

Satellites communicate with other terminals by using different polarizations. Consequently,dual circular polarization is preferable, since this allows sending and/or receiving twice as much

INTRODUCTION 4

data at the same time. A convenient characteristic is the orthogonality of left-hand circularpolarization (lhcp) and right-hand circular polarization (rhcp). Therefore, the proposed designshould live up to implementing circular polarization. The polarization converter will take careof injecting the correct polarization into the single antenna element.

A tracking unit will be responsible to apply the correct amplitudes and phase shifts to eachindividual element by means of low noise amplifiers (lnas) and/or power amplifiers (pas)and/or phase shifters, such that a multi beam radiation pattern can be generated for beamsteering. These units are all regulated by the controller, which is responsible for choosing thebest possible satellite for a good connection. The controller can use information from a globalpositioning system (gps), an altimeter and the speed of the vehicle to postulate the optimalsatellite connection. The received data is down converted and fed to the digital processing unit(dpu) for further processing. Everything described in this paragraph falls out of the scope ofthis thesis, but was discussed here for completeness.

Other than providing circular polarization, the single antenna element should be cost-effectiveand compatible with standard printed circuit board (pcb) processing, such that it can be massproduced. Developing a design that is lightweight, has a small footprint and a low profile is ofutmost importance for invisible and unobtrusive integration into a moving vehicle. Providing agood antenna platform isolation for close proximity integration of active components is a bonus.There are no itu regulations for the K/Ka-band downlink [12], thus some guidelines will beextracted from the literature review.

A concise literature review is presented in Chapter 2, in order to give more background onthe matter and to propose some guidelines regarding the antenna design. Chapter 3 continueswith the design of an antenna element with microstrip line feeding, while Chapter 4 discussesa substrate integrated waveguide (siw) feeding topology. Finally, Chapter 5 handles the designof the final antenna element, whereas Chapter 6 gives a brief conclusion.

LITERATURE REVIEW 5

Chapter 2

Literature Review

2.1 Concept

Antenna elements operating at Ka-band frequencies can be used for multiple purposes whenintegrated in an array, such as for local multipoint distribution systems (lmdss) or interactivehigh-data-rate satellite communications [23]. Fundamental in satellite communication is therealization of a small footprint, low profile, lightweight, cost-effective and robust antenna system[28]. The integration of user-terminals in high-mobility applications, such as aeroplanes, movingvehicles, etc., is more likely if the system meets the aforementioned requirements.

A lot of antenna elements or arrays found in literature are bulky, heavy and thus only suitableas base station equipment [25], [42], [43], [40], [39]. Moreover, reflect antennas or lens antennasare compelled to use mechanical beam steering, which becomes obsolete [45]. Therefore, it isinteresting to look into microstrip patch antennas or other low profile, compact solutions [24].

On the other hand, electrical beam steering allows for fast tracking of and communication withmultiple satellites at the same time. These antennas are called (adaptive) beam steering smartmulti-antenna systems [33].

2.2 State-of-the-Art

A Ka-band dual frequency shorted annular patch, fed by a microstrip line coupled through aslot, was proposed in [3]. The dimensions of the prototype with casing are larger than half awavelength λ/2, 40mm x 40mm x 11.5mm, it lacks robustness, the design is rather expensive tofabricate and the reserved civil frequency band is not completely covered. However, a maximumgain of 7.1 dBi is attained and the paper suggests an interesting array configuration.

LITERATURE REVIEW 6

A low profile, small footprint patch antenna for phased arrays was proposed in [31], targeting thefrequency spectrum around 30GHz. The maximum gain for a one-by-eight array was 8.6 dBi. Arather mediocre maximum gain and a limited scanning range of 60° was obtained, while requiringa bulky feeding structure.

A compact circular polarized cavity-backed stacked-patch antenna with four-port capacitivefeeding is illustrated in [41], covering the reserved civil and military K/Ka-band downlink.Subsequently, a compact low profile and circularly polarized K/Ka-band antenna with a verylarge bandwidth, covering approximately from 17.6GHz to 30.4GHz, is presented in [26]. Adual-band five-layer circularly polarized antenna element, that covers the reserved civil andthe military frequencies and exhibits a gain of about 8 dBi, is discussed in [27]. All of thesedesigns can be used as measures for the performance of the proposed solution of this master’sdissertation. However, their fabrication complexity increases, due to their printed circuit board(pcb) multi-layer structure.

In conclusion, it is possible to state that most of the proposed solutions make trade-offs. Often,the entire K/Ka-band downlink and/or uplink are not covered, or they suffer from high losses,large footprints, large profiles or high costs.

2.3 Proposed Solution

The initial developments of this thesis are based on a select number of papers. The first ideais based on a cost-effective compact planar feed structure for Ka-band antennas [7]. Somemore inspiration can be harvested from other papers [30], [46]. Consequently a more elaboratestudy was done on the feeding techniques. Differential feeding surely has advantages [1], butso has an air-filled substrate integrated waveguide (afsiw) feeding topology, where losses areminimized [8].

Basic requirements can be formulated as a guideline based on the literature review. A heightbelow 5mm and a footprint smaller than 15mm x 15mm seem realistic. Moreover, a gain ofmore than 4 dBi and a 3 dB beamwidth of more than 60° is desirable. The choice of fabricationmaterials should be based on creating a reliable and cost-effective antenna system.

ANTENNA ELEMENT WITH MICROSTRIP LINE FEED 7

Chapter 3

Antenna Element with MicrostripLine Feed

3.1 Proposed Solutions

This section elaborates on the design procedure in computer simulation technology microwavestudio (cst mws®) for four antenna elements with microstrip line feeding [6].

3.1.1 Antenna Topology

The proposed antenna topology consists of a resonator cavity, fed by microstrip lines, as shownin Figure 3.1. This design exhibits minimal backwards radiation, and as such, any componentson the back of the antenna will be sufficiently isolated. Furthermore, a high suppression of thesurface waves is attainable and the low profile allows for easy integration into surfaces of movingvehicles. Finally, a planar antenna array based on this single element can easily be constructed.

The resonator cavity is a half open cylindrical waveguide. This circular design enables easyexcitation of circular polarization modes, which is advised for proper K/Ka-band operation. Inaddition, the mode distribution is symmetric in the azimuthal plane, which is the plane shownin Figures 3.1b, 3.1c, 3.1d and 3.1e. A cylindrical cavity is more favourable for this application,even if the dimensions are larger than a rectangular cavity which excites the same propagatingmode.

The feeding plane consists of two microstrip lines at the opposite sides of the resonator cavity,as shown in Figure 3.1b. The lines are excited with a 180° phase shift, which results in ahigher polarization purity, since the delay from one side of the cavity to the other is minimized.Circular polarization is achieved, by adding an additional pair of microstrip lines orthogonal to


Feedplane

PCBstack

Through hole Through hole

wd

wh

hd

FR-4t

ROt

cutx

z

(a)

ROw

ROl

hd

mlinl mlinwwd

margin

x

y

(b)

ROw

ROl

hd

mlinl mlinwwd

margin

x

y

(c)

FR-4w

FR-4l

hd

wdplatingt

x

y

(d)

FR-4t

FR-4t

hd

x

y

(e)

Figure 3.1: Antenna topology with microstrip line feed. (a) Side view of the complete stack.Top view of (b) the linear, (c) the orthogonal feeding layer with RO4350B™ substrate, (d) thefirst, second and third drilled and plated standard pcb with fr-4 substrate and (e) the bottomstandard pcb with fr-4 substrate.


the original feedlines, as demonstrated in Figure 3.1c. Moreover, all the feedlines are excitedwith a 90° phase shift. Consequently, circular polarization is achieved by a sequential feedingtechnique and could be excited in both directions, depending on the feeding network : lhcp

and/or rhcp.

3.1.2 Fabrication Technology

The cavity-backed microstrip line fed antenna is partially fabricated by using standard pcb

manufacturing technologies, targeting the 4 lowest layers in Figure 3.1a. The production costsare greatly reduced by using this stacked standard pcb technology with flame retardant 4 (fr-4)as substrate. Remark that the relative permittivity εr of this material is not very reliable.However this is not an issue, since it is only used to realise the resonator cavity by milling outa cylinder and using edge plating to create the side walls, Figure 3.1d.

The feeding layer of the antenna topology requires a reliable relative permittivity εr. Therefore,the stacked standard pcb layers will have a pcb layer with RO4350B™ [13] substrate on top.This is represented by the top layer in Figure 3.1a. Besides having a reliable εr, the RO4350B™laminate is suitable for high frequency applications, featuring excellent electrical performanceand low losses.

The substrate losses inside the cavity can be greatly reduced by milling out some parts of theRO4350B™ substrate, as shown in Figures 3.1b and 3.1c. These milled out parts will not beedge plated, to avoid shorting the microstrip line feed with the ground plane.

The width of the microstrip line, mlinw, is essential for good impedance matching and thusoptimal power transfer [14]. Therefore, spacing is kept between the microstrip line edge and thesubstrate edge to deal with fabrication tolerances. The frequency region of operation shifts withthe effective permittivity of the entire resonator cavity. Consequently, it is possible to state thatthe more substrate is present, the higher the effective permittivity gets and thus the lower theresonance frequency will be. This fabrication disadvantage can be turned into a limited tuningfactor for the frequency range, together with the length of the microstrip line into the cavity.

The milling of the substrate is only possible with a limited number of drill diameters. Thereforetuning the milling radius milr has a limited value for optimizing the design. Another restriction isgiven by the height of the cavity, which should be an integer amount of standard pcb thicknesses,to profit from the cheaper manufacturing technology. Finally, some through holes are created,with diameter hd, to be able to hold the entire structure together with screws [21], [22].


3.1.3 Antenna Dimensions

In order to get an acceptable initial sizing of the antenna, some important features need to beconsidered. These characteristics are explained in this section.

Parameter Small design [mm] Large design [mm]cut 0.035 0.035FR-4t 1.55 1.55FR-4w 15.3 20.0FR-4l 15.3 20.0ROt 0.25 0.25ROw 15.3 20.0ROl 15.3 20.0wd 7.8 14.4wh 5.145 5.145hd 1.0 1.0mlinw 0.475 0.475mlinl 3.75 6.05margin 0.25 0.2milr 0.5 0.5

Table 3.1: Optimized dimensions for linear microstrip line feeding of the resonator cavity.

Cut-Off Frequency. The chosen frequency band of operation is crucial for the initial sizing.Most of the sizes depend on the excited modes of the circular cavity. Generally, these modescan be selected such that your antenna topology works optimally by positioning slots where thefields of the desired excited mode are the strongest [6].

However, a circular cavity is used here, where minimization is key. The excited modes arebased on roots of Bessel functions or their n-th order derivatives. The cut-off frequencies ofthe transverse electric (te) and transverse magnetic (tm) modes for a circular waveguide areexpressed by Equation (3.1) [6], [36].

fc,mnl =c

2π√µrεr

√(smn

a

)2+

(lπ

h

)2

(3.1)

Here a = wd/2, defined in Figure 3.1, represents the radius of the cylindrical waveguide. Theheight of the resonator h = wh, also defined in Figure 3.1, plays a role in the cut-off frequency forhigher order modes in the altitudinal direction. Concrete values for smn = χmn or smn = χ′

mn

can be found in Table 3.2 or Table 3.3 for tm or te modes respectively. Use the light speedas propagation speed v = c = 2.998 · 108 m/s, since the waveguide is quasi-hollow: µr ≈ 1 andεr ≈ 1. The influence of the RO4350B™ substrate is neglected here, since an actual calculation


of the effective relative permittivity is cumbersome.

m \n 1 2 30 2.405 5.520 8.6541 3.832 7.016 10.1742 5.136 8.417 11.6203 6.380 9.761 13.015

Table 3.2: Roots of the Bessel function Jm for tm modes : Xmn = n-th root of the m-th Besselfunction.

m \n 1 2 30 3.832 7.016 10.1731 1.841 5.331 8.5362 3.054 6.706 9.9703 4.201 8.015 11.346

Table 3.3: Roots of the Bessel function derivative J ′m for te modes : X ′

mn = n-th root of them-th Bessel function derivative.

The tm01 mode has the lowest tm cut-off frequency, Table 3.2. However, when looking at Table3.3, the most favourable excitation mode for this purpose will be the te11 mode, since it hasthe lowest cut-off frequency and can thus be realized with minimal dimensions. A safety marginon the lower frequency bound of 17.7GHz is introduced by setting the cut-off frequency fc to17.5GHz. Consequently, the corresponding resonator diameter is approximately wd ≈ 10mm,which will be the initial sizing of the cavity before optimization.

One should keep in mind that any introduction of substrate will result in the excitation ofmodes with different cut-off frequencies, since the effective relative permittivity εr,eff of thetotal structure changes. The cut-off frequency will decrease, since the effective permittivity forRO4350B™ substrates is substantially higher than for air. The propagation speed will in thatcase be different from the speed of light and will transform into the more general equation [50] :

v =c

√µr,effεr,eff

(3.2)

In case of a larger margin between the microstrip line edge and the substrate edge inside thecavity, as shown in Figures 3.1b and 3.1c, the effective relative permittivity εr,eff of the totalcavity rises. Consequently, the speed of the electromagnetic waves will decrease, shown byEquation (3.2), and the cut-off frequency will decrease as well, shown by Equation (3.1). Inconclusion, a smaller resonator cavity diameter than wd ≈ 10mm can be obtained by increasingthe margin, this is in line with the dimensions in Table 3.1.


Microstrip line feeding the cavity as shown in Figure 3.1b favours the tm11 mode. Consequently,the resonator cavity diameters for both designs, wd = 7.8mm and wd = 14.4mm, are based onthe tm11 excited mode, as shown later on. A theoretical waveguide diameter of wd = 21mm isobtained, when calculating the cut-off frequency of the tm11 mode by using the light speed andneglecting the RO4350B™ substrate, as previously done for the te11 mode.

Constructive Interference. The height of the resonator cavity must be a multiple of thethickness of a standard pcb. Nonetheless the height should also be approximately wh ≈ nλ/4,with n an odd integer value, since constructive interference occurs after a path length of kλ/2,with k an integer value. The footprint of this design should be as small as possible, so settingn = 1 is the best choice.

The calculation of the wavelength λ requires the effective relative permittivity εr,eff and preferablythe centre frequency of the desired K/Ka-band downlink fcentre = 18.95GHz [50]. However, asdiscussed previously, determining εr,eff is not straightforward. Therefore εr,eff is approximatedby εr = 1 of the quasi-hollow resonator cavity.

λ =c

fcentre · εr,eff(3.3)

By using three stacked pcbs the calculated value of wh ≈ 4mm is approached as close as possible,if one would neglect the RO4350B™ substrate. This is confirmed by the value shown in Table3.1.

Manufacturing Limits. The manufacturer imposes certain amount of fabrication limits,which are carefully documented in their design guidelines [20]. Straight corners in cut-outsin a pcb can not be manufactured, the milling tool has a certain radius, henceforth called milr.

3.1.4 CST Solver Remarks : Meshing

cst divides the structure into smaller elements, called meshing, on which the boundary conditionshave to apply when simulating the fields, to solve Maxwell’s equations. This meshing causesslight degeneracy of the geometry. The propagation speed deviates in each cell, because a verysmall part of the wave is reflected at each border. In other words, the boundary conditions causedispersion on a small scale.

Time domain solver uses a grid. This means that it divides a 2D space by means of non-equallyspaced horizontal and vertical lines into smaller rectangles as shown in Figure 3.2. It can be


easily seen that this kind of meshing neglects part of the geometry of the structure that is notorientated along the axes of this grid, e.g. slant parts and circle-like shapes. Consequently, theseshapes are modelled less accurate.

These fields will be especially susceptible to accumulations of small errors due to staircasing.Staircasing is demonstrated by the approximated (meshed) path marked by the dashed blue linein Figure 3.2, compared to the actual path shown by the solid black line. The deviations areclearly visible. Even when the mesh is refined, these errors will remain, nonetheless they can bemade extremely small.

Figure 3.2: Example of time domain meshing and staircasing.

These errors are accumulated over the entire path due to the anisotropy of the time domainmeshing, most pronounced in frequencies further away from the centre frequency of the simulation.In reality, a fully symmetric resonator cavity is used and should consequently result in equalscattering parameters, more commonly known as S-parameters. However, the S-parameters willnot be equal due to the staircasing phenomenon.

Frequency domain solver uses tetrahedrons. This enhances the degree of isotropy on a largescale, since the fields are scattered in all possible directions and some of the previously mentionederrors annihilate each other. This clearly is the preferred solver, although simulations take moretime and the calculation of the active S-parameters is more complicated.

3.2 Simulation Results

The cst simulation results, concerning the S-parameters and the realised gain radiation patternsfor two cross-sections of ϕ = 0° and ϕ = 90° in case of a microstrip line feeding topology, arediscussed in this section.


3.2.1 Linear Feeding

First of all, the cavity is simulated with two microstrip lines, with a 180° phase difference, as canbe seen from Figure 3.1b. The excited modes determined in Section 3.1.3 are confirmed by thesimulations of the electric field, shown in Figure 3.3. These field distributions can be identifiedby comparing them to Figure 3.4 [37]. The electric field lines are represented by solid arrows,while the magnetic field lines are represented by dashed arrows.

(a) (b) (c)

Figure 3.3: Excited tm11 mode at 18.95GHz for the large cylindrical waveguide (a) as magnitudeof the electric field (b) as vector plot in an altitudinal cut (c) as vector plot in an azimuthal cutalong the microstrip lines.

Figure 3.4: General overview of mode profiles for a circular cavity and their cut-off frequenciesw.r.t the lowest cut-off frequency of the te11 mode [37].

It is mandatory to look at the simultaneous excitation of both microstrip lines, since the interestgoes out to circular polarization. Therefore, the active S-parameters will be considered in thesimulations. This simulation will require the centre frequency fcentre = 18.95GHz in case of the


time domain solver.

S11,a = [S11(0°) + S12(180°)](18.95GHz) (3.4)

S22,a = [S21(0°) + S22(180°)](18.95GHz) (3.5)

The reflections at port 1 will be considered when port 1 and port 2 are excited simultaneously,but port 2 will possess a 180° phase shift. Combining the passive S-parameters with the correctphase shift, where only one port at the time is excited, results in the same outcome.

16 18 20 22−70

−60

−50

−40

−30

−20

−10

017.67 19.97

2.3GHz

Frequency [GHz]

|Sa|[dB]

|S11,a||S22,a|

(a) Time domain simulation.

16 18 20 22−70

−60

−50

−40

−30

−20

−10

017.5 19.83

2.33GHz

Frequency [GHz]

|Sa|[dB]

|S11,a||S22,a|

(b) Frequency domain simulation.

Figure 3.5: Active S-parameters for a resonator cavity with diameter wd = 7.8mm and a linearmicrostrip line feed.

16 18 20 22−40

−30

−20

−10

016.05 20.94

4.89GHz

Frequency [GHz]

|Sa|[dB]

|S11,a||S22,a|


16 18 20 22−40

−30

−20

−10

015.91 21.77

5.86GHz

Frequency [GHz]

|Sa|[dB]

|S11,a||S22,a|


Figure 3.6: Active S-parameters for a resonator cavity with diameter wd = 14.4mm and a linearmicrostrip line feed.


There are some differences between the time and frequency domain solver, as can be seen fromFigure 3.5 and from Figure 3.6, especially for frequencies further away from the centre frequencyfcentre = 18.95GHz. This is where the time domain modelling is less accurate. The solid blue boxin these figures represents the desired frequency band of operation of the K/Ka-band downlink.The bandwidth where the active S-parameters Sa < −10 dB is annotated by a black arrow.

The optimized design for a small resonator cavity diameter of wd = 7.8mm does not cover theentire downlink band from 17.7GHz to 20.2GHz, where the larger design with wd = 14.4mm

does. However, one of the key features is to miniaturize the antenna structure as much aspossible, so the design with the smaller diameter is more desirable. Since there are a couple ofoptions to enlarge the bandwidth in the future, both designs are fabricated.

The radiation patterns for the small resonator cavity diameter are represented in Figure 3.7and Figure 3.8 for time and frequency domain simulator respectively. An overview of the mostimportant specifications is given in Table 3.4. The side lobe level is defined w.r.t. the main lobelevel.

Considering the smaller design, both the time and frequency domain solver deliver arguably thesame results. The main lobe magnitude is always larger than 6 dBi. Moreover, side lobe levelsstay well below the main lobe magnitude at all times. Finally, the angular beamwidth is largerthan 68°.

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(a) At 17.7GHz for ϕ = 0°.

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[dBi]

(f) At 20.2GHz for ϕ = 90°.

Figure 3.7: Radiation patterns for a resonator cavity with diameter wd = 7.8mm and a linearmicrostrip line feed in case of the time domain solver.


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(a) At 17.7GHz for ϕ = 0°.

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[dBi]

(f) At 20.2GHz for ϕ = 90°.

Figure 3.8: Radiation patterns for a resonator cavity with diameter wd = 7.8mm and a linearmicrostrip line feed in case of the frequency domain solver.

Realised gain Time domain Frequency domainfmin = 17.7GHz ϕ = 0° ϕ = 90° ϕ = 0° ϕ = 90°Main Lobe Magnitude [dBi] 6.4 6.87Main Lobe Direction [°] 1 0 0 0Angular Width (3 dB) [°] 88.1 88.9 80 77.5Side Lobe Level [dB] -13.8 -20.2 -11.6 -17.4fcentre = 18.95GHz ϕ = 0° ϕ = 90° ϕ = 0° ϕ = 90°Main Lobe Magnitude [dBi] 6.95 6.78Main Lobe Direction [°] 0 0 0 0Angular Width (3 dB) [°] 81.5 85.7 102.7 78Side Lobe Level [dB] -10 -15.7 -8.0 -13.2fmax = 20.2GHz ϕ = 0° ϕ = 90° ϕ = 0° ϕ = 90°Main Lobe Magnitude [dBi] 6.82 6.07Main Lobe Direction [°] 0 0 0 0Angular Width (3 dB) [°] 75.5 76.2 123.9 68.4Side Lobe Level [dB] -9 -12.7 -6.2 -10.2

Table 3.4: Overview of radiation pattern characteristics for linear microstrip line feeding of thesmall resonator cavity.

The radiation patterns for the large resonator cavity diameter are plotted in Figure 3.9 andFigure 3.10 for the time and frequency domain solver respectively. An overview of the mostimportant specifications is given in Table 3.5.

Considering the smaller design, both the time and frequency domain solver deliver arguably thesame results. The main lobe magnitude is always larger than 8.3 dBi. Moreover, side lobe levelsstay well below the main lobe magnitude at all times. Finally, the angular beamwidth is larger


than 52°.

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(f) At 20.2GHz for ϕ = 90°.

Figure 3.9: Radiation patterns for a resonator cavity with diameter wd = 14.4mm and a linearmicrostrip line feed in case of the time domain solver.

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(a) At 17.7GHz for ϕ = 0°.

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(f) At 20.2GHz for ϕ = 90°.

Figure 3.10: Radiation patterns for a resonator cavity with diameter wd = 14.4mm and a linearmicrostrip line feed in case of the frequency domain solver.

In conclusion, an increase in gain, or main lobe magnitude, is related to a decrease in 3 dB

beam width. The scanning angle is reduced, since the antenna becomes more directive. This isconfirmed by comparing Tables 3.4 and 3.5.

The aforementioned trade-off can be theoretically expressed by Equation (3.6). In other words,the 3 dB-beamwidth (Φ1r, Φ2r) will decrease with increasing gain (G0,max) due to elevated


directivity (D0) [5].D0 =

4π

ΩA≈ 4π

Φ1rΦ2r

G0,max = e0D0

(3.6)

Realised gain Time domain Frequency domainfmin = 17.7GHz ϕ = 0° ϕ = 90° ϕ = 0° ϕ = 90°Main Lobe Magnitude [dBi] 8.8 8.78Main Lobe Direction [°] 4 0 0 0Angular Width (3 dB) [°] 60.1 65.7 61.7 59Side Lobe Level [dB] -14.2 -16.1 -12.9 -16.4fcentre = 18.95GHz ϕ = 0° ϕ = 90° ϕ = 0° ϕ = 90°Main Lobe Magnitude [dBi] 8.41 8.36Main Lobe Direction [°] 0 0 0 0Angular Width (3 dB) [°] 73.2 68.6 80.7 62.2Side Lobe Level [dB] -19.4 -21.9 -14.9 -22.9fmax = 20.2GHz ϕ = 0° ϕ = 90° ϕ = 0° ϕ = 90°Main Lobe Magnitude [dBi] 8.85 9.02Main Lobe Direction [°] -3 0 0 0Angular Width (3 dB) [°] 59.1 56.1 63.6 52.8Side Lobe Level [dB] -17.7 -21.3 -14.5 -20.1

Table 3.5: Overview of radiation pattern characteristics for linear microstrip line feeding of thelarge resonator cavity.

3.2.2 Orthogonal Feeding

The circular polarization is achieved by feeding four microstrip lines out of phase. The layoutcan be observed in Figure 3.1c. The excited mode is still the same as for the small linearly fedresonator cavity : tm11. The visualisation of the electric field is the same as in Figure 3.3, onlynow these fields rotate around the altitudinal axis.

The active S-parameters are simulated, since orthogonal polarization is desired. These activeS-parameters can be expressed by the same principle as before, like in the case of linearfeeding. Formulas below are valid when the numbering of the ports happens clockwise or counterclockwise.


S11,a = [S11(0°) + S12(90°) + S13(180°) + S14(270°)](18.95GHz) (3.7)

S22,a = [S21(0°) + S22(90°) + S23(180°) + S24(270°)](18.95GHz) (3.8)

S33,a = [S31(0°) + S32(90°) + S33(180°) + S34(270°)](18.95GHz) (3.9)

S44,a = [S41(0°) + S42(90°) + S43(180°) + S44(270°)](18.95GHz) (3.10)

Only the small diameter is of importance, as discussed in the previous section. This will alsobe confirmed by the measurement results further on. This paragraph will restrict itself to thefrequency domain solver, since the differences between time and frequency domain solver havebeen demonstrated earlier and since the frequency domain solver proved to be the more reliableoption.

A reduction in height is possible by optimizing the single antenna element with orthogonalfeeding, based on the small cavity single antenna element that had a linear feeding networkbefore. As a consequence, both resonator cavity heights of 2 layers and 3 layers fr-4 areinvestigated. The final dimensions of both designs can be found in Table 3.6. Comparing thetwo shows the trade-off between a small footprint and a low profile antenna.

Parameter 2 layers fr-4 [mm] 3 layers fr-4 [mm]cut 0.035 0.035FR-4t 1.55 1.55FR-4w 11.6 11.4FR-4l 11.6 11.4ROt 0.25 0.25ROw 11.6 11.4ROl 11.6 11.4wd 7.6 7.4wh 3.525 5.145hd 1.0 1.0mlinw 0.475 0.475mlinl 4.4 4.35margin 0.4 0.25milr 0.5 0.5

Table 3.6: Optimized dimensions for orthogonal microstrip line feeding of the resonator cavity.

The S-parameters of both designs are plotted in Figure 3.11, where the blue solid box representsthe desired frequency band from 17.7GHz to 20.2GHz. Since the design has a certain symmetry,the active S-parameters for each port are supposed to be the same, this can be easily verifiedby looking at Figure 3.11. These plots also have approximately the same bandwidth, but the


resonance for the 3 layered cavity (Figure 3.11b) is significantly better. However, when lookingat production cost, the design with one standard pcb layer less is preferable, even if that cavitydiameter is slightly larger.

16 18 20 22−60

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017.54 20.72

3.18GHz

Frequency [GHz]

|Sa|[dB]

|S11,a||S22,a||S33,a||S44,a|

(a) Resonator cavity height wh = 3.525mm byusing two plated layers fr-4 substrate.

16 18 20 22−60

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017.63 20.71

3.08GHz

Frequency [GHz]|S

a|[dB]

|S11,a||S22,a||S33,a||S44,a|

(b) Resonator cavity height wh = 5.145mm byusing three plated layers fr-4 substrate.

Figure 3.11: Active S-parameters for orthogonal microstrip line feeding of the resonator cavity.

16 18 20 22 24−70

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017.54 20.72

Frequency [GHz]

|Sa|[dB]

wd = 7.5mmwd = 7.6mmwd = 7.7mmwd = 7.8mmwd = 7.9mmwd = 8mm

Figure 3.12: Trade-off between small cavity diameter and optimized S-parameters.

Remember that the minimization of this design is an important key feature. Therefore theoptimized parameters for the resonator cavity consisting of 2 layers fr-4 substrate of Table 3.6are given for acceptable S-parameters in case of a waveguide diameter of wd = 7.6mm. TheseS-parameters could be optimized even more by allowing larger cylindrical cavity diameters, asshown in Figure 3.12.


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(c) At 20.2GHz for ϕ = 0°.

Figure 3.13: Radiation patterns for a resonator cavity based on two plated fr-4 substrates andorthogonal microstrip line feed.

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(c) At 20.2GHz for ϕ = 0°.

Figure 3.14: Radiation patterns for a resonator cavity based on three plated fr-4 substratesand orthogonal microstrip line feed.

The antenna’s feeding network of four microstrip lines results in approximately the same radiationpatterns for ϕ = 0° and ϕ = 90°. Therefore only the realized gain patterns for ϕ = 0° are plottedin Figures 3.13 and 3.14. An overview of the characteristics of the farfield realised gain patterncan be found in Table 3.7.

2 layers fr-4 3 layers fr-4fmin = 17.7GHz

Main Lobe Magnitude [dBi] 5.79 5.95Main Lobe Direction [°] 0 0Angular Width (3 dB) [°] 85.6 82.2Side Lobe Level [dB] -10.8 -9.9fcentre = 18.95GHZ

Main Lobe Magnitude [dBi] 5.98 6.33Main Lobe Direction [°] 0 0Angular Width (3 dB) [°] 93.4 87.7Side Lobe Level [dB] -12.4 -13.9fmax = 20.2GHz

Main Lobe Magnitude [dBi] 7.06 6.88Main Lobe Direction [°] 0 0Angular Width (3 dB)[°] 75.7 78.6Side Lobe Level [dB] -13.7 -14.4

Table 3.7: Overview of radiation pattern characteristics for orthogonal microstrip line feedingof the resonator cavity.


3.2.3 Sensitivity Analysis

It is important to check the design for fabrication tolerances, before sending the simulatedprototypes to the manufacturer. If the design is too sensitive, the purpose of having a reliableminimal cost antenna is completely annihilated. Therefore, the sensitivity to fabrication tolerancesis checked in this section. Remark that the simulations are done in the frequency domain, sincethis is the preferred solver for this type of circular structures.

The substrate thickness sensitivity is simulated with relative variations of 12.5% or more. Thisboils down to a variation of 50µm for the RO4350B™ substrate, since ROt = 250µm, and200µm for the fr-4 substrate, since fr-4t = 1.55mm. The value of the relative permittivity εr

is rarely exactly determined for these high frequencies. Therefore, deviations of more than 1.5%are simulated, which results in a variation of 0.06 with respect to the original εr,RO4350B = 3.66

in case of the RO4350B™ substrate. Variations on εr for the fr-4 substrate are not relevanthere, since the standard pcbs are merely used as buffer layers to construct the resonator cavityand thus do not influences the fields inside the cavity.

3.2.3.1 Linear Feeding

This section elaborates on the sensitivity analysis regarding the resonator cavity with a linearfeeding topology for both small and large cavity diameter.

Resonator Cavity with Small Diameter. The small cylindrical waveguide has a robustdesign, as can be seen from Figure 3.15. Furthermore, the sensitivity analysis shows that thecopper thickness cut, the relative permittivity for the RO4350B™ substrate εr,RO4350B and themilling radius of the manufacturer milr are the most robust parameters.

The analysis of the fr-4 substrate thickness fr-4t, shown in Figure 3.15c, shows that a smallerthickness results in a small shift in the active S-parameters towards higher frequencies. Incontrast, a thicker substrate causes a small shift towards lower frequencies. The cavity height isof great importance, as previously explained, it determines whether constructive or destructiveinterference occurs. Hence, a larger cavity height allows for constructive interference to occurat larger wavelengths λ, favouring smaller frequencies. Analogous results are obtained for thecase of a smaller cavity height.

Variations regarding the microstrip lines are the most sensitive. The resonance shifts towardshigher frequencies when the microstrip line is made smaller, while a shift towards lower frequenciesis observed for a larger microstrip line. This assumption is valid for the length of the microstrip


line mlinl (Figure 3.15e) as well as for the width mlinw (Figure 3.15f) and the inset inside thecavity mlininset (Figure 3.15g).

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Frequency [GHz]

|Sa|[dB]

cut− 5µmcut

cut+ 5µm

(a)

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0

Frequency [GHz]

|Sa|[dB]

εr − 0.06εr

εr + 0.06

(b)

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0

Frequency [GHz]

|Sa|[dB]

fr-4t− 200µmfr-4t

fr-4t+ 200µm

(c)

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0

Frequency [GHz]

|Sa|[dB]

milr− 50µmmilr

milr+ 50µm

(d)

Figure 3.15: Sensitivity analysis in case of the small resonator cavity for (a) cut, (b) εr,RO4350B,(c) fr-4t and (d) milr.

It is important to distinguish variations of the length of the microstrip line mlinl from thevariations in inset length into the cavity mlininset. Variations in mlinl are realised by decreasingor increasing the substrate width and length as well. In other words, the microstrip line lengthinside the cavity stays the same. In contrast, the inset length inside the cavity mlininset isadjusted, without altering the microstrip line length outside the cavity, such that the substratedimensions stay the same.

In order to be able to perform useful measurements and minimize reflections, the antenna elementis matched to the 50Ω of the network analyser. A smaller line width mlinw, as well as a larger


RO4350B™ substrate thickness, results in a higher line impedance, while a larger mlinw and asmaller ROt cause a smaller line impedance than 50Ω.

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Frequency [GHz]

|Sa|[dB]

mlinl− 50µmmlinl

mlinl+ 50µm

(e)

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Frequency [GHz]

|Sa|[dB]

mlinw− 50µmmlinw

mlinw+ 50µm

(f)

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0

Frequency [GHz]

|Sa|[dB]

mlininset− 50µmmlininset

mlininset+ 50µm

(g)

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Frequency [GHz]

|Sa|[dB]

margin− 50µmmargin

margin+ 50µm

(h)

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0

Frequency [GHz]

|Sa|[dB]

ROt− 50µm

ROtROt+ 50µm

(i)

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0

Frequency [GHz]

|Sa|[dB]

wd− 100µmwd

wd+ 100µm

(j)

Figure 3.15: Sensitivity analysis in case of the small resonator cavity for (e) mlinl, (f) mlinw,(g) mlininset, (h) margin, (i) ROt and (j) wd. (cont.)


Resonator Cavity with Large Diameter. The large cylindrical waveguide also has a robustdesign, as can be seen from Figure 3.16. This time, the sensitivity analysis shows that thecopper thickness cut, the relative permittivity for the RO4350B™ substrate εr,RO4350B, thefr-4 substrate thickness fr-4t, the milling radius of the manufacturer milr, the microstrip linelength mlinl, the margin between substrate edge and microstrip line edge, and the waveguidediameter wd are the most robust parameters.

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Frequency [GHz]

|Sa|[dB]

cut− 5µmcut

cut+ 5µm

(a)

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0

Frequency [GHz]

|Sa|[dB]

εr − 0.06εr

εr + 0.06

(b)

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−10

0

Frequency [GHz]

|Sa|[dB]


fr-4t+ 200µm

(c)

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0

Frequency [GHz]

|Sa|[dB]

milr− 50µmmilr

milr+ 50µm

(d)

Figure 3.16: Sensitivity analysis in case of the large resonator cavity for (a) cut , (b) εr,RO4350B,(c) fr-4t and (d) milr.

The most dangerous parameters, that could cause deterioration of the antenna with a largediameter, are the same as in case of the small antenna. This can be confirmed by looking atthe sensitivity analysis for the microstrip line width mlinw, the inset of the feed line into thecavity mlininset and the RO4350B™ substrate thickness ROt. These parameters influence theeffective relative permittivity of the cavity, that determines the cut-off frequency of the te- and


tm-modes as previously explained, and the proper matching to 50Ω in case of mlinw and ROt.

16 18 20 22 24−40

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0

Frequency [GHz]

|Sa|[dB]

mlinl− 50µmmlinl

mlinl+ 50µm

(e)

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−10

0

Frequency [GHz]

|Sa|[dB]

mlinw− 50µmmlinw

mlinw+ 50µm

(f)

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0

Frequency [GHz]

|Sa|[dB]


mlininset+ 50µm

(g)

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0

Frequency [GHz]

|Sa|[dB]


margin+ 50µm

(h)

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0

Frequency [GHz]

|Sa|[dB]

ROt− 50µm

ROtROt+ 50µm

(i)

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−10

0

Frequency [GHz]

|Sa|[dB]

wd− 100µmwd

wd+ 100µm

(j)

Figure 3.16: Sensitivity analysis in case of the large resonator cavity for (e) mlinl, (f) mlinw, (g)mlininset, (h) margin, (i) ROt and (j) wd. (cont.)


3.2.3.2 Orthogonal Feeding

This section elaborates on the sensitivity analysis regarding the resonator cavity with an orthogonalfeeding topology for a resonator cavity consisting of two and three layers standard pcb technology.

Two Layered Resonator Cavity. Corresponding to previous observations, the two layeredcylindrical waveguide also has a robust design, as can be seen from Figure 3.17.

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−20

−10

0

Frequency [GHz]

|Sa|[dB]

cut− 5µmcut

cut+ 5µm

(a)

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−10

0

Frequency [GHz]

|Sa|[dB]

εr − 0.06εr

εr + 0.06

(b)

16 18 20 22 24

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−10

0

Frequency [GHz]

|Sa|[dB]


fr-4t+ 200µm

(c)

16 18 20 22 24

−20

−10

0

Frequency [GHz]

|Sa|[dB]

milr− 50µmmilr

milr+ 50µm

(d)

Figure 3.17: Sensitivity analysis in case of the two layered cavity for (a) cut, (b) εr,RO4350B, (c)fr-4t and (d) milr.

These sensitivity plots are in line with the aforementioned critical parameters, being the microstripline width mlinw, the inset of the feed line into the cavity mlininset and the RO4350B™ substratethickness ROt, shown in Figure 3.17i.


16 18 20 22 24

−20

−10

0

Frequency [GHz]

|Sa|[dB]

mlinl− 50µmmlinl

mlinl+ 50µm

(e)

16 18 20 22 24

−20

−10

0

Frequency [GHz]

|Sa|[dB]

mlinw− 50µmmlinw

mlinw+ 50µm

(f)

16 18 20 22 24−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]


mlininset+ 50µm

(g)

16 18 20 22 24

−20

−10

0

Frequency [GHz]

|Sa|[dB]


margin+ 50µm

(h)

16 18 20 22 24

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

ROt− 50µm

ROtROt+ 50µm

(i)

16 18 20 22 24−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

wd− 100µmwd

wd+ 100µm

(j)

Figure 3.17: Sensitivity analysis in case of the two layered cavity for (e) mlinl, (f) mlinw, (g)mlininset, (h) margin, (i) ROt and (j) wd. (cont.)


Three Layered Resonator Cavity. The three layered cylindrical waveguide has a robustdesign, as can be seen from Figure 3.17, which is completely in line with the aforementionedsensitivity analysis.

16 18 20 22 24

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

cut− 5µmcut

cut+ 5µm

(a)

16 18 20 22 24−60

−50

−40

−30

−20

−10

0

Frequency [GHz]|S

a|[dB]

εr − 0.06εr

εr + 0.06

(b)

16 18 20 22 24

−70

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]


fr-4t+ 200µm

(c)

16 18 20 22 24

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

milr− 50µmmilr

milr+ 50µm

(d)

Figure 3.18: Sensitivity analysis in case of the three layered cavity for (a) cut, (b) εr,RO4350B,(c) fr-4t and (d) milr.

Notice the small glitch in the sensitivity analysis of the fr-4 substrate thickness plotted in Figure3.18c. This is probably caused by a solver issue, since the time domain simulation of the sameparameter with deviations of 200µm gives no glitch.

These sensitivity plots confirm once more the critical parameters, being the microstrip line widthmlinw, the inset of the feed line into the cavity mlininset and the RO4350B™ substrate thicknessROt.


16 18 20 22 24

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

mlinl− 50µmmlinl

mlinl+ 50µm

(e)

16 18 20 22 24

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

mlinw− 50µmmlinw

mlinw+ 50µm

(f)

16 18 20 22 24

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]


mlininset+ 50µm

(g)

16 18 20 22 24

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]


margin+ 50µm

(h)

16 18 20 22 24

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

ROt− 50µm

ROtROt+ 50µm

(i)

16 18 20 22 24

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

wd− 100µmwd

wd+ 100µm

(j)

Figure 3.18: Sensitivity analysis in case of the two layered cavity for (e) mlinl, (f) mlinw, (g)mlininset, (h) margin, (i) ROt and (j) wd. (cont.)


3.2.4 Loss Mechanisms

The loss mechanisms are shown in Figure 3.19 by plotting the maximal Institute of Electrical andElectronics Engineers (ieee) gain Gmax over the desired operating frequency region. Remarkthat the gain defined by ieee is not the same as the realised gain, which is used to generate theradiation patterns.

First of all, the two layered cavity antenna topology with orthogonal feeding is simulated withoutlosses, meaning that perfect electric conductor (pec) is used instead of copper, without anyconductor losses or dielectric losses or surface roughness K = 1. Subsequently, a simulation isdone by replacing all pec material to copper, targeting the conductor losses without dielectriclosses or surface roughness. Afterwards, some surface roughness 1 < K < 2 with an rootmean square (rms) value of 2µm is added to the lossy copper material. Finally, the dielectriclosses are simulated by using pec material, but by adding a loss tangent αd to the RO4350B™substrate. The total loss of the antenna, containing conductor losses, surface roughness anddielectric losses, is also simulated.

18 18.5 19 19.5 206.5

7

7.5

8

8.5

17.7 20.2

Frequency [GHz]

Gm

ax[dB]

No lossesConductor losses (CL)

CL + rough surfaceDielectric losses

Total losses

Figure 3.19: Loss mechanisms in case of the two layered resonator cavity with orthogonal feeding.

There is seemingly no difference between the lossless structure and the dielectric losses, this canbe easily seen by observing the different loss mechanisms in Figure 3.19. Consequently, thereis no difference between, on the one hand, the simulation with conductor losses and a roughsurface and, on the other hand, the total losses. This is explained by the negligible dielectriclosses. In conclusion, it is conceivable to state that the conductor losses are the dominant lossfactor, as expected.


3.2.5 Axial Ratio

The axial ratio represents the polarization purity. Circular polarization (cp) is still acceptableas long as the axial ratio stays below 3 dB. In case of the time domain solver, the axial ratio hasa range of 104° in the elevation plane at the centre frequency, in case the axial ratio is simulatedfor different elevation angles θ. This is plotted in Figure 3.20a. However, this becomes a rangeof 130° when using the frequency domain solver, which can be deduced from Figure 3.21a.

−150 −100 −50 0 50 100 1500

10

20

30

40−52 52

3

Theta []

AR

[f=18.95

GH

z][dB]

(a)

18 18.5 19 19.5 200

0.2

0.4

0.6

0.8

1

17.7 20.2

Frequency [GHz]

AR

[θ=0][dB]

(b)

Figure 3.20: Axial ratio for the small two layered resonator cavity with orthogonal feeding inthe case of the time domain solver.

By looking at θ = 0°, one can conclude that the axial ratio stays below 0.9 dB for the entirebandwidth of operation in case of the time domain solver, as can be seen from Figure 3.20b. Inaddition, the axial ratio stays below 0.07 dB in case of the frequency domain solver, as can beseen from Figure 3.21b.

−150 −100 −50 0 50 100 1500

10

20

30

40−65 65

3

Theta []

AR

[f=18.95

GH

z][dB]

(a)

18 18.5 19 19.5 200

0.2

0.4

0.6

0.8

1

17.7 20.2

Frequency [GHz]

AR

[θ=0][dB]

(b)

Figure 3.21: Axial ratio for the small two layered resonator cavity with orthogonal feeding inthe case of the frequency domain solver.


3.3 Measurement Results

The actual measurements of the S-parameters for the three prototypes are recorded here. Thecomparison with the simulations, performed in cst®, is also discussed in this section.

(a) (b)

Figure 3.22: Prototype with a small resonator cavity using microstrip line (a) linear (b)orthogonal feeding.

The measurement of the S-parameters is based on the de-embedding of the Southwest connectorsand part of the feed line structure, such that only the simulated element is measured [15].Thereto, a personal microstrip line thru-line-reflect (trl) calibration kit is designed for thecentre frequency of the desired operating region [44].

16 18 20 22 24

−20

−10

0

Frequency [GHz]

|Sa|[dB]

m1 : Sa,1

m1 : Sa,2

m2 : Sa,1

m2 : Sa,2

m3 : Sa,1

m3 : Sa,2

Figure 3.23: Measured S-parameters for three prototypes of the small resonator cavity withlinear feeding.


16 18 20 22 24

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

cst, time : Sa,1

cst, time : Sa,2

m2 : Sa,1

m2 : Sa,2


16 18 20 22 24

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

cst, freq : Sa,1

cst, freq : Sa,2

m2 : Sa,1

m2 : Sa,2


Figure 3.24: Comparison of one of the measured S-parameters with the simulated S-parametersfor the small resonator cavity with linear feeding.

The measured passive S-parameters of all three prototypes of the small resonator cavity antennawere transformed into active S-parameters with a matlab-script. The resulting active S-parametersare quite similar for all prototypes, as can be seen from Figure 3.23. Consequently, this confirmsthe robustness of the design, as clarified in the sensitivity analysis. Although the S-parametercurves are less steep than in the simulations, the benefit of covering the entire K/Ka-downlinkfrequency band is the most important feature.

In addition, one of the measured active S-parameters, m2, is compared to the cst simulationsfor time and frequency domain in Figures 3.24a and 3.24b. In both cases, a slight shift towardshigher frequencies is observed, but the trend is more or less stable.

The same matlab-script was used to transform the measured passive S-parameters into activeS-parameters for all three prototypes in case of the large resonator cavity antenna. The robustnessis once more validated, as can be observed from Figure 3.25. The entire desired frequency bandand more is covered, as expected from the simulations. However, in practice, this might turnout to be a disadvantage, since the additional bandwidth will cause the antenna to pick upundesired frequencies as well.

Moreover, one of the measured active S-parameters, m3, is compared to the cst simulations forthe time and frequency domain in Figures 3.24a and 3.24b respectively. It seems that the datais more in line with the time domain simulation, due to unequal S-parameters. An explanationcan be formulated by stating that the fabrication tolerances introduce errors that violate thesymmetric behaviour of the design. Hence, the S-parameters are in between the simulationresults for the time and frequency domain solver.


16 18 20 22 24

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

m1 : Sa,1

m1 : Sa,2

m2 : Sa,1

m2 : Sa,2

m3 : Sa,1

m3 : Sa,2

Figure 3.25: Measured S-parameters for three prototypes of the large resonator cavity withlinear feeding.

16 18 20 22 24−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

cst, time : Sa,1

cst, time : Sa,2

m3 : Sa,1

m3 : Sa,2


16 18 20 22 24−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

cst, freq : Sa,1

cst, freq : Sa,2

m3 : Sa,1

m3 : Sa,2


Figure 3.26: Comparison of one of the measured S-parameters with the simulated S-parametersfor the large resonator cavity with linear feeding.

The measured passive S-parameters for three prototypes with a two layered orthogonally fedresonator cavity are converted to active S-parameters. The resulting data is plotted in Figure3.27a. Subsequently, it can be observed that all prototypes cover the desired frequency band,despite the many variations for Sa < −10 dB. The fabrication tolerances have a more explicitinfluence on the active S-parameters when compared to the measurements of the linearly fedprototypes. A comparison between the measurement of a cherry picked prototype and thesimulated data is illustrated in Figure 3.28b.


18 20 22 24

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

m1 : Sa,1

m1 : Sa,2

m1 : Sa,3

m1 : Sa,4

m2 : Sa,1

m2 : Sa,2

m2 : Sa,3

m2 : Sa,4

m3 : Sa,1

m3 : Sa,2

m3 : Sa,3

m3 : Sa,4

(a)

18 20 22 24

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB] cst : Sa,1

cst : Sa,2

cst : Sa,3

cst : Sa,4

m3 : Sa,1

m3 : Sa,2

m3 : Sa,3

m3 : Sa,4

(b)

Figure 3.27: (a) Measured S-parameters for three prototypes in case of a two layered cavity. (b)Comparison of one of the measured S-parameters with the simulated S-parameters.

18 20 22 24

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

m1 : Sa,1

m1 : Sa,2

m1 : Sa,3

m1 : Sa,4

m2 : Sa,1

m2 : Sa,2

m2 : Sa,3

m2 : Sa,4

m3 : Sa,1

m3 : Sa,2

m3 : Sa,3

m3 : Sa,4

(a)

18 20 22 24

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB] cst : Sa,1

cst : Sa,2

cst : Sa,3

cst : Sa,4

m1 : Sa,1

m1 : Sa,2

m1 : Sa,3

m1 : Sa,4

(b)

Figure 3.28: (a) Measured S-parameters for three prototypes in case of a three layered cavity.(b) Comparison of one of the measured S-parameters with the simulated S-parameters.

The active S-parameters for three prototypes with a three layered orthogonally fed resonatorcavity are plotted in Figure 3.28a. Many variations can be observed once more. However, thebandwidth remains almost the same, covering the desired frequency band of operation. A generaltrend can be extracted, where the resonance shifts towards higher frequencies. The comparisonof active S-parameters for one prototype with the simulation results is shown in 3.28b.


3.4 Conclusion

Comparing the small and large resonator cavity with linear microstrip line feeding resultedin a preference for the smaller design. This smaller design, besides having a smaller footprint,also avoids picking up extra undesired frequencies. Alternative options to broaden the frequencybandwidth were not necessary, as it turned out from the measurements that the entire K/Ka-banddownlink was covered in a reliable way.

Proceeding with an investigation into the two and three layered resonator cavities with orthogonalmicrostrip line feed, showed that the two layered design is preferred. Since both designs coverthe entire K/Ka-band downlink and their farfield specifications are alike, the lower profile wasadvantageous for the purpose of this master’s dissertation. A trade-off was made between theheight and the cavity diameter, in other words, an increase of 200µm had to be tolerated inorder to gain a reduction of 1.62mm in height.

The disadvantages of this topology lie with the sensitivity to the inset of the microstrip line feedinto the cavity. This crucial parameter can deteriorate the entire system when the deviationfrom its optimal value is too large, as shown in Figures 3.15g, 3.16g, 3.17g and 3.18g.

Finally, microstrip lines are very lossy and should be avoided as much as possible to keepthe system efficient. Therefore the use of siw as feeding topology could be more beneficial.Another alternative is to minimize the length of the microstrip feed lines by transferring thefields downwards towards the backplane of the cavity.

ANTENNA ELEMENT WITH SUBSTRATE INTEGRATED WAVEGUIDE FEED 39

Chapter 4

Antenna Element with SubstrateIntegrated Waveguide Feed

4.1 Proposed Solutions

This section elaborates on the design procedure in cst mws for two antenna element with ansiw feeding topology [48], being a small and a large resonator cavity with linear feeding.


Although microstrip lines are more compact, they considerably suffer from parasitic radiation,making them a lossy feeding option. This is prevented by using an siw feeding topology, aspresented in Figure 4.1. Since siw can not be inserted into the resonator cavity, a transitionfrom microstrip to siw has to be designed as well [17], as shown in Figure 4.1b.

wd

wh

hd

FR-4t

ROt

cutx

z

(a)

Figure 4.1: Antenna topology with siw feed in (a) side view.


ROw

ROlhd

mlinl mlinwwd

margin

taperlsiwl taperw siww

x

y

(b)

Figure 4.1: Antenna topology with siw feed in (b) top view with linear feeding. (cont.)


Important features for the correct initial sizing of the siw and the tapering to the microstripline are explained in this section. The sizing of the rest of the antenna topology remains thesame as in section 3.1.3.

Larger widths for microstrip lines are allowed, since they are not connected directly to thenetwork analyser. A transition to siw is made, then half of the thru of the calibration kit isadded, where the siw transforms back into a microstripline with a width of 0.475mm to matchthe 50Ω port impedance.

Substrate Integrated Waveguide. The cut-off frequency of a rectangular siw can be calculatedby using equation (4.1), which is valid for a rectangular waveguide [5], [9], [15].

fc,m,n,0 =c

2π√εr

√(mπ

Weff

)2

+(nπ

h

)2(4.1)


Parameter Small design [mm] Large design [mm]cut 0.035 0.035FR-4t 1.55 1.55FR-4w 17.5 22.8FR-4l 17.5 22.8ROt 0.25 0.25ROw 17.5 22.8ROl 17.5 22.8wd 8 14.3wh 5.145 5.145hd 1.0 1.0mlinw 0.47 0.5mlinl 2.45 3mlininset 2.45 3margin 0.2 0.2milr 0.5 0.5siwl 2 2siww 6.05 6.05taperl 2.75 2.25taperw 0.9 1.4viad 0.25 0.25vias 0.5 0.5viae 0.15 0.15

Table 4.1: Optimized dimensions for linear siw feeding of the resonator cavity.

cut

h

viad

vias

W

L

Figure 4.2: Substrate integrated waveguide topology.

There are different methods to calculate the effective width Weff of an siw as discussed in theliterature [10], [38]. The width Weff as defined in Equation (4.2) is used here.

Weff = W −via2d

0.95vias; (4.2)

The width W represents the shortest distance between the two rows of vias, measured from thecentre of the vias, as shown in Figure 4.2. The spacing between the centres of two adjacent


vias vias is required, as well as their diameter viad, to calculate the effective width of the siw.The te10 mode can easily propagate when using a width of W = 5.8mm. A via diameterof viad = 250µm and a via spacing of vias = 2viad = 500µm is used, compliant with theliterature [10] and using the minimal manufacturable plated through hole (pth) size for rf poolby Eurocircuits [19]. The distance between the via edge and the edge of the top copper layer isviae = 150µm.

Taper from microstrip line to substrate integrated waveguide. Indicative startingvalues for the taper width and length can be found from literature [35], [16].

120π

ηH[

taperwH + 1.393 + 0.667 ln

(taperw

H + 1.444)] =

4.38

Wexp

−0.627εr

εr+12 + εr−1

2√

1+ 12Htaperw

(4.3)

Where η is the free space intrinsic impedance 377Ω, H is the height of the substrate and Wis defined as in Figure 4.2. The taper width found after solving Equation (4.3) that is smallerthan the siw width is taperw = 2.1mm.

taperl =nλg

4(4.4)

The guided wavelength λg is based on the centre frequency and the relative permittivity of theRO4350B™ substrate. To minimize the footprint of the antenna element, n = 1 will be assumed.The resulting length of the taper is taperl = 2mm.


The cst simulation results, concerning the S-parameters and the realised gain radiation patternsfor two cross-sections of ϕ = 0° and ϕ = 90° in case of a linear siw feeding topology, are discussedin this section.

The excited modes, as well as how they are created, as discussed in section 3.2.1 are still valid.Once more, the difference between the time and frequency domain simulators are illustrated inFigures 4.3 and 4.4.

The small resonator cavity does not cover the entire operational bandwidth, while the largercavity covers more than needed. This is in line with the simulation results of the antenna withmicrostrip line feed, a presented in Figures 3.5 and 3.6. Since the objective is to design an


efficient antenna with a small footprint, the small resonator cavity with siw feeding topology isstill in favour.

16 18 20 22

−30

−20

−10

017.65 20.09

2.44GHz

Frequency [GHz]

|Sa|[dB]

|S11,a||S22,a|


16 18 20 22

−20

−10

017.94 20.92

2.98GHz

Frequency [GHz]

|Sa|[dB]

|S11,a||S22,a|


Figure 4.3: Active S-parameters for a resonator cavity with diameter wd = 8mm and a linearsiw feeding topology.

16 18 20 22

−30

−20

−10

016.36 21.44

5.08GHz

Frequency [GHz]

|Sa|[dB]

|S11,a||S22,a|


16 18 20 22

−20

−10

015.58 22.76

7.18GHz

Frequency [GHz]

|Sa|[dB]

|S11,a||S22,a|


Figure 4.4: Active S-parameters for a resonator cavity with diameter wd = 14.3mm and a linearsiw feeding topology.

The time domain solver shows two unequal active S-parameters, which is most likely due tostaircasing in the meshing, creating a virtually slightly asymmetric design. The frequencydomain solver of the same design shows that the design is probably not optimized in the correctway, since the bandwidth is not centred around the centre frequency fcentre = 18.95GHz, asshown in Figures 4.3b and 4.4b. However, a realistic manufacturing will never result in acompletely symmetric design due to the fabrication tolerances on its production. Therefore,


optimizing for the centre frequency, as is done in the time domain simulations, might beadvantageous.

Radiation patterns for the small resonator cavity in case of linear siw feeding can be found inFigures 4.5 and 4.6 for the time and frequency domain solvers respectively. An overview of themost important characteristics can be found in Table 4.2.

030

60

90

120

150180

−150

−120

−90

−60

−30

−20

−10

0

10θ

gain

[dBi]

(a) At 17.7GHz for ϕ = 0°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−15−10−50510

θ

gain

[dBi]

(b) At 18.95GHz for ϕ = 0°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−20

−10

0

10θ

gain

[dBi]

(c) At 20.2GHz for ϕ = 0°.0

30

60

90

120

150180

−150

−120

−90

−60

−30

−15−10−50510

θ

gain

[dBi]

(d) At 17.7GHz for ϕ = 90°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−15−10−50510

θ

gain

[dBi]

(e) At 18.95GHz for ϕ = 90°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−15−10−50510

θ

gain

[dBi]

(f) At 20.2GHz for ϕ = 90°.

Figure 4.5: Radiation patterns for a resonator cavity with diameter wd = 8mm and a linearsiw feeding topology in case of the time domain solver.

030

60

90

120

150180

−150

−120

−90

−60

−30

−30−20−10

010

θ

gain

[dBi]

(a) At 17.7GHz for ϕ = 0°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−20

−10

0

10θ

gain

[dBi]

(b) At 18.95GHz for ϕ = 0°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−30−20−10

010

θ

gain

[dBi]

(c) At 20.2GHz for ϕ = 0°.0

30

60

90

120

150180

−150

−120

−90

−60

−30

−15−10−50510

θ

gain

[dBi]

(d) At 17.7GHz for ϕ = 90°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−15−10−50510

θ

gain

[dBi]

(e) At 18.95GHz for ϕ = 90°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−15−10−50510

θ

gain

[dBi]

(f) At 20.2GHz for ϕ = 90°.

Figure 4.6: Radiation patterns for a resonator cavity with diameter wd = 8mm and a linearsiw feeding topology in case of the frequency domain solver.


Considering the smaller design with siw feed, both the time and frequency domain solver deliverarguably the same result. The main lobe magnitude is always larger than 5.7 dBi. Moreover,side lobe levels stay well below the main lobe magnitude at all times. Finally, the angularbeamwidth is larger than 67°.

Realised gain Time domain Frequency domainfmin = 17.7GHz ϕ = 0° ϕ = 90° ϕ = 0° ϕ = 90°Main Lobe Magnitude [dBi] 6.27 5.7Main Lobe Direction [°] 0 0 0 0Angular Width (3 dB) [°] 93.6 79.1 94.5 78.1Side Lobe Level [dB] -9.6 -13.3 -9.6 -15.2fcentre = 18.95GHz ϕ = 0° ϕ = 90° ϕ = 0° ϕ = 90°Main Lobe Magnitude [dBi] 6.92 6.79Main Lobe Direction [°] 0 0 0 0Angular Width (3 dB) [°] 82.8 84.3 87.5 80.4Side Lobe Level [dB] -8.9 -12.0 -9.1 -2.31fmax = 20.2GHz ϕ = 0° ϕ = 90° ϕ = 0° ϕ = 90°Main Lobe Magnitude [dBi] 7.29 7.46Main Lobe Direction [°] 0 0 0 0Angular Width (3 dB) [°] 67.1 80.9 73.1 79.2Side Lobe Level [dB] -8.9 -10.3 -9.4 -12.7

Table 4.2: Overview of radiation pattern characteristics for linear siw feeding of the smallresonator cavity.

030

60

90

120

150180

−150

−120

−90

−60

−30

−40−30−20−10

010

θ

gain

[dBi]

(a) At 17.7GHz for ϕ = 0°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−30−20−10

010

θ

gain

[dBi]

(b) At 18.95GHz for ϕ = 0°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−40−30−20−10

010

θ

gain

[dBi]

(c) At 20.2GHz for ϕ = 0°.0

30

60

90

120

150180

−150

−120

−90

−60

−30

−20

−10

0

10θ

gain

[dBi]

(d) At 17.7GHz for ϕ = 90°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−20

−10

0

10θ

gain

[dBi]

(e) At 18.95GHz for ϕ = 90°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−20

−10

0

10θ

gain

[dBi]

(f) At 20.2GHz for ϕ = 90°.

Figure 4.7: Radiation patterns for a resonator cavity with diameter wd = 14.3mm and a linearsiw feeding topology in case of the time domain solver.

Radiation patterns for the large resonator cavity in case of linear siw feeding can be found in


Figures 4.7 and 4.8 for the time and frequency domain solvers respectively. An overview of themost important characteristics can be found in Table 4.3.

030

60

90

120

150180

−150

−120

−90

−60

−30

−40−30−20−10

010

θ

gain

[dBi]

(a) At 17.7GHz for ϕ = 0°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−30−20−10

010

θ

gain

[dBi]

(b) At 18.95GHz for ϕ = 0°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−30−20−10

010

θ

gain

[dBi]

(c) At 20.2GHz for ϕ = 0°.0

30

60

90

120

150180

−150

−120

−90

−60

−30

−30−20−10

010

θ

gain

[dBi]

(d) At 17.7GHz for ϕ = 90°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−20

−10

0

10θ

gain

[dBi]

(e) At 18.95GHz for ϕ = 90°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−20

−10

0

10θ

gain

[dBi]

(f) At 20.2GHz for ϕ = 90°.

Figure 4.8: Radiation patterns for a resonator cavity with diameter wd = 14.3mm and a linearsiw feeding topology in case of the frequency domain solver.

Realised gain Time domain Frequency domainfmin = 17.7GHz ϕ = 0° ϕ = 90° ϕ = 0° ϕ = 90°Main Lobe Magnitude [dBi] 9.08 9.1Main Lobe Direction [°] 0 0 0 0Angular Width (3 dB) [°] 54.6 59.9 54.6 59.9Side Lobe Level [dB] -17.5 -17.7 -19.0 -27.3fcentre = 18.95GHz ϕ = 0° ϕ = 90° ϕ = 0° ϕ = 90°Main Lobe Magnitude [dBi] 8.9 9.28Main Lobe Direction [°] 0 0 0 0Angular Width (3 dB) [°] 62.8 67.2 57.4 62.5Side Lobe Level [dB] -22.3 -17.6 -19.5 /fmax = 20.2GHz ϕ = 0° ϕ = 90° ϕ = 0° ϕ = 90°Main Lobe Magnitude [dBi] 9.2 9.74Main Lobe Direction [°] 0 0 0 0Angular Width (3 dB) [°] 56.0 70.4 52.2 62.3Side Lobe Level [dB] -24.4 -17.9 -20.1 -21.6

Table 4.3: Overview of radiation pattern characteristics for linear siw feeding of the largeresonator cavity.

Considering the larger design with siw feed, both the time and frequency domain solver deliverarguably the same result. The main lobe magnitude is always large than 8.9 dBi. Moreover, sidelobe levels stay well below the main lobe magnitude at all times. Finally, the angular beamwidthis larger than 52°.


The same trade-off can be observed as previously described in section 3.2.1. A larger realisedgain inevitably results in a smaller 3 dB beam width, which is undesired for this application.


The actual measurements of the S-parameters are presented in this section. The comparisonwith the simulations performed in cst® is also discussed here.

Figure 4.9: Prototype with a small resonator cavity using siw linear feeding.

A custom siw line trl calibration kit is designed for the centre frequency of the desired operatingregion [44]. This calibration kit allows the de-embedding of the Southwest connectors and partof the feed line structure.

16 18 20 22 24−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

m1 : Sa,1

m1 : Sa,2

m2 : Sa,1

m2 : Sa,2

m3 : Sa,1

m3 : Sa,2

Figure 4.10: Measured S-parameters for three prototypes of the small resonator cavity with alinear siw feeding topology.

The measured passive S-parameters from all three prototypes with siw feeding topology and


small resonator cavity are transformed to active S-parameters by means of a matlab-script.The resulting active S-parameters, plotted in Figure 4.10, resemble each other well.

16 18 20 22 24−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

cst, time : Sa,1

cst, time : Sa,2

m1 : Sa,1

m1 : Sa,2


16 18 20 22 24−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

cst, freq : Sa,1

cst, freq : Sa,2

m1 : Sa,1

m1 : Sa,2


Figure 4.11: Comparison of one of the measured S-parameters with the simulated S-parametersfor the small resonator cavity with a linear siw feeding topology.

However, the measurements are not in line with the simulations. This can be easily seen bycomparing Figure 4.10 to the simulated active S-parameters in case of the time and frequencydomain solver, Figure 4.11a and Figure 4.11b respectively.

16 18 20 22 24

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

m1 : Sa,1

m1 : Sa,2

m2 : Sa,1

m2 : Sa,2

m3 : Sa,1

m3 : Sa,2

Figure 4.12: Measured S-parameters for three prototypes of the large resonator cavity with alinear siw feeding topology.

The same matlab-script has been used to transform the measured passive S-parameters intoactive S-parameters for all three prototypes of the large resonator cavity with siw feedingtopology. The resulting S-parameters, plotted in Figure 4.12, are similar to each other.


16 18 20 22 24

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

cst, t : Sa,1

cst, t : Sa,2

m1 : Sa,1

m1 : Sa,2


16 18 20 22 24

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

cst, f : Sa,1

cst, f : Sa,2

m1 : Sa,1

m1 : Sa,2


Figure 4.13: Comparison of one of the measured S-parameters with the simulated S-parametersfor the resonator cavity with a linear siw feeding topology.

Once more, the measurements are not compliant with the simulations. This can easily be seenby comparing Figure 4.12 to the simulated active S-parameters in case of the time and frequencydomain solver, Figure 4.13a and Figure 4.13b respectively.

Since another calibration kit has been used than in the previous section with microstrip linefeeding, the faulty measurement results might be due to an incorrect calibration kit. Whendoing the simulations with an extended substrate, including part of the calibration thru, themeasurement results follow the simulation.

4.4 Conclusion

This path will not be pursued due to the unreliability of these measurements. A properinvestigation of these incompatibilities falls out of the scope of this master’s dissertation. Asecond reason not to pursue the less lossy siw feeding network is that it is too bulky for thisapplication.

An alternative feeding topology is required, preferably one that allows good isolation whenintegrating other components on the backplane of the resonator cavity. The footprint of theantenna element can be kept as small as possible by integrating all power and phase alteringelements on this backplane.

FINAL SINGLE ANTENNA ELEMENT 50

Chapter 5

Final Single Antenna Element

5.1 Proposed Solution

This section gives a solution to the problems encountered with the previous prototypes. Asolution to lossy microstrip lines or an unreliable siw calibration kit is provided by using acavity-backed afsiw topology. This section elaborates on the design procedure of the finalantenna element for this master dissertation in cst mws.


This design is based on the two layered resonator cavity that is orthogonally fed by fourmicrostrip lines from section 3.2.2. The proposed circular cavity-backed afsiw topology consistsof a feeding layer on top, some edge plated drilled fr-4 standard pcbs and a bottom layer forthe main feeding topology, as illustrated in Figure 5.1.

Using a transition from the top plane to the back plane of the antenna, allows for futureintegration of more components onto the single antenna element without increasing its footprint.If these components are small enough, they can be fit into the tight space beneath the antenna’sresonator cavity, shown in the centre of Figure 5.1c. If larger components would be used, anintermediate hidden layer could be introduced to integrate these components.

However, the microstrip lines on this layer should point inwards to access these components andto be able to simulate and manufacture the antenna array. Nonetheless, the design is optimizedfor the microstrip lines pointing outwards, for practical considerations. This way, measurementscan be performed in a straightforward manner and the system can be thoroughly validated.


wd

whhd

FR-4t

ROt

cut

afsiww afsiwh

cus

spacing mlinw,top mlinl,top

mlinl,bottom

x

z

(a)ROw

ROlhd

mlinl,top mlinw,topwd

margin

spacing

cus

x

y

(b)

ROw

ROlhd

mlinl,bottom mlinw,bottom

spacing

x

y

(c)

afsiwα

FR-4w

FR-4l

hd

wdplating afsiww

cus

x

y

(d)

Figure 5.1: Layout of the cavity-backed afsiw antenna topology in (a) side view (b) top view(c) bottom view (d) top view of both drilled and edge plated standard pcbs with fr-4 substrate.


Moreover, the transition from the front- to the backplane of the antenna ensures minimizationof the radiation losses due to the microstrip lines on the top feeding layer, shown in Figure 5.1b.The microstrip lines can be minimized in length and the stubs can be tuned to achieve optimaloperation of the antenna. Additionally, adding vias all around confines the fields carried by thefeeding structure even more. This creates a grounded coplanar waveguide (gcpw)-like structureat the end of the microstrip lines.

The transition makes use of afsiws, which are in this case edge plated milled holes in bothstandard pcb layers. These pths are filled with air, as illustrated in Figure 5.1d, resultingin low losses and high power handling capabilities. The coupling of the fields guided by themicrostrip lines to the afsiws happens more efficiently when providing a boundary of vias inboth RO4350B™ feeding layers, creating classic siw topology.


Important features for the correct initial sizing of the afsiw are explained in this section. Thesizing of the rest of the antenna topology remains the same as in section 4.1.2.

Parameter Dimensionscut 0.035mmFR-4t 1.55mmFR-4w 12mmFR-4l 12mmROt 0.25mmROw 12mmROl 12mmmilr 0.5mm

wd 8mmwh 3.56mmhd 1.0mm

afsiww 0.5mmafsiwα 82°

Parameter Dimensionsmlinw,top 0.5mmmlinl,top 3.9mmmlininset 2.4mmmlinw,bottom 0.475mmmlinl,bottom 2.2mm

margin 0.4mmspacing 0.1mmcus 0.2mm

viad 0.25mmvias 0.5mmviae 0.15mm

Table 5.1: Optimized dimensions for final single antenna element.

The radius around the resonator cavity with a margin of cus = 2mm, results in a minimalavailable circumference.

∆ = 2πR = 2π(wd

2+ cus

)= 26.4mm (5.1)

Four afsiw are present the proposed design, all with isolating vias in between, which mean thattheir maximal length for this minimal circumference is represented by afsiwl, which can also


be expressed as an angle afsiwα.

afsiwl =∆

4− 2viae − viad = 6mm (5.2)

Consequently, the cut-off frequency can be calculated by using Equation (4.1) and setting Weff =

afsiwl, h = afsiww and assuming that εr = 1, resulting in a value of fc,1,0,0 = 25GHz.Nonetheless, the effective relative permittivity εr,eff should be used to calculate the cut-offfrequency, therefore taking the amount of RO4350™ into account with respect to the amountof fr-4. When using two standard pcb layers, the te10 mode can propagate, since εr,eff willbe larger than one and thus the cut-off frequency will be lower.

However, the εr,eff will be smaller than in the previous case, but still larger than 1, when usingthree standard pcb layers and no mode will propagate in the afsiw. Consequently, optimizingthe three layered design was not possible.

The width of the bottom microstrip lines are optimized for a line impedance of 50Ohm, whilethe width of the microstriplines on the top can have a different width. Ideally, it is expectedthat the optimal width for the top microstrip lines is wider, since wider microstrip lines canmore easily couple to the fields of the afsiw.

A spacing between both edge plated drills is added as a safety for fabrication tolerances. Thevalue of cuc represents the closest distance between the edge of the afsiw walls and the resonatorcavity walls. Another margin is added between two etched copper planes, being the microstriplines and the grounded patch. This is represented by the variable spacing.

To cope with fabrication tolerances, an extra margin of 100µm is introduced between the inneredge of the afsiw and the edged copper on top, such that this grounded plane can not cover apart of the afsiw.


The cst® simulation results, concerning the S-parameters and the realised gain radiation patternsfor a single cross-section of ϕ = 0° for the circular cavity-backed afsiw topology, are discussedin this section.

The active S-parameters, simulated with the frequency domain solver, are plotted in Figure 5.2.The entire K/Ka-band downlink is covered without too much additional bandwidth on the sides,making them the best S-parameters of this master dissertation until now.


16 18 20 22−70

−60

−50

−40

−30

−20

−10

017.65 20.42

2.77GHz

Frequency [GHz]

|Sa|[dB]

|S11,a||S22,a||S33,a||S44,a|

Figure 5.2: Active S-parameters for the cylindrical cavity-backed afsiw topology.

The radiation patterns for the final cavity-backed afsiw antenna topology are plotted in Figure5.3, whereas an overview of the most important characteristics can be found in Table 5.2.

030

60

90

120

150180

−150

−120

−90

−60

−30

−10−50510

θ

gain

[dBi]

(a) At frequency 17.7GHz forϕ = 0°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−10−50510

θ

gain

[dBi]

(b) At frequency 18.95GHz forϕ = 0°.

030

60

90

120

150180

−150

−120

−90

−60

−30

−10−50510

θ

gain

[dBi]

(c) At frequency 20.2GHz forϕ = 0°.

Figure 5.3: Radiation patterns for the cylindrical cavity-backed afsiw topology.

fmin = 17.7GHz fc = 18.95GHz fmax = 20.2GHz

Main Lobe Magnitude [dBi] 5.05 4.83 4.52Main Lobe Direction [°] 0 0 0Angular Width (3 dB) [°] 81.0 97.1 106.0Side Lobe Level [dB] -6.7 -7.5 -5.8

Table 5.2: Overview of radiation pattern characteristics for the circular cavity-backed afsiwtopology.

The main lobe magnitude is always large than 4.5 dBi. Moreover, side lobe levels stay well belowthe main lobe magnitude at all times. More back radiation, and thus a higher side lobe level,should be expected due to the microstrip lines on the bottom plane. However, the additional viasproved to be a valid design choice, such that the fields are confined and thus the back radiationremains limited. Finally, the angular beamwidth is larger than 81°. Notice the trade-off between


the main lobe magnitude and the 3 dB angular beamwidth. The larger the realised gain, thesmaller the beamwidth.

A sensitivity analysis is performed to test the robustness of the design. The results are shownin Figure 5.4.

16 18 20 22 24

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

cut− 5µmcut

cut+ 5µm

(a)

16 18 20 22 24

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

εr − 0.06εr

εr + 0.06

(b)

16 18 20 22 24

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]


fr-4t+ 200µm

(c)

16 18 20 22 24

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

milr− 50µmmilr

milr+ 50µm

(d)

Figure 5.4: Sensitivity analysis for the cylindrical cavity-backed afsiw topology for (a) cut, (b)εr,RO4350B, (c) fr-4t and (d) milr.

The most sensitive parameters are the length of the microstrip lines mlinl,bottom and mlinl,top,the inset into the cavity mlininset, the thickness of the RO4350B™ substrate ROt and the fr-4substrate fr-4t, the waveguide diameter wd and the spacing. The thickness of the fr-4 substrateis, without a doubt, the parameter that could deteriorate the entire system when deviations causea larger thickness.


16 18 20 22 24 26

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

mlinl,bottom− 50µmmlinl,bottom

mlinl,bottom+ 50µm

(e)

16 18 20 22 24 26

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

mlinl,top− 50µmmlinl,top

mlinl,top+ 50µm

(f)

16 18 20 22 24 26

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]


mlininset+ 50µm

(g)

16 18 20 22 24

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]


margin+ 50µm

(h)

16 18 20 22 24

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

ROt− 50µm

ROtROt+ 50µm

(i)

16 18 20 22 24

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

wd− 100µmwd

wd+ 100µm

(j)

Figure 5.4: Sensitivity analysis for the cylindrical cavity-backed afsiw topology for (e)mlinl,bottom, (f) mlinl,top (g) mlininset, (h) margin, (i) ROt and (j) wd. (cont.)


16 18 20 22 24

−60

−50

−40

−30

−20

−10

0

Frequency [GHz]

|Sa|[dB]

spacing− 50µmspacing

spacing+ 50µm

(k)

Figure 5.4: Sensitivity analysis in case of the cavity-backed afsiw for (k) spacing. (cont.)

The axial ratio has a range of 88° in the elevation plane at the centre frequency, in case the axialratio is simulated for different elevation angles θ. This is plotted in Figure 5.5a. By looking atθ = 0°, one can conclude that the axial ratio stays below 0.033 dB for the entire bandwidth ofoperation, illustrated in Figure 5.5b.

−150 −100 −50 0 50 100 1500

10

20

30

40−44 44

3

Theta []

AR

[f=18.95

GH

z][dB]

(a)

18 18.5 19 19.5 200

0.2

0.4

0.6

0.8

1

17.7 20.2

Frequency [GHz]

AR

[θ=0][dB]

(b)

Figure 5.5: Axial ratio for the small two layered resonator cavity with orthogonal feeding in thecase of the frequency domain solver.



The prototypes provided by Eurocircuits were not manufactured correctly at the fr-4 level.Sadly, there will be no measurement data available for the comparison with the simulations andthe previous antenna designs.

(a) (b)

Figure 5.6: Prototype of the final antenna element (a) front view (b) back view.

5.4 Conclusion

The cavity-backed afsiw antenna topology seems like a very promising design, judging fromthe simulations. While the entire downlink frequency band is covered, the antenna’s main lobemagnitude stays above 4.5 dBi while the 3 dB beamwidth is larger than 81°. Nonetheless, apower divider and phase shifter as well as a way to switch from lhcp to rhcp still needs to beimplemented. Furthermore, the K/Ka-band uplink still has to be provided.

The design could have been improved by optimizing a gcpw feed to a line impedance of 50Ω,instead of using a microstrip line on the bottom layer. This could result in a better side lobelevel, since fields are more confined within the substrate.

CONCLUSION 59

Chapter 6

Conclusion

6.1 Evaluation of the Final Antenna

The initial requirements of a cost-effective, small footprint, low profile, light weight antenna havedefinitely been reached. Moreover, the high antenna platform isolation allows close proximityintegration of active electronics beneath the resonator cavity. However, to create an operationalbeam steering array that functions for both circular polarisations, lhcp and rhcp, and at bothfrequency bands, the downlink and uplink, some additional work is required.

Overall, the circular cavity-backed afsiw antenna topology has quite good characteristics. Firstoff, a trade-off can be made between larger gain and larger 3 dB beamwidth by adjusting theparameters of the circular resonator cavity. Depending on the application, the user might havethe desire to use the smaller or larger cavity diameter from section 3.2.1.

Proceeding, all antenna designs are robust and can withstand errors in the order of standard pcb

manufacturing tolerances. However, the more complex the design becomes, the more sensitiveit will be to fabrication tolerances, as can be concluded when comparing the sensitivity analysisfor the small two fr-4 layered antenna with orthogonal microstrip line feeding and the finalcircular cavity-backed afsiw topology.

6.2 Future Work

The proposed circular cavity-backed afsiw topology consists of a feeding layer on top, someedge plated drilled fr-4 standard pcbs in between, possibly a hidden layer and a back layer forthe main feeding topology. This hidden layer could be introduced to integrate phase shiftersand power dividers to apply proper amplitudes and phases to each signal, without enlarging the

CONCLUSION 60

footprint of the antenna. Consequently, an adaptive system can be created by the possibilitythat each element can be tuned individually.

Developing an array out of the the final single antenna element, will require that the microstriplines on the bottom have to be turned around, pointing towards the cavity and not the edge ofthe substrate. This way, the microstrip lines can be directly connected to the rest of the adaptivefeeding network beneath the resonator cavity. Even active electronics can be used, since a goodisolation is provided by the resonator cavity.

Each element will be fed by a single input signal, but feeding the proposed topology requiresfour input signals with a 90° phase shift. The use of hybrid couplers for this matter is stronglydiscouraged due to their bulky structure [11], [32], [52]. More suitable is the use of a ratrace [47], [34] or some novel, frequency stable and compact phase shifter [2].

A dual band antenna array for the K/Ka-band uplink and downlink is advised for a fullyoperational system. Therefore, an antenna layout that provides both frequency bands is desired.An interesting solution is given in the literature [4]. However, it requires the design of threedifferent single antenna elements, being a downlink antenna, an uplink antenna and a dual bandantenna.

BIBLIOGRAPHY I

Bibliography

[1] Grzegorz Adamiuk, Werner Wiesbeck, and Thomas Zwick. Differential feeding as a conceptfor the realization of broadband dual-polarized antennas with very high polarization purity.In Antennas and Propagation Society International Symposium, 2009. APSURSI’09. IEEE,pages 1–4. IEEE, 2009.

[2] Abdulrahman Alaqeel, Hussein Shaman, Sultan Almorqi, Osama Haraz, Saleh Alshebeili,and Abdel-razik Sebak. Butterfly-shaped slot coupled microstrip 90° hybrid couplers fork-and ka-band millimeter-wave radar applications. In Microwave Symposium (MMS), 201414th Mediterranean, pages 1–3. IEEE, 2014.

[3] E. Arnieri, L. Boccia, and G. Amendola. A ka-band dual-frequency radiator for arrayapplications. IEEE Antennas and Wireless Propagation Letters, 8:894–897, 2009.

[4] E. Arnieri, L. Boccia, and G. Amendola. A ka-band dual-frequency radiator f>or arrayapplications. IEEE Antennas and Wireless Propagation Letters, 8:894–897, 2009.

[5] Constantine A Balanis. Antenna Theory: Analysis and Design, third edition. John Wiley& Sons, 2005.

[6] Constantine A Balanis. Advanced Engineering Electromagnetics, second edition. John Wiley& Sons, 2012.

[7] Hendrik Bayer, Alexander Krauss, Ralf Stephan, and Matthias A Hein. A compactplanar feed structure for ka-band satcom-on-the-move tracking antennas. In Antennasand Propagation (EuCAP), 2015 9th European Conference on, pages 1–5. IEEE, 2015.

[8] Angel Belenguer, Juan Luis Cano, Héctor Esteban, Eduardo Artal, and Vicente EBoria. Empty substrate integrated waveguide technology for e plane high-frequency andhigh-performance circuits. Radio Science, 52(1):49–69, 2017.

[9] Dries Bosman, Igor Lima de Paula, and Laura Van Messem. A 24 ghz continuous-waveautomotive radar front-end. 2015.

BIBLIOGRAPHY II

[10] Maurizio Bozzi, Anthimos Georgiadis, and Kaijie Wu. Review of substrate-integratedwaveguide circuits and antennas. IET Microwaves, Antennas & Propagation, 5(8):909–920,2011.

[11] Yujian Cheng, Wei Hong, and Ke Wu. Novel substrate integrated waveguide fixed phaseshifter for 180-degree directional coupler. In Microwave Symposium, 2007. IEEE/MTT-SInternational, pages 189–192. IEEE, 2007.

[12] Jorn Christensen. Itu regulations for ka-band satellite networks. In 30th AIAA InternationalCommunications Satellite System Conference (ICSSC), page 15179, 2014.

[13] Rogers Corporation. RO4000® Series High Frequency Circuit Materials. https://www.

rogerscorp.com/documents/726/acm/RO4000-Laminates---Data-sheet.pdf, 2017.

[14] W. Alan Davis. Impedance Matching, pages 380–. Wiley-IEEE Press, 2010.

[15] Thomas Deckmyn. Design of a 60 ghz hybrid mode substrate integrated waveguidecavity-backed antenna array. 2015.

[16] Dominic Deslandes. Design equations for tapered microstrip-to-substrate integratedwaveguide transitions. In Microwave Symposium Digest (MTT), 2010 IEEE MTT-SInternational, pages 704–707. IEEE, 2010.

[17] Dominic Deslandes and Ke Wu. Integrated microstrip and rectangular waveguide in planarform. IEEE Microwave and Wireless Components Letters, 11(2):68–70, 2001.

[18] ESA. RO4000® Series High Frequency Circuit Materials. https://www.esa.int/Our_

Activities/Telecommunications_Integrated_Applications/Satellite_frequency_

bands, 2018.

[19] Eurocircuits GmbH. Online PCB prototype and small series specialist.

[20] Eurocircuits GmbH. PCB design guidelines. https://www.eurocircuits.com/

pcb-design-guidelines/, 2018.

[21] Fabory. Cilinderschroef met zaaggleuf DIN 84 Messing CU2/CU3 M1X8.https://www.fabory.com/nl-BE/nl-BE/bevestigingsartikelen/schroeven/

cilinderschroef-met-zaaggleuf-din-84-messing-cu2-cu3-m1x8/p/48020010008,2017.

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https://www.rogerscorp.com/documents/726/acm/RO4000-Laminates---Data-sheet.pdf

https://www.esa.int/Our_Activities/Telecommunications_Integrated_Applications/Satellite_frequency_bands



https://www.eurocircuits.com/pcb-design-guidelines/

https://www.eurocircuits.com/pcb-design-guidelines/

https://www.fabory.com/nl-BE/nl-BE/bevestigingsartikelen/schroeven/cilinderschroef-met-zaaggleuf-din-84-messing-cu2-cu3-m1x8/p/48020010008

https://www.fabory.com/nl-BE/nl-BE/bevestigingsartikelen/schroeven/cilinderschroef-met-zaaggleuf-din-84-messing-cu2-cu3-m1x8/p/48020010008

BIBLIOGRAPHY III

[22] Fabory. Zeskantmoer DIN 934 (1987) Messing CU2/CU3 Rechts M1. https://www.

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