>TMTT Manuscript 2959 1

Coherent Optical Vector Modulation for Fiber Radio Using Electro-Optic Microchip Lasers

Yifei Li, Maja Bystrom, Member, IEEE, David Yoo, Student Member, IEEE, Samuel M. Goldwasser, Member, IEEE, and Peter Herczfeld, Fellow, IEEE

Abstract—Future communication systems will require high

data rates and flexible modulation. Direct optical phase modulation of two microchip lasers by information-bearing signals allows for high-rate delivery via fiber to a basestation. At the basestation the coherent optical signals are combined with a reference in a photodetector to produce a microwave/millimeter wave carrier with arbitrary M-ary quadrature amplitude modulation which can then be transmitted over a wireless channel. Rapid tuning of the microwave/millimeter wave carrier, the modulation scheme, and the data rate is achievable through this method with no fixed oscillators at the basestation thus providing for flexible architectures. Results show a high-quality carrier and for 4- and 16-QAM with data rates to 200 Mbps. Extensions to higher data rates are discussed.

Index Terms— broad band wireless, coherent fiber optic link, fiber radio, optical vector modulation.

I. INTRODUCTION he next generations of broadband wireless local area networks (WLANs) will require wide frequency bands

and efficient signaling techniques in order to support wireless bridges and to supply high data rates to fixed or mobile users. For applications such as high-rate voice and video transmission, capacities per user on the order of hundreds of Mbps or even Gbps will be required [1,2]. To support the anticipated growth in usage and ubiquitousness of computing access, as well as these high-rate applications, provisions are being made worldwide to operate with carriers in the tens of GHz [3,4,5]. , Whether local area or metropolitan area, these networks will demand high-bandwidth delivery to the distribution points, such as by the fiber distribution system for delivery of millimeter-wave subcarriers to radio base stations as proposed by Novak, et al. [6], and Ogawa, et al. [7]. In addition, flexible digital communication systems will be needed to support variable-rate data and channel-adaptive forward error control coding and power adjustment. This implies low-cost, bandwidth-efficient, frequency-agile

modulation of microwave or millimeter-wave carriers, particularly for downlinks.

Manuscript received January 21, 2005. This work was supported in part by

the Office of Naval Research under grant N00014-00-1-0781. Y. Li, D. Yoo, S. Goldwasser and P. Herczfeld are with the Center for

Microwave / Lightwave Engineering, Drexel Univ, Philadelphia, PA 19104 phone: 215-895-2914; fax: 215-895-1092; e-mail: yifeili@io.ece.drexel.edu.

M. Bystrom, is with the Information Systems and Sciences Laboratory, Boston Univ, Boston, MA USA; e-mail: bystrom@bu.edu.

M-ary quadrature amplitude modulation (M-QAM) is an ideal signaling strategy for many of the high-carrier, high-rate applications indicated above. Indeed, as an example, 16- and 64-QAM have been adopted by the IEEE in the new 802.16 standard for carriers in the 10-66 GHz range [8]. The feasibility of distributing 256-QAM signals over fiber using forward error control block codes to compensate for signal attenuation was demonstrated by Ohtsuka, et al. [9], thereby illustrating the potential for delivery of millimeter-wave signals to basestations via fiber.

Conventional high-frequency communications systems generate a QAM signal by first selecting an intermediate frequency (IF) and then upconverting, possibly in multiple stages, to the millimeter-wave band. However, it is not yet cost effective to implement highly-stable, high frequency oscillators, thereby motivating work in both direct generation of QAM signals at the millimeter-wave subcarrier frequency as well as simultaneous transmission of carriers for purpose of avoiding use of oscillators during demodulation [10,11]. There have been significant results in direct modulation using two or more external Mach-Zehnder modulators (MZMs). Recently, Jemison, et al. demonstrated the generation of 4- and 16-QAM at slightly less than 1MSps through varying the bias of two MZMs [12]. The MZM outputs are then combined and a photodetector is used to generate the modulated subcarrier. A similar approach using a pair of external modulators was taken by Candelas, et al. [13]; however, in this proposed approach two fibers, one from the output of each MZM, were required for delivery of the modulated signal to the basestation, where each quadrature component is separately detected and then electrically combined to produce the modulated millimeter-wave subcarrier.

In this paper a method for direct laser modulation to yield QAM at millimeter-wave frequencies is described for purposes of supplying bandwidth-efficient, rate- and carrier-adaptable digital modulation for wireless networks and short wireless bridges. This new approach, made possible by the use of electro-optically tunable microchip lasers, satisfies the requirements stated above. The proposed technique is purely optical QAM generation of a millimeter-wave subcarrier, requiring no external modulators. The potential subcarrier range is wide, from the hundreds of MHz to the tens of GHz and is limited only by the availability of photodetectors for the selected subcarrier frequency; these are commercially

T

>TMTT Manuscript 2959 2

available beyond 60 GHz. Due to the use of rapidly-tunable electro-optic microchip lasers, the rapid and straightforward variation of the carrier frequency as well as adjustable power levels and QAM constellation points, makes this system ideal for adaptable communications networks. Prior experiments [14] utilized less optimal components, particularly the baseband sources and the driving circuits, thus limiting the modulation to 4-QAM at lower data rates (<10MHz) and resulting in significantly more spread in the constellation. In the following, the proposed optical vector modulation (OVM) system is described in detail with emphasis in identifying the performance potential, and results (both theoretical and experimental) are shown for selected QAM modulation orders. It is shown that high-quality, high-rate (on the order of 200 MSps) modulation is readily achieved at a range of carrier frequencies, with demonstrated results limited only by available test equipment. Straightforward extensions to higher data rates are discussed. Furthermore, the output of each laser is a phase-modulated optical carrier with high spectral quality, so that this modulation strategy can potentially implement a coherent optical communication system.

II. OPTICAL VECTOR MODULATION CONCEPT In this section, we address the generation of optical vector

modulation for QAM fiber radio using photonic down-conversion. This method uses direct modulation of single-frequency microchip lasers (Figure 1).

Each microchip laser contains an intracavity electro-optic frequency tuning element [15]. Laser 1 and laser 2 are locked to the same optical frequency but with a 90 degree phase offset, through the use of an optical phase locked loop (OPLL). Because each microchip laser (µL) is frequency modulated, differentiators are used to obtain phase modulation.

I modulation

Q modulation

OPLL 1

ωm ω0-ωm

ω0

ω0

OPLL 2

PD

µL1

µL2

µL3

OVM

d/dt

d/dt

Fig.1. A block diagram of optical vector modulation scheme for fiber-radio using three microchip (µL) lasers and optical PLLs.

In this paper, a simple RC differentiator is assumed. In the

regime where the baseband signal frequency ω << 1/RC, the output of the differentiator will approach

( )tIdtdRC and ( )tQ

dtdRC ,

where I(t) and Q(t) are the baseband in-phase and quadrature information-bearing signals, respectively. Therefore, (neglecting initial phase offsets and noise terms) the optical outputs of lasers 1 and 2 will be

( )( )

0

0

Laser 1: cos 2

Laser 2: - sin 2

A t RC I t

A t RC Q t

ω πβ

ω πβ

⎡⋅ +⎣⎡ ⎤⋅ +⎣ ⎦

⎤⎦ (2.1)

where A is the optical field amplitude (assumed to be equal in both lasers), ω0 is the optical frequency, β is the microchip laser modulation sensitivity, and I(t) and Q(t) are the information-bearing signals for the orthogonal in-phase and quadrature carriers. Laser 3 is locked to an offset of the optical frequency of laser 2, equal to a desired microwave or millimeter-wave frequency corresponding to the carrier in the wireless domain. Thus, the output of laser 3 is simply:

( )0Laser 3: - sin mB tω ω−⎡ ⎤⎣ ⎦

where B is the optical field amplitude of laser 3. The three optical signals are launched into the fiber and

their sum is incident at a photodetector at the basestation. By using a small angle approximation to convert the angular information into amplitudes, a QAM representation can be established, and the photodetector output current can be written as:

( ) ( )( )( )( )

1 2 cos

1 2 sin

rec PD m

PD m

s t R AB RCI t t

R AB RCQ t t

πβ ω

πβ ω

= ⋅ ⋅ +⎡ ⎤⎣ ⎦

+ ⋅ ⋅ − ⋅⎡ ⎤⎣ ⎦

(2.2)

where RPD is the responsivity of the photodetector.

Since this output is amplitude modulation of two signals in quadrature, this scheme is analogous to traditional QAM architectures. However, note the presence of an additional carrier term, not normally seen in QAM systems. This carrier is advantageous when designing a coherent receiver, as it can function as a pilot signal to injection lock a local oscillator. Alternatively, if this carrier is suppressed, the output current reverts to conventional QAM, and may be sent to legacy digital processing sub-systems. An additional advantage to this method of generation is that by varying either the information-bearing modulating signals or the power of the optical carrier, it is straightforward to adaptively modify the signaling constellation based on channel knowledge and other system conditions. At the same time, the wireless carrier frequency can be varied by changing ωm for frequency agile systems.

Compared with other RF over fiber systems, by imposing a baseband signal directly onto the phases of the two orthogonal optical carriers, this eliminates the need for any intermediate frequency (IF) signal. This elimination enables us to attain much higher and adaptable data rates. Flexible carrier frequencies are also obtained since the QAM is initially generated in optical domain and later down-converted though laser heterodyning. In addition, the proposed scheme employs direct laser modulation, thus removes the need for external optical modulator, which could be an added complexity and a source for link loss. Furthermore, it is important to note that although the specific example presented here is an optical

>TMTT Manuscript 2959 3

QAM system, the specific modulation scheme and therefore, constellation, is arbitrary. The larger, more general focus of this work is demonstration of optical vector modulation. In particular, the QAM system here is facilitated by the use of the small angle approximation to reduce the equations to an easily recognizable form. This need not be the case; use of a lookup table in the transmitter end can eliminate the need for such an approximation. A variety of technologies can be used to implement a high speed lookup table-based QAM driver. Data rates up to at least several hundred MSps can be achieved using readily available commercial digital memory ICs such as double data rate static random access memory (DDR SRAM) with separate digital-to-analog converter (DAC), or a video random access memory digital-to-analog converter (RAMDAC) which combines one or more high speed SRAM-based lookup tables with DACs in a single chip. Straightforward design techniques should enable multiple GSps systems to be developed.

III. ANALYSIS In this section, the performance of the OVM architecture

will be analyzed. First, the phase locking subsystem used to generate stable carriers in quadrature will be presented. Secondly, the noise and distortion terms will be examined to determine the effects on the transmitted signal. Finally, the ultimate theoretical performance in terms of bit error rate (BER) and data rate of this approach will be discussed.

A. Phase locking subsystem As seen in Figure 1, two OPLLs are present in the OVM-

QAM transmitter. OPLL 2 is similar to a digital microwave frequency synthesizer and was addressed in [16]. Such OPLLs have also been demonstrated by various groups – for example, see [17, 18]. OPLL 1 is an optoelectronic variant of an analog phase locked loop (PLL) as illustrated in Figure 2. In this system a photodetector is used as an optical phase detector. Since each microchip laser contains an electro-optic frequency tuning element, it takes the place of a voltage controlled oscillator in an electronic PLL.

µL 1

µL 2

PD

Loop Filter

ω0

Σ-Vbias

ω0

Fig. 2. A block diagram of the optical phased locked loop (OPLL).

This structure forces the two lasers to maintain a quadrature phase offset. Writing the output of each laser as:

( ) 2 1,i ,cos 0 =+ itA θω yields an output voltage at the photodetector:

dbiasbiasPD VVV θcos+=

where , in which Z2bias PD PDV R Z A= PD is the photodetector

impedance and . The constant is removed by applying a bias voltage of –V

12 θθθ −=d

bias leaving the error signal applied to the loop filter . Since the OPLL acts to force

dbiaserr VV θcos=

( ) 0=∞→tVerr , in steady state we have 2/πθ =d .

B. I / Q Interference and Noise At this point, noise effects and the in-phase and quadrature

interference are considered. The downlink employs optical fiber, therefore fading and interference in the wireless channels will be neglected. Modulating the phases of the two optical carriers may introduce coupling or interference between the I and Q channels. Such interference is nonlinear in nature, and intensifies as the baseband modulation index is increased. Reducing the modulation index minimizes the I / Q interference, but reduces the signal to noise ratio (SNR) and increases the bit error rate (BER).

Assuming lasers 1 and 2 acquire identical amplitudes and frequencies but have quadrature phase offset, the combined output of the three-laser array is given by:

( ) [ ]

[ ][ ]

1 0 1 1

2 0 2 2

3 0 3

1 ( ) cos ( ) ( )

1 ( ) sin ( ) ( )

1 ( ) sin ( ) ( )m

E t A n t t t t

A n t t t t

B n t t t

ω θ φ

ω θ φ

ω ω φ

⎡ ⎤= ⋅ + ⋅ + +⎣ ⎦⎡ ⎤− ⋅ + ⋅ + +⎣ ⎦⎡ ⎤− ⋅ + ⋅ − ⋅ +⎣ ⎦

(3.1)

where and are the frequency-dependent amplitude

and phase noise of the i

)(tni )(tiφth laser (i=1,2,3), and iθ (i=1,2)

represents the I and Q phase modulation as in Eq. 2.1. Photomixing of the combined outputs yields the following

photocurrent in the microwave range:

( ) [ ][ ]

1 3 1 1 3

2 3 2 2 3

( ) 1 ( ) ( ) sin ( ) ( ) ( )

1 ( ) ( ) cos ( ) ( ) ( )

rec ex PD m

PD m

s t n t R AB n t n t t t t t

R AB n t n t t t t t

ω θ φ φ

ω θ φ φ

⎡ ⎤= − ⋅ ⋅ + + ⋅ − − − +⎣ ⎦⎡ ⎤+ ⋅ ⋅ + + ⋅ − − − +⎣ ⎦

(3.2) where the bandpass Gaussian noise, , is the noise current induced by receiver thermal and shot noise. Re-organizing the terms and assuming

)(tnex

1θ and 2θ are small values yields an approximation to the received current which can be written as:

( ) 1 2

2 1

cos( ) [sin ( ) cos ( ) ( )]sin( ) [ sin ( ) cos ( ) ( )]

rec PD m I

PD m Q

s t R AB t t t n tR AB t t t n t

ω θ θω θ θ

= ⋅ ⋅ ⋅ + +

+ ⋅ ⋅ ⋅ − + + (3.3)

The terms and are the total noise for the I and

Q channels, respectively, and can be represented as

( )In t ( )Qn t

( )

2 3 3 1( )( ) ( ) ( ) ( ) ( )

Iex

IPD

n tn t n t n t t tR AB

φ φ= + + − +⋅

(3.4a)

( )

1 3 3 2( )

( ) ( ) ( ) ( ) ( )Q

exQ

PD

n tn t n t n t t t

R ABφ φ= − + + + −

⋅ (3.4b)

where and are the in-phase and quadrature-( ) ( )I

exn t ( ) ( )Qexn t

>TMTT Manuscript 2959 4

phase components of . For white Gaussian noise, we

have < >= and < >= . Note that Eq. 3.3 reduces to Eq. 2.2 with additional noise terms.

( )exn t2

exn ><2)( I

exn 2exn ><

2)(Qexn

The sources for I / Q channel interference and noises are clearly identified in Eq. 3.3. The I / Q channel interference is deterministic and can be reduced using post-compensation. Thus, the noise sources are the limiting factors for the overall system performance. In the OVM system, the I and Q channels are symmetric. Therefore, in the following discussion only the I channel will be addressed. According to Eq. 3.4, the I channel noise is a combination of laser amplitude (AM) and phase noises, as well as electronic thermal and shot noises.

The spectral shape of the laser amplitude noise is determined by the dynamics of the photon density inside the laser cavity. For solid state microchip lasers, the AM noise peaks at the laser relaxation oscillation frequencies (~1MHz), beyond which the noise gradually reduces to the quantum noise floor (~-150dB/ Hz ). In the OVM implementation, a relative intensity noise (RIN) suppression feedback loop is inserted for each laser element in order to significantly reduce the AM noise near the relaxation oscillation. In principle, by using an ideal RIN noise feedback, AM noise close to the quantum noise floor is attainable. Thus, for simplicity, the frequency dependence of the laser AM noise spectrum may be neglected, and the quantum noise floor is used in its place.

As shown in Figure 1, the OPLLs lock the relative phases (

21 φφ − ,31 φφ − ) between the pairs of lasers, so that within the

PLL loop bandwidth the power spectral density of the relative phases are reduced. However, the PLLs do not distinguish between phase noise and the phase modulation induced by baseband signals. As a result, the baseband signal within the PLL bandwidth is also suppressed. Ideally, the baseband signal should have as little energy as possible inside the PLL loop bandwidth. However, outside the loop bandwidth, the outputs from the lasers become uncorrelated and their relative phase differences are solely determined by the laser intrinsic phase noise. Therefore, external feedback does not alleviate phase noise within the signal band, where the laser intrinsic phase noise dominates. Fortunately, solid state microchip lasers have low intrinsic phase noise within the signal band.

Away from the optical carrier (>1MHz offset), the intrinsic phase noise of the microchip lasers is quantum noise limited and has a Lorentzian shape [19] with a slope of -20dB/dec. By measuring the phase noise of the beat tone signal between two identical microchip lasers, the optical phase noise at a 1MHz offset is found to be -110 dBc/Hz. The integrated phase noise can therefore be calculated as:

( ) )11(log1044 102

31MaxMin

nn

ff−⋅+−=>−< φφ dB rad

where and are the lower and the upper frequency limits (in MHz) for integration and are determined by the frequency band of the baseband signal.

Minf Maxf

Using Eq. 3.4, the root mean square (RMS) noise in the unit

of radian versus the channel bandwidth under a typical operation condition is calculated and plotted in Figure 3. It is assumed that the photodiode has responsivity of 0.6 mA/mW, a termination resistance of 50 ohm, and sees equal power (1mW) from each laser. When calculating phase noise, the lower end of the baseband signal is assumed to be 1MHz and thereby the integration of the noise starts from fMin=1MHz. As shown in Figure 3 the laser phase noise dominates the other noise sources. The total rms phase noise is not strongly correlated with channel bandwidth and is found to be approximately 0.006 rad. The dominance of the phase noise is expected, because it is the optical intrinsic phase noise that affects the system performance. In addition, since the optical phase noise rolls off at a rate of -20dB/dec, its effect can be significantly reduced when the lower end of the baseband signal is increased (see Figure 4). As is increased from 1MHz to 100MHz, the RMS noise contributed by optical phase noise is reduced from 0.006 to 0.0005. When

, the laser AM noise replaces the phase noise as the dominant noise source.

Minf

70MHzMinf >

0 50 100 150 20010

-5

10-4

10-3

10-2

Receiver bandwdith: MHz

I cha

nnel

RM

S n

oise

in ra

d Thermal noiseShot noiseLaser RIN Laser phase noise

Fig. 3. The rms noise (integrated from 1MHz) in the I channel vs. receiver bandwidth. The optical power fed into the photodiode from each laser is 1 mW, and the termination resistance and responsivity of photodiode are 50 ohm and 0.6mA/mW, respectively.

0 20 40 60 80 10010

-4

10-3

10-2

fmin: MHz

I cha

nnel

RM

S n

oise

in ra

d Thermal noiseShot noiseLaser RINLaser phase noiseTotal noise

Fig. 4. The rms noise vs. Minf for a system with 200MHz receiver bandwidth.

C. Performance potential In the previous subsection, it was found that the noise

sources (especially the optical phase noise) degraded the performance. The achievable performance of the OVM will now be discussed. For M-QAM the average in-phase or

>TMTT Manuscript 2959 5

quadrature energy is given [20] as:

21 ( 1)3av d sMξ θ τ= ⋅ − ⋅ ⋅ (3.7)

where 2 /[2(d p pRCV Mθ πβ −= 1)]− is half the distance

between adjacent in-phase or quadrature constellation points,

sτ is the symbol duration ,V is the peak to peak voltage of

baseband signal, and is the time constant for the differentiator. To assure differentiation, the following relationship must be satisfied:

p p−

RC

sRC τ<< . In practice, we let 10/sRC τ= . Accordingly, the average signal-to-noise ratio

(SNR) per symbol is:

( )( )

2

2 2

0.1 21 ( 1)12 1

s p pav

I s I

VSNR M

n M n

π β τξτ

−⋅ ⋅ ⋅ ⋅= = ⋅ − ⋅

< > ⋅ − ⋅ < > (3.8)

For Gray coding, where adjacent in-phase or quadrature

symbols differ by only one bit, the bit error rate can be calculated with the formula [20] for a Gaussian noise assumption,

2

2( 1) / 3log ( ) 1M MBER erfc SNR

M M⎛−

≈ ⋅ ⎜⎜ ⎟−⎝ ⎠

⎞⋅ ⎟ (3.9)

0 50 100 150 20015

20

25

30

35

40

45

50

Symbol rate: Msps

SNR

per

sym

bol:

dB

4-QAM16-QAM64-QAM256-QAM

0 200 400 600 800 1000

10-8

10-6

10-4

10-2

Symbol rate: Msps

Bit

erro

r rat

e

4-QAM16-QAM64-QAM256-QAM

Fig. 5. SNR (a) and BER (b) vs. symbol rates, assuming Vp-p=2V, β=20MHz/V, and the remaining parameters are identical to those of Fig. 3.

Using Eqs. 3.8 and 3.9, the SNR per symbol and BER were

calculated (see Figure 5). It was assumed that the Vp-p was 2V

and the laser FM sensitivity was 20MHz/V, which is the measured value for the 1mm thick laser crystal used in the experiments. The remaining laser parameters were identical to those in Figure 3. According to the previous discussion, for fixed baseband voltage, as the symbol rate is increased, the SNR per symbol is reduced (see Figure 5a) even though the RMS noise power barely changed (see Figure 3). This is expected since when the data rate is increased, the symbol duration will be reduced (RC time of the differentiator is reduced). For a fixed baseband voltage, according to Eq. 3.7 and the following discussion, the optical phase change (or QAM symbol separation) is consequently reduced, thus resulting in reduced SNR. This represents the fundamental limitation of the OVM for high data rate operation; however much higher data rates are achievable if the laser FM sensitivity is raised. The FM sensitivity is inversely proportional to the thickness the E-O section. If the laser is implemented in an integrated form with the waveguide thickness of the E-O section equal around 5 µm [20], the FM sensitivity will be several GHz/V. This is consistent with sensitivities of off-the-shelf Mach-Zehnder modulators. Multi-GHz/V modulation sensitivity thereby enables data rates in the tens of Gbps or higher.

In addition, according to Figure 5b, for a fixed BER, the lower order QAMs (4-QAM or 16-QAM) are more suitable for higher data rate operation. For instance, at a BER of 10-6, 4-QAM could operate at over 1100Mbps; while 64-QAM could only reach 500Mbps. This occurs because the system is limited by the optical phase noise, and the noise power is not a strong function of bandwidth. However, the situation will reverse (i.e. higher order QAM is preferred) if the baseband signal starts from higher frequencies and the white amplitude noise becomes the limiting factor.

(a) IV. EXPERIMENTAL VERIFICATIONS

In this section, the experimental results generating QAM for transmission over a microwave / fiber link are discussed. The experimental setup is depicted in Figure 6.

I modulation

PD OPLL2

ω0-ωm

ω0

Vector signal analyzer

Synthesizer

10MHz Ref

Spectrum analyzer

Communication signal analyzer

Carrier suppression

PD

ω0

OPLL1

µL1 Q

modulation

Phase shifter

Microchip laser module (µL)

µL2

µL3

d/dt

d/dt

Optical fiber

Optical vector modulator

RIN noise suppression

Tuning voltage

Pump diode

Laser crystal

(b)

Fig. 6 Experiment setup for testing OVM (Detailed block diagram of electro-optic microchip laser module shown in inset).

An array of three electro-optic (E/O) tunable microchip

>TMTT Manuscript 2959 6

lasers was employed. The E/O microchip lasers, fabricated by Casix, Inc., consisted of a 0.5mm gain section (Nd:YVO4) and a 1.0 mm electro-optic tuning section (MgO:LiNbO3), which were optically bonded for ruggedness and to facilitate low cost mass production. Two dielectric mirrors were deposited on each side of the laser crystal to form a plano-plano resonator. Each laser was butt-coupled to 808 nm pump diodes and operated in the 1064 nm wavelength, single frequency, and single transverse mode regime. The electro-optic microchip lasers had low threshold (<50mW) and high slope efficiency (>50%), with a maximum single mode output power of around 100mW and FM modulation sensitivity of 20MHz/volt. The fiber attenuation for 1064nm wavelength is around 1.5dB/km, which is usable for fiber radio applications where the fiber link is less than 10km. However, for longer links, a lasing wavelength at 1340 nm is preferred. By changing the laser mirror coatings, the microchip lasers can operate at 1342nm. The selection of 1064nm lasing wavelength for this experimental setup is just a matter of convenience.

The subcarrier frequency was set by changing the frequency difference between the lasers using OPLL2. The tuning range varied from 0.5 GHz to 40GHz (limited by the photodiode used). The loop bandwidths of both OPLLs were adjusted to 1MHz to minimize distortion to the baseband signals and to assure low phase noise of the microwave carrier.

At the receiver, the microwave optical subcarrier was converted to a microwave signal by a high speed photodiode. A communication signal analyzer (Tektronix CSA803) and a microwave spectrum analyzer (Agilent 8564E) monitored the time domain waveform and the signal spectrum. Ideally, a QAM demodulator specifically tailored to the requirements of the OVM scheme is needed to realize the full potential of the proposed system. However, since the main emphasis of this experiment was in generating QAM, a commercial unit (Agilent vector signal analyzer E8408A) was used for demodulation, which limited the maximum modulation index. Since the signal analyzer (VSA) required a conventional QAM input, with negligible carrier components, the carrier had to be suppressed. Unfortunately, carrier suppression introduces a small, but noticeable phase fluctuation that tends to degrade the demodulator (VSA) performance.

Fig. 7 Microwave carrier spectrum at 2.4GHz

A clean microwave carrier signal is vital for basestation implementation of a fiber radio system. Therefore, prior to discussion of the QAM results, the quality of the carrier signal is considered. The 2.4 GHz microwave carrier spectrum is depicted (in Figure 7). The measured phase noise was -95dBc/Hz at a 10kHz offset. The choice of the 2.4 GHz carrier frequency was simply a matter of convenience since QAM demodulation was performed by the VSA, which had a detection range below 2.5GHz. By setting the frequency difference between lasers 1 and 3, microwave carriers up to 40GHz can easily be generated [14].

Fig. 8 AM noise measurements for a 2.4 GHz carrier signal with and without noise suppression.

The AM noise was measured with and without AM noise

feedback control (Figure 8). Without the feedback, two noise spurs caused by the relaxation oscillations of lasers 1 and 2 (at 2MHz and 3MHz) were observed in the AM noise spectrum. With the feedback enabled the spurs disappear. It is seen that significant AM noise exists at frequency offsets below 1.8MHz; which is caused by the phase-to-amplitude noise conversion that occurs when the two quadrature carrier signals are combined. We conclude that the OVM scheme generates microwave carriers with low phase and amplitude noise.

(a)

(b)

Fig. 9 4-QAM (a) signal constellation and (b) error vector measurements.

>TMTT Manuscript 2959 7

Next, the quality of the QAM modulation is considered.

First, the results of the simplest scheme, 4-QAM, is discussed and presented. In the OVM scheme, 4-QAM is readily realized when two binary NRZ baseband signals modulate the phases of lasers 1 and 2. In the experiment, the baseband signals were taken from the de-serialized outputs of a digital pattern generator (Advantest D3173A). The baseband peak-to-peak voltage (Vp-p) was set to 2V. The RC time constant of the differentiation stage was set to 1/10 of the symbol period. The microwave subcarrier was tuned to 2.4GHz and the 4-QAM symbol rate was set to 30MSps. In addition, to minimize the signal distortion due to the phase locked loops, the baseband signals were programmed to have few spectral components below 1MHz. The signal constellation and the error vector measurements were generated by the VSA and are shown in Figure 9. The magnitude of the error vector was around 1.5% (~SNR per symbol: 36dB) consistent with the theoretical calculation in Section III.c). The very tight constellation is indicative of high-quality 4-QAM.

Fig. 10 (a) I channel eye diagram (b) microwave spectrum. 4-QAM with a 100MSps symbol rate was also generated

and evaluated. In this test, the peak-to-peak voltage, Vp-p, of the baseband signal remained 2V and the clock rate of the baseband sources was increased to 100MHz. In addition, the baseband data was a pseudorandom sequence with length (27-1). The RC time constant of the differentiation stage was also accordingly reduced to assure good differentiation. The 100MSps rate was beyond the range of the VSA. Therefore, the experimental results are presented in the time and frequency domains. At the receiver, the I / Q waveforms were recovered by the Tektronix communication signal analyzer. Only the I – “eye diagram” is shown (Figure 10a) since the I / Q channels are symmetric. The “eye” is wide open and suggests good signal quality (Figure 10a). It is noted that the lower rail of the eye diagram is wider than the upper rail. This

is attributed to signal distortion due to the finite bandwidth of the phase locked loops and to the fact that the baseband signal was not completely random. Such distortion can be greatly reduced by limiting the baseband signal energy within the PLL loop bandwidth. In addition, the microwave spectrum for the 100MSps 4-QAM modulated signal is shown in Figure 10b. As expected, a 2.4 GHz carrier coexists with the 4-QAM spectrum.

16-QAM modulation for a 2.4GHz subcarrier frequency at a 25MSps symbol rate (once again limited by the bandwidth of the VSA) was also successfully carried out. In this test, laser 1 and laser 2 were modulated by two independent quaternary baseband signals taken from the baseband output of a vector signal generator (Agilent E-4438C). The Vp-p of the baseband signals was set to be 2V. As for 4-QAM at 30MSps, the baseband waveforms were carefully programmed to minimize the spectral components within the PLL loop bandwidth. The measured signal constellation, plotted in Figure 11, appears satisfactory aside from a slight spread of the constellation points. This is a consequence of the phase instability inside the carrier suppression subsystem.

(a)

(b)

Fig. 11 16-QAM constellation Employing M-ary (M>4) baseband signals in the OVM

scheme, other QAM modulation orders (e.g., 64-QAM, etc.) can be similarly achieved. Furthermore, any vector modulation scheme can be implemented with OVM using proper baseband signaling.

Although the proposed OVM scheme is able to transmit high speed data over a single channel, it is sometimes desirable for a microwave / fiber optic link to transmit data over multiple frequency channels with low data rates. An attractive approach, currently used by most fiber radio links, is IF subcarrier modulation (IF/SCM), where IF signals modulate the microwave optical subcarrier. The transmitter in the OVM scheme is compatible with IF/SCM. In Figure 6, if the modulation to laser 2 is removed and the baseband signal to laser 1 is replaced with an IF signal, IF/SCM will be generated. Compared with other IF/SCM methods, the use of the OVM transmitter has the benefit of directly generating widely tunable microwave carrier frequencies, while being low cost (by eliminating the expensive optical modulator for generating high data rate and high carrier frequencies). To verify this backward compatibility, a +3 dBm, 8 MSps 16-

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QAM modulated IF signal at 40MHz (taken from an Agilent vector signal generator) was used to modulate laser 1 while leaving laser 2 idle. Again, these experiment parameters were chosen based on available test equipment. The microwave spectrum of the resulting IF modulated microwave carrier was captured and is depicted in Figure 12a. A clean microwave carrier at 2.4GHz and two 16-QAM modulated sidebands at a 40MHz offset are clearly apparent. One of the sidebands was demodulated by the VSA, which captured the signal constellation (Figure 12b). Slight spread in the constellation points due to phase error was observed. It should be noted that although the IF modulation generates two sidebands, single sideband modulation can be obtained if laser 2 is modulated by the inverse of the IF signal instead of being left idle. Single sideband modulation is attractive in terms of increasing efficiency and reducing the impact from wireless channel dispersion.

Fig. 12 16-QAM subcarrier modulation. (a) microwave spectrum and (b) signal constellation.

V. CONCLUSION It is desirable for the next generation of WLAN systems to

be able to provide high data rates, potentially exceeding Gbps per channel, while limiting the receiver and transmitter cost. These requirements imply the need for very low-noise, ultra-wide bandwidth performance of source components and for simple transmitter and receiver design with a minimum number of components integrated into a compact form factor. An optical vector modulation scheme was introduced for fiber-radio and results were demonstrated for 4- and 16-QAM.

The experimental results illustrated generation of both a high-quality carrier (phase noise -95dBc/Hz @10 KHz off-set) and QAM signal, and showed a data rate of 200 Mbps for 4-QAM, Upgrading the electronics resident in the transmitter and receiver would raise the data rate to 1 Gbps or higher,

while thinning the laser crystal could further improve the rate to 5 Gbps. Finally, an integrated optic implementation of the laser would increase the modulation sensitivity by orders of magnitude, yielding potential data rates in the tens of Gbps.

We have shown that the proposed OVM system is ideal for generating adaptable QAM modulation on single or multiple microwave/millimeter-wave subcarriers. Since the modulated subcarriers are directly generated by heterodyning electro-optic microchip lasers, the system is frequency-agile, that is, the subcarrier frequency can be rapidly varied. This allows for applications such as rapid frequency hopping, which may be desired as a security measure in the next generation of networks. Finally, we note that although this paper is concerned with wireless communication, specifically fiber radio, the experiments also clearly demonstrate the viability of this approach for coherent optical communications

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