120

 

CHAPTER 4

BANDWIDTH ENHANCEMENT OF THE ANTENNA

One of the important characteristics of microstrip patch antennas is its impedance

bandwidth which can be significantly improved by using multilayer dielectric

configuration. To achieve this, the emphasis is mainly laid on bandwidth

enhancement techniques of a multilayer patch antenna, designed for X-band

applications. The conformal transformation of multidielectric microstrip antenna,

discussed in the last chapter, has been employed for performance analysis. Wheeler’s

transformation, employed to map the complex permittivity of a multilayer substrate to

a single layer, leads to a simpler approach of analysis.

In the design of microstrip patch antenna presented in this chapter, the antenna losses

are therefore contained by controlling those quality factors which can have significant

impact on bandwidth for given permittivity and substrate thickness. The gain

bandwidth product is a constant, therefore an effort has been made to improve the

bandwidth of the patch antenna while ensuring desired radiation pattern. Next, the

effect of cover layer on impedance matching, Q factor hence bandwidth and

frequency correction is discussed. The Method of Moments and Finite Difference

Time Domain approach have been used for computation of the results presented in the

chapter.

4.1 Need for Bandwidth Enhancement

Primary barriers to implement patch antennas in modern broadband communication

system applications are their narrow bandwidth. The microstrip antennas are often

realized with bandwidth of the order of 1% to 5%. Bandwidth enhancement

technique is one of the areas of research in the field of microstrip antennas. Basically

the bandwidth is defined more concisely as a percentage (f/ f0) × 100%, where f and f0

respectively represent the width of the range of acceptable frequencies and the

resonant frequency of the antenna. The parameters such as radiation efficiency, return

loss, and voltage standing wave ratio (VSWR) [88] are often used to define the

bandwidth of a microstrip antenna.

121

 

In literature a popular measure of bandwidth of an antenna, is determined in terms of

range of VSWR as shown below:

1 ≤ VSWR ≤ 2

For broadband antenna design the following considerations are necessary in antenna

geometry.

• Larger substrate thickness or lower permittivity of the dielectric to obtain low

Q.

• Feed impedance must be matched.

• Optimization of patch geometry.

• Suppression of surface waves in a thick substrate.

The impedance bandwidth of microstrip antennas can be improved by using the

substrate of lower dielectric constant or by increasing the substrate thickness. The

substrate thickness and lower permittivity increases surface wave resulting in low

gain, low efficiency of the antenna and distorts the radiated field pattern. Thus, as the

dielectric substrate height increases beyond 2% of the wavelength these effects

become significant [78]. In the case of single layer dielectric microstrip antenna, the impedance bandwidth

may be enhanced by increasing the thickness of the substrate but at the expense of

efficiency of the antenna due to surface wave factor. Alternately multilayer microstrip

antenna may be designed to achieve desired bandwidth or the quality factors (Q) at

the required frequency of operation [78].

Since it is easy to analyze a single dielectric microstrip antenna, parameters for a

single layer patch are obtained by keeping frequency and height of substrates same.

As discussed in Section 3.2.1 the effective permittivity of the transformed multilayer

microstrip antenna may be calculated using Wheeler’s transformation [89]. Using this

transformation, all the losses i.e. radiation and dielectric, conduction and surface wave

losses, and quality factors associated with it may therefore be calculated for

computing the impedance bandwidth. It is therefore necessary to consider the factors

that affect these losses. In the cavity model, the radiation pattern of the microstrip

antenna is modelled by an effective loss tangent.

122

 

EEjJtDH dσωε +=+∂∂

=×∇ … (4.1)

)1(]1[ EjjEjj d −=−= ωεωεσ

ωε                     ... (4.2)

Therefore losses associated with a microstrip antenna may be obtained by determining

effective loss tangent using a cavity model. The loss tangent, which represents

dielectric loss, may be defined as the ratio of the imaginary part to real part of the

dielectric constant. Following Maxwell’s equation may be used to determine the

complex dielectric constant, where ( )djδε −1 and dσ respectively represent the

complex dielectric constant and bulk conductivity of the dielectric medium.

ωεσ

δ dd ==

constant dielectriccomplex theofpart realconstant dielectriccomplex theofpart imaginary … (4.3)

dd

dre P

PPδδ

+= … (4.4)

Therefore the loss tangent may be expressed by dδ as given in equation (4.3). The

effective loss tangent δec that accounts for the power radiated from the microstrip

patch Pr and power dissipated in the dielectric substrate Pd may therefore be

expressed as shown in equation (4.4).

dd

cdrec P

PPPδδ

++= … (4.5)

Further, in order to account for the conductor (Ohmic) loss, the expression for the

effective loss tangent may be modified as given in equation (4.5), where Pc represents

the Ohmic loss in the patch conductor.

)}1({0 ecr jkk δε −= … (4.6)

The complex wave number k for a lossy medium is defined in terms of effective loss

tangent δec as given in equation (4.6).

dcrTec QQQQ

1111++==δ                                                                                      ... (4.7) 

The effective loss tangent δec is related to the total quality factor QT for the patch as

expressed in equation (4.7), where Qr, Qc and Qd respectively account for radiation,

conductor and dielectric losses.

123

 

SWdcrTec QQQQQ

11111+++==δ                      ... (4.8)  

To account for the surface wave loss, the expression for the total quality factor, given

by equation (4.7), may be expressed as shown in equation (4.8). 1

1111−

⎟⎟⎠

⎞⎜⎜⎝

⎛+++=

SWdcrT QQQQ

Q ... (4.9) 

The total quality factor is therefore given by equation (4.9).

QT = W {E/ ( SWcdr PPPP +++ )} ... (4.10)

The total quality factor QT may also be expressed in terms of the average energy

stored and average power loss per second as in equation (4.10), where W represents

the width of the cavity.

Qd = 2 π fr (E / Pd ) ... (4.11)

Quality factor Qd that accounts for the dielectric loss is given by equation (4.11). The

Quality factor Qd as expressed by the equation (4.11) depends on the Power Pd

dissipated (or absorbed) and the energy E stored in the substrate.

E = ∫∫ =VV

Zr dvHdvE 20

2

0 21

21 µεε ... (4.12)

∫∫ =AA

Zr daHhdaEh 20

2

0 22µεε                  ... (4.13)           

The expressions for E may be obtained in terms of expressions for the peak energy

stored by the electric or the magnetic field distribution within the substrate using the

equations (4.12) and (4.13), where Ez and H represent respectively the magnitude of

the electric field component in the z-direction and the magnetic field.

                                   … (4.14)

The power Pd dissipated in the substrate may be determined in terms of Ez, and h is

the height of the substrate.

d

rr

AZd

AZrr

df

daEh

daEhfQ

σεεπ

σ

εεπ0

2

2

0 2

2

22

==∴

∫

∫ … (4.15)

daEhdvEPA

ZdV

Zdd

22

221

∫∫ == σσ

124

 

The quality factor Qd may therefore be determined using equations (4.11), (4.12) and

(4.14) as given in equation (4.15).

Qc = 2πfr (E/Pc) … (4.16)

Qr = 2πfr (E/Prad) ... (4.17)

Similarly the Quality factor Qc that accounts for the conductor loss and the quality

factor Qr that accounts for the total power radiated, Prad, over the solid angle at a

distance r are respectively given by the expressions (4.16) and (4.17).

                                 … (4.18)

Using the expression for Energy stored E, the quality factor Qc may be obtained as

given in equation (4.18).

∫×=A

sc daJRP 2||212 = cr

s

r fhR

hfσµπ

µπ0

0 = … (4.19)

The conductor (Ohmic) loss in the conductor may be expressed in terms of Rs the

conductor resistance and J the current density as in equation (4.19), where

c

rs

fR

σµπ 0= . We choose h such that excitation of TE11 mode is avoided and there

is elimination of other higher order surface wave modes for a given operating or

resonant frequency fr of the patch.

     … (4.20)

⎟⎠⎞

⎜⎝⎛

⎟⎠⎞

⎜⎝⎛−

=b

yna

xmhbkVnj

H xπππωε

coscos20

00               … (4.21) 

… (4.22)

Energy stored and power radiated can be obtained from the knowledge of field

components. Ez, Hx, and Hy in the cavity corresponding to TMmn mode as given by

equations (4.20), (4.21) and (4.22) respectively, where Vo corresponds to the peak

value of V due to the field components.

daHR

daEhfQ

As

zrA

r

c 2

20

212

22

∫

∫

×=

εεπ

⎟⎠⎞

⎜⎝⎛

⎟⎠⎞

⎜⎝⎛=

byn

axm

hakVmj

H yπππωε

cossin20

00

.coscos0 ⎟⎠⎞

⎜⎝⎛

⎟⎠⎞

⎜⎝⎛=

byn

axm

hV

Ezππ

125

 

V=EZ L = ⎟⎠⎞

⎜⎝⎛

⎟⎠⎞

⎜⎝⎛

byn

axmV ππ coscos0                              … (4.23)

The boundary condition for the cavity model of Figure 4.1 may be used to determine

the components V(x) and V(y) using the expression in equation (4.20)  for Ez as in

equation (4.23).

… (4.24)

The components V(x) and V(y) may be determined by applying the boundary

conditions at the edges of Figure 4.1. Thus, the boundary conditions at the edge 1: x=

0 and 0 ≤y ≤ b, lead to equation (4.24).

Figure 4.1: Cavity model of the microstrip patch antenna

( ) ⎟⎠⎞

⎜⎝⎛=

bynmVyV ππ coscos)( 0 … (4.25)

Applying the boundary conditions at edge 2: x= a and 0≤ y≤ b, V(y) may be expressed

as in equation (4.25).

)cos()cos()( 0 axmnVxV ππ=               … (4.26)

Similarly with the boundary conditions at the edges 3 and 4 respectively

corresponding to y = 0 and 0≤ x≤ a and y = b and 0 ≤ x≤ a, we obtain )(xV .

⎟⎠⎞

⎜⎝⎛=

bynVyV πcos)( 0

126

 

E  ∫ ∫ ⎟⎠⎞

⎜⎝⎛

⎟⎠⎞

⎜⎝⎛=

a br dxdy

byn

axm

LV

0 0

222

00 coscos2

ππεε             … (4.27)  

The expression for the energy stored E given by equation (4.12) may now be

expressed as the volume integral over the limits along the length and the width of the

cavity, which is equation (4.27).   

E h

abVr

4

200εε

=                 ... (4.28)      

Further, the energy stored E in TM10 mode may be obtained as in equation (4.28).            

ϕθθπ

ϕ

π

θ

ddrPP rrad sin22/

0

2

0∫ ∫= =

= = eGV 202

12× … (4.29)

The total power radiated power Prad over the solid angle at a distance r may be

determined using the power radiated Pr from the microstrip patch as in equation

(4.29), where Ge is the transconductance of the patch..

e

rerr hG

abfQ

20εεπ

=

Using the expression given in equation (4.28), Qr given by equation (4.17) may be

written as above. 2

0901

⎟⎟⎠

⎞⎜⎜⎝

⎛=

λbGe for  1

0

<<λb

Ignoring radiation from the non-radiating edges and considering only the radiation

from the radiating edges, Ge may be expressed by the following empirical relation.

Taking 2

02

0re

2or and

22⎟⎠

⎞⎜⎝

⎛⎟⎠

⎞⎜⎝

⎛===c

ffca r

r

λλλ

ελ ,

2

0

00

9012

22

⎟⎟⎠

⎞⎜⎜⎝

⎛

⎟⎟⎠

⎞⎜⎜⎝

⎛

==

λ

εεπεεπ

bh

bfcf

hGabf

Q rrer

e

rerr                ...  (4.30) 

2

33

r

rer hbf

cQ εα⇒

127

 

Hence we obtain rQ as expressed in equation (4.30), on the basis of this equation it

may be inferred:

• As h increases Qr reduces due to increase in Pr.

• Similarly as b decreases Qr reduces due to increase in Pr. @This has been substantiated by the simulated results shown in Table 4.1, 4.2 and 4.3.

Computation of Q factors may repetitively be carried out for various standard

substrate heights and permittivity and data so obtained may be plotted and analyzed.

Suitable range of quality factors may be found by minimizing return loss. The gain,

directivity, and power radiated may then be determined. @The work reported in this chapter is based on the following research paper contributions [94]: Gupta S.D., Srivastava M.C., “Multilayer Microstrip Antenna Quality Factor Optimization for Bandwidth Enhancement”, Journal Of Engineering Science & Technology (JESTEC)” School Of Engineering, Taylor’s University, Malaysia, vol. 8, Issue 1, Feb 2013.

4.1.1 Measures for Impedance Bandwidth Improvement

Impedance bandwidth depends on the feed network design and optimization of the

quality factors (Q). There is a need to analyze individual quality factors and optimize

them to achieve a significant trade-off between gain and losses. Use of thick substrate

and low permittivity dielectric, enhances the antenna bandwidth. The losses in

dielectric are due to surface wave excitation resulting in lower gain and thereby

reducing the efficiency. Choosing a substrate with low loss tangent reduces dielectric

loss. Surface waves propagate from the patch downward to the substrate and are

reflected from the ground plane. More surface wave’s modes exist as substrate

becomes electrically thicker. A major drawback of these waves is it introduces

spurious coupling between antenna elements and it therefore necessitates suppression.

The microstrip antenna under consideration consists of a rectangular patch of width W

and length L with two dielectric layers εr1, εr2 and height h1, h2 respectively as shown

in Figure 4.2. With a cover layer introduced as shown in (Figure 3.1(b) Chapter 3)

there is a significant improvement on the performance indices of the microstrip

antenna. In various subsections of the present chapter, the effect of cover layer on

impedance matching, Q factor and hence bandwidth and frequency correction are

discussed.

128

 

Figure 4.2: Multilayer Microstrip Antenna

4.1.2 Role of Cover Layer on Impedance Matching, Quality Factor and

Bandwidth

The cover layer plays a significant role in impedance matching and affects the quality

factor and the bandwidth. The expressions for the affected impedance, quality factor

and bandwidth are therefore presented in the following subsections. Enhancement of

bandwidth is at the expense of gain, therefore an effort has been made to improve the

bandwidth of the patch antenna while ensuring desired radiation pattern with reduced

losses.

4.2 Effect on Impedance Matching

The input admittance or impedance of a microstrip antenna must be matched with 50

ohms feed line. Proper matching of the feed line is an important factor determining

the bandwidth of an antenna. The discontinuity between the line feed and an antenna

can be modelled by a shunt combination of conductance G and capacitance jB.

1

0

22

222 )2sin()(sin)(cos2)(

−

⎥⎦

⎤⎢⎣

⎡+

++= z

YBz

YBGzGzY

o

βββ … (4.31)

Input impedance at an offset z may therefore be expressed by an equivalent formula

[95], as given by equation (4.31). As discussed earlier, the bandwidth has been

considered to lie in the frequency range for which the VSWR ≤ 2. Antenna is

matched when the impedance locus is as close as possible to centre of Smith chart that

result in a low return loss at the resonant frequency. In order to obtain a large

bandwidth it is required to shift impedance locus inside bandwidth circle. A smoother

return loss over the frequency is desired instead of compromising with a minimum

return loss. Broad bandwidth of the antenna may be achieved with proper selection of

design parameters.

129

 

4.2.1 Effect on Quality Factors Quality factor determines the frequency selectivity as well as bandwidth of a

microstrip antenna. The efficiency of a microstrip antenna can be calculated from

cavity model in terms of its quality factors.

HWL

cpQ

e

e

r

rRad

0

1163 λε

= … (4.32)

A microstrip antenna has dielectric, conductor and surface wave losses and therefore

its quality factor for desired radiation into the space may be expressed by equation

(4.32), where Le and We are effective lengths and width of the patch accounting for

fringing length and width, the terms pr and c1 are constants [96].

⎟⎟⎠

⎞⎜⎜⎝

⎛−

= swr

swr

radsw eeQQ

1 ... (4.33)

Surface wave excitation represents a loss mechanism with associated quality factor

Qsw. For thin substrates the same can be neglected, as its contribution is negligible

towards the total quality factor. Assuming the surface is infinite, the quality factor

Qsw, may therefore be expressed in terms of Qrad and the radiation efficiency of the

patch ersw as given by equation (4.33).

( )2

211

0111

431

1

⎟⎟⎠

⎞⎜⎜⎝

⎛−⎟⎟

⎠

⎞⎜⎜⎝

⎛⎟⎠⎞

⎜⎝⎛+

=

nchk

e

r

swT

µπ ... (4.34)

The radiation efficiency of the patch that depends on Psp the power radiated into

space, Psw the power launched as the surface wave, and QT/QR, may be calculated by

neglecting surface wave loses QT/QSW and dielectric losses for thin substrates. It is

expressed as given by equation (4.34).

⎟⎟⎠

⎞⎜⎜⎝

⎛⎟⎠⎞

⎜⎝⎛=

s

orc R

hkQ

20ηµ … (4.35)

⎟⎠⎞

⎜⎝⎛≈

δtan1

dQ … (4.36)

Following [96] the quality factor Qc associated with conductor losses, may be

expressed in terms of Rs, average of the two resistances of ground plane and patch

metal, given by equation (4.35). Further dielectric loss Qd may be expressed in terms

of loss tangent given by equation (4.36).

130

 

Substrate thickness plays a crucial role in deciding the quality factor of the antenna.

With the increase in number of interfaces among the substrate, surface wave

excitation also increases thereby reducing the radiation efficiency. These waves also

cause coupling between small microstrip circuits and are very difficult to measure.

The contribution of loss mechanisms of a multilayer rectangular microstrip antenna

with respect to substrate thickness h, the individual quality factors and their

efficiencies evaluated for various values of dielectric constants are shown in Table 4.1

& Table 4.2. The calculations in Table 4.1 & Table 4.2 correspond to heights of both

the layers (h1, h2) being same and the antenna being resonant at frequency of 7.5GHz

by keeping S11 > -25 dB and VSWR between 1 and 1.5. The values in these tables

correspond to multilayer microstrip antenna first designed and then the antenna is

transformed to single layer using transformation technique to obtain the quality

factors and bandwidth.

Table 4.1: Quality factors and bandwidth of a multilayer

microstrip antenna for low permittivity.

Height(mm) effε Qsw Qrad QT BW (%)

3.0 2.04 21.36 4.75 3.8 4.07

1.5 2.112 40.2 6.34 5.42 5.82

0.787 2.126 154.6 12.69 10.7 2.98

0.508 2.144 345.7 19.66 16.23 2.12

0.254 2.158 735.5 38.812 29.3 2.09

0.127 2.171 1158.3 77.624 45.19 1.87

131

 

Table 4.2: Quality factors and bandwidth of a multilayer

microstrip antenna for high permittivity

Height(mm) effε Qsw Qrad QT BW (%)

3.0 6.702 19.23 8.7 5.5 4.86

1.5 6.81 39.87 14.8 10.71 4.45

0.787 6.956 169 27.7 18.13 3.12

0.508 7.09 311.32 40.3 24.79 2.28

0.254 7.281 678.117 90.72 52.55 2.32

0.127 7.444 1334.69 181.44 102.98 2.14

4.2.2 Effect on Radiation Efficiency and Surface Wave Losses

The radiation efficiency for low and high permittivity substrates are shown in Figure

4.3 and Figure 4.4 respectively.

Figure 4.3: Radiation efficiency in low permittivity substrates

132

 

Figure 4.4: Radiation efficiency in high permittivity substrates

As shown in these figures the radiation efficiency increases from 71% to 85% with

change in height from 0.127mm to 1.5mm. In the case of a low permittivity multilayer

microstrip antenna, the radiation efficiency increases with the height and after an

optimum point starts decreasing. With height greater than 1.5mm the contribution of

surface waves start dominating which reduces the bandwidth of the antenna. For

better clarity of the simulated results the radiation efficiency & surface wave losses

for low and high permittivity antenna are shown in Table 4.3. Table 4.3: Radiation efficiency and Surface wave loses for low and high permittivity microstrip

antenna

Height

(mm)

Low Permittivity High Permittivity

Radiation Efficiency

Surfacewave Losses

Radiation Efficiency

Surfacewave Losses

3.0 80% 17.79% 63.21% 28.6%

1.5 85.48% 13.48% 72.365% 26.86%

0.787 84.31% 6.92% 65.45% 10.72%

0.508 82.55% 4.69% 61.51% 7.965

0.254 81.82% 3.98% 57.92% 7.74%

0.127 71.33% 3.905% 56.755 7.765%

133

 

Figure 4.5 shows the percent contribution of surface waves and total bandwidth in a

low permittivity (2.04 ≤εeff ≤ 2.171) multilayer microstrip antenna with respect to

height. Since surface waves arise due to change in medium and die out along the

depth of the substrate, its contribution is very less (3-4 %) at low heights. The same

pattern is followed in the case of the bandwidth, but the losses go up to 17% at

increased height.

Figure 4.5: Losses due to surface waves in low permittivity substrates

Figure 4.6: Losses due to surface waves in high permittivity substrates

134

 

The surface wave losses for high permittivity cases are shown in Figure 4.6. The

pattern is similar to the one for low permittivity case. However the losses increase up

to 28% at a slower rate with height changing from 1.5 to 3mm. As seen from Figure

4.4 and Figure 4.6 for high permittivity dielectric, the bandwidth increases with

increase in the height with surface waves loses increasing up to 28% but radiation

efficiency decreasing to 63%.

Figure 4.7: Total Quality factor and Bandwidth

An optimum point of setting up of height and quality factor can be inferred from

Figure 4.7 which shows the crossover of bandwidth and total quality factor. Fixing of

quality factor above the cross over point will result in more losses. When the losses

are low but not lowest the radiation efficiency is also found to be optimum. For low

permittivity substrates the intersection is approximately at 1.5mm hence optimum

bandwidth of 5.8% is obtained in this region with the quality factors set above the

cross over points. Whereas for high permittivity structures the substrate thickness

should be close to 0.5mm or more than 3mm as the contribution of quality factor

through conductor may be neglected in view of its being very low compared to

contribution due to dielectric losses.

135

 

4.3 Simulation Results for Multidielectric Layer Patch Antenna

Parameters based on Method of Moments

Designed antenna is simulated using Momentum (ADS) software which is based on

Method of Moments. In the following subsection simulation results for determining

parameters and characteristics of multidielectric layer antenna have been discussed.

The results have subsequently been compared with those obtained after

transformation of the dielectric antenna using conformal mapping technique.

4.3.1 Antenna Parameters for Multidielectric Layer Patch

Simple multilayer microstrip patch antenna shown in Figure 4.8, operating at

frequency of 7.5 GHz designed as per algorithm [87] is simulated with parameters

width W = 11mm, length L = 10.65mm with length of the microstrip feed line is taken

as 10mm. Permittivity of substrate layers 1 and 2 are taken as εr1 = 2.2, εr2 = 2.32 and

heights h1=h2 =1.5mm.

Figure 4.8: Basic patch layout on Momentum

Figure 4.9 shows a return loss (S11) of 34.6 dB and bandwidth of 5.82 percent

obtained with the antenna resonant at 7.5 GHz. As per antenna parameters in Table

4.4, a power radiated of 8.05 mW is obtained with a gain of 6.5 and directivity of 7.6

dB. Since the gain and bandwidth are inversely related, increasing bandwidth reduces

the gain.

136

 

Figure 4.9: S11 Return Loss of a Multilayer Microstrip Antenna

Table 4.4: Antenna Parameters

4.3.2 Parameters of Multidielectric Layer Antenna using Conformal

Transformation Technique

The conformal mapping approach derives closed form expressions for the resonant

frequency for the general case of multi dielectric layers. After the transformation [78]

and frequency correction [87], the patch antenna with design parameters of length L=

9.65mm, width W= 11mm, εeff = 2.211, length and width of feed line 10mm and

1.66mm respectively, a resonant frequency of 7.505 GHz is obtained as shown in

Figure 4.10. Thus, a minor deviation of 5 MHz is observed which is acceptable at

high frequencies.

137

 

Figure 4.10: Simulation results showing return loss (S11).

Table 4.5: Antenna parameters showing power radiated as 9.37 mW & gain equal to 7.07dB

As shown in Table 4.5 a radiated power of 9.37 mw and a gain of 7.07 dB are

obtained. For substrates of high permittivity with heights of both the layers being

3mm, permittivity of multidielectric layers 1 and 2 is taken εr1= 6.15 and εr2= 10.2,

the result obtained are shown in Table 4.6. It can be seen from the table that the power

radiated is reduced to 4.55mW, a 50% reduction as compared to result obtained using

low permittivity substrate based designs. It can therefore be inferred that by

increasing heights further we may enhance the bandwidth but at the expense of

radiated power. Figure 4.11 shows polar plot of the antenna radiation in φ plane.

138

 

Table 4.6: Antenna parameters for high permittivity

Figure 4.11: Polar plot of E Field in φ plane

Momentum (ADS) software has been employed based on MoM to analyze

transformed rectangular patch corresponding to the given multidielectric antenna. For

accurate analysis of the transformed patch Finite Difference Time Domain (FDTD)

method is also employed and it is observed that the results match with those obtained

using Method of Moments (MoM).

139

 

4.4 Simulation of Antenna Parameters using Finite Difference

Time Domain (FDTD) Approach

FDTD provides both electric and magnetic fields in 3D model providing Maxwell

equations solutions without complexity. This analysis is well suited for microstrip

antennas, as they provide very accurate results and can be used to predict the response

over a wide range of frequencies. In this technique spatial as well as time grid for the

electric and magnetic fields are generated over which the solution is calculated. The E

cells are aligned with the boundary of the configuration and H fields are located

between the E cells. Each cell contains information about material characteristics and

the cell excitation is based on Gaussian functions which propagate along the structure.

In this approach the discrete time variations of Maxwell’s equations are calculated at

the single input port of the network. FDTD based model for the multilayer microstrip

antenna is shown in Figure 4.12 and with Gaussian pulse input shown in Figure 4.13.

The field and the formation of surface wave of cells in the x and y directions are

shown in Figure 4.14 and Figure 4.15 respectively.

Figure 4.12: FDTD modelled multilayer patch layout top view

140

 

Figure 4.13: Gaussian Pulse at time t=35pico seconds

In Figure 4.15, the reflected pulse is absorbed by the source and forms the surface

waves, which ultimately dies out with the depth of the medium. After transformation

bandwidth of a multilayer antenna is reduced by nearly 50% and S11 improves as

shown in Figure 4.16.

Figure 4.14: Showing electric field reflected from the upper substrate at t=39ps

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Figure 4.15: Formation of surface waves at the interface at t=141ps

Figure 4.16: Comparison of S11 Return loss of multilayer and its transformation

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4.5 Fabrication and Measurements of Prototype Multidielectric

Microstrip Patch Antenna Parameters Figure 4.17 shows the prototype multidielectric microstrip patch antenna fabricated as

per the designed parameters to resonate at 7.5 GHz.

Figure 4.17: Prototype of Multidielectric Patch Antenna Designed to Resonate at fr= 7.5 GHz

Figure 4.18: Multidielectric Patch Antenna Resonating at a Frequency fr=7.499 GHz with Return Loss S11= -22 dB.

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Figure 4.18 shows the experimental set up used to determine the resonating frequency

the return loss S11 and the impedance bandwidth of the fabricated prototype antenna.

The antenna is seen to resonate at 7.499 GHz very close to the designed frequency,

with measured return loss S11=-22 dB. Increase in the return loss is attributed to the

enhancement of surface wave in the practical antenna. Figure 4.19(a) and Figure

4.19(b) shows the experimental set up for the measurement of frequencies f1 and f2 at

either side of the resonant frequency fr= 7.499 GHz. The frequencies f1 and f2

measured corresponds to return loss S11=-10 dB. The corresponding frequencies

measured are f1= 7.222 GHz and f2 = 7.694 GHz shown respectively in the Figure

4.19(a) and 4.19(b). Hence the measured impedance bandwidth is found to be

% Bandwidth = r

12

f )f-(f x 100 = ≈⎟

⎠⎞

⎜⎝⎛ − 100

499.7222.7694.7 x 6.3%

Hence the prototype microstrip patch antenna validates the process of bandwidth

enhancement based on the Q factor optimization. The bandwidth practically observed

has improved compared to that obtained based on the ADS based Momentum

simulation result.

Figure 4.19 (a): Impedance Bandwidth Measurement at S11=-10 dB (f2)

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Figure 4.19 (b): Impedance Bandwidth Measurement at S11=-10 dB (f1)

4.6 Antenna Performance Improvement using Cover Layer

Microstrip antenna operating for high frequency aircraft applications needs cover

layer to protect it from wear and tear due to harsh environmental conditions. The

biggest advantage of such a superstrate layer is that it offers protection while

maintaining the conformability and low profile ability of the antenna. The substrate

layer may prove advantageous or detrimental to the antenna radiation characteristics

depending on the thickness of the substrate as well as the relative dielectric

permittivity [97]. It can also be shown that by using proper combination of materials

and antenna dimensions with impedance matching, surface waves can be eliminated.

Cover layer plays a significant role in improving gain or bandwidth. A major

improvement in bandwidth was observed by optimizing quality factor and using a

cover layer of the same thickness with low permittivity.

Results obtained after using cover layer displayed in the return loss S11 shown in

Figure 4.20, which indicates an impressive increase in the impedance bandwidth and

power radiated as 6.97 mW & gain equal to 6.32 dB as shown in Table 4.7. Increase

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in the bandwidth is 7.5% of the centre frequency keeping all the parameters

optimized.

Figure 4.20: S11 Return Loss of a Multilayer microstrip antenna with a cover layer of thickness 1.5mm

and permittivity 2.2

Table 4.7: Antenna parameters with cover layer showing power radiated as 6.97 mW & gain equal to

6.32 dB

4.7 Conclusion In this Chapter bandwidth enhancement procedure is proposed which emphasizes the

importance of quality factor optimization as a parameter to realize broadband

communication. Evaluation of reflection and surface wave losses for low permittivity

substrate based on FDTD analysis has been carried out by using MATLAB.

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Realization of improved bandwidth with minimization of surface wave losses is due

to the contribution of offset impedance matching employed in feeding technique.

After a series of simulation studies based on MoM, FDTD and result obtained study

of a prototype multilayer microstrip antenna, an impedance bandwidth of 5.8% has

been obtained with low permittivity substrates. The bandwidth further improves to

7.5% by using a cover layer which is an improvement over a conventional

multidielectric layer antenna by approximately 30%. The prototype fabricated and the

experimental result very closely matches the designed and the simulated parameters.

The increase in surface wave is correlated by the increase in the return loss S11 and

enhancement of the practical antenna bandwidth.