application note apn1012: vco designs for wireless handset and

Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.com200322 Rev. A • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • July 21, 2005 1

APN1012: VCO Designs for Wireless Handset and CATV Set-Top Applications

APPLICATION NOTE

IntroductionVoltage Controlled Oscillators (VCOs) have come to the forefrontof RF designs together with the first PLL circuits. In the erabefore the PLL, oscillators were mostly free running, and only inrare cases were varactors used for modulation or temperaturecompensation. Nowadays, we rarely see free running oscillators,instead they have become varactor-controlled oscillators. This isbecause most RF applications require band coverage, which canbe realized through the PLL circuit requiring two sources of RFpower. The reference source frequency is often a VCXO or TCXO,while the other frequency is controlled by the PLL phase detector.

Usually, both VCXO/TCXO and the RF VCO are voltage controlledoscillators. The difference between a reference oscillator and atuned VCO is that the former usually has a very high Q resonator,

which allows for very stable oscillation, while the latter has alower Q resonator, allowing a relatively high tuning range. In ref-erence oscillators, varactors are used for fine tuning ortemperature compensation (TCXO). In tunable oscillators, varac-tors are used to change (tune) the frequency. In some VCOs,varactors may be used also for modulation, for example in aDECT system where modulation is used to generate a constantenvelope GMSK signal.

Although it is a small part of the RF design, the VCO is often amajor headache for designers. The goal, in this application note,is to show how Skyworks products and services may help you toovercome VCO concerns and help to make your design amongthe best products on the market.

Skyworks Solutions, Inc. • Phone [781] 376-3000 • Fax [781] 376-3100 • [email protected] • www.skyworksinc.comJuly 21, 2005 • Skyworks Proprietary Information • Products and Product Information are Subject to Change Without Notice. • 200322 Rev. A

APPLICATION NOTE • APN1012

2

VCOs in Digital Wireless PhonesConsider the hypothetical wireless handset phone. Today, thehandset is a dual-band (cellular/PCS) and multimode systememploying many VCO functions. There are many ways to realizethese functions, making it virtually impossible to specify the fre-quency and tuning range for all designs. However, there arecertain common features that are outlined in Figure 1.

In a typical receiver, dual conversion superheterodyne solutionsare usually employed. They convert either 900 MHz (cellular) or1.8 GHz (PCS) down to the SAW frequency range, which may bebetween 90–400 MHz. Further, this signal is either downcon-

verted or demodulated into a digital I/Q signal using a lower fre-quency IF VCO. The transmitter path is either directly modulatedat 900 MHz or uses a dual conversion scheme requiring at leasttwo VCOs.

When dual-band requirements are needed, up to eight or moreVCOs may be required to satisfy specific frequency plans. This isoften a technically and economically restrictive solution. Manydesigners try to solve this over-VCOed problem using both smartfrequency planning and multiband VCOs, as shown in Figure 1.

RF VCO Ranging:400–1900 MHz

T/RSwitch

I

IF VCO

RF VCO

BPF

BPF

BPF BPF

LNA

RFDetector

PABPF

PLLControl

PLL

PLLControl

PLL

π/2ΣBPF

PLLControl

π/2

PLLControl

PLL

I

Q

Q

RF VCO

IF VCORanging:

100–400 MHz

PLL

Coupler

Figure 1. VCOs in a Digital Wireless Phone



Fundamental Low Noise Colpitts VCOThe characteristic feature of the Colpitts VCO is that it uses acapacitive divider for the feedback consisting of C1 and C2, andan inductive branch including a parallel resonator and seriescapacitor C3. The parallel resonator includes inductive elementM1 (that may be a discrete inductor for lower frequencies or alength of microstrip line for RF) and a capacitive branch, con-sisting of a varactor and a series capacitor(s). The entireinductive branch should have inductive impedance at the fre-quency of oscillation, otherwise there will be no oscillation. Thismeans that the resonant frequency should be higher than theoscillation frequency.

Note that the resonator current circulates through the varactor,series capacitor C11 and inductor M1 and is the largest current inthe tank circuit. Because of this, losses introduced in this currentpath are the crucial ones with respect to phase noise.

Without delving deeply into phase noise theory, we note thatphase noise is inversely proportional to the power bypassedthrough the feedback loop, and the loaded Q of the tank circuit.Thus, the more power lost on the way to the transistor base, thehigher the noise. It is clear that varactor loss plays a crucial rolein the phase noise property of the VCO. If phase noise is an issue,the varactor series resistance should be carefully considered.

There is an additional concern because phase noise is not only afunction of varactor loss. The varactor capacitance voltage char-acteristic has a crucial impact on phase noise as well. With ahigher capacitance ratio, the varactor’s coupling to the resonatoris reduced, resulting in lower resonator current. Therefore, ahyperabrupt varactor having higher series resistance is often abetter choice than a lower capacitance ratio abrupt varactorhaving lower series resistance.

Q1

2SC5007 (34)

Q2

2SC5008 (44)

C12 p

C21.5 p

C3

0.75 p

C7

0.5 p (0402)

R1150

R2

2.7 k C4 0.75 p

C8

R3

200

R4

3.6 kC5

0.75 p

C6

0.75 p

M1

4 x 04 mm (Trim)

D1SMV1493

C10

15 p

C11

1.5 p (0603)

C9

C12

1 p

M2

6.5 x 0.2 mm

M3

4 x 0.35 mm

VTUNE

M4

2 x 0.2 mm

M6

5 x 0.2 mm

M5

2 x 0.2 mm

RF Out

VCC

Low RS, Low VoltageHyperabrupt Varactor

Figure 2. Low Noise, High-Performance Colpitts VCO



4

Differential VCO for Integration with an RF ICDesigns based on RF IC solutions, with built-in VCOs, oftenemploy a differential VCO configuration. One possible differentialVCO configuration is shown in Figure 3. In this case, the tank cir-cuit is formed by C3, C4 and a resonator C8, C9, D1, M1. Hereagain, the resonator current plays a decisive role in phase noisedefinition. Thus, phase noise is strongly dependent on resonatorloss. Capacitors C3 and C4 help establish the correct phase shiftvalue in the feedback loop moving oscillations closer to the reso-nant frequency. This is the principal difference between a Colpitts

and a differential VCO. In the Colpitts case, the resonant fre-quency is always higher than the oscillation frequency; in thedifferential VCO the resonant and oscillation frequencies maycoincide. Thus the loaded Q of the circuit becomes significantlyhigher, and the feedback loop losses are increased due to thehigher resonant currents. When this happens, the differential VCOis more vulnerable to resonator loss than the Colpitts VCO andusually shows 5–10 dB higher noise if compared to an equivalentColpitts case.

V1 V2

V3

R1

47R23 k

R39.1 k

R43 k

R53 k

R63 k

R73 k

R820

R93.9 k

L133 n

C1

100 p

C2 2 p

C34 p

C41.5 p

D1

SMV1493-079

C51.5 p

C6100 p

C7

100 p

VCC 3 V

VVAR

RF Out

C82 p

C96 p

M14 x 0.5 mm

M26.5 x 0.2 mm

R1051


RF IC

Figure 3. Differential VCO for the Integration with the RF IC



Dual-Band Switchable VCO SchematicOne way to improve design economics in the multi-VCO require-ment is to employ band switching in the VCO. If the frequencyswitching required isn’t very large (say within 20%), it may usu-ally be realized within the same tank circuit, by switching “on” or“off” an additional capacitor or inductor. However, if the requiredswitching is more than 30%, it becomes very difficult to satisfyboth wideband and low noise requirements in a single design.One possible solution is to use two separate tank resonator cir-cuits switched with two PIN diodes. In this case, the feedbackneeds to be optimized to fit both band requirements at the same

time. Thus, a trick is used — connecting a capacitor C11 in par-allel with C6 when a lower band resonator is selected. Thisprovides significant improvement in phase noise since C6 maythen be optimized for the best performance at high band, and C11at the lower band.

Another important feature of this switching scheme is that thePIN diodes are not in the resonator current path. Because of this,phase noise is not sensitive to the PIN diode resistance. This isfortunate, since it means less forward current is needed. In addi-tion, any noise on the PIN diode bias current (common for thenoisy digital environment of today’s phones) would not cause sig-nificant modulation noise.

L112 nH

D1

C18 pF

C2

470 pF

C3

100 pFC6

20 pF

C715 pF

R1300

R2

R3

P1

L256 nH

D2

SMV1139-011

C810 pF

R4

1.5 k

P2

SMP1320-011

1

J1

VTUNE

1J2

VSW_High

1J3

VSW_LowC4

100 pF

C5

100 pF

Q1NE68519

C9100 pF

C1020 pF

R5

100

R6

3 k

R7

6.8 k

1

J4

VCC 3 V

1J5

RF

1.5 k

1.5 k

C1130 pF

CCC

100 pF

SMV1408-011

SMP1320-011

Low Current SwitchingPIN Diodes


Figure 4. Dual-Band Switchable VCO Schematic



6

Dual-Band Switchable RF VCOAs mentioned before, relatively small (less than 20%) frequencyswitching may be achieved inside the same tank circuit by con-necting or disconnecting capacitors (and sometimes inductors).The PIN diode D2 performs a tricky task — it adds more capaci-tance in parallel with the existing parallel capacitance of theresonator, and also adds more capacitance in parallel with the

existing series capacitor. This technique is used to overcome theproblem of increased resonator Q, when connecting additionalparallel capacitance, by decreasing it with higher series capaci-tance. It allows D2 to keep phase noise near its optimum at bothbands. Another PIN diode in the output matching circuit tunes thebuffer to a frequency doubler mode when working in PCS band.

VCC (3 V)

RF Out

D3D2

D1

VVAR

VCTL1

VCTL2

D1: SMV1493-079

D2, D3: SMP1322-079

PINs are Used in Both Resonator Tankand Output Matching Circuits

Figure 5. Dual-Band Switchable RF VCO



VCOs in a Set-Top Cable Downconverter The typical set-top downconverter is a dual-conversion receiveremploying upconversion and downconversion techniques toovercome image problems in a wideband environment of50–1000 MHz. In the dual up/downconversion scheme, theproblem of image channel and input filtering virtually does notexist because there is no signal at the image channel. Theimage channel is always higher than the highest frequency ofthe cable signal.

Two RF VCOs are required for dual downconversion. The first is awideband VCO tuned from 1100–2000 MHz with a control voltagefrom 1–20 V. The other is a narrow-band VCO, which may use aCDR, coaxial dielectric resonator, at 1144 MHz. In a digitalsystem, the second IF signal may be further demodulated,requiring an additional 44 MHz VCO.

The specific action of the wideband VCO is its wideband tuningrequirement. Let us consider some possible solutions for thewideband VCO.

BPF

PLL PLLLPF

HPF LPF

Upstreamfilter

BPF

UpstreamPLL

ControlPLL

Control

AGC

1st Mixer75 Ω

54–860 MHz 40 dB45.75/44 MHz

1100 MHz

1154–1960 MHz 1144/1145 MHz

2nd Mixer

Low-Distortion PIN DiodeAttenuator

Wideband VCO Tuning in1100–2000 MHz Range

Narrow-Band VCO@ 1145 MHz

dB

Figure 6. VCOs in a Set-Top Cable Down-Converter



8

Wideband Colpitts VCO SchematicThe unique action of the wideband Colpitts VCO is in its tank cir-cuit design, which uses an inductor with a varactor connected inseries and no parallel capacitor, in contrast to the low noiseColpitts VCO described in Figure 7. The feedback capacitors areoptimized to the best power flatness over the entire frequency

band. Back-to-back varactors are often used to minimize para-sitic mounting capacitance (between mounting pads andadjoining components). This circuit is usually designed to mini-mize any parasitic parallel capacitance that may be caused bycomponent pads or transmission lines close to the inductive path.

VCC = 5 VICC = 9 mA

NE68519

SMV1265-011

3.3 k

9.1 k

200

3 k3 k

320 x 30 mils

1 p

560 p

1.62 p

300 p

2 p

5.6 nH

SMV1265-011

RF Output

VTUNE

High C (V) RatioHyperabrupt Varactors

Useful TuningRange:

980–2120 MHz

0.91.01.11.21.31.41.51.61.71.81.92.02.12.2

0 5 10 15 20 25 30

Varactor Voltage (V)

Freq

uenc

y (G

Hz)

POUT (dBm

)

0

1

2

3

4

5

6

7

Fexp Fmodel POUT_exp POUT_model

A carefully designed layout with minimum parasitic capacitancesmay show large frequency coverage, for example 980–2120 MHzas the performance indicates. The varactor selection is a crucial

part of the design. Skyworks new SMV1265-011 varactor isspecifically designed to fit this wideband application.

Figure 7. Wideband Colpitts VCO Schematic

Figure 8. Wideband Colpitts VCO Performance



Wideband Balanced VCO SchematicAn even wider tuning range may be achieved with a balancedVCO configuration. The reason for its wider tuning performance isthat the phase response of this VCO’s active element is flatter

over the range of tuning compared to a Colpitts VCO. This allowsthe tank circuit more control over the oscillation frequency.

The best results are achieved with back-to-back connectedSMV1265 varactors, where there is 820–2120 MHz coverage.

V1

NE68119

V2

NE68119

T1

16 x 0.4 mm

T2

15 x 0.7 mm

C1

10

C2

10

L1

33 nH

L2

33 nH

R1

33

R2

33

R3120

R4120

R5

820

R7

51

R81 k

R6

820

R10

2.4 k

D1

R9

1 k

R11

1000C3

100VVAR1

VCC1

5–8 V

C4

100

C5100

R12

50

A

D2

SMV1265SMV1265

T3

3 X 0.7 mm

C6

1000

High C (V) RatioHyperabrupt Varactors

0 5 10 15 20 25 30

Varactor Voltage (V) Varactor Voltage (V)

0.8

1.0

1.2

1.4

1.6

1.8

2.0

2.2

Freq

uenc

y (G

Hz)

Frequency Tuning

0 5 10 15 20 25 30-8

-6

-4

-2

0

2

4

6

8

Pow

er (d

Bm)

Power Response

Useful TuningRange:

820–2120 MHz

Measured @ 7 V

Simulated @ 7 V

Measured @ 5 V

Measured

Simulations

Figure 9. Wideband Balanced VCO Schematic

Figure 10. Wideband Balanced VCO Performance



10

Varactor FundamentalsLet us consider some fundamental properties of varactors. A var-actor is a specially designed P-N junction diode, whosecapacitance changes significantly in reverse bias mode. Thereare three important parameters characterizing varactors. The firstis the capacitance ratio at two reverse voltages; this value char-acterizes the tuning ability of the varactor capacitance and is oneof the most important parameters. The second is the value ofcapacitance at a given voltage. The third is the series resistanceof the varactor.

The structure of the basic varactor, called an abrupt junction var-actor, is shown in Figure 11. Generally, it is built as a P++ - N -N+ structure, using epitaxial N-growth on the N+ substrate witha constant doping level in the N-region. The lower doped N-region is the active area where the electron concentrationchanges, depending on the reverse voltage applied between theanode and cathode of the varactor. There are certain limitationson the level of doping in the N-region, which is usually defined bythe required capacitance ratio of the varactor. Because of this,

the conductance of the N-area is a major contributor to the varactor’s series resistance. Note that as the reverse voltage isincreased, the series resistance (due to the N-area) will decreasealong with the capacitance.

The hyperabrupt junction varactor has a more complicateddoping profile. Because of much higher doping on the P++border, the electron concentration changes much more abruptlycompared to an abrupt junction. As a result, the capacitance ofthe hyperabrupt diode at zero bias is much higher than for theabrupt diode. Therefore, the capacitance change vs. reverse biasbecomes significantly higher for hyperabrupt diodes. The trade-off for this better capacitance ratio is increased series resistance.The reason is that the doping level of the N-area has beenreduced to keep average doping level over the N-region the sameas the abrupt diode level. There are many ways to bring theseries resistance in the hyperabrupt diode to as low a level aspossible. Modern state-of-the-art hyperabrupt diodes for lownoise VCOs have series resistance almost as low as discreteceramic capacitors.

P++DopingLevel

N

Depth from Anode

V0

V0

V1V1 V2V2 VSAT

VSAT

Elec

tron

Conc

entra

tion

P++DopingLevel

N+

N

Depth from Anode

Elec

tron

Conc

entra

tion

Abrupt Junction Hyperabrupt Junction

N+

Figure 11. Varactor Fundamentals



24 m

il (0

.62

mm

)

62 mil (1.58 mm)

SC-79

Varactor PackagingMost high-volume discrete applications require varactors in low-cost, small surface mount plastic packages. Skyworks provides a

large variety of both plastic and ceramic packages. The recent,most advanced, miniature plastic package, SC-79, shown inFigure 12, is as small as 0402 discrete components.

Figure 12. Varactor Packaging



12

The coupling coefficient may be derived from the known (or typ-ical) values of the tuning frequency and varactor capacitancevariation. Note that the total temperature drift in this case isabout 0.5%, as compared to 1% maximum drift caused by tem-

perature compensated discrete ceramic capacitors. Even thosenumbers are extremely small when compared to the temperaturedrifts caused by a VCO transistor.

For the Typical Wireless Case:f = 1.6 ± -0.04 GHz

Using SMV1235-011 Varactor:

The Total Temperature DriftsDue to Varactor in the

-40 to +85°C Becomes:

LRES

CRES CDIV1

CDIV2

CBE

CCE

CCB

CSER2

CVAR

0.248

1.6 GHz0.08 GHz

22 ≈ = x x∆C VAR

=3.4 pF

pFCf

∆fK

Compare to the Discrete Capacitor!

f∆fT %54.0≈

Relative Capacitance Change vs. Temperature Figure 13 shows typical relative capacitance variations vs. tem-perature for different reverse voltages. It indicates a totalcapacitance change of 5–6% in the range of -40°C to +80°C. Incomparison, a temperature compensated, ceramic capacitor

residual variation bar is shown for a typical ±100 ppm device.This has a possible total capacitance change of over 2%. Whencomparing the overall effect of temperature on varactors andceramic capacitors, the coupling of the devices to the resonatorcircuit should be considered.

-4

-3

-2

-1

0

1

2

3

4

5

-40 -20 0 20 40 60 80

Temperature (°C)

Perc

enta

ge o

f Var

iatio

n (%

)

ConsiderVaractor

Coupling!!

VVAR = -4 V

VVAR = -1 V

VVAR = -2.5 V

Deviation Range for a TypicalTemperature Compensated

Discrete CeramicMultilayer Capacitor

Figure 13. Relative Capacitance Change vs. Temperature for Hyperabrupt Varactors

Figure 14. Varactor Temperature Effect on the Oscillation Frequency



Varactor SPICE ModelTo model a varactor in most commercial simulators, we recom-mend the available PN-junction diode SPICE model. We specifythe barrier junction capacitance parameters CGO, VJ, and M,instead of the default parameters. In addition, we add a value ofCP, in parallel with the junction capacitor, which is not the

package capacitance. For ideal abrupt junction varactors, theparameters are constant and may be defined from physicaltheory. However, for actual abrupt or hyperabrupt varactors, thesevalues are not constant. In these cases, we use the same equa-tion, to fit its parameters, for the best compliance with measuredcapacitance vs. voltage response.

M+ CP

VJ

VVAR

CGOCV =

1 +

SPICE Model for SMV1142-011

⎟⎠⎞

⎜⎝⎛

Because of formalization, parameters describing the junctioncapacitance of hyperabrupt varactors may be significantly dif-ferent from the default values used in the SPICE simulators forthe ideal silicon PN-junction.For example, typical hyperabrupt varactor SMV1235 was fittedwith M = 4 as opposed to 0.5, which follows silicon PN diode

theory. Note that some SPICE simulators offer fixed default valuesof M = 0.5 which can’t be changed. In this case, a diode modelmay not be used, however, direct nonlinear capacitance may beused as defined in the given formula.

0 2 4 6 8 10 12

Varactor Voltage

0

5

10

15

20

Capa

cita

nce

(pF)

SMV1235Approximation

SMV123516.13/(1-Vv/8)^4 + 2

81+

16.13

4+ 2 =

VVAR

CV

⎟⎠

⎞⎜⎝

⎛

Figure 15. Typical Varactor SPICE Model

Figure 16. C (V) Curve Fitting for Typical Hyperabrupt Varactor



14

Super-hyperabrupt Varactor ModelingTo overcome limitations of the “standard” PN-junction SPICEmodel for hyperabrupt and super-hyperabrupt devices, such asthe SMV1265, an interleaving technique is used. In this tech-nique, the entire capacitance reverse voltage range is broken intoseveral subranges. These subranges are small enough for the

formula to provide good approximation, not only within a givensubrange, but also for certain extensions beyond it. The extensionmargin is defined from previously estimated RF signal amplitude.Such interleaving ensures that the formula would work well, notonly in terms of DC bias, but for large signal RF analysis as well.

M = pwl (VVAR 0 2 2.5 2 2.500009 25 6.5 25 6.50009 7.3 11 7.3 11.0009 1.8 40 1.8)

VJ = pwl (VVAR 0 4 2.5 4 2.500009 68 6.5 68 6.50009 14 11 14 11.0009 1.85 40 1.85)

CP = pwl (VVAR 0 0 2.5 0 2.500009 0 6.5 0 6.50009 0.9 11 0.9 11.0009 0.56 40 0.56)

CJO = pwl (VVAR 0 22.5 2.5 22.5 2.500009 21 6.5 21 6.50009 20 11 20 11.0009 20 40 20)*10 -12

RS = pwl (VVAR 0 2.4 3 2.4 4 2.3 5 2.2 6 2 7 1.85 8 1.76 9 1.7 10 1.65 11 1.61 12 1.5 40 1.5)

0 5 10 15 20 25 30

Varactor Voltage

0.1

1.0

10

100

Capa

cita

nce

(pF)

0

0.2

0.4

0.6

0.8

1.0

ApproximationMeasured

SMV1265

Interleaving of Splines

Figure 17. Piece-Wise Curve Fitting for High C (V) Ratio Varactors



VCO Modeling ConceptFor the purpose of modeling and analysis, a VCO design may besimulated as an amplifier with parallel feedback. This analysisinvolves measuring loop gain using a specific idealized direc-

tional coupler called OSCTEST in Libra IV. (For Harmonica usersthere is an application note showing how to implement OSCTESTfunction using S-parameters file. Refer to your Harmonica vendorfor more information).

RL

L

CC1

C2

I1 I2

V1

V2

Simplified Colpitts VCO

Feedback Model of Colpitts VCO

Amplifier

Tank Circuit

Loop GainObservation

Plane

Figure 18. VCO Modeling Concept



16

The major goal of the large signal, open loop VCO analysis is toobserve the magnitude (defined in dB) and the phase of the openloop voltage gain Ku, to identify particular features of thedesigned VCO.

First, we need to establish the optimum conditions for the oscilla-tions in a given tuning range. Second, we need to find out

whether there are possibilities for parasitic oscillations both inthe lower and higher frequency ranges. If there are parasiticoscillations, some preventive measures should be taken. Third,we need to find ways to make both QL and the loop power (PIN)as high as possible to facilitate phase noise performance. Finally,other features of the VCO need to be addressed, among themload pulling and VCC pushing.

1.0 1.5 2.0 2.5 3.0 3.5

Frequency (GHz)

-4

-3

-2

-1

0

1

2

3

4

Mag

(Ku)

dB

-200

-100

0

100

200

OscillationPoint

Arg (Ku)

Mag (Ku)

0.5 1.0 1.5 2.0 2.5

Frequency

-250

-200

-150

-100

-50

50

0

100

VVAR = 0 V

Transitor X at PIN = 10 dBm

Resonator at DifferentVaractor Voltages

Parallel Resonance:

When it nears the oscillationpoint, tank circuit losses

increase; noise increase andpower decrease follow

Oscillation Happens at the 0 dB Gainand 0 Loop Phase Shift

VVAR = 30 V

Figure 19. Typical Loop Gain Results for the Colpitts VCO



Wideband Colpitts VCO ModelThe OSCTEST component interrupts the oscillator feedback,allowing the designer to analyze the VCO as an ordinary two-portcircuit (amplifier). To observe the loop response, we define theopen loop voltage gain Ku. For more details, please refer to the

VCO application notes listed in the References section. The var-actor model is defined as a PN-junction diode SPICE model forlarge signal, harmonic balance analysis. The transistor isdescribed by the Gumel Poon SPICE model with parameters pro-vided by the vendor.

TransistorSubcircuit

SeamlessVCO LoopOpener

VaractorModel

Figure 20. Wideband Colpitts VCO Model



18

Differential VCO FundamentalsThe differential VCO utilizes paired transistors in common emitterand common base configurations. The phase balance conditionfor sustaining oscillations requires significantly lower phase shiftin comparison to a Colpitts design (ideally 0 degrees vs. 180

degrees). This makes it possible to use a resonator tuned to theexact resonant frequency. However, the feedback losses may behigher because the higher resonating currents will causeincreased ohmic losses.

Resonator Works at itsParallel Resonance, Giving

Best Phase Slope Performance

Two Transistors in theLoop Give Advantageof Higher Loop Gain

Common-collector -Common-base Phase

Shift Ideally Is 0

Figure 21. Concept of Differential VCO



Balanced VCO FundamentalsFundamental properties of the balanced VCO are more clearlyunderstood using the simplified circuit diagram shown in Figure22. The transistors are in common collector configuration. Thisis characterized by high input impedance, looking from thetransmission line and referenced as LB. Capacitor CBP simulatesthe transmission lines and the grounding effect of the mountingpads. Coupling current ICPL circulates between the transistor

bases to drive them with a 180° phase shift. The emitter cur-rent IFB forms the feedback loop, carrying an amplified energysurplus that is needed to sustain resonant current IRES and cou-pling current ICPL through the emitter base path. Unlike aColpitts VCO, this circuit does not require frequency dependentfeedback to match the internal transistor, high frequency phaseshifts. When properly compensated for wideband performancewith interbase inductor, LB, this circuit will be more broadbandthan a Colpitts VCO.

LPAR LSER

DVAR

Re2

Le2

CBP

CBP

Re1

Le1

LB

Rcol CBP

CBP

Rcol

ICPL

IFB

IRES

Low pass Matching Serves to Improve

High-frequency Performance

Collector Currents are Shifted 180° In Phase. That’sWhy We Call It“Balanced”

Figure 22. Balanced VCO Fundamentals

References“Varactor SPICE Models for RF VCO Applications.” ApplicationsNote APN1004, Skyworks Solutions Inc., 1998.

“A Colpitts VCO for Wideband (0.95 GHz–2.15 GHz) Set-Top TVTuner Applications.” Applications Note APN1006, SkyworksSolutions Inc., 1998.

“A Balanced Wideband VCO for Set-Top TV Tuner Applications.”Applications Note APN1005, Skyworks Solutions Inc., 1998.

“Switchable Dual-Band 170/420 MHz VCO For Hand-Set CellularApplications.” Applications Note APN1007, Skyworks SolutionsInc., 1998.

“A Wideband General Purpose PIN Attenuator.” Applications NoteAPN1003, Skyworks Solutions Inc., 1999.

“Wideband VCO for Set-Top Applications.” Microwave Journal,April 1999.

“Circuit Models for Plastic Packaged Microwave Diodes.”Applications Note APN1001, Skyworks Solutions, Inc.

“Design with PIN Diodes.” Applications Note APN1002, SkyworksSolutions, Inc.

For the availability of the above materials, visit the Skyworks Website at: www.skyworksinc.com.



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application note apn1012: vco designs for wireless handset and

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