an944-1 microwave transistor bias considerations

8/10/2019 AN944-1 Microwave Transistor Bias Considerations

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Microwave TransistorBias Considerations

Application Note 944-1

Introduction

Often the least considered factor in microwave transistorcircuit design is the bias network. Considerable effort isspent in measuring s-parameters, calculating gain, andoptimizing bandwidth and noise figure, while the sameresistor topology is used to bias the transistor. Since thecost per dB of microwave gain or noise figure is so high, thecircuit designer cannot afford to sacrifice RF performanceby inattention to DC bias considerations.

Microwave transistor amplifier design requires biasingthe transistor into the active region of performance andholding this bias or quiescent point constant over varia-tions in temperature. At low frequencies, emitter resistorstabilization with negative current feedback is used for DCstability. At microwave frequencies, the by-pass capacitorbecomes a problem since a good RF bypass at the designfrequency often introduces low frequency instability andgives rise to bias oscillations. Figure 1 shows this effect.

In low noise amplifier applications, even if a capacitorcould be chosen to provide effective RF and low frequencyemitter bypass, any small series emitter impedance at theoperating frequency would reflect in a large noise figuredegradation. Most microwave circuit designs for best gainor lowest noise figure require that the emitter lead be DCgrounded as close to the package as possible so that theemitter series feedback is kept at an absolute minimum.

It has been found that the principal dependent variablein DC stability analysis is the collector current (IC). Thetransistor parameters that are temperature sensitive andinfluence ICare examined along with some passive resis-tive circuits that give stable DC operation and allow for

trimming due to variations in transistor types.

Analysis

In order to best select an optimum bias network, a methodof comparison has to be developed. Analysis of transistorbias network instability involves writing a collector currentequation in terms of the transistor equivalent circuit andthe external bias circuitry. Partial derivatives of the collec-tor current, with respect to the temperature dependentvariables, are calculated individually and the resultantstability factors can then be considered simultaneouslyto predict collector current temperature behavior.

Figure 1.

BYPASS

CAPACITOR

100 pF

EMITTER

BIAS

RESISTOR

S11(AT 4 GHz) = 0.52 154

S11(AT 0.1 GHz) = 0.901 14.9

S-PARAMETERS OF GROUNDED EMITTER

TRANSISTOR AT 4 GHz AND 0.1 GHz

S'11(AT 4 GHz) = 0.52 154 UNCHANGED

AT 4 GHz

S'11(AT 0.1 GHz) = 1.066 -8.5 |S11|>1

AT 0.1GHz

S-PARAMETERS OF SAME TRANSISTOR WITH

GOOD 4 GHz EMITTER BYPASS CAPACITOR.

S'11AT 4 GHz REMAINS UNCHANGED WHILE

|S11| AT 0.1 GHz IS GREATER THAN 1

INDICATING CONDITIONAL STABILITY.

S11


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2

A look at the typical bias circuit ofFigure 2a and its DC equivalentcircuit in Figure 2b identifies theinternal parameters that affectcollector current. Since the ex-

ternal resistors have negligible tem-perature change compared to thetransistor, the temperature sensitiveparameters are found to be V'BE,ICBO, and hFE.

Figure 2.

Temperature Sensitive DC Parameters

Base-Emitter Voltage (VBE)

V'BEis the base-emitter voltage. VBEis internal to the transistor and has anegative temperature coefficient of

-2 mV/C. Figure 3a shows a typicalsilicon PN junction current-voltagerelation. Notice the negative shiftin threshold voltage for increasedtemperatures. A bias circuit thatfixes a constant voltage on the base,independent of temperature, findsthe collector current increasing for

Figure 3.

increasing temperature. Fortunately,negative voltage feedback will helpcompensate for V'BEchanges as illus-trated in Figures 3b and 3c.

RB1

VBE

VCE

VCC

RB

RB2

RC

IB

IB

IBB+ IB IC+ IBB+ IB

B

E

C

A. BIAS CIRCUIT SHOWING ONLY THE DC COMPONENTS

B. THE EQUIVALENT CIRCUIT OF FIGURE 2A IS USED IN

DC STABILITY ANALYSIS.

IBB

IBB

IC

RB2

RBIC

VBE hFE IB ICBO (1 + hFE)V'BE

VCE

VCC

RB1

hie RC

B

E

C

IBB+ IB

NOTES:V'BEIS INTERNAL TO THE TRANSISTOR AND HAS A -2 mV/C TEMPERATURECOEFFICIENT.

hieIS THE HYBRID INPUT IMPEDANCE AND FOR AVAGO TECHNOLOGIESMICROWAVE TRANSISTORS IS APPROXIMATELY 500 OHMS. THE EXACT VALUEIS UNIMPORTANT SINCE hieIS ALWAYS IN SERIES WITH THE LARGER BASE BIAS

RESISTOR RB. IT IS ONLY INCLUDED FOR COMPLETENESS.

100

10

1.0

0.1

0.010 0.2 0.4 0.6 0.8 1.0 1.2

BASE-EMITTER VOLTAGE VBE(VOLTS)

COLLECTORCURRENTIC(mA)

100C

-50C25C

VBE

VCE

VCCRC

RB

IB

IC

VBE

VCE

VCC

RCRB

IB

IC

B. BIAS NETWORK THAT USES NO FEEDBACK

AND IS SENSITIVE TO CHANGES IN V'BE,

ICBO, AND hFE.

A. TYPICAL BASE-EMITTER VOLTAGE (VBE) vs.

COLLECTOR CURRENT (IC) CURVE SHOWINGTHE THRESHOLD SHIFT DUE TO SENSITIVITY

OF V'BE= -2 mV/C.

C. BIAS NETWORK THAT USES VOLTAGE

FEEDBACK AND IS LESS SENSITIVE TO

CHANGES IN V'BE, ICBO, AND hFE.


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3

Reverse Collector Current (ICBO)

ICBOis the current flowing througha reverse biased PN junction.Classically, this leakage current is

expected to double for every 10Ctemperature rise in a silicon semi-conductor junction.

Microwave transistors have a morecomplicated reverse current flow.A small component of this currentflow is a conventional ICBOterm, butthe major contributor is a surfacecurrent that flows across the top ofthe silicon crystal lattice. This surfacecurrent is a more linear function oftemperature than the ICBOcurrent.

The total reverse current, made upof ICBOand surface components,increases at a rate much less thanthat which would be expected froman ICBOcurrent alone of the samemagnitude. A typical reverse currentversus temperature relationship isshown for several Avago Technolo-gies microwave transistors in Figure4. The data applies to a collector-base voltage of 10 volts.

In general, the slope of the ICBOcurve for Avago Technologiesmicrowave transistors remains

unchanged with reverse bias. Forstability calculations, a family ofcurves, at specified collector tobase voltages, can be considered tofollow the slope of the curve withintercepts at 25C corresponding tothe data sheet or measured value ofICBO.

DC Current Gain (hFE)

The typical characteristic curve ofFigure 5 shows the collector currentversus collector to emitter voltage

for a constant base current.

The hfeis defined as the ratio of thechange of collector current to thechange in base current.

Figure 4.

hFE, the DC value of the current gain,is defined as the ratio of the collec-tor current (IC) to base current (IB).

Figure 5.

Figure 6.

It is this DC value of hFEthat is usedin the stability calculations; it isfound to typically increase linearlywith temperature at the rate of0.5%/ C. Figure 6 shows the tem-perature dependence of hFE.

1000

100

10

1.0-55 -35 -15 5 4525 65 85

TEMPERATURE (C)

ICBOR

EVERSE

CURRENT(nA)

TYPICAL REVERSE CURRENT vs. TEMPERATUREFOR AVAGO TECHNOLOGIES MICROWAVETRANSISTORS. FOR A MICROWAVE TRANSISTOR

ICBO= IS(SURFACE CURRENT) + Ij(JUNCTIONCURRENT)

AVAGO 35850 SERIES

AVAGO 35820 &35860 SERIES

CLASSICALICBOSLOPE

16

12

8

4

00 10 20COLLECTOR-EMITTER VOLTAGE VCE(VOLTS)

COLLECTORCURRENTIC

(mA)

A TYPICAL COMMON EMITTER CHARACTERISTIC

CURVE SHOWING THE RELATIONSHIPS BETWEEN

BASE CURRENT AND COLLECTOR CURRENT.

IC

IB

IB= 0.24 mA

IB= 0.16 mA

40

20

0

-20

-40-55 -35 -15 5 25 45 65 85

TYPICAL PERCENT CHANGE IN hFEVERSUS

TEMPERATURE (NORMALIZED TO 25C).

TEMPERATURE (C)

SLOPE = +0.5%/C

hFE

(PERCENTCHANGE)

hfe= =ICIB VCE= constant

hFE= DC=ICIB VCE= constant


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RF Sensitivity of Avago Technologies

Microwave Transistors

A look at the four s-parameters andnoise figure of Avago Technologiesmicrowave transistors reveals that

|S21 |2

and noise figure stand outas the most sensitive parameters tosmall changes in bias. Also, both ofthese parameters are stronger func-tions of collector current (IC) thanof collector to emitter voltage (VCE).

This means that if we know some-thing about how |S21 |2and noisefigure change with collector currentand with temperature, then someconstraints can be placed on thebias network to minimize changesin RF performance over a specified

temperature range.

The typical data shown in Figure7a is normalized to percentagechanges in both gain and collectorcurrent. The noise figure change isplotted in dB. Although the absolutegain and noise figure are frequencydependent, their sensitivity withrespect to collector current (IC) canbe considered frequency indepen-dent. Notice that a 20% increase incollector current has a very small ef-

fect on either a transistor biased forminimum noise figure or a transistorbiased for maximum gain.

Next, a look at some typical changesin noise figure and gain as a func-tion of temperature (Figure 7b)shows that both NF and gain de-grade with increasing temperature.We see, for example, that a biasnetwork that can hold the quiescentpoint, such that the current does notincrease more than 20% to 60C, hasa 5% degradation in gain or a 0.3 dBincrease in noise figure at 60C dueto transistor changes alone. Sometemperature compensation couldbe designed into the bias circuitryby using lower values of collector

Figure 7.

current at 25C and allowing thetemperature sensitivity of the biasnetwork to offset the temperaturesensitivity of the transistor.

It should be pointed out that eachamplifier function has a differentbias requirement. In other words,transistors used in gain stages inwhich the noise figure or the satu-rated output power are not criticalhave a much more relaxed bias sta-

bility requirement than a low noisefront-end transistor. This can beseen in Figure 7a, since |S21 |2has a

broad maximum compared to noisefigure. A transistor biased for highlinear output power must hold itsquiescent point such that the 1 dBcompression point is not degradedwith temperature, and so that themaximum power dissipation in thedevice is not exceeded with increas-ing temperature.

Suggested Bias Circuits

Three bias circuit topologies areshown in Table 1 along with a

general expression for the collec-tor current and the calculated DCbias stability factors for V'EB, ICBO,and hFE. Most microwave circuitdesigns, for reasons of noise figure,gain, and RF stability, require a DCgrounded emitter, Therefore, emit-ter resistor stabilized circuits are notconsidered. The grounded emitternon-stabilized bias circuit (Table 1a)receives very little usage in micro-wave circuit design since it exhibitsthe least DC bias stability.

Both circuits in Tables 1b and 1c findwidespread usage as bias networks.

The voltage feedback circuit usesfewer components and is almostas temperature stable as Table 1c.

The addition of RB1and RB2to thevoltage feedback circuit does twothings. First, it makes all the elementvalues lower in resistance and thismakes it more compatible with thin/thick film resistor values. In the volt-age feedback circuit, the value of RBwould typically be in the range of30 kto 100 k. These values aredifficult to achieve in hybrid inte-grated circuits. Second, the circuitof Table 1c can be considered tohave a constant base current source,

10

5

0

-5

-10-40 -20 0 20 40 60

A. TYPICAL CHANGE IN PERFORMANCE AS A

FUNCTION OF COLLECTOR CURRENT VARIATION

FOR A TRANSISTOR BIASED AT MINIMUM NOISE

FIGURE AND A TRANSISTOR BIASED AT

MAXIMUM GAIN.

PERCENT CHANGE FROM A

QUIESCENT COLLECTOR CURRENT

PERCENTCHANG

E

|S21|2

1

0

-1NOISEFIGURECHAN

GE(db)

|S21|2

NOISE FIGURE

10

5

0

-5

-10-35-55 -15 5 25 45 65 85

B. TYPICAL CHANGE IN PERFORMANCE AS AFUNCTION OF TEMPERATURE VARIATION

FOR A TRANSISTOR BIASED AT MINIMUM NOISE

FIGURE AND A TRANSISTOR BIASED AT

MAXIMUM GAIN (NORMALIZED TO 25C).

TEMPERATURE (C)

PERCENTCHANGE|S21|2

1

0

-1

NOISEFIGURE

CHANGE(db)

|S21|2

NOISE FIGURE


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through RB. This allows for trimming,on a production basis, to initially setthe collector current to the desired

value. The collector current can-not be measured directly since thecurrent in RCis made up of base cur-rent, base bias network current, andcollector current. However, since ICisproportional to VCE, monitoring VCEwhile adjusting RBaccommodatesany value of hFEencountered inAvago Technologies transistors.

Differences in collector currentstability for each topology are com-pared in Figure 8. It is important

to point out that, for the sake ofcomparison, each circuit was usedto bias the transistor to a commonquiescent point. This data is typicalfor frequently encountered micro-wave bias circuits and is valuable inrelative comparisons. Notice thatfor each of the circuits, the collectorcurrent is a positive linear functionof temperature. And from Figure 7a,we see that noise figure degradationand gain are negative functions ofboth collector current and junction

temperature.

Bibliography

1. Chirlian, P.M.Analysis and Designof Electronic Circuits. McGraw-Hill,Chapters 2, 4, pp. 58-61.

2. Corning, John J. Transistor CircuitAnalysis and Design. Prentice-Hall,1965, pp. 91-115.

3. Cutler, Philip. Electronic CircuitAnalysisVolume 2. McGraw-Hill,1967, Chapters 4-11, pp. 220-228.

4. G.E. Transistor ManualSixth Edi-tion. 1962, Chapter 7, pp. 101-109.

5. Joyce, Maurice V. and Clark, Ken-neth K. Transistor Circuit Analysis.Addison-Wesley, 1961, pp. 66-81.

Figure 8.

60

20

40

-20

0

-40

-60-55 -35 -15 5 25 45 65 85

TYPICAL DC STABILITY PERFORMANCE

OF EACH BIAS NETWORK (TABLE 1)

USED TO BIAS AVAGO TECHNOLOGIESMICROWAVE TRANSISTORS. GRAPH

SHOWS THE PERCENT CHANGE FROMA NOMINAL QUIESCENT COLLECTOR

CURRENT AS A FUNCTION OF

TEMPERATURE (NORMALIZED TO 25C)

TEMPERATURE (C)

PERCENTCHANGEFROMA

QUIESCENT

COLLECTOR

CURREN

T

A. NON-STABILIZED

B. VOLTAGE FEEDBACK

C. VOLTAGE FEEDBACK &

CONSTANT BASE

CURRENT SOURCE

RB1

RB

RB2

RC

VCC

RB RC

VCC

RB RC

VCCVBB

NON-STABILIZED

VOLTAGE FEEDBACK

VOLTAGE FEEDBACK AND CONSTANTBASE CURRENT SOURCE


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Table 1. Microwave Transistor Stability Factors

Bias networks for DC groundedemitter operation. For each circuit

the general expression for thecollector current along with thestability factors are given.

The voltage feedback circuit of1B and the voltage feedback andconstant base current circuit of 1Cprovides for temperature stable DCoperation and complements the RFperformance of the following Avago

Technologies transistor series:Avago Technologies 35820, Avago

Technologies 35850, and AvagoTechnologies 35860.

The non-stabilized circuit of 1Ais not recommended and is onlyshown for comparison.

Table 1A.

Collector current at any temperature (IC)

ICBOstability factor

ICBO=IC

ICBO hFE,V'BE= constant

(hie+ RB)

hFE(VBB V'BE)+ ICBO(1 + hFE)

1 + hFE

V'BEstability factor

V'BE=

IC

V'BE hFE,ICBO= constant-h

FEhie+RB

hFEstability factor

hFE=IC

hFE hFE, V'BE= constantVBB V'BE

hie+RB+ ICBO

A. Non-Stabilized

RB RC

VCCVBB


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Use of Stability Factors

First calculate the stability factors for V'BE, ICBO,and hFE. Then, to find the change in collectorcurrent at any temperature, multiply the changefrom 25C for each temperature dependent vari-

able with its corresponding stability factor andsum.

It would appear to be an easy task to further ana-lyze the individual stability factors for minimumsin terms of the external circuit resistor values.

This is not easily done since all the factors areinter-related. The stability factors must be con-sidered simultaneously since an optimum set ofresistor values to minimize one parameter couldgrossly increase another.

hFE(VCC V'BE) + ICBO(1 + hFE)(hie+RB+RC)

hie+RB+RC(1 + hFE)

hie+RB+RC(1 + hFE)(1 + hFE) (hie+RB+RC)

hie+RB+RC(1 + hFE)

- hFE

where: K = hie+RB+ RC and D = hFERC+ RB+hie+ RC

(hFERC+RB+hie+ RC)(VCC V'BE+ KICBO)

D2

(hFE[VCC V'BE+ KICBO]+ KICBO)

D2-RC

B. Voltage Feedback

Collector current at any temperature (IC)

ICBOstability factor

ICBO= ICICBO hFE,V'BE= constant

V'BEstability factor

V'BE=IC

V'BE hFE,ICBO= constant

hFEstability factor

hFE=IC

hFE hFE, V'BE= constant

RB RC

VCC

SICBO=

IC=

IC= SICBO ICBO+ SV'BE V'BE

ICBO+ V'BE+ hFE

ICICBO hFE, V'BE= constant

SV'BE=IC

V'BE ICBO, hFE= constant

ShFE=

+ ShFE hFE

IChFE ICBO, V'BE= constant

ICICBO

ICV'BE

IChFE

Table 1B.


8/11


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Design Examples

Examples of how to calculate theresistor values for the two mostcommonly used grounded emit-

ter bias circuits are given below.The nearest standard 1% toleranceresistor values are also shown. De-pending upon the exact application,other tolerance resistors can be usedwith little or no difference. Also list-ed, in tabular form, are other resistorvalues that have been calculated

for the indicated quiescent pointsat a 20 volt supply voltage. Othersupply voltages can be calculated

accordingly.

For bias circuit designs with AvagoTechnologies microwave transistors,assume hFE= 50 and neglect hie.

1. Determine supply voltage available (VCC= 20 V) and transistor operatingbias point (10 V, 10 mA). Assume ICBO= 0, VBE= 0.7 volts.

2. Knowing the measured value of hFE, (or assume 50), calculate basecurrent IB(mA) = IC(mA) / hFE= 10/50 = 0.2 mA.

3. Calculate RBknowing VBE= 0.7 V and VCE= 10 V.

4. Calculate RC.

Note that a 20% error in assuming a value of hFEwill only changeIC 1 mA.

RB(K) =VC VBE

IB= = = 46.5 K

(Volts) 10 0.70.2(mA)

(use RB= 46.4 K)

RC(K) =CC CIC

+ IB

= =

= 0.98 Ko s

(use RC= 1 K)

(mA)20 1010 + 0.2

VBE

VCE

VCC

RC

RB

IC + IB

ICIB

BIAS

CHOKE

EXAMPLE 1

(continues)


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5. The general equation for collector current can now be used to check thedesign using actual calculated resistor values.

6. After the transistor circuit is constructed, a quick measurement of VCEwillensure that the device is biased correctly. Since VCEis proportional to 1/hFE, the base resistor (RB) can be adjusted accordingly to compensate formanufacturing variations in hFE. This is easily done by varying RBwhilemonitoring VCEto properly obtain the desired collector current.

Example 1. (continued)

1. Determine supply voltage available (VCC= 20 V) and transistor biasoperating point (10 V, 10 mA). Assume ICBO= 0, VBE= 0.7 V.

2. Select VBBto be 2 V to ensure constant base current source.

3. Knowing the measured value of hFE, (or assume 50), calculate basecurrent IB(mA) - IC(mA) / hFE= 10 / 50 = 0.2 mA.

4. Calculate RBknowing VBE = 0.7 V and VBB= 2 V.

Table of Typical Resistor Values for Indicated Bias (VCC= 20 V)

Bias hFE RB RC

10 V, 5 mA 50 90.9K 1.96K

10 V, 10 mA 50 46.4K 1K

15 V, 15 mA 50 46.4K 348

VBE

VCE

VCC

VBB

RC

IC + IBB + IB

IC

BIAS

CHOKE

EXAMPLE 2

RB

IB

RB1

IBB + IB

RB2 IBB

RB(K) =VBB VBE

IB= = = 6.5 K

(Volts) 2 0.70.2(mA)

(use RB= 6.81 K)

(continues)


11/11

For product information and a complete list of distributors, please go to our web site: www.avagotech.com

Avago, Avago Technologies, and the A logo are trademarks of Avago Technologies, Limited in the United States and other countries.

Data subject to change. Copyright 2007-2010 Avago Technologies, Limited. All rights reserved. Obsoletes 5952-8376

5988-0424EN - May 11, 2010

5. Calculate RB2assuming IBB= 1 mA.

6. Now calculate RB1knowing IB, IBB, VBBand VCC.

7. Calculate RCknowing IC, IBB+ IB, VCCand VCE.

8. The general equation for collector current can now be used to check thedesign using actual calculated resistor values.

9. After the circuit is designed, RBmay be adjusted to obtain an exact valueof IC.

Example 2. (continued)

CC CE

IC+ IBB+ IBRC(K) = = = = 0.893 K

Vo ts

(use RC= 909)

(mA)20 10

10 + 1.2

VCE VBB

IBB+ IB

RB1(K) = = = = 6.66 K (Volts)

(use RB1= 6.19 K)

(mA)10 2

1.2

RB2(K) =VBBIBB

= =

= 2 K(Volts)

(use RB2= 1.96 K)

(mA)21

Table of Typical Resistor Values for Indicated Bias (VCC= 20 V)

Bias hFE RC RB RB1 RB2

10 V, 5 mA 50 1.62K

12.1K

7.5K

1.96K

10 V, 10 mA 50 909 6.81K 6.19K 1.96K

15 V, 15 mA 50 316 4.22K 10K 1.96K

an944-1 microwave transistor bias considerations

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