a zero voltage transition bidirectional

8/18/2019 A Zero Voltage Transition Bidirectional

1/11

3152 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 62, NO. 5, MAY 2015

A Zero-Voltage-Transition BidirectionalDC/DC Converter

Serkan Dusmez, Student Member, IEEE , Alireza Khaligh, Senior Member, IEEE , and

Amin Hasanzadeh, Member, IEEE

Abstract —A three-level (TL) bidirectional dc/dc con-verter is a suitable choice for power electronic systems witha high-voltage dc link, as the voltage stress on the switchesis half and inductor current ripple frequency is twice theconverter’s switching frequency. This study proposes azero-voltage transition (ZVT) TL dc/dc converter to en-able operation with higher switching frequency in orderto achieve higher power density and enhance efficiency.Two identical ZVT cells, each one composed of two res-onant inductors, a capacitor, and an auxiliary switch, are

integrated with the conventional TL topology to enablesoft switching in all four switches in both buck and boostoperation modes. In addition, a variable dead-time con-trol is proposed to increase the effective duty ratio atheavy loads. The proposed soft-switching feature has beendemonstrated under different loading conditions. A 650-Wprototype is designed and fabricated, which exhibits 95.5%at full load.

Index Terms —Bidirectional three-level (TL) converter,nonisolated dc/dc converter, power electronics, zero-voltage transition (ZVT) cell.

I. INTRODUCTION

IN bidirectional nonisolated buck/boost dc/dc converter ap-plications, a two-quadrant nonisolated buck/boost converter

has been preferred in many studies due to its simple two-switch

structure, low cost, and high efficiency [1]–[6]. However, in

high-voltage (400–600 V) and medium- to high-power appli-

cations such as the dc/dc bidirectional converter in electric

vehicles, the two-switch structure exhibits several shortcom-

ings: 1) low efficiency at light load due to high-power losses

Manuscript received June 22, 2014; revised September 22, 2014;accepted October 12, 2014. Date of publication February 19, 2015; dateof current version April 8, 2015. This work was supported in part bythe National Science Foundation Division of Electrical, Communicationsand Cyber Systems under Grant 1307228 and in part by Genovation

Cars, Inc.S. Dusmez was with the Power Electronics, Energy Harvesting and

Renewable Energies Laboratory, Electrical and Computer EngineeringDepartment, University of Maryland, College Park, MD 20742 USA. Heis now with the Electrical and Computer Engineering Department, TheUniversity of Texas at Dallas, Richardson, TX 75080 USA.

A. Khaligh is the Director of the Power Electronics Energy Harvestingand Renewable Energies Laboratory, Institute for System Research,and Electrical and Computer Engineering Department, University ofMaryland, College Park, MD 20742 USA (e-mail: [email protected]).

A. Hasanzadeh was with the Power Electronics, Energy Harvestingand Renewable Energies Laboratory, Electrical and Computer Engineer-ing Department, University of Maryland, College Park, MD 20742 USA.

Color versions of one or more of the figures in this paper are availableonline at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TIE.2015.2404825

associated with the parasitic capacitances of the switches;

2) utilization of insulated-gate bipolar transistors (IGBTs)

instead of metal–oxide–semiconductor field-effect transistors

(MOSFETs) in high-voltage applications, whereas the switch-

ing frequency of the converter is limited by the IGBT’s max-

imum switching frequency under hard switching. As a result,

the passive components such as the inductors and capacitors

render a large volume. To increase the input current ripple fre-

quency, interleaved structures are studied [7]–[9]. Interleavingreduces the size of the passive components but still requires

high-voltage rating switches. In order to equally distribute the

voltage stress among the switches and to use MOSFETs at

high voltages while still increasing the inductor current ripple

frequency to twice the switching frequency, a three-level (TL)

output voltage buck/boost dc/dc converter has been proposed

and suggested for use in high-voltage applications [10].

To increase power density through higher switching frequen-

cies without sacrificing converter efficiency, soft-switching

techniques are employed [11]–[13]. A common way to achieve

soft switching is to use interleaving circuits where the stored

energy in the interleaving inductor is used to discharge the

parasitic capacitances of the switches [14]–[17]. Similarly, an

auxiliary inductor can be coupled to the main inductor to reduce

the cost of the circuit [18]–[22].

Another method to effectively achieve soft switching is

through employing an auxiliary switch to store the switching

energy in the auxiliary capacitor, which is then used to soft-

switch the main switch either during turn-on or turn-off in-

stants. Soft-switching cells, consisting of an auxiliary switch,

a resonant inductor, and a capacitor, are common [23]–[26]. In

[23], the soft-switching cell is used to turn off the switch under

zero current and turn on the auxiliary switch under near-zero

current. Another ZCS was introduced in [24] for the boost con-

verter, which is then generalized for other converter types suchas buck, buck/boost, SEPIC, and Cuk. In [25], unified analysis

for these soft-switching cells was explored. In [26], a similar

soft-switching cell used in this paper has been proposed for

bidirectional nonisolated buck/boost dc/dc converter. However,

no soft-switched nonisolated TL converter has been explored in

the literature.

Reducing the switching losses is critical to achieve higher

efficiency. In this regard, this paper proposes a zero-voltage

transition (ZVT) TL bidirectional dc/dc converter for a wide

load range, as illustrated in Fig. 1. Two identical soft-switching

cells are deployed for each pair of switches, ensuring that all

four switches are turned on under zero voltage in both boost

0278-0046 © 2015 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.


2/11

DUSMEZ et al .: ZERO-VOLTAGE-TRANSITION BIDIRECTIONAL DC/DC CONVERTER 3153

Fig. 1. Proposed bidirectional TL dc/dc converter with an auxiliaryZVT cell.

and buck modes. Thus, the proposed ZVT TL converter can

be operated at higher switching frequencies, the input current

ripple frequency can be doubled, and the size of the inductor

can be significantly reduced.

The common issue with the soft-switching converters is

the limited soft-switching operation range due to the outputcurrent dependence of the soft-switching operation. When the

ZVT cell designed for a light-load condition operates under

heavy load, the effective on-time of the switches becomes less

than the reference. To partially compensate this negative effect

on the duty ratio of the main switch, the auxiliary switch

is controlled through adjusting the dead time with respect to

peak inductor current. The improvement on the effective duty

ratio using the proposed variable dead-time control is analyzed

and compared with the conventional fixed dead-time approach

within the content of this paper. Although the improvement on

the duty ratio is analyzed in this paper, the proposed technique

is a possible solution for secondary control objectives, such as

capacitor voltage balancing.

II. PROPOSED B IDIRECTIONAL Z VTTL DC/DC CONVERTER

The proposed converter achieves ZVT for all the switches

during turn-on instants in all operation modes. The ZVT cell

contains two resonant inductors, one resonant capacitor, and an

auxiliary switch to eliminate the turn-on losses of switches by

creating a resonance between the inductor and the capacitor.

The energy stored in the auxiliary inductor during normal

operation is transferred to the capacitor when the main switch is

turned off. This energy is then recycled through the body diodeof the main switch before it is turned on. An identical ZVT cell

TABLE ISUMMARY OF THE ZVT OPERATION

is placed between switches S 3 and S 4 to soft-switch the bottomtwo switches. The basic operation principle of the ZVT cell is

identical in each transition mode for both boost and buck op-

erations. Table I summarizes the ZVT cell operation depending

on the buck and boost modes, used resonant inductor, auxiliary

switch, and soft-switched MOSFET.

The equivalent circuits and operation waveforms are given

in Figs. 2 and 3, respectively. Here, the operation principle

will be given for the turning-on instant of S 2 in boost modewhen d > 0.5. It should be noted that Lr3 is not involvedduring this switching-state transition and can be assumed as

an additional contributing inductance to Lin. For the sake of symmetrical operation, the resonant inductors are chosen equal,

i.e., Lr1 = Lr2 = Lr3 = Lr4 = Lr.Operation Interval 1 [t < t0]: Before t0, both S 2 and S 3

are on, L in stores energy, and the converter operates in boostoperation mode of the TL converter. In this mode, iLr1(t0) = 0.If the on-time of the auxiliary switch is equal to the quarter of

the resonance period, the resonant capacitor voltage becomes

zero at t0, (vCr1(t0) = 0). This will be explained in lattersections. The current of Lr2 is equal to

iLr2(t) = iLr2(t7) + V in

Lin + 2Lrt. (1)

This time interval is determined by the effective on-time duty

ratio of S 2.Operation Interval 2 [t0 ≤ t < t1]: At t = t0, S 2 is turned

off. The current of Lr2 begins to resonate in the resonance tank composed of Lr1 − Lr2 − C r1. The sum of the voltages of theinductors vL1 + vL2 is equal to vcr1. During this short timeinterval, the parasitic capacitance of the upper switch, i.e., C 1,discharges over the output capacitor and transfers its energy

while C 2 is charged as vS 1(to) = V o/2, and vS 1 + vCr1 +

vS 2 = V o/2 during t01. The initial conditions can be noted as

iLr2(t0) = iLr2(t7) +

V in

Lin + 2Lr

t70 (2)

iLr1(t0) = 0 (3)

vCr1(t0) = 0. (4)

The natural frequency of the resonant tank is expressed as

ω = 1√ 2LrC r1

. (5)

Operation Interval 3 [t1 ≤ t < t2]: C 1 is completely dis-charged at the end of operation interval 2, as current begins


3/11


Fig. 2. Operation intervals and equivalent circuits. (a) Operation interval 1: t < t0. (b) Operation interval 2: t0 < t < t1. (c) Operation interval 3:t1 < t < t2. (d) Operation interval 4: t2 < t < t3. (e) Operation interval 5: t3 < t < t4. (f) Operation interval 6: t4 < t < t5. (g) Operation interval 7:t5 < t < t6. (h) Operation interval 8: t6 < t < t7.

commutating from Lr2 to Lr1, and D1 conducts. C 2 continuesto get charged. This interval occurs very quickly as C 1 is

typically on the order of several hundreds of picofarads andis not included in the calculations for simplicity. The energy

stored in Lr2 is transferred to resonant capacitor C r1 throughDa1. For this interval, the resonant inductor currents can beexpressed as

iLr2(t) = iLr2(t0)cos(ωt) (6)

iLr1(t) = iLr2(t0) (1 − cos(ωt)) . (7)

The resonant capacitor voltage can be written as

vCr1(t) = iLr2(t0)C r1ω

sin(ωt). (8)

At t = t2, iLr2 becomes zero as it transfers all its energy to theresonant capacitor

iLr2(t2) = 0. (9)

From this final state condition, t02 can be found as

t02 = π

2ω. (10)

Operation Interval 4 [t2 ≤ t < t3]: At t = t2, the resonantinductor current iLr1 is the same as iLr2(t0). The resonant ca-pacitor voltage can be found from the energy transfer between

vcr1 and iLr2 as

vCr1(t2) =

2LrC r1

i2Lr2(t0). (11)


4/11


Fig. 3. Switching scheme and circuit waveforms of the proposed ZVTconverter.

The energy stored in vcr1 will be used to achieve soft switchingin the latter stage. In this time interval, the resonant inductor

current iLr1 can be written as

iLr1(t) = iLr1(t2) −

V o/2 − V inLin + Lr

t. (12)

Operation Interval 5 [t3 ≤ t < t4]: Before S 2 is turned on,the auxiliary switch, i.e., S a1, is turned on to transfer the energystored in the capacitor to the resonant inductor, i.e., iLr2. S a1is turned on under zero current; thus, there is no switching

turn-on loss for the auxiliary switches. vLr2 is half the vCr1,

and iLr2 sinusoidally increases. C 2 is first quickly charged tovCr1 + V o/2 and then discharged as vCr1 decreases. At t = t3,the initial conditions are as follows: iLr2(t3) = 0, vcr1(t3) =vcr1(t2). In this interval, the resonant inductor currents andcapacitor voltage are

vCr1(t) = vCr1(t3)cos(ωt) (13)

iLr2(t) = − C rvCr1(t3)ω sin(ωt) (14)

iLr1(t) = iLr1(t3) + C rvCr1(t3)ω sin(ωt) (15)

t34 = π

2ω

. (16)

At the end of this mode, vCr1(t4) = vCr1(t0) = 0.

Operation Interval 6 [t4 ≤ t < t5]: When the energy storedin the resonant capacitor is completely transferred via S a1,C 2 quickly discharges to zero. This time interval can beexpressed as

t45 =

8Lr iLr2(t4) − iLr2(t4)2

− C 2V 2o8L

r V o

. (17)

At t = t5, the current of D1 increases to approximately twicethe load current.

Operation Interval 7 [t5 ≤ t < t6]: When vS 2 reaches to−0.7 V, the body diode of S 2 conducts. This time period isthe ZVT period. S 2 can be turned on at any time during thisinterval. At t = t5, the initial conditions are

iLr2(t5) = −

C r1v2

Cr1(t2)

2Lr(18)

iLr1(t5) = iLr1(t4) + C r1vCr1(t4)ω sin(ωt4) (19)

where vLr2 is equal to V o/4, and iLr2 linearly decreases, i.e.,

iLr2(t) = iLr2(t5) + V o4Lr

(20)

iLr1(t) = iLr1(t5) − V o4Lr

t. (21)

Operation Interval 8 [t6 ≤ t < t7]: This interval starts afterthe main switch S 2 is turned on. At t = t6, the main switch S 2is turned on while its body diode is conducting. The ZVT can

be achieved at this instant. With the assumption of negligiblet45 in comparison to t56, t46 ∼= t56, the ZVT time interval canbe expressed as

t56 = 4Lr (iLr2(t6) − iLr2(t5))

V o. (22)

First, the current of the body diode of S 2 reaches to zero. In thistime interval, S 2 begins to conduct, and input current commutesfrom Lr1 to Lr2. At t = t6, iLr1 is

iLr1(t6) = iLr1(t5) − V o4Lr

t56. (23)

The resonant inductor currents for this interval can be

expressed as

iLr1(t) = iLr1(t6) − V o4Lr

t (24)

iLr2(t) = iLr2(t6) + V o4Lr

t. (25)

At t = t7, the input current commutation from Lr1 to Lr2 iscompleted, and iLr1 becomes zero. This time interval can beexpressed as

t67 = 4Lr (iLr2(t7) − iLr2(t6))

V o. (26)


5/11


III. DESIGN C ONSIDERATIONS FOR ZVT FEATURE

The time between the turning-off action of S a1 and theturning-on action of the main switch S 2, which is denoted byt46, should be as short as possible to achieve ZVT with thesmallest amount of energy stored in the resonance tank. This

would minimize the size of the resonant tank elements and

allows achieving higher efficiency as the circulating current in

the resonant tank would be less. To achieve ZVT, the auxiliary

switch should be turned on at least t36 before the main switchis turned on. It should be highlighted that t34 is the timeduration in which the stored energy in the auxiliary capacitor is

transferred to the auxiliary inductor, and t45 is the time requiredto discharge the parasitic capacitance of the switch. For the

ideal case, the minimum required time is t35. t56 is the timeduration added to guarantee the soft switching of the main

switch taking the mismatch between the actual and datasheet

values of the switch. Considering this added time margin, the

auxiliary switch should be turned on at least t36 before the

main switch is turned on. Once t46 is determined, Lr can becalculated accordingly.

A. Constant Power Loads

The minimum value of Lr can be determined for constantpower loads with predefined P o, V o, V in, and L in and steady-state duty ratio (D). For simplicity in calculations, it is assumedthat iLr2 does not vary during t45, and |iLr2(t5)| = |iLr2(t0)|,which yields

Lr ≥ V o4iLr2(t0)

t46. (27)

In order to have zero voltage across C r1 from t = t4 to t =t0, the reverse resonance time, i.e., t34, should be equal toone fourth of the natural resonance period, i.e., t02, which isinitiated by turning on S a1. If t34 is forced to be shorter thant02, vCr1(t0) would be higher than zero. C r1 can be calculatedfrom (5) and (10), by equalizing t34 and t02 time periods, as

C r1 = 2

Lrπ2t202. (28)

Since two quarter resonance periods take place during the off-

time of the switch, their sum should be shorter than (1 − D)T s,where T s represents the switching period

t02 < T s(1 − D)

2 . (29)

As the quarter resonance period t02 increases, the requiredcapacitance also increases while the voltage stress on S 2 re-duces. The additional voltage stress across S 2 is equal to that of vcr1(t2) and can be expressed as

vcr1(t2) = iLr2(t0)

2LrC r1

. (30)

B. Variable Power Loads

From (27), it can be observed that the ZVT feature is depen-dent on the load current as it determines the stored energy in the

Fig. 4. Illustration of ZVT operation with fixed and variable dead timesfor evaluation of t67. (a) Fixed dead time. (b) Variable dead time.

resonant inductor. Therefore, for wide-load-range applications,

the minimum input current at which the ZVT feature is desired

should be determined for the given application, i.e.,

tZV T ≥ tdt + tm (31)where tdt denotes the selected dead time, and tm is the timemargin, representing the time from turn-on instant of the main

switch until iLr2 becomes zero, and should be included in

the calculations ensuring that the switch is switched underzero voltage. Minimum Lr for variable power loads can bedetermined as

Lr ≥ V o4iLr2(t0)

tZV T . (32)

In case the auxiliary switch is turned off tdt seconds beforethe main switch is turned on, D1 would conduct for the timeduration of tZV T .

The selected Lr should be sufficiently large to achieve softswitching for S 2 at minimum load current. However, the se-lected Lr would cause losing effective on-time of the switch

as the load gets heavier. This issue is illustrated in Fig. 4(a).Here, it is assumed that the load current is larger than I _min,and Lr is found using (27) for the ideal case, assuming tdt = 0.As shown in the figure, the reduction in the duty ratio, which

is denoted by DL, increases under heavy load. The reductionin the duty cycle should be compensated similar to dead-time

compensation applied in three-phase inverters [27]. To reduce

this effect, the phase of the gating signal of the auxiliary switch

should be varied such that the main switch is turned on right

before the current of the main-switch body diode reaches zero.

In this regard, a variable dead-time control is proposed in this

paper, as shown in Fig. 4(b). The gating signals for the auxiliary

switches are practically achieved by right-aligned pulsewidth

modulation (PWM) signal with inserted fixed dead time. Atheavy loads, instead of turning off the auxiliary switch with


6/11


TABLE IIDESIGN PARAMETERS

fixed tdt seconds, before the main switch is turned on, thedischarging of the resonant capacitor can be initiated earlier

by adjusting the dead time between the main and auxiliary

switch signals. Hence, the resonant inductor current crosses

zero tm seconds after the main switch is turned on. At the idealcase, i.e., tm = 0, this could reduce DL to DL/2, as illustratedin Fig. 4(b).

It is important to note that the proposed variable dead-time

approach can be used for achieving supplementary control

objectives such as dc-link capacitor voltage balancing. The vari-

able dead time has not been considered in any soft-switchingliterature because it may not be necessary in two-level convert-

ers. One of the difficulties in TL converters is the unbalanced

dc-link capacitor voltages, which may occur due to the un-

symmetrical circuit layout, differences in the equivalent series

capacitance of capacitors, delays in the gate driver circuit, etc.

The unbalanced voltage problem could be solved by controlling

the duty cycles of the main switches independently, letting ca-

pacitors charge/discharge more or less with respect to the other

dc-link capacitors. However, the common approach is to use

the same PWM duty-cycle generator for both of the switches

due to its simple control and, more importantly, for stable

operation. In this regard, developed secondary variable dead-

time control can be adopted to balance the dc-link capacitors aswell as contribute to the duty-cycle compensation, correcting

the unbalanced ZVT current injection due to unsymmetrical

ZVT circuit parameters and so on. However, in this paper, only

partial duty-cycle compensation has been studied, and capacitor

voltage balancing is out of the scope of this paper.

C. Example Application

This section provides a ZVT cell design procedure for a

variable power load. The example design specifications are

provided in Table II. The converter operates in CCM at full-

load range.The first design parameter to be determined is tZV T . As pre-viously discussed, t45 is relatively short, on the order of severalnanoseconds. The variation of t45 at minimum load current ac-cording to Lr is plotted in Fig. 5(a) for maximum and minimumoutput voltages. As the output voltage increases, more energy

is stored in the parasitic capacitances of the switches, which

extends the discharge time to 5 ns. At P o = 250 W, the deadtime, i.e., tdt, is chosen as 30 ns. This would be 5 ns for an idealcase; however, it is reasonable to insert a dead time to ensure

that soft switching will be robust to the mismatch of the actual

and datasheet values as well as to compensate the previous

simplifications. Since the switching period is 5 µs and consid-

ering the turn-on and turn-off time of MOSFETs, tm is chosenas 20 ns to ensure that the auxiliary switch turns off 30 ns

before the main switch is turned on. This way, the resonant

current will become zero after tm + tdt = 50 ns. The ZVTtime period corresponds to 1% of the switching period. The

reduction in the duty cycle, i.e., t67/DT s, can be expressed as

DL = (tdt + 2tm)

DT s. (33)

To find Lr, the worst condition should be considered; thatis, P o = P o,min, and V o = V o,max. Assuming that Lin is largeenough and input current is constant, iLr2(t0) in (32) can beapproximately equal to the input current. For this example, Lris calculated as 1 µH. The highest ZVT time, i.e., t57/2, is50 ns for minimum load at V o = V o,max. At maximum load,the ZVT period increases to 130 ns. For V o = V o,min, the ZVTperiod for minimum load is 55 ns, and at maximum output

power, it increases up to 145 ns. The same amount of time is

required for the main switch current to reach the input current,

i.e., t57/2. For a fixed tdt of 30 ns, t67 can be calculatedusing t57

−tdt as 70 and 230 ns for the lightest and heaviest

load conditions at V o = V o,max, respectively. Similarly, whenV o = V o,min, t67 becomes 80 and 260 ns for the lightest andheaviest load conditions, respectively.

In CCM, the duty ratios for V o = V o,min and V o = V o,maxare 44% (corresponding to 2.22 µs) and 50% (corresponding to2.5 µs). Using (33), DL is calculated as 3.6% and 11.6% forlight- and heavy-load conditions at V o = V o,min, respectively.Similarly, for V o = V o,max, it is calculated as 2.8% and 9.6%,respectively. Using the proposed variable dead-time approach,

DL can be effectively reduced as shown in Fig. 5(b). Thedead-time curve used to accomplish DL reduction is given inFig. 5(c).

The acquired data are summarized in Table III. It can be seenthat DL is reduced from 11.6% to 7.4% for low output voltageand heavy-load conditions. If tm is chosen smaller at the initialstep, DL can be reduced even further, to half the DL valuewith fixed dead time at the ideal case, when tm = 0. However,further duty-cycle compensation is still required.

Once tdt and Lr are determined, C r can be determinedfrom (28) and (29). As mentioned before, the pulsewidth of

S a1 should be equal to t02 to have zero voltage across vcr att = t0. When a dead time is inserted, (29) should be modifiedto incorporate tdt. Thus,

t02 <

T s(1

−D)

2 − tdt. (34)The highest tdt is achieved when output power is maximum

and output voltage is lowest. However, for this case, T s(1 −D)/2 − tdt for the V o = V o,max condition is smaller; hence,t02 should be determined based on maximum output voltage.This determines the upper bound for choosing t02, which iscalculated as 1.14 µs. The variation of the capacitance valueand voltage stress of the resonant capacitor for various t02values from 0.1 to 1.14 µs is given for the heaviest loadcondition in Fig. 6. Here, C r can be determined based on thetradeoff between the capacitance value and the voltage stress.

In addition, there should be sufficient margin for duty-cycle

compensation. Once t02 is determined, the duty ratio of S a1can be found by dividing t02 by T s.


7/11


Fig. 5. Curves for the ZVT operation. (a) Switch parasitic capacitance discharge time according to various Lr . (b) Reduction in effective dutyratio for various output power values when Lr = 1 µH. (c) Required dead time for the proposed variable dead-time control at various outputpower values.

TABLE IIIACQUIRED DATA FOR ZVT OPERATION

Fig. 6. Capacitance and voltage stress variation of the resonant ca-pacitor at various t02 values.

IV. SIMULATION AND E XPERIMENTAL R ESULTS

Here, simulation and experimental results are provided toverify the operation of the converter. The switching frequency

of the main and auxiliary switches is 200 kHz, where the

frequency of the inductor and output capacitor current ripples

is 400 kHz. The resonant inductors, i.e., Lr , are 1 µH, andthe resonant capacitors, i.e., C r, are 100 nF each. Only boostoperation mode is presented here, as buck operation is similar

to boost operation.

The simulation waveforms for d = 0.53 and d = 0.4 usingideal switches are given in Fig. 7(a) and (b), respectively. In

boost operation, only S 2 and S 3 are switched while body diodesof S 1 and S 4 conduct. The additional voltage stress on themain switches can be clearly observed. The duty ratio of the

auxiliary switch calculated from the analysis match with onefourth of the resonant period, such that the resonant capacitor

voltage is almost zero. The simulation waveforms agree with

the theoretical analysis.

The simulation waveforms for boost operation utilizing non-

ideal switches when d = 0.4 are presented in Fig. 7(c). In

the figures, it can be clearly seen that C oss of the switches,particularly that of the auxiliary switches, causes to form a

resonant tank between Lr − Lr − C r − C oss. Without includ-ing the internal resistances, this oscillation keeps circulating

back and forth. As seen from the input inductor voltage, this

oscillation causes inductor current to have a small oscillating

current in addition to the linearly increasing current.

A 650-W proof-of-concept prototype, shown in Fig. 8, is

designed to serve as a proof-of-concept and to show ZVT oper-

ation of the circuit. The circuit parameters are the same as those

of the simulation. The current ratings of the switches are chosen

based on the root mean square (rms) value. As the ZVT circuit

does not impose additional current stress on the main switches,the main-switch rms current can be found as in a traditional

TL converter [10]. The voltage rating of the main switches is

the sum of half the output voltage and the resonant capacitor

voltage, which was given in (30). It can be expressed as

V main = V o

2 + iLr2(t0)

2LrC r1

. (35)

On the other hand, the current rating of the auxiliary switch

is equal to

I aux,rms = iLr2(t0) 2 + π8ωT s

. (36)

The component parameters used in the experiments are given

in Table IV. Fig. 9 presents the input inductor current waveform

and gate–source voltages of S 2 and S 3. This figure illustratesthe boost mode operation of the TL converter, where the current

ripple frequency is twice the switching frequency.

The ZVT operation of the converter has been presented in

the experimental results provided in Figs. 10 and 11. Fig. 10

presents the ZVT operation during boost mode of operation

for d


8/11


Fig. 7. Simulation waveforms for boost operation mode. (a) d = 0.53 with ideal switches. (b) d = 0.4 with ideal switches. (c) d = 0.4 with nonidealswitches.

Fig. 8. Photograph of the designed experimental prototype.

TABLE IVCOMPONENT PARAMETERS


9/11


Fig. 9. Experimentalwaveforms of normal boostoperation; gate–sourcevoltages S 2 and S 3 and inductor current for (a) d < 0.5 and (b) d > 0.5.

voltage waveforms are presented, from which ZCS operationcan be clearly seen. The experimental waveforms agree with

the simulation results presented in Fig. 7(c). Different than

simulation, the oscillations are damped due to the internal

resistances of the components in the experiments. Likewise,

ZVT operation for boost mode for d


10/11


Fig. 11. Experimental waveforms of ZVT operation boost mode whend < 0.5 (high current). (a) Gate–source and drain–source voltages andcurrent of S 2 and gate–source voltage of S a1 for two switching periods.

(b) Zoomed-in snapshot. (c) Gate–source and drain–source voltagesand current of S a1 and gate–source voltage of S 2.

In addition, the very high frequency oscillation observed on

the gate signals is due to the high parasitic capacitance of the

differential probe. Since the grounds of the probes should be

isolated, a regular and a differential voltage probe are used

for sensing gates of the signals. To prove this claim, the

regular and differential probes have been connected to the gate–

source terminal of S 2 (purple waveform) and S a1 (green wave-form) in Fig. 10(b), respectively. As it can be seen, the very

high frequency oscillations are only on the S a1 gate signal.

In Fig. 10(c), these two probes are replaced. This time, theoscillations are observed on the S 2 gate signal. Thus, the high-

Fig. 12. Efficiency curve of (a) the proposed topology at V o = 180 Vand V o = 220 V and (b) the hard-switched TL converter.

frequency oscillations on the gate signals are completely due to

the differential probe parasitics.

The efficiency curves for two different output voltage levels,

along with hard-switched converter efficiency, are presentedin Fig. 12. It can be seen that the designed prototype ex-

hibits 95.5% at full load, whereas the peak efficiency of the

hard-switched converter is 91%. Considering the typical effi-

ciencies of industrial nonisolated bidirectional dc/dc converters,

which are between 90% and 95%, the proposed ZVT converter

provides competitive efficiency to its counterparts. In addition,

it should be noted that the proposed ZVT converter inherently

reduces the required input boost inductance and output filter

capacitance by half due to the TL structure, in comparison to

the state-of-the-art converters. Moreover, it allows using low-

voltage rated switch.

V. CONCLUSION

In this paper, a ZVT bidirectional TL dc/dc converter, em-

ploying two identical ZVT cells to fully soft-switch all four

switches in bidirectional power flow during turning-on instants

of the main switches, has been introduced. The design proce-

dures of the ZVT cell components were provided. Furthermore,

an actively controlled variable dead-time approach has been

introduced to minimize the reduction in the duty ratio due

to the soft-switching period, during the converter’s operation

under heavy loads. A 650-W prototype has been designed to

demonstrate the operation of the converter. The peak efficiency

at 200-kHz switching frequency is recorded as 95.5%.

REFERENCES

[1] A. Khaligh, A. Miraoui, and D. Garret, “Guest editorial: Special sectionon vehicular energy-storage systems,” IEEE Trans. Veh. Technol., vol. 58,no. 8, pp. 3879–3881, Oct. 2009.

[2] A. Khaligh and Z. Li, “Battery, ultracapacitor, fuel-cell, hybrid energystorage systems for electric, hybrid electric, fuel cell, plug-in hybridelectric vehicles: State-of-art,” IEEE Trans. Veh. Technol., vol. 59, no. 6,pp. 2806–2814, Jul. 2010.

[3] O. Onar and A. Khaligh, “A novel integrated magnetic structure baseddc/dc converter for hybrid battery/ultra-capacitor energy storage systems,”

IEEE Trans. Smart Grid , vol. 3, no. 1, pp. 296–307, Mar. 2012.[4] A. A. Ferreira, J. A. Pomilio, G. Spiazzi, and L. de Araujo Silva, “Energy

management fuzzy logic supervisory for electric vehicle power suppliessystem,” IEEE Trans. Power Electron., vol. 23, no. 1, pp. 107–115,Jan. 2008.


11/11


[5] S. Dusmez and A. Khaligh, “A supervisory power splitting approach fora new ultracapacitor-battery vehicle deploying two propulsion machines,”

IEEE Trans. Ind. Informat., vol. 10, no. 3, pp. 1960–1971, Aug. 2014.[6] S. Dusmez and A. Khaligh, “Generalized technique of compensating low-

frequency component of load current with parallel bidirectional dc/dcconverter in electric vehicles,” IEEE Trans. Power Electron., vol. 29,no. 11, pp. 5892–5904, Nov. 2014.

[7] H. Wang, “Highly efficient SiCbased onboard chargers for plug-in electric

vehicles,” Ph.D. dissertation, Elect. Comput. Eng. Dept., Univ. Maryland,College Park, MD, USA, Jul. 2014.

[8] H. Kosai, J. Scofield, S. McNeal, B. Jordan, and B. Ray, “Design andperformance evaluation of a 200 C Interleaved Boost Converter,” IEEE

Trans. Power Electron., vol. 28, no. 4, pp. 1691–1699, Apr. 2013.[9] M. B. Camara, H. Gualous, F. Gustin, A. Berthon, and B. Dakyo, “DC/DC

Converter design for supercapacitor and battery power management inhybrid vehicle applications–Polynomial control strategy,” IEEE Trans.

Ind. Electron., vol. 57, no. 2, pp. 587–597, Feb. 2010.[10] P. J. Grbovic, P. Delarue, P. L. Moigne, and P. Bartholomeus, “A bidirec-

tional three-level DC–DC converter for the ultracapacitor applications,” IEEE Trans. Ind. Electron., vol. 57, no. 10, pp. 3415–3430, Oct. 2010.

[11] J.-W. Yang and H.-L. Do, “High-efficiency bidirectional dc–dc converterwith low circulating current and ZVT characteristic throughout a fullrange of loads,” IEEE Trans. Ind. Electron., vol. 61, no. 7, pp. 3248–3256,Jul. 2014.

[12] A. K. Rathore and U. R. Prasanna, “Analysis, design, experimental results

of novel snubberless bidirectional naturally clamped ZCS/ZVS current-fed half-bridge DC/DC converter for fuel cell vehicles,” IEEE Trans. Ind.

Electron., vol. 60, no. 10, pp. 4482–4491, Oct. 2013.

[13] U. R. Prasanna and A. K. Rathore, “Extended range ZVT active-clampedcurrent-fed full-bridge isolated dc/dc converter for fuel cell applications:Analysis, design and experimental results,” IEEE Trans. Ind. Electron.,vol. 60, no. 7, pp. 2661–2672, Jul. 2013.

[14] W. Yu, H. Qian, and J.-S. Lai, “Design of high-efficiency bidirectionalDC–DC converter and high-precision efficiency measurement,” IEEE Trans. Power Electron., vol. 25, no. 3, pp. 650–658, Mar. 2010.

[15] L. Ni, D. J. Patterson, and J. L. Hudgins, “A high power, current sen-sorless, bi-directional, 16-phase interleaved, DC–DC converter for hybridvehicle application,” in Proc. IEEE ECCE , 2010, pp. 3611–3617.

[16] J. Zhang, J.-S. Lai, R.-Y. Kim, and W. Yu, “High-power density design of asoft-switching high-power bidirectional DC–DC converter,” IEEE Trans.Power Electron., vol. 22, no. 4, pp. 1145–1153, Jul. 2007.

[17] B.-R. Lin and C.-H. Chao, “Analysis, design, implementation of a soft-switching converter with two three-level PWM circuits,” IEEE Trans.Power Electron., vol. 28, no. 4, pp. 1700–1710, Apr. 2013.

[18] H.-L. Do, “Nonisolated bidirectional zero-voltage-switching DC–DCconverter,” IEEE Trans. Power Electron., vol. 26, no. 9, pp. 2563–2569,Sep. 2011.

[19] S. D. G. Jayasinghe, D. M. Vilathgamuwa, and U. K. Madawala, “Diode-clamped three-level inverter-based battery/super-capacitor direct integra-tion scheme for renewable energy systems,” IEEE Trans. Power Electron.,vol. 26, no. 12, pp. 3720–3729, Dec. 2011.

[20] R. K. Singh and S. Mishra, “A magnetically coupled feedback-clampedoptimal bidirectional battery charger,” IEEE Trans. Ind. Electron., vol. 60,no. 2, pp. 422–432, Feb. 2013.

[21] H. Wu, J. Lu, W. Shi, and Y. Xing, “Nonisolated bidirectional DC–DCconverters with negative-coupled inductor,” IEEE Trans. Power Electron.,vol. 27, no. 5, pp. 2231–2235, May 2012.

[22] S.-S. Lee, “Step-down converter with efficient ZVT operation with load

variation,” IEEE Trans. Ind. Electron., vol. 61, no. 1, pp. 591–597,Jan. 2014.

[23] H. Bodur and A. F. Bakan, “An improved ZCT-PWM DC–DC converterfor high-power and frequency applications,” IEEE Trans. Ind. Electron.,vol. 51, no. 1, pp. 89–95, Feb. 2004.

[24] H.-S. Choi and B. H. Cho, “Novel Zero-Current-Switching (ZCS) PWMswitch cell minimizing additional conduction loss,” IEEE Trans. Ind.

Electron., vol. 49, no. 1, pp. 165–172, Feb. 2002.[25] J. Abu-Qahouq and I. Batarseh, “Unified steady-state analysis of soft-

switching DC–DC converters,” IEEE Trans. Power Electron., vol. 17,no. 5, pp. 684–691, Sep. 2002.

[26] P. Das, B. Laan, S. A. Mousavi, and G. Moschopoulos, “A nonisolatedbidirectional ZVS-PWM active clamped DC–DC converter,” IEEE Trans.Power Electron., vol. 24, no. 2, pp. 553–558, Feb. 2009.

[27] Z. Zhang and L. Xu, “Dead-time compensation of inverters consideringsnubber and parasitic capacitance,” IEEE Trans. Power Electron., vol. 29,no. 6, pp. 3179–3187, Jun. 2014.

Serkan Dusmez (S’11) received the B.S.(Hons.) and M.S. degrees in electrical engineer-ing from Yildiz Technical University, Istanbul,Turkey, in 2009 and 2011, respectively. He re-ceived the M.S. degree in electrical engineeringfrom Illinois Institute of Technology, Chicago, IL,USA, in 2013. He is currently working towardthe Ph.D. degree at The University of Texas atDallas, Richardson, TX, USA.

During 2012–2013, he was a Faculty Re-search Assistant with the Power Electronics,

Energy Harvesting and Renewable Energies Laboratory, Electrical andComputer Engineering Department, University of Maryland, CollegePark, MD, USA. He is the author/coauthor of over 35 journal andconference papers. His research interests include design of power

electronic interfaces and energy management strategies for renewableenergy sources, integrated power electronic converters for plug-in elec-tric vehicles, and real-time fault diagnosis of power converters.

Alireza Khaligh (S’04–M’06–SM’09) is the Di-rector of the Power Electronics, Energy Har-vesting and Renewable Energies Laboratoryin the Electrical and Computer Engineer-ing Department and the Institute for Sys-tems Research at the University of Maryland(UMD), College Park,MD,USA. He is an author/ coauthor of over 130 journal and conferencepapers.

Dr. Khaligh is a recipient of various awards

and recognitions, including the 2013 GeorgeCorcoran Memorial Award from the Electrical and Computer Engineer-ing Department of UMD, the 2013 Best Vehicular Electronics Award fromthe IEEE Vehicular Technology Society, and the 2010 Ralph R. TeetorEducational Award from the Society of Automotive Engineers. He is theProgram Chair of the 2015 IEEE Applied Power Electronics Conferenceand Exposition. He was the General Chair of the 2013 IEEE Transporta-tion Electrification Conference and Exposition and the Program Chairof the 2011 IEEE Vehicle Power and Propulsion Conference. He is anEditor of the IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY andan Associate Editor of the IEEE T RANSACTIONS ON T RANSPORTATIONELECTRIFICATION. He was a Guest Editor or a Guest Associate Editorof various IEEE Transactions including the IEE E TRANSACTIONS ONPOWER ELECTRONICS and the IEEE TRANSACTIONS ON INDUSTRIALELECTRONICS.

Amin Hasanzadeh (M’11) received the B.Sc. degree from Sharif Uni-versity of Technology, Tehran, Iran, the M.Sc. degree from Khajeh NasirToosi University of Technology, Tehran, Iran, and the Ph.D. degree fromSharif University of Technology, in 1999, 2001, and 2009, respectively,all in electrical engineering.

He was a Post-Doctoral Research Associate with the Power Electron-ics, Energy Harvesting and Renewable Energies Laboratory, Electricaland Computer Engineering Department, University of Maryland, CollegePark, MD, USA.

a zero voltage transition bidirectional

Documents