we rely on standard methods for matrix fill. While we are able tomaintain a memory complexity which scales approximately asO(N log N) for targets smaller than about 25�, the reliance on astandard implementation of the matrix-fill makes the CPU cost ofthe algorithm scale as O(N2). We are currently investigatingmodifications of the LOGOS algorithm which may remove thislimitation.

It is expected that the algorithms reported above can be furtherimproved to reduce the complexity of the resulting LOGOS rep-resentation at low frequencies from O(N log N) to O(N) byallowing some overlap between the LOGOS modes. Finally, weare also considering the possibility of reducing the complexityscaling of the LOGOS algorithm for electrically large targetsthrough the use of a complementary set of expansion functionsreferred to as radiating LOGOS modes.

REFERENCES

1. A.F. Peterson, S.L. Ray, and R. Mittra, Computational methods forelectromagnetics, IEEE Press, New York, 1998.

2. W.C. Chew, J.M. Jin, E. Michielssen, and J. Song (Eds.), Fast andefficient algorithms in computational electromagnetics, Artech House,Boston, 2001.

3. R.J. Adams, F.X. Canning, F. Mev, and B.A. Davis, Beam-transformmethod for plane wave response matrices, Progress Electromagn Res2005 (to appear).

© 2005 Wiley Periodicals, Inc.

A NOVEL SURFACE-WAVE ANTENNADESIGN USING A THIN PERIODICALLYLOADED GROUND PLANE

Fan Yang,1,2 Amir Aminian,1 and Yahya Rahmat-Samii11 Electrical Engineering DepartmentUniversity of California at Los AngelesLos Angeles, CA 900952 Electrical Engineering DepartmentUniversity of MississippiUniversity, MS 38677

Received 30 April 2005

ABSTRACT: This paper presents a novel surface-wave antenna thatrealizes a low-profile configuration using a thin patch-loaded groundedslab, which has an in-phase reflection coefficient for plane waves, butno band gap for surface waves. When a dipole antenna is horizontallypositioned near this artificial ground plane, it works like a transducerthat converts the energy from the feeding probe into surface waves inthe ground plane. The propagation and diffraction of the surface wavesgenerate a monopole-type radiation pattern. The attractive antennaheight, which is less than 10% of a typical monopole antenna, is desir-

able for wireless-communication systems. The performance of the sur-face-wave antenna is demonstrated through both the FDTD simulationsand the measured results. © 2005 Wiley Periodicals, Inc. MicrowaveOpt Technol Lett 47: 240–245, 2005; Published online in Wiley Inter-Science (www.interscience.wiley.com). DOI 10.1002/mop.21136

Key words: artificial ground plane; dipole antenna; low profile; mono-pole antenna; surface waves

1. INTRODUCTION

There has been an increasing interest in periodic surfaces such assoft/hard surfaces, artificial magnetic conductors, and electromag-netic band-gap structures [1, 2]. Due to their inherent periodicity,these artificial surfaces exhibit unique electromagnetic character-istics which cannot be obtained by a conventional perfect electricconductor (PEC) surface. They have been applied in various an-tenna designs to improve antenna performances such as increasedgain, decreased back lobes, reduced mutual coupling, and theachievement of low profiles [3–5].

The mushroomlike EBG surface [6] is a useful artificial surfacethat provides a distinct stop-band for surface waves and an in-phase reflection coefficient for plane waves. It is observed thatwhen the vertical vias in a mushroomlike structure are removed,the band-gap for surface waves disappears, whereas the reflectioncoefficient remains the same for normally incident plane waves.Thus, this artificial surface, a thin grounded slab loaded withperiodic patches, is capable of providing novel functionality forantenna designs.

This paper presents an interesting application of the patch-loaded grounded slab in antenna engineering. When the artificialsurface is used as a ground plane for a horizontal dipole, becauseof its in-phase reflection coefficient, the antenna can achieve agood return loss even with a very low-profile configuration. How-ever, since the patch-loaded grounded slab does not have a bandgap for surface waves, strong surface waves can be excited in theground plane. Therefore, the dipole here operates more like atransducer rather than a radiator, and converts the energy from thefeeding probe to the surface waves. The propagation and diffrac-tion of the surface waves generate a radiation pattern similar to thatof a vertical monopole antenna. Thus, this radiating structure, ahorizontal dipole near a patch-loaded grounded slab, can be iden-tified as a surface-wave antenna. Compared to a typical monopoleantenna, the height is significantly reduced while the radiationpatterns remain similar. The radiation mechanism and the antennaperformance are validated by both the FDTD simulations andexperiments.

2. CHARACTERIZATIONS OF COMPLEX ARTIFICIALSURFACES

Figure 1 shows two complex artificial surfaces: a mushroomlikeEBG surface and a grounded dielectric slab loaded with periodicpatches. The important difference between these two structures isthe existence of vertical vias in the EBG surface that connect thepatches to the ground plane.

To identify the electromagnetic properties of these structures,the finite-difference time-domain (FDTD) method is used to sim-ulate their performance. Two electromagnetic properties of thesestructures are of special interest: the surface-wave property and theplane-wave property. A spectral FDTD method [7] is developed tocharacterize the former feature and a periodic standard FDTD codewith a plane-wave excitation [8] is used to analyze the latter one.

The dimensions of the analyzed surface are

W � 0.10�, g � 0.02�, h � 0.04�, �r � 2.94, (1)

Figure 6 Matrices Q (left) and R (right) obtained from a (numerically)exact QR factorization of the matrix Z depicted in Fig. 3

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where W is the width of the square patch, g is the gap width, h isthe substrate thickness, and �r is the dielectric constant of thesubstrate. The vias’ radius in the EBG structure is 0.005 � and � �75 mm, the free space wavelength at 4 GHz, is used as a referencelength to define the physical dimensions of artificial surfaces and

antennas studied in this paper. The selection of these parametersfollows the guidelines in [9].

Figure 2 shows k � � modal diagrams of the two artificialsurfaces characterized using the spectral FDTD method. The ver-tical axis shows the frequency and the horizontal axis representsthe values of transverse wave numbers (kx, ky). In the spectralFDTD method [7], each simulation outputs the frequencies of thesurface-wave modes for a given combination of wavenumbers (kx,ky). The simulation is repeated for 30 combinations of kx and ky inthe Brillion zone, and the corresponding frequencies of surfacewave modes are extracted and plotted in Figure 2. Each point in themodal diagram represents a certain surface-wave mode. For themushroomlike EBG structure, a band gap is observed between 3.5GHz and 5.9 GHz, which means the surface waves inside thisfrequency range will be suppressed. In contrast, for the patch-loaded grounded slab, since the vertical vias are removed, the bandgap disappears and the first surface-wave mode (TMz dominant)can exist in above frequency range.

Figure 2 K � � modal diagrams of two artificial surfaces: (a) a mush-roomlike EBG structure and (b) a patch loaded grounded slab. Along thehorizontal axis, �: kx � 0, ky � 0; X: kx � �/a, ky � 0; M: kx � �/a,ky � �/a; a � W � g. It is observed that the frequency band-gapdisappears when the vias in the EBG structure are removed

Figure 1 Geometry of two complex artificial surfaces: (a) a mushroom-like electromagnetic band-gap (EBG) structure and (b) a grounded dielec-tric slab loaded with periodic patches. The difference between (a) and (b)is the existence of vias connecting the patches and the ground plane in theEBG case

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The reflection phases of these two surfaces for a normallyincident plane wave are compared in Figure 3. The reflection phaseis defined as the phase of the reflected electric field normalized tothe phase of the incident electric field at the reflecting surface. Theincident plane wave travels along the �z direction, normallyilluminating the complex surfaces. It is known that a perfectelectric conductor (PEC) has an 180° reflection phase and a perfectmagnetic conductor (PMC) has a 0° reflection phase. The reflec-tion phases of the two artificial surfaces change continuously from180° to �180° as the frequency increases. A 90° reflection phaseis achieved around 4.6 GHz and a 0° reflection phase is realizedaround 5.8 GHz. It is observed from the comparison that removingthe vias has little effect on the reflection phase feature and bothsurfaces have very similar reflection-phase characteristics for nor-mal incidence.

In summary, the patch-loaded grounded slab has a similarreflection phase as the mushroomlike EBG surface for normallyincident plane waves, but it does not have a band gap for surfacewaves.

3. LOW-PROFILE SURFACE-WAVE ANTENNA

The mushroomlike EBG surface was successfully used as theground plane for a horizontal dipole antenna to radiate efficientlywith a low-profile configuration [6, 9, 10]. After discovering thesimilarities and differences between the patch-loaded groundedslab and the EBG surface, it is interesting to examine the perfor-mance of a horizontal dipole near the patch-loaded grounded slab.

3.1. Performance of a Horizontal Dipole Near a Patch-LoadedGrounded SlabFigure 4(a) shows the geometry of a horizontal dipole antenna neara patch-loaded grounded slab. The dipole is fed by a 50� coaxcable. One arm of the dipole is connected to the center conductorof the cable, and the other arm is connected to the outside con-ductor of the cable, which is soldered to the lower PEC of thecomplex surface. This is a simple feeding structure and experi-ments have demonstrated its applicability. To realize a low-profileconfiguration, the horizontal dipole is put very close to the artificialground plane.

The FDTD method is used to simulate the behavior of thisradiating structure. A finite ground with a size of 2� � 2� is usedin the analysis, including 16 � 16 patches. The patch dimensionsand dielectric substrate properties are the same as given in Eq. (1).The dipole is positioned only 0.02� above the top surface of theartificial ground plane and the radius of the dipole is 0.005�.

To obtain a good return loss, dipoles with different lengths aresimulated and the results are presented in Figure 4(b). The dipolelength is increased from 0.06� to 0.38�. It is observed that whenthe length of the dipole is increased, the resonant frequency of theantenna decreases. The return-loss value at the resonant frequencyalso changes with the dipole length. When the dipole length is

Figure 4 A horizontal dipole near a patch-loaded grounded slab: (a)antenna geometry and (b) FDTD simulated return-loss results of theantenna with different dipole lengths

Figure 3 Reflection-phase characterizations of the mushroomlike EBGstructure (with vias) and the patch-loaded grounded slab (no vias) for thenormally incident plane wave. Removing the vias has little effect on thereflection-phase feature for normal incidence

242 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 47, No. 3, November 5 2005

0.26�, the antenna achieves a good return loss around �30 dB. Itis important to point out that the length of the dipole is muchsmaller than a half-wavelength at the operating frequency. Incontrast, when a dipole is located near an EBG ground plane andresonates at the same frequency, the length of the dipole is 0.48�[9], which is close to a half-wavelength.

The FDTD simulation result is verified by experiments. Figure5(a) shows a photograph of the fabricated antenna prototype. ART/Duroid 6002 high-frequency laminate (�r � 2.94 � 0.04)with 120-mil (3.048-mm) thickness is used as the substrate and a2-mm-wide strip dipole is mounted 1.5 mm above the artificialsurface. The ground plane and the dipole dimensions are the sameas the FDTD simulation. Figure 5(b) shows the measured returnloss result. As expected from the FDTD simulations, although thedipole is very close to the ground plane, it still achieves a goodreturn-loss result (�25 dB) at the resonant frequency of 4.22 GHz.

The bandwidth of the fabricated antenna (S11 � �10 dB) is 6.0%.A slight frequency shift is noticed from the comparison, whichmay result from the numerical and fabrication errors.

The radiation patterns of the antenna are also computed andmeasured at its resonant frequency, and the results are presented inFigure 6. Several interesting properties are observed from theradiation patterns. The first observation is that the antenna showsa small radiation power in the broadside direction (� � 0°). Thisis different from the dipole antenna on an EBG ground plane [9]that has a maximum radiation power in the broadside direction.The main beam of this antenna points to � � 50° direction with again of 4.4 dB.

Second, E� is the co-polarized field in both the x–z and y–zplanes. Similar observations are also noticed in other � cut planessuch as the diagonal planes, which means that every � cut plane isan E-plane. In contrast, if the dipole is near an EBG ground plane,the x–z plane is the E-plane but the y–z plane is the H-plane [9].Therefore, this antenna has vastly different polarization feature toa horizontal dipole on an EBG ground plane. In addition, the

Figure 6 Measured radiation patterns (4.22 GHz) of the surface-waveantenna compared with the FDTD simulation results: (a) the x–z planepattern and (b) the y–z plane pattern. The antenna gain is 4.4 dB

Figure 5 A fabricated surface-wave antenna: (a) antenna photograph and(b) measured return loss. The fabricated antenna resonates at 4.22 GHzwith a bandwidth of 6.0%

MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 47, No. 3, November 5 2005 243

antenna exhibits similar co-polarization pattern shapes in the x–zand y–z planes.

However, due to the unbalanced feed of this antenna structure,the x–z plane pattern shows some asymmetries. In addition, arelatively high cross-polarization level in the y–z plane is alsoobserved, which is contributed by the direct radiation of the dipole.

3.2. Radiation Mechanism of the Surface Wave AntennaAlthough horizontal dipole antennas can obtain good return-lossresults both near the EBG and near the patch-loaded ground slab,different radiation performances are observed due to their distinctradiating mechanisms. When a dipole is positioned near an EBGground plane, no surface wave can be excited because the EBGstructure has a band gap for surface waves. The radiation iscontributed by the dipole, thus the dipole length is close to ahalf-wavelength for resonance. The radiation of the dipole dictatesthe antenna-beam direction and polarizations.

In contrast, for the patch-loaded grounded slab, since the ver-tical vias that suppress the TM mode are removed, the band gapdisappears and surface waves can propagate along the groundplane. When a dipole is positioned near a patch-loaded groundedslab, it excites strong surface waves. The dipole works more likea transducer rather than a radiator. Therefore, the length of thedipole does not necessarily have to be a half-wavelength, and theFDTD simulation shows that a proper dipole length is around0.26�.

Due to the thin dielectric layer and PEC boundary conditionfrom the top patch and bottom conductor, the surface waves aredominated by the TMz modes and the electric field is verticallypolarized. When the TM surface waves propagate and diffract atthe boundary of the ground plane, the radiation pattern can bedetermined. For example, the diffractions of surface waves at theedge of the ground plane are hard diffractions; hence, diffractionrays from opposite edges will cancel each other in the broadsidedirection, resulting in a radiation null. The vertically polarizedsurface waves also determine that the diffraction field must bepolarized along the � direction. Thus, a monopole-type radiationpattern is generated, as observed in Figure 6. Therefore, thisantenna structure can be identified as a surface-wave antenna(SWA), which realizes a monopole-type pattern with a low-profileconfiguration.

Compared to other surface-wave antenna structures [11, 12],this antenna design use the novel artificial ground plane to realizea low-profile configuration, which is usually desired in modernwireless-communication systems.

3.3. Comparison Between the SWA and the Vertical MonopoleAntennaA careful evaluation on Figure 6 reminds us of the monopoleantenna, which has a null in the broadside direction and is polar-ized along the � direction as well. Thus, it is instructive to comparethe radiation patterns of the SWA to an actual vertical monopoleantenna in order to appreciate how they resemble each other. Tothis end, a vertical monopole on a finite PEC ground plane isdesigned and its radiation patterns are calculated for comparison.The ground plane has the same size of 2� � 2�, and the monopolelength is tuned to 0.22� so that it has a resonant frequency similarto that of the surface-wave antenna. The radius of the monopole is0.005� and the antenna bandwidth is 20.6%.

Figure 7 compares the radiation patterns of both antennas in they–z plane, as E� polarization is of interest in this plane. It is notedthat the two antennas have similar radiation patterns. For example,the vertical monopole has the same beam direction at � � 50° asthe SWA. Both the monopole and surface-wave antennas have E�

as the co-polarized fields. The back lobes of the two antennas arealso close to each other: both the field levels and null positions.Similar observations are also obtained in other planes, such as thex–z plane and the diagonal planes. Compared to the monopoleantenna, the SWA has a relatively higher cross-polarization leveland a slightly lower directivity. These deficiencies can be furtherimproved using a symmetric feeding structure such as a circulardisk to replace the dipole.

The attractive feature of the SWA is its low-profile configura-tion. The height of the vertical monopole antenna is 0.22�,whereas the height of the horizontal dipole over the artificialground plane is only 0.02�. Thus, the dipole height is less than10% of the monopole antenna. If the thickness of the substrate isconsidered, then the overall height is 0.06�, which is still muchsmaller than that of the monopole. Therefore, the low-profile SWAwith a monopole-type radiation pattern has a promising potentialin wireless communications such as satellite-radio systems forvehicles.

4. CONCLUSION

This paper has presented a novel surface-wave antenna (SWA) thatcan radiate a monopole-type radiation pattern with a low-profileconfiguration. The low-profile property has been realized by re-siding a dipole near an artificial ground plane: a patch-loadedgrounded slab. The dipole can successfully transfer the energyfrom the feeding probe to the surface waves, and the propagationand diffraction of surface waves form a radiation pattern similar tothat of a monopole antenna. A prototype of the SWA was de-signed, fabricated, and measured in order to verify the radiationmechanism. The antenna resonates at 4.22 GHz with a 6.0%bandwidth, and a monopole-type radiation pattern is realized witha gain of 4.4 dB. The advantage of the SWA is its attractiveantenna height: only 1.5 mm above the artificial ground plane. Thissurface-antenna design has a great potential for wireless-commu-nication systems when a low-profile configuration and an omnidi-rectional pattern are desired.

Figure 7 Radiation-pattern comparisons between the surface-wave an-tenna and a vertical monopole antenna. The SWA has a radiation patternsimilar to that of the monopole antenna for co-polarized fields. Note that asthe cross-polarization of the monopole is lower than �40 dB, it does notappear in the figure

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REFERENCES

1. Special Issue on Meta-materials, IEEE Trans Antennas Propagat 51(2003).

2. Special Issue on Artificial Magnetic Conductors, Soft/Hard Surfacesand Other Complex Surfaces, IEEE Trans Antennas Propagat 53(2005).

3. R. Gonzalo, P. de Maagt, and M. Sorolla, Enhanced patch antennaperformance by suppressing surface waves using photonic-bandgapsubstrates, IEEE Trans Microwave Theory Tech 47 (1999), 2131–2138.

4. F. Yang and Y. Rahmat-Samii, Microstrip antennas integrated withelectromagnetic band-gap (EBG) structures: a low mutual couplingdesign for array applications, IEEE Trans Antennas Propagat 51(2003).

5. T.H. Liu, W.X. Zhang, M. Zhang, and K.F. Tsang, Low-profile spiralantennas with PBG substrate, Electron Lett 36 (2000), 779–780.

6. D. Sievenpiper, L. Zhang, R.F.J. Broas, N.G. Alexopoulos, and E.Yablonovitch, High-impedance electromagnetic surfaces with a for-bidden frequency band, IEEE Trans Microwave Theory Tech 47(1999), 2059–2074.

7. A. Aminian and Y. Rahmat-Samii, Spectral FDTD: a novel computa-tional technique for the analysis of periodic structures, 2004 IEEE APSInt Symp, Monterey, CA, 2004.

8. H. Mosallaei and Y. Rahmat-Samii, Periodic bandgap and effectivedielectric materials in electromagnetics: characterization and applica-tions in nanocavities and waveguides, IEEE Trans Antennas Propagat51 (2003), 549–563.

9. F. Yang and Y. Rahmat-Samii, Reflection phase characterizations ofthe EBG ground plane for low profile wire antenna applications, IEEETrans Antennas Propagat 51 (2003), 2691–2703.

10. W.E. Mckinzie III and R. Fahr, A low-profile polarization diversityantenna built on an artificial magnetic conductor, 2002 IEEE AP-S IntSymp, San Antonio, TX, 2002, pp. 762–765.

11. R. Hougardy and R.C. Hansen, Scanning surface wave antennas—oblique surface waves over a corrugated conductor, IRE Trans Anten-nas Propagat 6 (1958), 370–376.

12. G. Fikioris, R.W.P. King, and T.T. Wu, Novel surface wave antennas,IEE Proc Microwave Antennas Propagat 143 (1996), 1–6.

© 2005 Wiley Periodicals, Inc.

WIDEBAND CPW-FED UNIPLANARSLEEVE-SHAPED MONOPOLEANTENNA

Sheng-Bing Chen,1 Yong-Chang Jiao,1 Wei Wang,2 andQi-Zhong Liu1

1 National Laboratory of Antennas and Microwave TechnologyXidian UniversityXi’an, Shaanxi 710071, P.R. China2 East China Research Institute of Electrical EngineeringHefei, Anhui 230031, P.R. China

Received 23 April 2005

ABSTRACT: A novel compact and wideband coplanar waveguide(CPW)-fed sleeve-shaped monopole antenna is proposed. The uniplanarmonopole antenna consists of a CPW-fed monopole and two equalground planes, including rectangular stubs, according to the concept ofthe sleeve monopole antenna. The proposed antenna has a measuredoperating-frequency range from 2.05 to 5.14 GHz for a return loss ofless than �10 dB. Good radiation characteristics of monopolelike pat-terns are obtained. © 2005 Wiley Periodicals, Inc. Microwave OptTechnol Lett 47: 245–247, 2005; Published online in Wiley InterScience(www.interscience.wiley.com). DOI 10.1002/mop.21137

Key words: monopole antenna; coplanar waveguide (CPW); bandwidth

1. INTRODUCTION

CPW-fed antennas [1–3] have received much attention due totheir many attractive features such as wide bandwidth, simpleuniplanar structure, no need for via holes, and easy integrationwith active devices and MMICs. Recently, dual-band strip-sleeve monopole antennas have been reported [4 – 6], and dual-frequency operation has been achieved by adding two parasiticsleeves to a monopole.

In this paper, a compact and wideband CPW-fed monopoleantenna is proposed. Two rectangular stubs serving as a sleeveare connected to the ground planes and situated symmetricallyon each side of the monopole, according to the concept of thesleeve monopole antenna. The proposed antenna is configuredto be a uniplanar sleeve-shaped monopole antenna. The antennaperformance is investigated numerically, including the fre-quency bandwidth and radiation pattern. A test antenna ismanufactured and measured in order to verify the validity of thedesign.

2. ANTENNA GEOMETRY AND DESIGN

The geometrical configuration of the proposed antenna is depictedin Figure 1. The antenna has a simple structure with only one layerof metallization, and is etched on Rogers substrate with thicknessof 2 mm and relative permittivity of 10.2. The substrate size of theantenna is fixed at 50 � 50 mm2. A 50� CPW transmission line,which consists of a signal strip width of 2 mm and a gap distanceof 0.8 mm between the signal strip and the coplanar ground plane,is used for feeding the monopole. The length of the CPW line istaken as 25 mm. The main antenna structure connecting with theend of the CPW feedline is a monopole with dimensions of 19.8 �10 mm2 and a gap distance of 1.2 mm between the pole andground.

Two equal ground planes of the same size are arranged sym-metrically on each side of the CPW line. Two parasitic rectangularstubs serving as a sleeve are arranged symmetrically on each sideof the monopole and connected to the ground planes. To achieveoptimal antenna performance, the dimensions and positions of the

Figure 1 Geometry of the proposed uniplanar sleeve-shaped CPW-fedmonopole antenna (all dimensions in mm)

MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 47, No. 3, November 5 2005 245