IEEE TRANSACTIONS ON NUCLEAR SCIENCE, VOL. 44, NO. 3, JUNE? 1997 557

A Novel Electronic Chain for High Level Neutron Coincidence Counters

A. Fazzi, Member, IEEE, C. Pirovano and V. Varoli, Member, IEEE Politecnico di Milano, Dip. Ingegneria Nucleare, via Ponzio 34/3, Milano 20133 Italy

Abstract This paper describes a counting chain (charge

preamplifier, shaping amplifier and threshold discriminator) devoted to 3He neutron detectors.

The preamplifier adopts an innovative circuit with the gate of the input JFET floating and a DC feedback loop that stabilizes the output voltage acting on the input cascade second transistor. Static and dynamic analysis, including the effects of the detector bias network, is developed and reported.

The shaping amplifier is a fifth-order quasi-gaussian OPAMP active filter that minimizes the number of components. Shaping time constants ranging from 1 ps down to 100 ns are explored in order to maximize the count rate capability, maintaining nevertheless neutron discrimination. Experimental signals and spectra show that 100 ns shaping time is a lower limit for the usual 3He proportional detector.

Since this counting chain is characterized by single 5V supply operation, high counting rate, small size and low cost, it is well suited for high efficiency neutron well detectors where a large number (10 - 100) of counting tubes are used (High Level Neutron Coincidence Counter HLNCC).

I. INTRODUCTION The verification of fissile material stocks (plutonium) is

usually obtained by detecting the neutrons generated by the spontaneous fission of 240h and analyzing the time correlation of their arrival, or by performing more sophisticated analyses of their time of arrival distribution [ 11. Since each fission produces a burst of neutrons while other nuclear reactions (e.g. a,n) generate a single neutron, these analyses effectively discriminate between fission and non- fission sources.

The global detection efficiency and the detector dead time are key parameters in these measurements. High efficiency is obtained using a large number of neutron counting tubes (usually 3He filled proportional counters) inserted in a moderating material that surrounds the sample. The detector outputs, after analog processing and amplitude discrimination, are "summed" in a single pulse train and sent to a time correlation analyzer. A technique to achieve a "lossless sum" of the pulses is described in [2]. With such a circuit all the count losses are only due to the analog processing chain. Analog dead time can be reduced to the intrinsic limit of the counting tube when each detector is connected to a fast counting chain, but this solution is seldom used for cost and

size reasons. Depending on the maximum count rate, three or more tubes are usually connected in parallel to the input of a single electronic chain.

The aim of this paper is to present a complete electronic counting chain cheap, small and fast to make it possible to minimize the counting losses due to the analog dead time.

This chain operates with a single 5 V power supply, thus simplifying the cabling in the detector head and reducing the cost of the power supply. The use of such a single low voltage supply, uncommon in nuclear analog electronics, is practically a standard in HLNCC systems.

The following sections describe the different parts of the counting chain. Experimental results obtained with a HLNCC system are reported.

11. CHARGE PREAMPLIFIER Most of the charge preamplifier circuit diagrams are

derived from the simplified schematic of Figure la. The DC operating point is stabilized by the feedback resistor Rf and the gate-source junction of the input RET is reversed biased. Such an operating point is not possible if only a positive supply is available, as in our application.

---&--I I

from detector

-I--+ Q 1 p +vm

Active load

output stage

ping amp.

Figure la: Split supply preamplifier scheme. It was shown in [3], [4] and [SI that it is possible to

remove the feedback resistor and operate the JFET with floating gate. However these circuits, devoted to high resolution X-ray spectrometry, require selected components, a thermally controlled environment and split power supplies.

0018-9499/97$10.00 0 1997 IEEE

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For the present application, component selection is The AC analysis of the “forward block”, that is the voltage unacceptable for cost reasons, a wide operating temperature amplifier between the gate of J1 and the preamplifier output range must be ensured and a single supply of 5v must be vo, supplies the fo l low~g frequency response: used.

A novel design has been developed. The simplified circuit vo I VgJl = - gmJl R1 I (gmJ2 Rmt) ( l 4- cmt Qnt) / + as + bs2)

diagram of the new charge preamplifier is shown in Figure lb. where:

a = Rl Cmt / gmj2 RA and b a RA CA (4)

+vRef CA is the capacitance of the dominant pole due to the active load.

The forward block has a very low DC gain, one zero at w = I / Cint Rh t due to the feedback integrator pole and two poles at higher frequencies. The gain-bandwidth product is gmJ1 I cA.

For obvious safety reasons, counter tubes are always operated with the cathode (the outer wall) at ground potential and the anode at high voltage. Hence a detector bias network, as shown in Figure 2, is required. This network becomes part of the outer capacitive loop of the charge preamplifier (Figure 2) and may impair its stability.

Figure lb: Single supply preamplifier scheme.

The input stage is a JFET - BJT (Jl, Q1 ) folded cascade, with the J1 gate floating. In this way the gate potential stabilys at about 0 V and precisely at the small positive value at which drain to gate and gate to source leakage currents are equal. The drain current is thus about IDSS. Since large unit to unit spread and temperature dependence of IDSS are expected, a DC stabilizing feedback is mandatory. This last consists in the active integrator (op 1, Cint, Rht, R1 ) and the JFET J2 . The transistor J~ has a current rating larger than J1 and operates in

Figure 2: Charge preamplifier feedback loop scheme. The High Voltage filtering network is part of the capacitive Feedback Block.

the ohmic region as a voltage controlled resistor. It’s gate voltage, that is the output voltage of the feedback integrator, is such as to set the drain-source resistance RDSR to the value that satisfies the equation:

RDSJ2 = vDSL12 I IDS32 =

= ( Vcc - t VBB i- VEBQ~ 1 11 ( IDSSJ~ +- IEQI 1 (1) where VEBQl is Q1 emitter-base voltage drop and I,,, is Q1

Disregarding the emitter impedance of Q1, the very high emitter current set by the active load.

gain of the DC stabilizing loop is:

GLDC = &ntl R1 mJ2 RA (2) where Rint/R1 is the DC gain of the feedback integrator, gmJ2 is the transconductance ( in ohmic region ) of J, and RA is the DC impedance of the active load. The preamplifier output voltage is therefore strongly stabilized at the reference voltage Vref.

In fact it can be easily shown that the “feedback block” contributes to the loop gain with three poles and three zeroes. A simplified analysis shows that the pole with the highest angular frequency is at Q = 1/EC where & is the parallel between Rfill and the input resistance of the forward block and C is the sum of the detector capacitance CDET , the feedback capacitance C f and the input capacitance of the forward block. To avoid low frequency oscillations in the closed loop response the loop gain at must be low enough (e.g. less than -6 dB).

A surface mount prototype of the charge preamplifier has been mounted and fully characterized. The frequency response of the forward block is shown in Figure 3 (continuous line) together with the experimental measurements. The loop gain function, affected by the detector bias network, is also shown on the same figure (dashed line).

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The pulse response of the preamplifier has a rise time of about 10 ns, an exponential decay of time constant of about 150 ps and a longer negative exponential tail of about 1 ms. This longer exponential tail is due to the pole in the loop gain at 0. All these values fit very well the foreseen closed loop singularities.

8o 1

d 1 c P)

.-

E E o -80 Forward gain L

POLE 4th ORDER VCVS CELL ZERO

1 Figure 4: Shaping amplifier scheme. A differentiator with pole-zero cancellation feeds a fourth order VCVS cell.

With the short shaping time constant that this circuit uses, the finite bandwidth of the operational amplifier is the major source of errors and, in addition, speed and DC accuracy are conflicting requirements.

Two quad rail-to-rail operational amplifiers were tested: a medium speed one (about 28 MHz bandwidth) with excellent DC performance (LT1214) and a high speed one (150 MHz) with poorer DC characteristics (AD8044). The large offset and drift of the latter dictated the use of the slower one, properly compensated. One OPAMP is used for the active filter cell, two other for the gain stages and the last for the preamplifier feedback integrator.

Frequency (Hz)

- 1 Figure3: Forward block gain (continuous line) and charge

preamplifier loop gain (dashed line) vs. frequency.

111. SHAPING AMPLIFIER The two most important requirements to our shaping

amplifier are a high count rate capability and a very good DC performance. In order to reduce the pile-up, a quasi-gaussian impulse response and a short peak time are suggested.

The normalized poles of the filter are determined by series expansion up to the fifth order of the gaussian and denormalized to the required time constant [6]. As shown in Figure 4 the real pole is realized with a CR differentiator with pole-zero cancellation and the two complex pole pairs with a single fourth order Voltage Controlled Voltage Source (VCVS) cell.

High order VCVS cells are not very popular since they require complex calculations to find the component values and are quite sensitive to component tolerances. The first drawback is Overcome by implementing the computing algorithm in a computer program ["I. In regard to the component tolerance sensitivity, we found by experience that I% tolerance components are adequate only if none of the pole pairs has a Q factor higher than about 1.3 - 1.5. The fifth order gaussian filter meets this condition.

2

Figure: 5 Trace 1 test input 50 mV/div., trace 2 output 100 mV/div., her. o.2 ps/div.

Figure 5 shows the Pulse response ofthe shaping amplifier with shaping time constant T equal to 100 ns where 7 is defined as the inverse of the cutoff angular frequency 0-3dB 161.

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IV. THRESHOLD DISCRIMINATOR This section is the easiest to design since fast comparators

fully characterized for 5 V operation are commercially available. Care has been taken to minimize crosstalk of this section with the analog ones.

V. RESULTS

40000

30000

v) C .c-'

=J 20000 s 10000

0

T = l ps

100 200 300 Channel number

Figure 6a: Pulse height distribution at z = 1 ps.

20000 7 J

3 c 7

8

~

Y v

1 400

T = 145 ns 16000 41 11 \ J

1 12000

4000 41 i I i

'; j 0 1 I I 1 - 7 I I I 100 200 300 400

Channel number

Figure 6b: Pulse heirrht distribution at T = 145 ns.

The complete electronic chain has been tested connected to a Xeram 150NH100 3He tube detecting thermalized neutrons from an AmBe source surrounded with paraffin. The neutron discrimination has been investigated at shaping time constants T spanning from 1 ps to 100 ns.

Figures 6a, 6b and 6c show the pulse height differential spectra obtained. At 1 ps the 764 keV peak is well resolved, while at time constant shorter than the detector charge collection time, a distorted spectrum is obtained. However a sharp discrimination of neutrons from low-energy event (e.g. electronic noise and y-rays) is achieved down to 100ns shaping time constant. The details on the shape of the spectra and on their modification changing the shaping time constant depend on the behavior of neutron cylindrical gas proportional detectors [8].

30000 1

I f

1 a0 200 300 400 Channel number

Figure 6c: Pulse height distribution at T = 100 ns.

The circuit has been compared with the standard circuit generally adopted for High Level Neutron Coincidence Counter (based on Amptek A1 11) [9]. This last performs a CR-RC shaping with peak time 150 ns. In a HLNCC used by safeguards inspectors, three 3He tubes connected in parallel fed the new circuit and three other tubes fed the standard one. Measurements were made with 252Cf neutron sources of different activity located at the center of the HLNCC.

At the maximum count rate of 50 kcps no count losses was detected.

Spectra at low rate (800 cps) and high rate (40 kcps) were taken and are shown in Figure 7 and Figure 8. (The shaping time-constant for the new circuit was 100ns).

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3000

Neutron count rate: 800 cps

New circuit

1 u) c w

8 2000 0

IOoo ‘i 0 200 400 600 800 1000

Channel number

Figure 7: Neutron spectra measured with standard HLNCC system. Comparison between the new circuit and the standard one (A1 1 1) at low count rate (800 cps).

High count rate

AmptekAll l

0 400 800 1200 1600 Channel number

Figure 8: Comparison between the new circuit and the A1 1 1 at high count rate (40 kcps).

The shapes of the spectra are almost equal except for the presecence at high rate of a tail due to pile-up.These tails are expanded in Fig. 9 which shows the same spectra in a logarithmic format. Even if the gain of the new circuit is higher, the pile-up is lower,

High count rate (40 kcps)

-+$%--- New circuit -*- Amptek A I 11

10 100 1000 Channel number

Figure 9: Comparison between the new circuit and the A1 1 1 at high count rate (40 kcps) in logarithmic scale.

This improvement is due to the different pulse response of the two circuits. The quasi-gaussian response is more symmetric than the CR-RC one, with a much faster decay.

Finally, the noise level of the new circuit is good; better than the noise of the standard one, as shown in Figure 9 by the smaller low-energy events tail.

VI. CONCLUSIONS A compact and cheap electronic chain for high rate 3He

neutron counters operating with a single 5 V supply has been designed built and tested.

It adopts innovative solutions in the preamplifier section and minimizes component count in the amplifier one.

Neutron counting capability up to 50 kcps has been successfully verified.

VII. ACKNOWLEDGMENTS The support of A. Boffa, B.Pedersen and R.Jaime during

the measurements is gratefully acknowledged.

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VIII. REFERENCES [I] Zucker, “Neutron correlation counting for the

nondestructive analysis of nuclear materials,” NBS 528, 1980.

[2] Fazzi, V. Varoli, D. Rozzi, V. Vocino, R. Jaime, “A cost- effective circuit for dead-time reduction in neutron multiplicity counters, “ I 7th Annual Esarda Meeting Working Document, pp.567-570, 1995.

[3] Bertuccio, P. Rehak, D. Xi, “A novel charge sensitive preamplifier without the feedback resistor,” Nucl. Instr. and Meth., vol. A326, pp.71-76, March 1993.

[4] Bertuccio, L. Fasoli, C. Fiorini, M. Sampietro, “Spectroscopy Charge Amplifier for Detectors with Integrated Front-End FET,” IEEE Trans. Nuc. Sci., vol. 42 N. 4, pp.1399-1405, August 1995.

[5] F.Olschner, J.C.Lund, “Low Noise Charge Sensitive Preamplifier Using Drain Current Feedback,” I992 IEEE Nuclear Science Symposium Conference Record, vol. 1,

[6] J. Blinchikoff, A. I. Zverev “Filtering in the time and frequency domains,” John Wiley & Sons, 1976.

[7] Yu Jen Wong , W. E. Ott, “Function circuits,” McGraw- Hill, 1976.

[SI R.Cervellati, A.Kazimierski, “Wall Effect in BF, Counters,” Nucl. Instr. and Meth., vo1.60, pp. 173-178, 1968.

[9] Swansen et al., “Shift-register coincidence electronics system for thermal neutron counters,“ Nucl. hstr. and Meth., ~01.176, 1980.

pp.378-380.