7-we2p-01
TRANSCRIPT
-
8/2/2019 7-WE2P-01
1/4
Fully Substrate-Integrated High-Gain Thin Fabry-Perot Cavity
AntennasBo Zhu
#1,*2, Zhi Ning Chen
*2, Yijun Feng
#3
#Department of Electronic Science and Engineering, Nanjing University, Nanjing, 210093 China
*RF and Optical Department, Institute for Infocomm Research, Singapore [email protected]
ABSTRACT Fabry-Perot cavity (FPC) antennas are fully
integrated onto a printed circuit board. The antenna consists of
a partially reflective sheet formed by finite-size periodic array of
metal patches printed on a grounded dielectric substrate and a
feeding microstrip patch embedded in the substrate. The field
distribution reveals the basic features of operating modes and
the distribution of power loss. Simulated and measured results
of the proposed antennas operating at 10 GHz show the realized
gain of 15 dBi with an aperture of 22 the operating wavelength
in free space. Moreover, the proposed antenna realizes
circularly polarized radiation. The study shows that such fullydielectric-integrated planar FPC antennas are able to provide
high gain and circular polarization operation function with low
profile, easy integration onto circuit board, and mechanical
robustness, suitable for low-cost mass production.
Index Terms Circularly polarized, Fabry-Perot cavity, high-gain antenna, substrate-integrated.
I. INTRODUCTIONWith the miniaturization of wireless electronic devices, the
demands for the high-gain antennas which can be readily
integrated to circuit boards with mechanical robustness are
increasing significantly. The fully substrate integration of
antennas is one option because of the merits such as
compactness with low profile, mechanical robustness,
stability of installation and fabrication, low-cost fabrication,
as well as less insertion loss. However, the fully dielectric
integration will cause several design challenges such as
increase in dielectric loss and lowered directivity caused by
small volume of antennas.
Leaky-wave antennas (LWAs) have been well known as an
option to enhance the directivity of antennas for years [1]. An
LWA usually consists of a waveguided structure in which fast
wave modes with a complex wave number or leaky waves
propagate and radiate. The planar LW structures have been
used to design two-dimensional LWAs.Fabry-Perot cavity (FPC) antennas are usually constructed
by a partially reflective surface (PRS) suspended above a
ground plane with a half-wavelength gap. The reflectivity
amplitude of the PRS should be close to unity in order to
achieve high-gain performance [1, 2]. By combining the PRS
with for example, an artificial magnetic conductor (AMC),
the profile of FPC antennas can be greatly reduced but with
high losses [3, 4]. Circularly polarized radiation can be
achieved through embedding a circularly polarized feeding
source into the cavity of FPC antennas [5]. Due to the
existence of air-gap between the PRS and ground plane these
configurations suffer bulky volume, in particular, at higher
operating frequencies, severely the tolerance of fabrication
and installation.
In this paper, we first propose fully substrate-integrated
FPC antennas. A high-gain FPC antenna is fully integrated
onto a dielectric substrate with an array of patches as the
PRS, a ground plane, and a microstrip patch source embedded
into the substrate using the printed circuit board technique.Distribution of electric fields for the finite-size configurations
is examined. Parametric study provides the information about
the design and fabrication tolerance. After that, a circularly
polarized FPC antenna is designed. All simulation is carried
out by CST Microwave Studio 2009.
II. SUBSTRATE-INTEGRATED HIGH-GAIN FPCANTENNAS
A. Antenna DesignAccording to plane wave expansion approximation, for a
high directivity at a given operating frequency with a
wavelength 0 (0 is the operating wavelength in free space),the thickness of the FP cavity which is fully air filled between
its PRS and ground plane is determined by [4],
04 / 2 , 0, 1, 2,......,prs h N N + = = (1)
whereand prs are the reflection phase of the ground plane
and the PRS, respectively. h indicates the thickness of the FP
cavity. When an array of single-layered metallic patches or
slots in the metallic screen is used as the PRS, the height of
the cavity, h is about 0/2. Recent research has showed thatthis height can be reduced to 0/4 or even smaller by
exploiting a double-layered metamaterial structure as the PRS
with the required reflection phase determined by Eq. (1) [3,
4]. However, the double-layered structure suffers high ohmicloss when its reflection phase tends to zero so that the antenna
radiation efficiency is lowered.
Fig. 1(a) shows the proposed FPC antenna operating at 10
GHz. A single layer of 77 metallic square patches is printed
onto a dielectric substrate with a periodicity ofp=9 mm. The
size of square patch is w=8.2 mm. The dielectric substrate is
formed by multi-layered dielectric boards (Rogers 4003, r=3.38, tan= 0.0027). The overall dimension of the antenna is
64648.43 mm (or 220.30). The cavity thickness is 0.30
comparable to that of the air-filled AMC-loaded cavity design
Proceedings of the Asia-Pacific Microwave Conference 2011
978-0-85825-974-4 2011 Engineers Australia 602
-
8/2/2019 7-WE2P-01
2/4
[3]. A 68-mm microstrip patch is e
substrate at the center ofx-y plane at a hei
above the ground plane. The patch is
microstrip line (c=1.55 mm) through a Su
A (SMA) connector as shown in Fig. 1(b).
The reflectivity of the PRS is optimize
directivity with a given aperture of the
reflection coefficient of the PRS can be esti
simulation of a single unit cell using
conditions as shown in Fig. 2. The cell
metallic patch printed on the interface betsubstrate and free space. A waveguide por
bottom of the dielectric substrate to g
incident plane wave illuminating on the P
Fig. 2, the magnitude and phase of reflec
and 147o
at 10 GHz forw=8.2 mm andp=
Fig. 3(a) compares the simulated and me
as the realized gain at boresight. The meas
than 10 dB over 10.4-10.8 GHz. The mea
up to 15.5 dBi at 10.5 GHz. The peak of
9.75 10.00 10.25 10.500.80
0.85
0.90
0.95
1.00
Amplitude
Refle
ctionMagnitude
Frequency (GHz)
w = 8.0 mm
w = 8.2 mm
w = 8.0 mm
w = 8.2 mm
Phase
w = 8.4 mm
Fig. 2 The simulated reflection coefficient of PRS wi
p = 9 mm. Inset shows the unit cell in simulation.
(a)
(b) (c)
Fig. 1 Profile of (a) proposed FPC antenna (s = 64 m
the microstrip feeding patch (a = 6 mm, b = 8 mm) an
bedded into the
ght ofd=0.81 mm
fed by a 50-
Miniature version
to obtain a high
PC antenna. The
mated through the
eriodic boundary
is composed of a
een the dielectricis attached to the
nerate a normal
RS. As shown in
ed waves are 0.9
9 mm.sured |S11| as well
ured |S11| is lower
ured gain reaches
he measured gain
shifts down by 150 MHz c
which may be caused by the
dielectric sheets. The theoretic
antenna isDmax = 10log10 (4A
area ofA=6464 mm at 10
antenna is 86% of the theoreti
drop of gain is due to the di
amplitude and phase distributidue to much short distance bet
The radiation patterns in E a
3(b). The measured and sim
compared at 10.5 and 10.65 G
that radiation patterns are sy
planes. The measured ratio of
levels is higher than 15 dBi ev
B. Field Distribution andFig. 4(a) illustrates the elect
plane atz=4.2 mm at 10.65 G
field is strong around the
gradually along radial directi phase distribution of electric
the phase descends along the r
gradient of phase variation wh
can be regarded as evenly dis
radiation. Therefore, the leaky
descends along radial direction
radiates simultaneously so th
complex while the standing wa
The distribution of the ov
longitudinal section of the ante
10.0 10.2 10.4-40
-30
-20
-10
0
|S11
|(dB)
Freque
Mea
Simu
(
(
Fig. 3 Measured and simulated (a)
radiation patterns in E and H-planes.
10.75
ReflectionPhase
-170o
-160o
-150o
-140o
th the unit cell period
m, h = 8.43 mm), (b)
d (c) a PRS patch.
mpared with the simulation,
air-gap between the stacked
al maximum directivity of the
02) =18 dBi with the aperture
Hz. The realized gain of the
ical maximum directivity. The
electric loss and non-uniform
ns of the field on the apertureeen the source and PRS.
nd H-planes are plotted in Fig.
ulated radiation patterns are
z, respectively. It can be seen
metric in both the E and H-
co to cross-polarized radiation
n in the E-planes at 10.5 GHz.
oss Analysis
ric field distribution in thex-y
Hz. It is seen that the electric
feeding patch and decreases
n. In Fig. 4(b), the simulatedield component Ey shows that
adial direction. With the small
re the field is strong, the phase
tributed for a highly directive
wave mode is excited, which
s inside the FPC antenna, and
at the radial wavenumber is
ves exist in the cavity.
rall power loss density in a
nna is plotted in Fig. 4(c). It is
10.6 10.8 11.00
4
8
12
16
BoresightGain(dBi)
cy (GHz)
ured
lated
a)
b)
|S11| and gain at boresight, and (b)
603
-
8/2/2019 7-WE2P-01
3/4
observed that the majority of power loss a
central of the FPC due to the dielectric l
strongest electric field in this portion as sho
Table I compares the directivity and ga
with/without dielectric loss and ohmic loss.
dielectric loss reduces the gain by 1 dB w
of PRS reduce the gain by 0.1 dB due to the
TABLEICOMPARISON OF GAIN FOR
Antenna Loss D, dBi G-IEEE
ideal No 18 1
proposed Dielectric loss*
and ohmic loss
16.49 15.
proposed ohmic loss only 16.66 16.
proposed dielectric loss only 16.50 15.
*Dielectric loss tangent: tan= 0.0027
C. Parametric StudyWith simulation a parametric study is c
the effect of fabrication tolerance on anten
the study, only one parameter is varied at a
Substrate Permittivity: Fig. 5 shows
substrate dielectric constant lowers the r
where the peak gain is achieved. However
dielectric constant distorted radiation patt
and H-planes because the condition of i
waves in the FP cavity is degraded. The des
for r=3.38 at 10.65 GHz. The distorvariation in beamwidth and sidelobe
permittivity higher than optimal 3.38. Th
permittivity tolerance of dielectric substrate
Substrate Thickness: Fig. 6 shows that ithickness lowers the resonant frequency
circle), reduces the gain, and degrade
matching. Increasing the thickness from the
h = 8.43 mm causes higher sidelobe levels
H-planes. Such effects stem from the si
change which distorts the reflected wave ph
the FP cavity. Therefore, a particular attenti
to the substrate thickness in fabrication.
(a)
(c)
Fig. 4 Distribution of (a) magnitude, (b) phase ofEymm and (c) power loss density in a longitudinal s
10.65 GHz.
ppears around the
ss caused by the
wn in Fig. 4(a).
in for the designs
It is clear that the
hereas the patches
ohmic loss.
ANTENNAS
, dBi -radiation, %8 100
68 83
59 98
78 85
onducted to study
a performance. In
ime.
that the higher
sonant frequency
a 3% change in
rns in both the E
in-phase of leaky
ign was optimized
ion includes the
levels for the
erefore, the 3%
is acceptable.
creasing substrate(indicated by the
s the impedance
optimal thickness
in both the E and
nificant thickness
ase distribution in
ion should be paid
PRS Patch Dimension: Fig
matching and resonant frequen
by the dimensions of PRS pat
phase of reflection of PRS in
reflection decreases as the pwith the fixed periodicity of
size from w=8.2 mm widens t
The larger patches increase t
gain slightly due to the reduc
from the PRS. Moreover, the
of PCB fabrication is suggeste
10.0 10.5 11.0 11.50
4
8
12
16
BorsightGain(dBi)
Frequency (GHz)
|S11
|(dB)
w=7.8 mm
w=8.2 mm
w=8.6 mm
Fig. 7 Effects of PRS patch si
10.0 10.5 11.0 11.50
4
8
12
16
-
-
-
-
BoresightG
ain(dBi)
Frequency (GHz)
|S11|
(dB)
h=8.01 mm
h=8.43 mm
h=8.85 mm
Fig. 6 Effects of substrate thickn
10.0 10.5 11.0 11.50
4
8
12
16
-4
-3
-2
-1
r=3.28
r=3.38
r=3.48
BoresightGain(dBi)
Frequency (GHz)
|S11
|(dB)
Fig. 5 Effects of substrate permittivit
(b)
inx-y plane atz=4.2ection of antenna at
. 7 shows that the impedance
cy but gain are greatly affected
ches. As shown in Fig. 2, the
creases and the amplitude of
tch of PRS becomes smallerp=9 mm. Reducing the patch
e beamwidth in the E-planes.
e sidelobe levels and reduce
tion of leaky waves radiating
study shows that the tolerance
to be less than 0.2 mm.
10.0 10.5 11.0 11.5-40
-30
-20
-10
Frequency (GHz)
e on |S11| and boresight gain.
10.0 10.5 11.0 11.540
30
20
10
0
Frequency (GHz)
ss on |S11| and boresight gain.
10.0 10.5 11.0 11.50
0
0
0
0
Frequency (GHz)
y on |S11| and boresight gain.
604
-
8/2/2019 7-WE2P-01
4/4
III. CIRCULARLY POLARIZED D
The CP operation of the antenna can
replacing the linearly polarized feeding
antenna presented in Section II with a corn
ofa=6 mm and b=4 mm as illustrated in Fi
ofl= 6 mm of the microstrip feeding line
width c = 1.55 mm to c= 0.4 mm. All oth
parameters are kept the same as those of thpolarized antenna. Fig. 8(a) shows that th
below 10 dB over the frequency range o
The discrepancy between the simulated an
is mainly caused by the air gap between die
measured axial ratio (AR) at boresight is
frequency range of 10.4 to 10.8 GHz and
achieves 15 dBic at 10.5 GHz as illustrate
9 plots the simulated AR profile for the CP
the feeding patch rotated by 45o. As can
property is achieved from 10.6 to 10.8 GH
similar to the previous one without any rota
patch. This suggests that the FP cavity is i plane and the CP performance is in
polarization direction of each orthogonal m
IV.CONCLUSION
Two fully substrate-integrated high-gain
antennas have been presented. Circularly p
with the gain of 15 dBi under antenna ape
been realized at X-band. Parametric stu
thickness, permittivity and PRS size have
the effects on the antenna performanc
fabrication tolerance. The field distributio
basic features of the leaky wave mode in a
loss distribution. The study has shown that
dielectric-integrated FPC antennas feature
integration onto circuit boards and mech
suitable for cost-effective mass production.
ACKNOWLEDGEMENT
This work is partially supported by th
Research Program of China (2004CB
National Nature Science Foundations of
60990320, 60671002 and 60801001) and
Metamaterial Program: Meta-Antennas (09
REFERENCES
[1] G. von Trentini, Partially reflecting sheetAntennas Propag. , vol. 4, no. 4, pp. 666-671,
[2] S. N. Burokur, R. Yahiaoui, and A. Lumetamaterial based resonant cavities fed bhigh directivity, Microw. Opt. Technol. Le
1883-1888, Aug. 2009.
[3] P. Feresidis, G. Goussetis, S. Wang, anArtificial magnetic conductor surfaces and t
ESIGN be achieved by
atch of the FPC
er truncated patch
. 8(a). A segment
is narrowed from
er dimensions and
previous linearlymeasured |S11| is
10.05-11.0 GHz.
measured results
lectric layers. The
elow 3 dB over a
he measured gain
in Fig. 8(b). Fig.
FPC antenna with
be seen, the CP
and the profile is
tion of the feeding
sotropic in the x-ysensitive to the
de.
abry-Perot cavity
olarized operation
rture of 220 has
dies of substrate
een done to show
and acceptable
has revealed the
cavity and power
the proposed fully
low-profile, easy
anical robustness,
e National Basic
19800) and the
China (60990322,
A*STAR SERC
154 0097).
arrays, IEEE Trans.
ct. 1956.
trac, Subwavelength
multiple sources fort., vol. 51, no. 8, pp.
J. C. Vardaxoglou,eir application to low-
profile high-gain planar antenvol. 53, no. 1, pp. 209215, Ja
[4] A. Ourir, A. de Lustrac, and J.sub-wavelength cavities (/6Appl. Phys. Lett., vol. 88, no 8
[5] Ederra, R. Gonzalo, A. GoshGavrilovic, Y. Demers, and Penhancement of a circularly p
Antennas Propag. Soc. Int. Sy
2993-2996.
10.4 10.60
3
6
9
AxialRatio(dB)
Frequenc
Fig. 9 Simulated AR profile for a
10.0 10.2 10.4-40
-30
-20
-10
0
Measured
Simulated
|S11
|(dB)
Freque(
10.0 10.2 10.40
3
6
9
AxialRatio(dB)
Freque
MeasuredSimulated
(
Fig. 8 Measured and simulated (a) |S
nas, IEEE Trans. Antennas Propag.,. 2005.
-M. Lourtioz, All-metamaterial-based
0) for ultrathin directive antennas,, Feb. 2006, paper 084103.
, J. Laurin, C. Caloz, Y. Brand, M.
. de Maagt, EBG superstrate for gainlarized patch antenna, inProc. IEEE
mp., Albuquerque, NM, Jul. 2006, pp.
10.8 11.0
(GHz) 45o rotated feeding patch.
10.6 10.8 11.0
cy (GHz) a)
10.6 10.8 11.00
4
8
12
16
cy (GHz)
BoresightGain(dB
ic)
b)
11| and (b) AR and boresight gain.
605