2.2 2.4 ghz phased array

Final Project 2.2-2.4 GHz Phased-Array Conceptual

Redesign

by Russell Hofer

May 2005

Table of Contents

ACKNOWLEDGEMENTS ............................................................................................. 1

INTRODUCTION............................................................................................................. 2

BACKGROUND OF EXISTING ANTENNA ARRAY................................................ 2

REDESIGN OBJECTIVES ............................................................................................. 6

INCREASE ANTENNA BANDWIDTH......................................................................... 7

Increase of Antenna Element’s Bandwidth...................................................................................... 7

Increase of Bandpass Filters’ Bandwidth ...................................................................................... 13

CONCEPTUAL REDESIGN OF ANTENNA ELEMENT PCB ............................... 18

Steering Redesign ......................................................................................................................... 18

Schematic Redesign...................................................................................................................... 24

TRANSMIT CAPABILITIES ....................................................................................... 29

TRANSMIT CAPABILITIES ....................................................................................... 30

CONCLUSIONS ............................................................................................................. 39

FURTHER WORK......................................................................................................... 40

REFERENCES................................................................................................................ 41

APPENDICES................................................................................................................. 42

Appendix I: Matlab Code .............................................................................................................. 42 Figure 9.................................................................................................................................................... 42 Figure 17, 18............................................................................................................................................ 42

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Acknowledgements I would like to acknowledge Dr. Christopher Allen for his guidance through this project. In addition, I would like to thank Mr. Dan Depardo for his willingness to answer questions and share his valuable experience of antenna and microwave design. Lastly, Dr. Jim Stiles for consultation on general microwave principles and providing what has proven to be a very useful course in microwave design. I would like to extend a thanks to Dr. Christopher Allen, Dr. Prasad Gogineni, and Dr. Kenneth Demarest for participation in my committee. Last, but not least, I would like to extend my gratefulness to my wife, Kristie Hofer, for her support and patience.

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Introduction In 2004, Kansas University and Honeywell FM&T Kansas City Plant jointly built a 2.3-2.4 GHz Phased-Array Prototype. Mr. Dan Depardo, RF electronics engineer, was responsible for the design, construction, and preliminary testing. The antenna has proven to perform adequately per its original design criteria. However, Dwayne Brown, KCP (Kansas City Plant) product engineer, has extended requirements that are to be incorporated into the second design iteration. In this report, the results of the conceptual redesign of the antenna array are discussed. Included is a brief background of the existing antenna array, redesign objectives, new antenna design, simulation results, and further work/conclusions.

Background of Existing Antenna Array

The following discussion gives a limited overview of the original antenna array design. The interested reader is referred to [1] for a detailed overview. The original antenna consists of eight patch elements positioned in a linear array. The array is capable of performing electronic steering in azimuth. The bandwidth of the antenna encompasses 2.3-2.4 GHz. Figure 1 shows an image of the original 2.3-2.4 GHz phased-array antenna prototype [1]. Figure 1: 2.3-2.4 GHz Phased-Array Antenna Prototype Each patch antenna element consists of a low-noise RF amplifier, two-stage RF bandpass filter (100 MHz B/W), two variable voltage phase shifters, and further small-signal RF amplification. Figure 2 shows a basic block diagram of the antenna element. Notice that each antenna element must be given a steering signal. This signal controls the amount of phase shift incurred by the RF signal at each element.

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Output

Patch Antenna

5th-Order BPF (Bandpass Filter)

Φ1

Φ2

Small-Signal Amplifier

SC (Steering Control Voltage)

LNA (Low Noise Amplifier)

Phase Shifters

3rd-Order BPF (Bandpass Filter)

Figure 2: Block Diagram of an Antenna Element

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Ideally, each antenna element will have a constant delta phase shift with respect to its neighboring elements given by Equation 1 [2]:

0cos*/2 θλπθ d=∆ (1) d = interelement spacing λ = wavelength θ0 = steering angle For a broadside beam, there is no relative phase shift per element. The steering control voltage, depicted in Figure 2, is generated by an analog op-amp array. This circuit generates a maximum delta steering voltage between elements of approximately 1 volt. Assuming the phase shifters’ behavior is roughly linear, this results in a maximum delta phase shift between elements of approximately 30 degrees. Also, the distance between elements with respect to wavelength is approximately 0.23. Using this information in Equation 1, the maximum obtainable steering angle is +/- 20 degrees from broadside. The signals from all antenna elements are combined with two RF power combiners and a 180o hybrid coupler. These signals, one given the designation ‘SUM RF’ and the other ‘DELTA RF’ are routed onto the RF Signal Processor board. Additional functions of the RF signal processing board are to provide additional variable gain, sum level, and delta level detection as follows. Both the ‘DELTA RF’ and the ‘SUM RF’ signal are further amplified with a variable gain amplifier. Both signals are then sampled using a bi-directional coupler. The sampled signal is used for level detection. The detected signals are given the name ‘SUM LEVEL’ and ‘DELTA LEVEL’. The purpose of the ‘SUM LEVEL’ and ‘DELTA LEVEL’ signals are to determine the strength of the received signal and the direction of the transmitter. The ‘DELTA LEVEL’ signal has the desirable attribute that it provides a sharp null when the antenna is steered in the direction of the transmitter. This allows for accurate tracking. The RF signal processing board interacts with a user supplied digital control system as depicted in Figure 3. Figure 3 shows an overview of the original system. As can be seen from Figure 3, the user supplied digital control system provides the RF signal processor with a steering control voltage. The RF signal processor subsequently provides each element with a steering control voltage. The control voltage is directly applied to each element’s set of phase shifters. The digital control system is also responsible for interpreting the ‘SUM LEVEL’ and ‘DELTA LEVEL’ signals. Based upon its interpretation, it may vary the gain of the RF Signal Processor’s variable gain amplifier and/or adjust the steering control.

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RF Signal Processor

Digital Control System“user supplied”

RF Signal to S-band Receiver

Steering Control (1.0 – 8.0 V)

Delta Level (270mV – 3.0V)

Sum Level (270mV – 3.0V)

Antenna Array

Figure 3: Top

Gain Control (VGA)(0-2.5V)

Level Block Diagram

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Redesign Objectives The following list details the redesign objectives:

• Maximize antenna gain / efficiency • Increase antenna bandwidth to 200 MHz covering 2200-2400 MHz • Maximize steering angle • Variable beamwidth (controllable) • Increase dynamic range of antenna • Add dual mode capabilities – transmit & receive at 10 W • Transmission and reception must operate in duplex mode

The following list details personal objectives to be gained at the end of this project:

• Expertise in antenna design • Learn antenna design CAD software (HFSS) • Learn RF microwave design software (ADS) • Perform simulations in both antenna design software and RF microwave design

software • Increase familiarity with RF microwave design terminology

Initially, the following schedule was adopted to perform the described objectives: Research and determine best CAD software to use, obtain license if Oct 2004 needed Attend course to learn CAD software, pending availability of course Dec 2004 Attend course to learn RF microwave design software Dec 2004 Literature search and review of the topic Feb 2005 Modify element design to include increased bandwidth April 2005 Modify array design & create transceiver design May 2005 (Deliverable is a proof of concept / virtual prototype; conceptual schematic & geometry) Produce a hardware prototype (if possible) May 2005

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Increase Antenna Bandwidth In order to increase the antenna bandwidth, two things must be realized. The antenna patch element bandwidth must operate at 2200-2400 MHz; all electronic components must satisfy the increased bandwidth requirement. This means that the bandwidth of both the first and second stage bandpass filters must be increased. Increase of Antenna Element’s Bandwidth Each antenna element consists of a single layer patch antenna. A number of different approaches are available to increase the bandwidth of patch antennas. The most common involve the use of cavities and ssubstrate techniques [3]. However, these techniques increase the complexity, size, and weight of the design. The simplest approto increase the height of the substrate. Thdrawback to this approach is that if the heighis made too tall, an increase in the production of surface waves will decrease the antenna’s efficiency.

tacked

ach is e

t

[4] y

The strategy to be employed is to moderately increase the height of the patch antenna in order to achieve the desired increase in bandwidth. Figure 4 shows the geometry of a typical patch antenna. Figure 4 is close to the original configuration, with the exception that the patch antenna was fed using a coaxial probe. More discussion with regards to Figure 4 will be given in the proceeding paragraphs. The basic design equations used are given by [5]. A is included in this report. For more detailed informat[5]. The model used throughout the design process cline model. Some of the concepts of the cavity modetransmission-line model assumes that the patch antenline with an open circuit at each end of the transmissiradiating apertures of the patch antenna. Since the enare lossy, this is only a rough approximation. The lenthat the transmission line transforms the total admittaimpedance. It can be shown from transmission line tIt is actually somewhat less to account for fringing/ra

Figure 4: Patch Geometr

brief overview of the design strategy ion, the interested reader is referred to onsists of the 1st order transmission-l are also deployed. The na may be treated as a transmission on line. These ‘open circuits’ are the d apertures radiate and subsequently gth of the patch antenna is chosen so nce of the open circuit ends to a real heory that this is approximately λ/2. diating effects.

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With the correct patch length, a real impedance is obtained. Now, the real input impedance can be matched to the line impedance feeding the patch. In the case of a coaxial feed, this is best illustrated by observing Figure 4. When the patch is in resonance, the electric field will be as depicted. As can be seen, the E-field is a maximum at the two edges of the patch antenna. The current will be close to zero at the two edges of the patch and a maximum at the center. Consequently, the impedance will be close to 0 at the center and relatively large at the ends. It can be shown that an approximate relation is given by the following equation [5]:

⎟⎠⎞

⎜⎝⎛=== 0

2cos*)0()( yL

yRinyoyRin π (2)

Thus, if one moves the coaxial feed lengthwise with respect to the patch, a resonance point may be found. The design strategy employed is as follows. Initially, the patch height is increased. The coaxial feed is centered along the width of the patch antenna. The coaxial feed is moved lengthwise along the patch until resonance is achieved at any arbitrary frequency. The patch length is adjusted to achieve resonance at the desired frequency of 2.3 GHz. An iterative procedure is necessary, as second order effects relate all design parameters; for example, adjusting the length of the patch antenna will slightly alter the required feed location.

In the original design, the patch height was set to 0.15”. The height was chosen due to the availability of excess material from another research project. The original dielectric material was Rogers Dielectric 5870, with a relative dielectric constant of 2.33. Rogers Corporation only carries standard sizes up to 0.125” thick. After contacting the manufacturer, they were able to verify availability of Dielectric 5880, 0.25” thick. This material has a slightly lower loss factor compared to 5880, with a relative dielectric constant of 2.2. It was decided that this would be a good candidate for the design. HFSS was used to model the antenna’s performance. The original model was developed using an example from [6]. The design model is depicted in Figure 5. The initial model used idealizations in order to improve simulation speed. These assumptions were later relaxed during final simulations. Assumptions include all conductors being

Figure 5: Patch Geometry

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PEC (perfect electric conductors), the ground plane being infinite in extent, and a reduced size radiation boundary. The simulated antenna was totally enclosed in an air box, as depicted in Figure 5. The walls of the box are assigned as the radiation boundary. A larger radiation boundary will yield more accurate results. According to HFSS documentation, the following requirements should be met for the radiation boundary to accurately simulate open space: 1) the radiation boundary must be greater than ¼ wavelength from any radiating surface; 2) boundary orientation must be set perpendicular to incident radiation. These requirements were met to a fair degree; during final prove-in of the simulation, they were improved. The patch height was increased to 0.25” using Dielectric 5880 as the substrate. The coaxial feed was centered along the width of the patch antenna. A failed attempt was made to move the coaxial feed lengthwise along the patch to reach resonance. After a great deal of consideration, it was determined that the increased probe length was introducing too much inductance into the feed. See Figure 6. Probe Inductance

Probe Inductance

Equivalent Circuit

Figure 6: Probe Inductance Two simple methods are available to counteract the introduced inductance. The first method is that a gap may be introduced between the probe and the patch. This will add a capacitance that negates the inductive reactance at the design frequency. Second, the probe width may be increased. This minimizes the inductance of the probe. The first method was too sensitive to design deviations. The second technique was found to be adequate. It was verified that a probe radius of 0.15 cm could achieve a reasonably good 50-ohm match. A probe radius of 0.15 cm corresponds to a 3 mm wide probe. This may be difficult to solder onto the patch due to heat dissipation, but construction should still be feasible. In addition to increasing the height, it can be shown that increasing the patch width lends towards an increase in bandwidth. Unfortunately, it was experimentally determined that an increase in patch width also increases inductance. Consequently, the patch width was also reduced from 6.239 cm to 6 cm in order to minimize inductance to achieve an acceptable impedance match. The final patch dimensions can be summarized as follows. The dielectric height is 0.25”; the patch width is 6 cm; the probe radius is 0.15 cm; the patch length is 3.913 cm. The patch is fed using an inset of 0.75 cm from the edge (this is the distance from the patch’s edge to the radial center of the coaxial feed). The final simulation was made more realistic by using the patch’s true conductor (finite thickness copper). The simulation volume was also expanded to include ~2 wavelength width radiation boundary (compared

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to ¼ wavelength in original simulation). The ground plane was reduced from being infinite in extent. The resulting simulation took ~30 minutes per iteration. The simulated results were not drastically different from the ideal simulations. We now proceed to describe the simulation results. Figure 7 shows the impedance match as a function of frequency vs. VSWR.

Figure 7: Antenna VSWR As can be seen from Figure 7, the resonant frequency is centered at 2.3 GHz. The 2:1 VSWR band extends from 2.2-2.4 GHz. This will allow ~ 90% of the incident power to be transmitted into a 50-ohm matched transmission line. Figure 8 shows a 3D plot of the radiation pattern in dB. As to be expected, the beam pattern is fairly symmetrical with respect to the z-axis. There is ~5 dB attenuation at ~60 degrees compared to the field calculated at broadside. This field is calculated with 1 W of incident power on the antenna’s input port.

Figure 8: 3-D Polar Plot from Patch Antenna Element

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In order to obtain a better estimate of the antenna’s expected performance in a receiver, the realized gain is plotted vs. theta for phi=0,90 degrees. These results are shown in Figure 9. The realized gain is defined as follows:

PincidentUgainrealized *4_ π= (3)

U = the radiation intensity in the specified direction (watts per steradian) Pincident = the total incident power on antenna’s input port (watts) This definition of gain indicates the actual power/steradian radiated compared to that of an ideal lossless isotropic radiator. The realized gain pattern shown in Figure 9 does not include coupling effects between elements. Figure 7 & 8 were also created with no regards to coupling effects. Results shown in Figures 7, 8, and 9 will differ from actual results due to the presence of other neighboring antenna elements. Coupling effects are dependent upon frequency as well as the scan angle of the patch antenna [9]. It is possible that coupling effects could reduce the realizable steering angle of the patch antenna array. We will briefly consider the effects of coupling as follows. The patch elements are positioned collinearly in the E-plane. The distance between patch edges is ~3.6 cm. This yields a normalized edge separation with respect to wavelength of ~0.27. According to [5, p 765], this will yield an |S12|2 of ~-20 dB. This result shows that coupling effects are not expected to be a dominate factor. All subsequent results assume that coupling effects may be ignored. However, it is recommended that a simulation be performed to verify that coupling effects do not significantly alter the obtained results. A simulation should be performed to test the complete antenna array performance while the array is steered both at broadside and at a maximum steering angle.

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‘Broadside’

Radiation Pattern of One Element, psi = 0, 90 degrees

Figure 9: Realized Antenna Gain

In order to more easily model the gain of the system in subsequent discussion, we will proceed to calculate an approximate relationship for the antenna gain with respect to θ. A good first order approximation for gain will have the following form [4, p 213]:

))sin(*cos(max*_ thetaalphagainrealized = (4) alpha=constant max=maximum realized gain Curve fitting to Equation 4 was accomplished by setting the maximum value to 6.32 dB as read from Figure 9. In order to obtain alpha, an arbitrary reading from θ=60 degrees was used. Figure 10 was obtained by plotting the described expression in Matlab (see Appendix I for the actual code).

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2

4

6

8

30

210

60

240

90

270

120

300

150

330

180 0

Approximate Radiation Pattern of One Patch Element

‘Relative Orientation of Element’

‘Broadside’

3rd-Order Filter

5th-Order Filter

Figure 11: PCB Layout

Figure 10: Approximate Antenna Gain

Comparison of Figure 9 and Figure 10 reveal that this approximation is reasonable. This approximation will be used in subsequent discussion of the overall system performance to include the effects of the array.

Increase of Bandpass Filters’ Bandwidth In order to increase the antenna bandwidth, all electronic components must satisfy the increased bandwidth requirement. This means that both the first and second stage bandpass filters must be modified. The original PCB layout contains two bandpass filters. The first bandpass filter is of 3rd-order, placed directly after the antenna input, while the other bandpass filter is of 5th-order directly after the low noise amplifier. It might be noted that placement of a filter before the low noise amplifier does not follow conventional design. The purpose

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of the 3rd-order filter is to attenuate signals in the 2400-2500 MHz band (802.11 WLAN traffic) to an acceptable level. This type of filter is referred to as an Interdigital Connect Filter. The interested reader is referred to [7] for a more detailed discussion of this filter type.

m1freq=dB(S(2,1))=-5.142

2.330GHz

1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.81.0 3.0

-45

-40

-35

-30

-25

-20

-15

-10

-5

-50

0

dB(S

(2,1

))

m1

B on

n. It

Similar

he original filter design was made using a

. p,

R

The first design strategy was to use the built-in wizard contained within ADS design software. The process involves specifying both the upper and lower pass and stop bands. The maximum allowable passband attenuation and the minimum allowable stopband attenuation are also specified. The filter order may be specified; if it is not, the wizard will calculate the minimum order needed to meet the requirements. Lastly, the response type may be specified as maximally flat, or Chebychev. An example of the best obtainable output for a 5th-order filter is presented in Figure 12. This result was obtained after considerable tuning of the input parameters. As can be seen, the resulting filter is not acceptable. There is a 5 dnotch in the center of the passband. The reasfor this unacceptable behavior is unknowappears that Agilent may not have thoroughly tested this design guide for robustness. results were obtained with the 3rd-order filter. Tproduct from Eagleware. Unfortunately, thissoftware was unavailable for immediate use. An alternative solution was to make use of optimization features contained within ADSThis solution proved to be a bit tedious to setubut once setup it is extremely flexible and effective.

freq, GHz

Figure 12: 5th Order Filter esponse using Built-in ADS

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Figure 13 shows the optimization circuit for a 5th-order filter. As can be seen, there are a total of four goal objects. The goal objects make reference to ‘stop1’,’stop2’,’avg’, and ‘delta’. ‘Stop1’ refers to the stopband attenuation found at the lower stopband frequency. ‘Stop2’ refers to the stopband attenuation found at the upper stopband frequency. ‘Avg’ refers to the average attenuation within the passband. ‘Delta’ refers to the maximum-minimum passband attenuation. Specific definitions of these variables are defined underneath the Meas Eqn object in the lower right hand corner. The optimization object, ‘Optim’, makes references to all of the defined goal objects, determines the search algorithm, weighting, and other options. All goals are each assigned an acceptable level. For instance, if the lower stopband is less than 40 dB, no penalty is incurred. If all goals are found to be within their acceptable level, ‘perfect’ optimization is achieved and optimization stops. In most situations, the designer is trying to obtain the ‘best’ possible solution with conflicting goals; consequently, acceptable levels may be set in such a way that the optimization does not reach an optimal value. In this scenario (this also describes the current filter optimization), the ability to weight goals relative to their importance level is a critical feature. This was taken advantage of in order to obtain a steep rolloff in the upper stopband edge of the 3rd-order filter (to remove as much 802.11 interference as possible). When the optimization cannot reach a ‘perfect’ optimization, the optimizer may be set to a maximum number of iterations. Lastly, another interesting feature of ADS’s optimization is the ability to see the results within a data display in real-time as the optimizer performs iterations. This facilitated troubleshooting/tuning of parameters to obtained the desired results.

Figure 13: 5th-Order Optimization Circuit

GoalOptimGoal4

RangeMax[1]=RangeMin[1]=RangeVar[1]=Weight=Max=-40Min=SimInstanceName="SP1"Expr= "stop2"

GOAL

GoalOptimGoal3

RangeMax[1]=RangeMin[1]=RangeVar[1]=W eight=Max=-40Min=SimInstanceName="SP1"Expr= "stop1"

GOALMLSUBSTRATE2Subst1

LayerType[2]=groundLayerType[1]=signalCond[2]=1.0E+50T[2]=34 umCond[1]=1.0E+50T[1]=34 umTanD=0.0021H=20 milEr=3.38

Di e le c tri c -i : ER[i ] , H[i ], TAND[i ]M e ta l -i : T[i ], COND[i ], TYPE[i ]

M e ta l -2

M e ta l -1

Di e l e c tri c -1

MeasEqnMeas1

stop2=dB(S(2,1)[freq_LSB])stop1=dB(S(2,1)[freq_USB])avg=mean(dB(S(2,1)[freq_lower::freq_upper]))delta=abs(max_dev-min_dev)max_dev=max(dB(S(2,1)[freq_lower::freq_upper]))min_dev=min(dB(S(2,1)[freq_lower::freq_upper]))freq_LSB= find_index(freq,2.1e9)freq_USB=find_index(freq,2.5e9)freq_upper= find_index(freq,2.4e9)freq_lower= find_index(freq, 2.2e9)

EqnMeas

GoalOptimGoal2

RangeMax[1]=RangeMin[1]=RangeVar[1]=W eight=1Max=Min= -1SimInstanceName="SP1"Expr= "avg"

GOAL

GoalOptimGoal1

RangeMax[1]=RangeMin[1]=RangeVar[1]=Weight=1Max=1.5Min=SimInstanceName="SP1"Expr= "delta"

GOAL

S_ParamSP1

Step=0.02 GHzStop=40 GHzStart=2.0 GHz

S-PARAMET ERS

OptimOptim1

UseAllGoals=yes

UseAllOptVars=yesSaveAllI terations=noSaveNominal=yesUpdateDataset=yesSaveOptimVars=yesSaveGoals=yesSaveSolns=yesSeed= SetBestValues=yesNormalizeGoals=yesFinalAnalysis= "SP1"StatusLevel=10DesiredError=0.0MaxIters=999OptimType=Random

OPT IM

TermTerm2

Z=50 OhmNum=2

TermTerm1

Z=50 OhmNum=1

VARVAR1

L2=660 mil opt{ 550 mil to 700 mil }L1=126 mil opt{ 80 mil to 160 mil }S2=54 mil opt{ 20 mil to 80 mil }S1=43 mil opt{ 20 mil to 80 mil }W 1=37 mil opt{ 20 mil to 120 mil }

E qnVar

ML5CTL_VCLin1

ReuseRLGC=noRLGC_File=Layer[5]=1

Layer[4]=1Layer[3]=1Layer[2]=1Layer[1]=1W[5]=W 1S[4]=S1W[4]=W 1S[3]=S2W[3]=W 1S[2]=S2W[2]=W 1S[1]=S1W[1]=W 1Length=L2Subst= "Subst1"

ML5CTL_VCLin2

ReuseRLGC=noRLGC_File=Layer[5]=1

Layer[4]=1Layer[3]=1Layer[2]=1Layer[1]=1W[5]=W 1S[4]=S1W[4]=W 1S[3]=S2W[3]=W 1S[2]=S2W[2]=W 1S[1]=S1W[1]=W 1Length=L1Subst= "Subst1"

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The results obtained for the 5th-order filter after optimization are shown in Figure 14. For brevity, the 3rd-order optimization results are not shown (final simulation results for both filters will be included at the end of this section). As can be seen from Figure 14, the maximum dip in the passband is 1.5 dB—a dramatic improvement over that obtained with the wizard. Also, the rolloff of the filter is steeper. As a final verification o erformed using

omentum. Momentum is a 2.5d RF electromagnetic simulator packaged with ADS. This

slightly lengthened to compensate. The final simulation results of both the 3rd-order and 5th-order filters are given by Figures 16 and 17, respectively. These results are acceptable, and this completes the discussion of the filter design.

m1freq=dB(S(2,1))=-1.52

f the filter design, a more accurate simulation was pMallows for relatively quick simulation of the designed filters by importing the filter design into Momentum. Figure 15 shows an example layout of the 5th-order filter imported into Momentum. Some necessary changes were made to make the layout realizable; for example, the ground vias were made circular. As a result, the lengthwise dimensions had

to be

22.330GHz

1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.81.0 3.0

-60

-50

-40

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0

freq, GHz

dB(S

(2,1

))m1

Figure 14: 5th-Order Filter Response using ADS Optimization

Figure 15: 5th-Order Filter Layout in Momentum

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S21

Frequency Figure 16: 3rd-Order Filter Frequency

Response in Momentum

2.1 2.2 2.3 2.4 2.5 2.6 2.72.0 2.8

-50

-40

-60

0

-30

-20

-10 M

ag [d

B]

2.1 2.2 2.3 2.4 2.5 2.6 2.72.0 2.8

-60

-50

-40

-30

-20

-10

-70

0

Frequency

Mag. [dB]

S22

Figure 17: 5th-Order Filter Frequency Response in Momentum

Frequency

Mag

[dB

]

S21

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Concept design of Antenna Element PCB Steering Redesign The following design objectives will be addressed in this sub-section:

• Maxim ering angle • Variable beamwidth (controllable)

The original configuration obtained steering through the use of two phase shifters. Each phase shifter only allows for a maximum phase shift ~110 degrees. This yields a total maximum phase shift of 220 degrees. To increase the steering angle, a larger phase shift is needed. Upon consultation with the manufacturer, SVMicrowave, by utilizing three phase shifters in series, a full 360 degrees of phase shift should be obtainable. The only sacrifice being made is an extra i ti Another way to maximiz rol the phase shift. The original configuration used uit in order to control each phase shifter’s phase shift. This limresult, the maximum steering angle is ~20 degrees. Also, the analog control assumes that the delta phase shift is linear with respect to voltage. Unfortunately, the linearity of the phase shifter’s voltage response curve is mediocre at best (see [1] for details). In order to overcom ility in beam control, the phase shifters in the new design will be individually control the digital control system. This will allow for digital linearity compensation and added flexibility via arbitrary phase control. The second objective listed is to add variable beamwidth control. In order vary the beamwidth, a tape tribution of weights is required. (Each weight is multiplied by the incoming signal of each element, and then summed together.) The comparison made in most antenna textbooks is that a tapered distribution is mathematically analogous to windowing of a Fourier series. Therefore, a tapered distribution will have a wider beamwidth and lower sidelobes. The original configuration only changed the phase of each received signal, with no tapering. Tapering of the weights is to be obtained in the new configuration through the utilization of variable gain amplification on each antenna element. These will be controlled by the digital control system. Figure 18 shows the modified block diagram of the antenna element’s PCB to include an additional phase shifter, digital control of each phase shifter, and variable a ifi on. Figure 19 shows a top system level diagram. La nce, Figure 20 shows a block diagram of the RF signal proces

sor. stly, for refere

catimplgain

red dis

e these weaknesses, and add more flexle

ibd by

e the steering angle is to more accurately contan analog voltage divider circ

ited the maximum delta phase shift to ~30 degrees; as a

on loss. nser 1.3 dB of

ize ste

ual Re

19 / 44

Figure 18: Block Diagram of a Modified Antenna Element

Φ1

Φ2

Φ3

SC (Steering Control Voltage)

VGA (Variable Gain Amplifier)

Phase Shifters

BS (Beam Shaping Voltage)

LNA (Low Noise Amplifier)

Output

5 -Order BPF (Bandpass Filter)

th

r3 d-Order BPF (Bandpass Filter)

Patch Antenna

20 / 44

RF Signal Processor

Digital Control System“user supplied”


Steering Control Bus (0-10V)

8

Beam Shaping Bus (0-2.5V)

8

Gain Control (VGA) (0-2.5V)

Delta Level (270mV – 3.0V)

Sum Level (270mV – 3.0V)

Figure 19: Block Diagram of a Modified System

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Antenna Element Circuit #1

SC1 BS1


SC2 BS2


SC3 BS3


SC4 BS4


SC5 BS5


SC6 BS6


SC7 BS7


SC8 BS8

4-in

put p

ower

com

bine

r

Power Combiner

“Σ”

Hybri

Degree

d

180

“∆”

RF

Pres

soc

ing

Circ

uit

VGA


evel

Delta Level

Sum L

n

Figure 20: Block Diagram of RF Signal Processor

4-in

put p

ower

com

bine

r

Key: SC = Steering Control BS = Beam Steering VGA = Variable Gain Amplificatio

22 / 44

We are now in a position to simulate the generated beam pattern obtained from both the ante t and the weighted summer. It can be shown that both the element beam pattern and the array beam pattern may be multiplied together to form the composite beam pattern. In the simulations to follow, the array beam pattern has been normalized to yield 0 dB in the direction of arrival. One may envision a constant 9 dB added to the simulated plot to take int unt the array gain associated with adding all elements toge would de the overall antenna gain with respect to an isotropic antenna. Nevertheless, the follow ots accurately show the shape of the beam pattern. The first simu se a tapered distribution. The beam pattern will be calculated at steering angles (relative adside) of 0, 30, 60, and 90 degrees. Figure 21 shows the results. As can be seen f igure 21, when the beam is steered to 90 degrees, the elem patterns starts to attenuate the composite gain. Also, the peak radiation is not at the steering angle since the element’s beam pattern is strongly distorting the com pattern. For comparison, a maximum gain of 2.1stee f 60 deg The actual realized steering angle is 48 degrees. On the other hand, a maximum of gai 6.25 dB is achieved for a steering angle of θ=0 degrees (relative to broadside). Values for all an bulated in Table 1.

nna elemen

s in order ther. This

o accoscribeing pl

to brorom F

rees. n of ~

lation will not u

ent’s beam

posite beamring angle o

dB is achieved for a

gles are ta

‘Relative Orientation of Phased Array’

AB

pproxima diation Pattern of Composite eam at V Steering Angles

te Raarious

Broadside’ ‘

Fi Simulated gain for various steering angles, using auniform distribution

gure 21:

23 / 44

Steering Angle (deg)

3-dB Beamwidth (deg)

Maximum Gain (dB)

Realized Steering Angle (deg)

Sidelobe Level (dB)

0 27.5 6.3 0 -10.5 30 30.2 5.0 25.9 -7.0 60 42.1 2.1 47.9 -6.5 90 49.1 0.35 57.8 -6.7

Table 1: Tabulated parameters of simulated gain for various steering angles,using a uniform distribution
The second simulation will use a tapered distribution. The tapered distribution to be used is the Kaiser distribution, with Beta=3 [8, p 107]. A larger value of beta will increamain beamwidth and decrease the sidelobe level. As in Figure 17, the beam pattern wilcalculated at steering angles (relative to broadside) of 0, 30, 60, and 90 degrees. Figure 22 shows the results. As one would expect, the element’s beam pattern still distorts the arbeam pattern. By comparison to Figure 21, the beamwidth has widened. lso, one
bserves substantially lower side lobes. It should be noted that the peak gain is lower.

se the l be

ray’s A

sequence of the variable gain amplifiers being turned down to obtain a tapered distribution. Still, by close observation, the gain is greater than that obtained in Figure 21 over a broader range of angles. Table 2 contains tabulated values of various parameters for comparison with Table 1.

oThis is a con

Figure 22: Simulated gain for various steering angles, using a Kaiser tapered distribution

‘Broadside’

‘Relative Orientation of Phased Array’

Approximate Radiation Pattern of Composite Beam at Various Steering Angles

24 / 44

S T

Idbrvo

Steering Angle (deg)

3-dB Beamwidth (deg)

Maximum Gain (dB)

Realized Steering Angle (deg)

Sidelobe Level (dB)

0 32.9 3.2 0 -26.8 30 35.6 2.0 24.5 -21.6 60 46.4 -0.76 44.5 -20.590 55.8 -2.3 52.9 -20.5

(m FcpVacDsCapBCOinth

Table 2: Tabulated parameters of simulated gain for various steeringangles, using a tapered distribution

chematic Redesign

he following design objectives will be addressed in this sub-section: • Increase dynamic range

n this subsection, the Antenna Element’s schematic will be modified to incorporate theiscussed changes. Minimal changes will be necessary for the new RF Signal Processor oard, so no discussion of the RF Signal Processor board is given (notably, only the emoval of the analog phase shifter voltage section will be necessary). The addition of a ariable gain amplifier will improve the dynamic range of the system. The implementation f the schematic into ADS

will be briefly discussed. Some components were simplified with the proper use of engineering judgment) in order to allow implementation in a timely anner.

iguonnected directly to port 1. Both the 3rd- and 5th-order filters are modeled using the reviously discussed design. AMP1 and AMP2 use a manufacturer-supplied model. The P242D phase shifters are modeled using their scattering parameters. There is

pproximately 1.3 dB of insertion loss for each. The phase shift may be manually ontrolled for each phase shifter. Unfortunately, it was not possible to easily convert the C supplied phase shift control voltage into a phase shift due ns of ADS

oftware. The only feasible method for doing this would have involved custom -programming within the ADS environment. Lastlymplifier is modeled using a system level amplifier. arameters are incorporated from the datasheet. The GA2031 described its small-scale nonlinear distortihannel Power Rejection). Unfortunat I (Third rder Intercept) value for insertion into the system level amplifier. (The third order terce ual to the ird h

re 23 shows a top level view of the antenna element’s schematic. The antenna is

to limitatio

, the BGA2031 variable gain The P1dB point and scattering amplifier specification sheet for on using the term ACPR (Adjacent

ely, this is not easily convertible to a TO

pt is the theoretical output power of the desired signal when it would be eqarmonic.)

25 / 44

MLSUBSTRATE2Subst1

LayerLayerType[1]=signalCond[2]=174000T[2]=34 umCond[1]=174000T[1]=34 umTanD=0.0021H=20 milEr=3.38

Type[2]=ground

Dielect r ic- i : ER[ i] , H[ i] , TAND[ i]M et al- i : T[ i] , CO ND[ i] , TYPE[ i]

M et al- 2

M et al- 1

Dielect r ic- 1

MSUBMSub1

Rough=.0014 meterTanD=.0021T=34 umHu=3.9e+034 milCond=174000Mur=1Er=3.38H=20 mil

MSub

S_ParamSP1

Step=10 MHzStop=3.0 GHzStart=2.0 GHz

S-PARAMETERS

TermTerm1

Z=50 OhmNum=1

PortVGA_ControlNum=1

CCX5C=100 pF

LLX1

R=L=22 nH

Port+3_vDCNum=2

DC_FeedDC_Feed3

CCX10C=100 pF

DC_FeedDC_Feed5

RRX2R=0

RRX1R=0

CC22C=0.01 uF

BGA2031X4

PortRF_OutputNum=4

LLX2

R=L=22 nH

DC_FeedDC_Feed4

CCX7C=1000 pF

CCX6C=0.1 uF

CCX9C=1000 pF

CCX11C=0.1 uF

TermTerm2

Z=50 OhmNum=2

RRXR=22 Ohm

CCX8C=36 pF

CCX4C=36 pF

PortAntenna_InputNum=5

5th_orderX6

CCX12C=36.0 pF

DC_Feed_Feed1DC

CC21C=100 pF

3rd_orderX7

CC2C=36.0 pF

LL1

R=L=4 nH

RR1R=68 Ohm

CC19C=1000 pF

CC1C

8=0.1 pF

LFB1

R=L=100 nH

V_SR

DCC2

dc=5.0 VV

CC3C=36 pF

sa_hp_MGA-86576_19930601AMP1

OFFSET=0 mil SMT_Pad="Pad1"

sa_hp_MGA-86576_19930601AMP2SMT_Pad="Pad1" OFFSET=0 mil

DC_FeedDC_Feed2

CC20C=1 uF

RR2R=150 Ohm

LL2

R=L=4 nH

CC5C=22.0 pF

CC4C=36.0 pF

D dD d6

C_FeeC_Fee

VP242DX2

VP242DX1

VP242DX3

CCX2C=0.1 uF

CCX3C=1000 pF

CC15C=0.1 uF

CC14C=1000 pF

CC17C=0.1 uF

CC16C=1000 pF

CCX1C=36 pF

PortPhase_Shift_Control

CC6C=36 pF

Num=3

in

ors.

The following set of tests was executed. Results are described within the corresponding figures:

• Gain. The overall system gain was measured. Figure 24. • Gain Compression. The power level for the system experiencing 1 dB of ga

compression was computed. Figure 25. • Noise Figure. The overall noise figure of the system with the various contribut

Figure 26. • Intermodulation. The system receives two tones at maximum power of -35 dBm.

The simulator calculates the resulting mixer terms within the system bandwidth. Figure 27.

Figure 23: Top-Level Schematic of Antenna Element

26 / 44

The65dB of attenuation occurs for a signal falling outside of the passband by 200 MHz. It shopro s

overall system gain is ~65 dB within the passband of the receiver. Approximately

uld be noted that an additional 20 dB of variable gain is available in the RF signal ces or.

Figure 24: Overall System Gain vs Frequency

2.2 2.4 2.6 2.82.0 3.0

-40

-20

0

20

40

60

-60

80

freq, GHz

dB(G

ain_

Tota

l..S

(2,1

))

Antenna Element PCB, S21 Gain

Antenna Element PCB, S21 Phase

2.2 2.4 2.6 2.82.0 3.0

-100

0

100

-200

200

freq, GHz

pG

ain_

Tota

l..S

(2,1

))ha

se(

27 / 44

m

t to its minimum setting, Pin can be as high as –35 dBm without gain mpression.

The P1dB output power is 11.69 dBm. With the 3rd-stage VGA amplifier set to maximusetting, Pin can be as high as –53 dB without gain compression. With the 3rd-stage amplifier seco

Figure 25: Gain Compression of System

P1dB

q=m(HB1.

2.300

output power

m2fredB HB.Vout)=11.685

GHz

0.5 1.0 1.5 2.00.0 2.5

2

4

6

8

10

0

12

freq, GHz

O/P

Pow

er a

t 1dB

com

p (d

Bm)

m3Pin=linear=13.071

-53.000

m4Pin=dBm(HB2.HB.Vout[::,1])=11.971

-53.000

-90 -80 -70 -60 -50-100 -40

-20

0

20

-40

40

Pin (dBm)Po

ut (d

Bm)

m3m4

Output Pow er vs Pin

-90 -70 -60 -50 -40 -30-100 -20

P1dB compression calculated using ADS Gain Compression Simulator (P1dB = 11.685 dBm)

P1

-80

-80

-60

-40

-20

0

-100

20

Pin (dBm)

Pout vs 1st, 2nd, and 3rd Stage Amplifiers

With a setting of -33dB gain f or 3rd stage amplf ier, Pin can be as highas -35 dB without gain compression.

dB compression calculated by manually varying of Pin

Output Power Linear Fit

2nd Stage Amplifier 1st Stage Amplifier (LNA)3rd-Stage Amplifier (Variable Gain)

With a setting of -33 dB gain for the 3rd stage VGA amplifier, Pin can be as high as –35 dBm without gain compression.

Pin for

)dB

mPo

ut (

28 / 44

nffreq

2.200GHz2.225GHz2.250GHz2.275GHz2.300GHz2.325GHz2.350GHz2.375GHz2.400GHz

nf(1) nf(2)3000.0003000.0003000.0003000.0003000.0003000.0003000.0003000.0003000.000

2.4942.4452.7052.4122.1212.4082.8532.6722.867

Noise_Figure1..port1.NC.freq2.

Noise_Figure1..port1.NC.name300GHz

AMP1.a1AMP2.a1

R1X1.S2P2X6.CLin1X6.CLin2X7.CLin1X7.CLin2

_total

Noise_Figure1..port1.NC.typeAmplifierAmplifier

RS_Port

Lineine

Noise_Figure1..port1.NC.vnc

PC_PC_LPC_LinePC_Line

_total

200.1pV1.132pV85.55fV16.53fV917.5fV544.9fV

129.0pV41.55pV241.7pV

Figure 26: Noise Figure of System

Noise Figure Contributions

The system noise figure is 2.12 dB at 2.3 GHz. As to be expected the major contributor is

e first stage LNA. The noise figure of the LNA is ~2 dB. th

29 / 44

f req

0.0000 Hz50.00MHz100.0MHz150.0MHz200.0MHz250.0MHz300.0MHz350.0MHz400.0MHz450.0MHz500.0MHz1.950GHz2.000GHz2.050GHz2.100GHz2.150GHz2.200GHz2.250GHz2.300GHz2.350GHz2.400GHz2.450GHz2.500GHz2.550GHz2.600GHz

MixMix(1) Mix(2)

0-1-2-3-4-5-6-7-8-9

-1010

9876543210

-1-2-3

0123456789

10-9-8-7-6-5-4-3-2-101234

m1freq=dBm(Vout)=16.099

2.400GHz

m2freq=dBm(Vout)=-10.703

2.350GHz


2.400GHz


2.350GHz

-120

-100

-80

-60

-40

-20

0

-140

20

2.20 2.25 2.30 2.35 2.40 2.45 2.50 2.55 2.602.15 2.65

f req, GHz

dBm

(Vou

t)

m1

m2

m signal level for the receiver).t.

The input signals are 2.4 and 2.45 Ghz tones at -35 dBm (the maximuAs can be seen, there is ~26 dB of attenuation in the third order produc

Figure 27: Intermodulation System Test

Mixing Products Included in Simulation

Mixing Product In-Band

Signal Output at Edge of Passband

The input signals are 2.4 and 2.45 GHz tones at –35 dBm (the maximum signal level for the active antenna). As can be seen, there is ~26 dB of attenuation in the third order product.

30 / 44

Trans The following design objectives will be addressed in this sub-section:

• Add dual mode capabilities – transmit & receive at 10 W • Transmission and reception must operate in duplex mode

The ideal scenario would be to use the same antenna for both transmit and receive. Unfortunately, this is not possible. This is a result of the requirement that the antenna be able to send and receive simultaneously at the same frequency. The closest realization would have made use of an isolator. Due to the relatively poor mismatch of the antenna, reflected signal from the transmit circuit would overwhelm the receiver. No filtering could be used to attenuate the reflected signal since it is the same frequency as the receive section. Upon consultation with the customer, it was decided to make the transmitter separate from the receiver. Consequently, the device would only be able to transmitthat 8 W of transmit power was adequate. An additional requirement to trrange of 1 mW thru 8 W was added. Fortunately, the receive antenna would be able to reuse a large portion of receiver design. Figures 28 and 29 show a block diagram of the preliminaconcept. Notice that the composite output ranges from 0.23 mW thru 8 W. The low side of 0.23 m will allow tapering of the elements when operating at the minimal power of 1 mW. Unfortunately, the same is not true when operating at 8 W. A brief description of the signal flow in Figures 28 and 29 follows. The modulated signal is amplified to a level such that the phase shifters in each antenna element will receive a signal level of ~0 dB of signal power after going through a splitter. This is a requirement of the phase shifters for optimal performance. The signal is then filtered to remove any spurious signals. Finally, the signal is amplified to 1 W (or the desired transmit power level) at each antenna element. As a result, the total radiated power is ~8 W (not including losses in the antenna structures). The c r is to provide a ‘driver amplifier’ that meets the listed specification: 50 dB dynamic range, -10 dB minimum gain, 40 dB maximum gain, 1 W maximum output. Further requirements of the ‘driver amplifier’ are listed in subsequent ragra To verify the calculations in Figures 28 and 29, a simplified schematic was incorporated into ADS. The implementation of the schematic into ADS will be briefly discussed. Some omponents were simplified in order to allow implementation in a timely manner. These

simulate insertion loss. Although not explicitly used in the simulation, the power splitter’s model has an isolation of 20 dB. It is recommended that the cables connecting the splitter to the active elements be close to the same length (~3/100*λ or ~3 mm). This is not crucial since path length differences may be accounted for by proper adjustment of the phase

mit Capabilities

. It was also decided ansmit through a

the existing ry design

W

ustome

pa phs.

cwill be briefly discussed. The power splitter was modeled by configuring multiple two-way splitters in series. An attenuator is placed within the power splitter circuit in order to

31 / 44

shifters. The power amplifier was reviously discussed amplifier prop

modeled using a system amplifier. In addition to the erties, the following were also incorporated. The

f 30.2 dBm, and a 40 by comparison of a

sim rmance, these spe ic er filter and VP242D

ha s na design.

pamplifier has a NF (Noise Figure) of 5 dB, a 1 dB compression point odBm TOI (Third Order Intercept). These specifications were obtained

ilar off the shelf component. In order to ensure adequate perfocif ations should be met or exceeded. Lastly, both the 5th-ord

p

se hifters use the same model as that used in the receive anten

32 / 44

Figure 28: Top-Level Block Diagram of Active Antenna Array

Driver Amplifier1 W Maximum Power40 dB Maximum Gain50 dB Dynamic Range-10 dB Minimum Gain

Composite Pow-6.4 dBm (0.2339 dBm (8 W)Dynamic Range

Signal Level = 0 dBm thru 30 dBmAdjust Signal to 9.7 dBm

Signal Level = -9.7 dBm through 20.3 dBmMust Limit Signal Level to exactly 0 dBmfor Active_Antenna Element (Phase Shifter limitation)

Signal Level -13.1 dBm thru 30.0 dBm (1 W)'Active Antenna Array'

'Transmitter'

Signal Level = -25 dBm

er Radiated mW)

= 43.1 dB

S1P_EqnActive_Antenna_Element1








SplitterX1

Isolation=20 dBInsertion_Loss=0.7 dB

2

3

4

6

5

7

8

9

1

S1P_EqnModulator

AmplifierDriver_Amplifier

33 / 44

Figure 29: Top-Level Schematic of Active Antenna Element

May add optional 5th order filterto remove spurious signals outsideof 2.2 - 2.4 GHz

Insertion Loss = ~1.5 dB

Signal Level -15.4 dBm thru 30.0 dBm (1 W)

Driver Amplifier1 W Maximum Power40 dB Maximum Gain50 dB Dynamic Range-10 dB Minimum Gain

Signal Level -5.4 dBmSignal Level -3.9 dBmSignal Level 0 dBm

'Active Antenna Element'

RF Power = 0 dBm NominalInsertion Loss = 1.3 dB TypicalReturn Loss = 18 dB



AmplifierDriver_Amplifier1

5th_order5th_order_filter

S1P_EqnPhase_Shifter3



34 / 44

in the corresponding figures:

• • Gain Comp e system experiencing 1 dB of gain

comp• with the various

• Interm the modulator at maximum power, -28 dBm ixer terms within the system

The following set of tests was executed. Results are described with

Gain. The overall system gain was measured. Figure 30. ression. The power level for th

ression was computed. Figure 31. Noise Figure. The overall noise figure of the systemcontributors. Figure 32.

odulation. The system receives two tones from. The simulator calculates the resulting m

bandwidth. Figure 33.

35 / 44

With a maximum gain setting, the overall system gain is ~55 dB. With a modulated signal of –25 dBm, this will produce an output signal of 30 dBm, or 1 W. This is the output of one element. The composite power output will be 8 W. Similarly, for minimum gain setting, a modulated signal of –25 dBm will produce an output signal of -15 dBm, or 0.03 mW. This is the output of one element. The composite power output will be 0.24 mW.

2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.92.0 3.0

10

20

30

40

50

0

60

freq, GHz

dB(S

(2,1

))

2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.92.0 3.0

-150

-100

-200

-50

0

50

100

150

200

freq, GHz

,1)

phas

e(S(

2)

2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.92.0 3.0

-40

-30

-20

-10

0

10

-50

20

freq, GHz

dB(M

in_G

ain_

Tran

smitt

er..S

(2,1

))

2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.92.0 3.0

-150

-100

-50

0

50

100

150

-200

200

freq, GHz

phas

e(M

in_G

ain_

Tran

smitt

er..S

(2,1

))

Figure 30: Overall Active Antenna Gain vs Frequency

Maximum Gain Setting

Minimal Gain Setting

36 / 44

The input power from the modulator is set to a constant of 25 dBm. The VGA (Variable

is he

Gain Amplifier) of the patch is varied. With the VGA set to 35 dB, the output power ~30 dBm (1 W). Under this scenario, the transmitter is outputting maximum power. Tamplifier is experiencing 0.844 dB of gain compression.

9.970 10 20 400 30-1

-10

0

10

20

30

-20

40

VGA (dB)

Pout

(dBm

)

m3m4Output Power vs Pin

Element_Gain35.000

compression0.844

Gain of VGA in Transmitter Element

Figure 31: Gain Compression of itter Element Antenna Transm

37 / 44

freq

2.200GHz2.225GHz2.250GHz2.275GHz2.300GHz2.325GHz2.350GHz2.375GHz2.400GHz

nfnf(1) nf(2)

0.0000.0000.0000.0000.0000.0000.0000.0.

5.0345.0365.0355.0325.0355.0375.0365.0355.036

000000

...itter_No e_Figure1..port2.NC.freqis

2.300GHz...ter_Noise_Figure1..port2.NC.name

_totalAMP1

X2.AMP1X2.X3.S2P2

X1.PWR13.CMP1X2.X2.S2P2X2.X1.S2P2X2.X4.CLin1

X1.PWR11.CMP1X2.X4.CLin2

X1.PWR10.CMP1X1.ATTEN1.CMP1

X3.X1.S2P2X6.X1.S2P2

...itter_Noise_Figure1..port2.NC.type_total

AmplifierAmplifier

S_PortS_PortS_PortS_Port

PC_LineS_Port

PC_LineS_PortS_PortS_PortS_Port

...itter_Noise_Figure1..port2.NC.vnc428.6nV426.1nV38.74nV12.06nV10.44nV10.39nV8.912nV8.586nV7.110nV4.819nV4.234nV2.262nV1.625nV1.625nV

Noise Figure of Antenna Transmitter

Noise Figure Contributions

Figure 32:

The system noise figure is 5 dB. The primary contributor is the first stage driver amplifier before the power splitter. Since the system signal level is at 25 dBm, the noise

gure should not introduce a significant amount of noise. fi

38 / 44

f req

0.0000 Hz50.00MHz100.0MHz150.0MHz200.0MHz250.0MHz300.0MHz350.0MHz400.0MHz450.0MHz500.0MHz1.950GHz2.000GHz2.050GHz2.100GHz2.150GHz2.200GHz2.250GHz2.300GHz2.350GHz2.400GHz2.450GHz2.500GHz2.550GHz2.600GHz

MixMix(1) Mix(2)

0-1-2-3-4-5-6-7-8-9

-1010

9876543210

-1-2-3

0123456789

10-9-8-7-6-5-4-3-2-101234


2.400GHz


2.350GHz


2.400GHz


2.350GHz

The input signals are 2.4 and 2.45 GHz tones at -28 dBm (the maximum signal level for the transmitter). As can be seen, there is ~33 dB of attenuation in the third order product.

2.20 2.25 2.30 2.35 2.40 2.45 2.50 2.55 2.602.15 2.65

-120

-100

-80

-60

-40

-140

-20

0

20

40

f req, GHz

dBm

(Vou

t)

m1

m2

Figure 33: Intermodulation of Active Antenna

Mixing Products Included in Simulation

Mixing Product In-Band

Signal Output at Edge of Passband

39 / 44

onclusio

or the presented conceptual redesign, it has been shown that all major redesign bjectives have been met. The antenna bandwidth was increased to operate over the 200-2400 MHz band. Simulations show that the new patch element meets this andwidth yielding a VSWR less than or equal to 2 in the passband. This corresponds to pproximately 90% of the power being transmitted at the edge of the passband. The andpass filters were modified to accommodate the extended bandwidth. The antenna eamforming circuit has been modified to include digital control over both arbitrary hase and attenuation of each element. Simulations incorporating both the element and rray beam pattern show that the beam may be steered to a realizable angle of 60 degrees om broadside. At a steering angle of 60 degrees, however, the main return axis is ttenuated by ~6 dB. This is due to the limitations of the patch element’s beam pattern. he receive antenna circuit was modified to accommodate an increased dynamic range. simulation of the antenna revealed the capability of receiving a signal power as high as 35 dBm without overdriving the system. The gain of the receive systeme ~65 dB with an additional 20 dB of variable gain within thether tests include a noise figure (2.1 dB) and an intermodulation test (~26 dB of ppression). Next, a block diagram design for an active transmit antenna was presented. iven that the discussed driver amplifier specification is met, the a

apable of transmitting within a range of 0.25 mW thru 8 W. As ae block diagram, a simulation in ADS was performed. The circuit demonstrated

cceptable performance in a suite of tests to include gain, gain compression, noise figure nd intermodulation. In conclusion, these results show that both a new receive antenna nd an ive transmit antenna should be capable of meeting the proposed redesign bjectiv .

C ns Fo2babbpafraTA– was shown to

RF signal processor. bOsuG ctive antenna will be

second validation of cthaaa act

eso

40 / 44

Further Work The following outline briefly summarizes anticipated future work with the seconditeration of the antenna array.

1. Build prototype of antenna element a. Verify / Order required components b. Calibration strategy c. Complete board layout

i. Antenna element ii. Signal processing board

iii. Patch antenna d. Fabricate antenna e. Calibrate and test antenna f. Interconnect user supplied digital controller g. Create desired adaptive beamforming algorithm

2. Build prototype of active antenna element (transmitter)

a. Complete design of driver amplifier b. Verify / order required components c. Calibration strategy d. Complete board layout

i. Antenna element ii. Signal processing board

iii. Fabricate antenna

design

e. Calibrate and test antenna f. Interconnect user supplied digital controller g. Create desired adaptive beamforming algorithm

3. Future enhancements a. Possible to use same antenna for transmit and receive

i. Use different operating frequencies ii. Isolate by using time division multiple access

b. Increase number of elements / size of antenna i. May not be feasible given design constraints

41 / 44

References

ive, Volume I, Norwood, MA: Artech House, 1981, p 160. ] Radiation Patterns Using

[4] Stutzm enna Theory & Design. New Sons, Inc., 1998.

[5] Bal s nalysis and Design, John Wiley and Sons, New h Frequency Structure Simulator v9 Us orporation, Pittsburgh, PA, March 2004.

[7] Mat e icrowave Filters, Impedance-Matching Networks,

[8] Van John Wiley & Sons, Inc.,

] l. AP-13, pp. 342-345,

[1] Depardo, Dan, “2.3-2.4 GHz Phased-Array Prototype”, 2004. [2] Ulaby, Fawwaz; Moore, Richard; Fung, Adrian, Microwave Remote Sensing

Active and Pass[3 Huie, C. Keith, Microstrip Antennas: Broadband

Photonic Crystal Substrates, 2002. an, Warren L. & Gary A. Thiele. AntYork, NY: John Wiley &

ani , C.A. Antenna Theory: A York, 1997.[6] Ansoft Hig

r’s Guide, Rev 2.0, Ansoft CeYoung and Jones, Mtha i,

and Coupling Structures, Artech House, 1980, pp. 614. Trees, Harry L. Optimum Array Processing, Part IV of Detection,

Estimation, and Modulation Theory. New York:2002.

[9 P. W. Hannan and M. A. Balfour, “Simulation of a phased-array antenna in waveguide,” IEEE Trans. Antennas Propagat., voMay 1965.

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Appendices Appendix I: Matlab Code

Figure 9

%Written by:

point1_db=6.

maximum=10^(

alpha=acos(p

zero=20*log10(element)>0 element_db=element_db.*zero polar(theta,element_db)

Figure 17, 18

%Written by: Russell Hofer %March 2005 %Program Description: Consummate beam pattern at various steering % angles Uniform or Tapered Weighting N = 8; % Elements in array d_cm = 2.9 % Element spacing in centimeters d_m = d_cm/100; % Element spacing in meters c = 3e8; % Speed of light f = 2.3e9; % Nominal frequency of array lambda = c/f; d = d_m/lambda; % spacing wrt wavelength steering = 90 % steering angle in degrees (90 = broadside) Beta = 0 % Beta = 0 => not tapered, positive value => tapered D=d*[-(N-1)/2:1:(N-1)/2]; % element locations ang = pi*[-1:0.001:1]; u = cos(ang); %calculate an approximate beam pattern for an element

Russell Hofer %March 2005 %Program Description: Approximate Element Beam Pattern theta=-pi/2:.01:pi/2;

32 %maximum value of realized gain (db) point2_db=1.17 %realized gain at theta=60 degrees

point1_db/20) point2=10^(point2_db/20)

oint2/maximum)*2/(3^.5)

element=maximum*cos(alpha*sin(theta)); element_db=20*log10(element)

43 / 44

theta=pi*[-1/2:.001:(1/2-.001);] %temporary variable used to calculate ern aximum value of realized gain (db)

oint2_db=1.17 %realized gain at theta=60 degrees

^(point2_db/20); s(point2/maximum)*2/(3^.5);

(theta)); t]; %necessary to align coordinate system

0] %steering relative to broadside

degrees = broadside

ng/180*pi)*D') %Steering, phase portion %Beamforming, amplitude

%Use bessel distribution if

1-(2*n_tilda/N)^2) AB(i+1)=besseli(0,expression)

= AS.*AB %Weight vector including steering

%Normalize weights so max is 1/N

rn to form

element beam pattpoint1_db=6.32 %mp maximum=10^(point1_db/20) point2=10alpha=aco

element=maximum*cos(alpha*sin001:1]*0,elemenelement=[[0:.

of element count=0 r steering=[0,30,60,9fo

steering=90-steering %transform steering, 90

unt=count+1 co AS = exp(j*2*pi*cos(steeriAB = ones(N,1)/N rtion po

Beta~=0 if

tapered for i=0:(N-1) n_tilda=i-(N-1)/2; expression=Beta*sqrt(

end end

W W=W/max(abs(AB))/N Au = exp(j*2*pi*D'*u); B = W'*Au B = B.*element %Incorporate element beam patteconsummate beam pattern G = 20*log10((abs(B))); figure(1) if count==1 h=polardb(ang,G,-50,'r'); elseif count==2 h=polardb(ang,G,-50,'g'); elseif count==3 polardb(ang,G,-50,'b'); h=

elseif count==4 ; h=polardb(ang,G,-50,'c')

end ld on ho

end

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legend('','','','','','','','','','','','steering = 0 deg','steering = 30 deg','steering = 60 deg','steering = 90 deg')

2.2 2.4 ghz phased array

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