15a step-down regulator with internal switches

General DescriptionThe MAX15035 pulse-width modulation (PWM) controllerprovides high efficiency, excellent transient response,and high DC-output accuracy. Combined with the inter-nal low on-resistance MOSFETs, the MAX15035 pro-vides a highly efficient and compact solution for smallform factor applications that need a high-power density.

Maxim’s proprietary Quick-PWM™ quick-response,constant on-time PWM control scheme handles wideinput/output voltage ratios (low-duty-cycle applications)with ease and provides 100ns instant-on response toload transients while maintaining a relatively constantswitching frequency. The output voltage can be dynam-ically controlled using the dynamic REFIN, which sup-ports input voltages between 0V to 2V. The REFINadjustability combined with a resistive voltage-divideron the feedback input allows the MAX15035 to be con-figured for any output voltage between 0V to 0.9VIN.

The controller senses the current across the synchro-nous rectifier to achieve a low-cost and highly efficientvalley current-limit protection. External current-limit con-trol is provided to allow higher current-limit settings forapplications with heatsinks and air flow, or for lowercurrent applications that need lower current-limit set-tings to avoid overdesigning the application circuit. Theadjustable current limit provides a high degree of flexi-bility, allowing thermally compensated protection orfoldback current-limit protection using a voltage-dividerpartially derived from the output.

The MAX15035 includes a voltage-controlled soft-startand soft-shutdown to limit the input surge current, pro-vide a monotonic power-up into a precharged output,and provide a predictable soft-start time. The controlleralso includes output fault protection—undervoltage andovervoltage protection—as well as thermal-fault protec-tion.

The MAX15035 is available in a small 40-pin, 6mm x 6mm,TQFN package.

ApplicationsServer Computers

GPU Core Supplies

DDR Memory—VDDQ or VTT

Point-of-Load Applications

Step-Down Power Supplies

Storage Power Supplies

Features 4.5V to 26V Input Voltage Range

Fast Transient Response

Monotonic Power-Up with Precharged Output

Supports Any Output CapacitorNo Compensation Required with Polymers/ Tantalum

Stable with Ceramic Output Capacitors Using External Compensation

Dynamically Adjustable Output Voltage0.5% VOUT Accuracy Over Line and Load

Adjustable Valley Current-Limit ProtectionThermal Compensation with NTCSupports Foldback Current Limit

Programmable Switching Frequency

Overvoltage Protection

Undervoltage Protection

Voltage Soft-Start and Soft-Shutdown

Power-Good Window Comparator

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15A Step-Down Regulator with Internal Switches

________________________________________________________________ Maxim Integrated Products 1

MAX15035

THIN QFN(6mm x 6mm)

TOP VIEW

35

36

34

33

12

11

13

EN V DD LX

PGND

PGND

14

N.C.

N.C.

IN INAGND

TON

BST

IN IN

1 2

REFIN

4 5 6 7

27282930 26 24 23 22

REF

SKIP

IN

N.C.

LX

PGND

AGND

IN

3

25

37VCC PGND

38

39

40

PGOOD

N.C.

N.C.

PGND

PGND

PGND

ILIM

32

15

INFB

31

16

17

18

19

20 IN

PGND

PGND

PGND

IN

8 9 10

21

N.C.

+

EP1 EP3

AGND

IN

LX

EP2

Pin Configuration

19-4123; Rev 2; 10/08

For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,or visit Maxim’s website at www.maxim-ic.com.

EVALUATION KIT

AVAILABLE

Quick-PWM is a trademark of Maxim Integrated Products, Inc.

Ordering Information

+Denotes a lead-free/RoHS-compliant package.

*EP = Exposed pad.

PART TEMP RANGE PIN-PACKAGE

MAX15035ETL+ - 40°C to + 85°C 40 TQFN - E P *

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ABSOLUTE MAXIMUM RATINGS

ELECTRICAL CHARACTERISTICS(Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = 0°C to +85°C, unless otherwise spec-ified. Typical values are at TA = +25°C.) (Note 1)

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functionaloperation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure toabsolute maximum rating conditions for extended periods may affect device reliability.

IN to PGND.............................................................-0.3V to +28VTON to GND ...........................................................-0.3V to +28V VDD to GND..............................................................-0.3V to +6VVCC to GND................................................-0.3V to (VDD + 0.3V)EN, SKIP, PGOOD to GND.......................................-0.3V to +6VREF, REFIN to GND....................................-0.3V to (VCC + 0.3V)ILIM, FB to GND .........................................-0.3V to (VCC + 0.3V)GND to PGND .......................................................-0.3V to +0.3VLX to PGND ...............................................................-1V to +28VBST to PGND...............................................(VDD - 0.3V) to +34VBST to LX..................................................................-0.3V to +6V

BST to VDD .............................................................-0.3V to +28VREF Short Circuit to GND...........................................Continuous IN RMS Current Rating (continuous)......................................15APGND RMS Current Rating (continuous) ...............................20AContinuous Power Dissipation (TA = +70°C)

40-Pin, 6mm x 6mm Thin QFN (T4066-MCM)(derate 27mW/°C above +70°C)................................2162mW

Operating Temperature Range ...........................-40°C to +85°CJunction Temperature Range ..........................................+150°CStorage Temperature Range .............................-65°C to +150°CLead Temperature (soldering, 10s) .................................+300°C

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

PWM CONTROLLER

Input Voltage Range VIN 4.5 26.0 V

Quiescent Supply Current (VDD) IDD + ICC FB forced above REFIN 0.7 1.2 mA

Shutdown Supply Current (VDD) ISHDN EN = GND, TA = +25°C 0.1 2 μA

VCC Undervoltage LockoutThreshold

V U V LO( V C C ) Rising edge, PWM disabled below thislevel; hysteresis = 100mV

3.95 4.2 4.45 V

VDD-to-VCC Resistance RCC 20 ΩRTON = 97.5kΩ (600kHz) 123 164 205

RTON = 200kΩ (300kHz) 275 303 331On-Time tON

VIN = 12V,VFB = 1.0V(Note 3) RT ON = 302.5kΩ ( 200kH z) 379 442 505

ns

Minimum Off-Time tOFF(MIN) (Note 3) 225 350 ns

TON Shutdown Supply CurrentEN = GND, VTON = 26V,VCC = 0V or 5V, TA = +25°C

0.01 1 μA

REFIN Voltage Range VREFIN (Note 2) 0 VREF V

REFIN Input Current IREFIN TA = +25°C, REFIN = 0.5V to 2V -50 +50 nA

FB Voltage Range VFB (Note 2) 0 VREF V

TA = +25°C 0.495 0.5 0.505VREFIN = 0.5V,m easur ed at FB,V I N = 4.5V to 26V ,SKIP = VDD

TA = 0°C to +85°C 0.493 0.507

TA = +25°C 0.995 1.0 1.005VREFIN = 1.0V

TA = 0°C to +85°C 0.993 1.007

FB Voltage Accuracy VFB

VREFIN = 2.0V TA = 0°C to +85°C 1.990 2.0 2.010

V

FB Input Bias Current IFB VFB = 0.5V to 2.0V, TA = +25°C -0.1 +0.1 μA

FB Output Low Voltage ISINK = 3mA 0.4 V

Load-Regulation Error SKIP = VDD 0.1 %

Line-Regulation Error VCC = 4.5V to 5.5V, VIN = 4.5V to 26V 0.2 %

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Soft-Start/Soft-Stop Slew Rate SSSR Rising/falling edge on EN 0.4 1.2 2.2 mV/μs

Dynamic REFIN Slew Rate DYNSR Rising edge on REFIN 3 9.45 18 mV/μs

REFERENCE

No load 1.990 2.00 2.010Reference Voltage VREF

VCC = 4.5Vto 5.5V IREF = -10μA to +50μA 1.98 2.02

V

FAULT DETECTION

With respect to the internal target voltage(error comparator threshold); rising edge;hysteresis = 50mV

250 300 350 mV

Dynamic transitionVREF +

0.30

Output Overvoltage-ProtectionTrip Threshold

OVP

Minimum OVP threshold 0.7

V

Output OvervoltageFault-Propagation Delay

tOVP FB forced 25mV above trip threshold 5 μs

Output Undervoltage-ProtectionTrip Threshold

UVPWith respect to the internal target voltage(error comparator threshold) falling edge;hysteresis = 50mV

-240 -200 -160 mV

Output UndervoltageFault-Propagation Delay

tUVP FB forced 25mV below trip threshold 100 200 350 μs

UVP falling edge, 25mV overdrive 5

OVP rising edge, 25mV overdrive 5PGOOD Propagation Delay tPGOOD

Startup delay 100 200 350

μs

PGOOD Output-Low Voltage ISINK = 3mA 0.4 V

PGOOD Leakage Current IPGOODFB = REFIN (PGOOD high impedance),PGOOD forced to 5V, TA = +25°C

1 μA

Dynamic REFIN Transition FaultBlanking Threshold

Fault blanking initiated; REFIN deviationfrom the internal target voltage (errorcomparator threshold); hysteresis = 10mV

±50 mV

Thermal-Shutdown Threshold TSHDN Temperature rising, hysteresis = 15°C 160 °C

CURRENT LIMIT

ILIM Input Range 0.4 VREF V

ILIM Input Bias Current TA = +25°C, ILIM = 0.4V to 2V -0.1 +0.1 μA

VILIM = 0.4V, VGND - VLX 18 20 22Current-Limit Threshold VILIMIT

ILIM = REF (2.0V), VGND - VLX 92 100 108mV

Current-Limit Threshold(Negative)

VINEG VILIM = 0.4V, VGND - VLX -24 mV

Current-Limit Threshold(Zero Crossing)

VZXVILIM = 0.4V,VGND - VLX, SKIP = GND or open

1 mV

Ultrasonic Frequency SKIP = open (3.3V); VFB = VREFIN + 50mV 18 30 kHz

ELECTRICAL CHARACTERISTICS (continued)(Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = 0°C to +85°C, unless otherwise spec-ified. Typical values are at TA = +25°C.) (Note 1)

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4 _______________________________________________________________________________________

ELECTRICAL CHARACTERISTICS (continued)(Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = 0°C to +85°C, unless otherwise spec-ified. Typical values are at TA = +25°C.) (Note 1)


U l tr asoni c C ur r ent- Li m i tThr eshol d

SKIP = open (3.3V); VFB = VREFIN + 50mV,VGND - VLX

-35 mV

Internal BST SwitchOn-Resistance

RBST IBST = 10mA, VDD = 5V 4 7 Ω

INPUTS AND OUTPUTS

EN Logic-Input Threshold VEN EN rising edge, hysteresis = 450mV (typ) 1.20 1.7 2.20 V

EN Logic-Input Current IEN EN forced to GND or VDD, TA = +25°C -0.5 +0.5 μA

High (5V VDD)VCC -0.4

Open (3.3V) 3.0 3.6

Ref (2.0V) 1.7 2.3

SKIP Quad-Level Input LogicLevels

VSKIP

Low (GND) 0.4

V

SKIP Logic-Input Current ISKIP SKIP forced to GND or VDD, TA = +25°C -2 +2 μA

ELECTRICAL CHARACTERISTICS(Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = -40°C to +85°C, unless otherwisespecified.) (Note 1)

PARAMETER SYMBOL CONDITIONS MIN MAX UNITS

PWM CONTROLLER

Input Voltage Range VIN 4.5 26 V

Quiescent Supply Current (VDD) IDD + ICC FB forced above REFIN 1.2 mA

RTON = 97.5kΩ (600kHz) 115 213

RTON = 200kΩ (300kHz) 270 336On-Time tON

VIN = 12V,VFB = 1.0V(Note 3) RT ON = 302.5kΩ ( 200kH z) 368 516

ns

Minimum Off-Time tOFF(MIN) (Note 3) 400 ns

REFIN Voltage Range VREFIN (Note 2) 0 VREF V

FB Voltage Range VFB (Note 2) 0 VREF V

VREFIN = 0.5V 0.49 0.51

VREFIN = 1.0V 0.99 1.01FB Voltage Accuracy VFB

M easur ed at FB,V I N = 4.5V to26V , SKIP =VDD VREFIN = 2.0V 1.985 2.015

V

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ELECTRICAL CHARACTERISTICS (continued)(Circuit of Figure 1, VIN = 12V, VDD = VCC = VEN = 5V, REFIN = ILIM = REF, SKIP = GND. TA = -40°C to +85°C, unless otherwisespecified.) (Note 1)

PARAMETER SYMBOL CONDITIONS MIN MAX UNITS

REFERENCE

Reference Voltage VREF VDD = 4.5V to 5.5V 1.985 2.015 V

FAULT DETECTION

Output Overvoltage-ProtectionTrip Threshold

OVPWith respect to the internal target voltage(error comparator threshold) rising edge;hysteresis = 50mV

250 350 mV

Output Undervoltage-ProtectionTrip Threshold

UVPWith respect to the internal target voltage(error comparator threshold);falling edge; hysteresis = 50mV

-240 -160 mV

Output UndervoltageFault-Propagation Delay

tUVP FB forced 25mV below trip threshold 80 400 μs

PGOOD Output-Low Voltage ISINK = 3mA 0.4 V

VCC Undervoltage LockoutThreshold

V U V L O( V C C ) Ri si ng ed g e, P W M d i sab l ed b el ow thi s l evel ,hyster esi s = 100m V

3.95 4.45 V

CURRENT LIMIT

ILIM Input Range 0.4 VREF V

VILIM = 0.4V, VGND = VLX 17 23Current-Limit Threshold VILIMIT

ILIM = REF (2.0V), VGND - VLX 90 110mV

Ultrasonic Frequency SKIP = open (3.3V), VFB = VREFIN + 50mV 17 kHz

INPUTS AND OUTPUTS

EN Logic-Input Threshold VEN EN rising edge hysteresis = 450mV (typ) 1.20 2.20 V

High (5V VDD)VCC -0.4

Mid (3.3V) 3.0 3.6

Ref (2.0V) 1.7 2.3

SKIP Quad-Level InputLogic Levels

V SKIP

Low (GND) 0.4

V

Note 1: Limits are 100% production tested at TA = +25°C. Maximum and minimum limits over temperature are guaranteed bydesign and characterization.

Note 2: The 0 to 0.5V range is guaranteed by design, not production tested.Note 3: On-time and off-time specifications are measured from 50% point to 50% point at the unloaded LX node. The typical 25ns

dead time that occurs between the high-side driver falling edge (high-side MOSFET turn-off) and the low-side MOSFET turn-on) is included in the on-time measurement. Similarly, the typical 25ns dead time that occurs between the low-side driverfalling edge (low-side MOSFET turn-off) and the high-side driver rising edge (high-side MOSFET turn-on) is included in theoff-time measurement.

Typical Operating Characteristics(MAX15035 Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, RTON = 200kΩ, TA = +25°C, unless otherwise noted.)

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1.5V OUTPUT EFFICIENCYvs. LOAD CURRENT

MAX

1503

5 to

c01

LOAD CURRENT (A)

EFFI

CIEN

CY (%

)

1010.1

30

40

50

60

70

80

90

100

200.01 100

SKIP MODEPWM MODE

7V20V

12V


MAX

1503

5 to

c02

LOAD CURRENT (A)

EFFI

CIEN

CY (%

)

1010.1

30

40

50

60

70

80

90

100

200.01 100

SKIP MODE

PWM MODE

ULTRASONIC MODE

1.5V OUTPUT VOLTAGEvs. LOAD CURRENT

MAX

1503

5 to

c03

LOAD CURRENT (A)

OUTP

UT V

OLTA

GE (V

)

1413121110987654321

1.495

1.505

1.515

1.4850 15

SKIP MODE

PWM MODE

ULTRASONIC MODE


MAX

1503

5 to

c04

LOAD CURRENT (A)

EFFI

CIEN

CY (%

)

1010.1

30

40

50

60

70

80

90

100

200.01 100

SKIP MODEPWM MODE

7V

20V20V

12V

12V


MAX

1503

5 to

c05

LOAD CURRENT (A)

EFFI

CIEN

CY (%

)

1010.1

30

40

50

60

70

80

90

100

200.01 100

SKIP MODE

PWM MODE

ULTRASONIC MODE


MAX

1503

5 to

c06

LOAD CURRENT (A)

OUTP

UT V

OLTA

GE (V

)

1413121110987654321

1.05

1.06

1.040 15

ULTRASONIC MODE

SKIP MODE

PWM MODE


MAX

1503

5 to

c07

LOAD CURRENT (A)

EFFI

CIEN

CY (%

)

1010.1

30

40

50

60

70

80

90

100

200.01 100

20V

12V

7V

SKIP MODEPWM MODE


MAX

1503

5 to

c08

LOAD CURRENT (A)

EFFI

CIEN

CY (%

)

1010.1

30

40

50

60

70

80

90

100

200.01 100

SKIP MODE

PWM MODE

ULTRASONIC MODE


MAX

1503

5 to

c09

LOAD CURRENT (A)

OUTP

UT V

OLTA

GE (V

)

1413121110987654321

3.380

3.365

3.350

3.320

3.305

3.290

3.275

3.260

3.245

3.335

3.2300 15

ULTRASONIC MODE

PWM MODE

SKIP MODE

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SWITCHING FREQUENCYvs. LOAD CURRENT

MAX

1503

5 to

c10

LOAD CURRENT (A)

SWIT

CHIN

G FR

EQUE

NCY

(kHz

)

10.1

50

100

150

200

250

300

350

400

00.01 10

SKIP MODEULTRASONICMODE

PWM MODE

PWM MODE SWITCHING FREQUENCYvs. INPUT VOLTAGE

MAX

1503

5 to

c11

INPUT VOLTAGE (V)

SWIT

CHIN

G FR

EQUE

NCY

(kHz

)

14 1810

350360370

320330340

290300310

260250

270280

380

2406 22 2412 168 20

ILOAD = 5A

NO LOAD

Typical Operating Characteristics (continued)(MAX15035 Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, RTON = 200kΩ, TA = +25°C, unless otherwise noted.)

MAXIMUM OUTPUT CURRENTvs. AMBIENT TEMPERATURE

MAX

1503

5 to

c14

TEMPERATURE (°C)

MAX

IMUM

OUT

PUT

CURR

ENT

(A)

806040200-20

7

9

11

13

15

5-40 100

0 LFM

100 LFM

300 LFM

FOUR-LAYER PCB WITH 2oz COPPER USED


_______________________________________________________________________________________ 7

SWITCHING FREQUENCYvs. TEMPERATURE

MAX

1503

5 to

c12

TEMPERATURE (°C)

SWIT

CHIN

G FR

EQUE

NCY

(kHz

)

20 40 60 80-20 0

360

380

370

390

350-40 100

ILOAD = 10A

ILOAD = 5A

MAXIMUM OUTPUT CURRENTvs. INPUT VOLTAGE

MAX

1503

5 to

c13

INPUT VOLTAGE (V)

MAX

IMUM

OUT

PUT

CURR

ENT

(A)

15 18 219 12

15.00

14.80

15.60

15.40

15.20

14.40

14.20

14.60

16.00

15.80

14.006 24

NO-LOAD SUPPLY CURRENT (IBIAS)vs. INPUT VOLTAGE

MAX

1503

5 to

c15

INPUT VOLTAGE (V)

I BIA

S (m

A)

10 12 14 16 18 20 228

4

2

8

6

10

12

06 24

SKIP MODE

PWM MODE

ULTRASONIC MODE

REF OUTPUT VOLTAGEvs. LOAD CURRENT

MAX

1503

5 to

c17

LOAD CURRENT (µA)

REF

OUTP

UT V

OLTA

GE (V

)

10 20 30 400

2.001

2.002

2.003

2.005

2.004

2.000-10 50

NO-LOAD SUPPLY CURRENT (IIN)vs. INPUT VOLTAGE

MAX

1503

5 to

c16

INPUT VOLTAGE (V)

I IN (m

A)

10 12 14 16 18 20 228

0.1

1

10

100

0.016 24

SKIP MODE

PWM MODE

ULTRASONIC MODE

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SOFT-START WAVEFORM(HEAVY LOAD)

MAX15035 toc18

200µs/divC. VOUT, 1V/divB. INDUCTOR CURRENT, 10A/div

1.5V

8A

05V

0

0

0

5V A

B

C

D

A. EN, 5V/divB. PGOOD, 5V/div

IOUT = 8A

SOFT-START WAVEFORM(LIGHT LOAD)

MAX15035 toc19


1.5V

05V

0

0

01A

5V A

B

C

D


IOUT = 1A

SHUTDOWN WAVEFORMMAX15035 toc20


A

B

C

D


IOUT = 6A

1.5V

0

5V

0

0

0

8A

5V

LOAD-TRANSIENT RESPONSE(PWM MODE)

MAX15035 toc21

20µs/divB. VOUT, 20mV/divC. INDUCTOR CURRENT, 5A/div

8A

1A

1.5V

8A

0A

A

B

C

A. IOUT, 10A/div

IOUT = 1A TO 8A TO 1A

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(MAX15035 Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, RTON = 200kΩ, TA = +25°C, unless otherwise noted.)

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LOAD-TRANSIENT RESPONSE(SKIP MODE)

MAX15035 toc22

20µs/divB. VOUT, 20mV/divC. INDUCTOR CURRENT, 5A/div

8A1A

1.5V

8A

0A

A

B

C

A. IOUT, 10A/div

IOUT = 1A TO 8A TO 1A

OUTPUT OVERCURRENT WAVEFORMMAX15035 toc23

200µs/divB. VOUT, 1V/divC. PGOOD, 5V/div

5V

20A

0

1.5V

0

0

A

B

C

A. INDUCTOR CURRENT, 10A/div

IOUT = 2A TO 20A

OUTPUT OVERVOLTAGE WAVEFORMMAX15035 toc24

200µs/divB. PGOOD, 5V/div

0

1.5V

5V

0

A

B

A. VOUT, 1V/div

IOUT = 2A TO 20A

NO-LOAD BIAS CURRENTvs. FREQUENCY

MAX

1503

5 to

c25

FREQUENCY (kHz)

I BIA

S (m

A)

550500400 450300 350250

10

12

14

16

18

20

22

24

26

28

30

8200 600

PWM MODE

OUTPUT CURRENT LIMITvs. ILIMIT VOLTAGE

MAX

1503

5 to

c26

ILIMIT VOLTAGE (mV)

CURR

ENT

LIM

IT (A

)

900800700600

10

12

14

16

18

20

8500 1000

PREBIAS STARTUP-OUTPUT VOLTAGEMAX15035 toc27

200µs/div

500mV/div

1.5V

1.2V


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Pin DescriptionPIN NAME FUNCTION

1, 17, 27, 31,39, 40

N.C. No Connection. Not internally connected.

2 EN

Shutdown Control Input. Connect to VDD for normal operation. Pull EN low to put the controller into its2μA (max) shutdown state. The MAX15035 slowly ramps down the target/output voltage to groundand after the target voltage reaches 0.1V, the controller forces LX into a high-impedance state andenters the low-power shutdown state. Toggle EN to clear the fault-protection latch.

3, 28 AGND Analog Ground. Internally connected to EP1.

4 VDDSupply Voltage Input for the DL Gate Driver. Connect to the system supply voltage (+4.5V to +5.5V).Bypass VDD to power ground with a 1μF or greater ceramic capacitor.

5, 16 LXInductor Connection. Internally connected to EP2. Connect LX to the switched side of the inductor asshown in Figure 1.

6–15 PGND Power Ground

18–26 IN Power MOSFET Input Power Source. Internally connected to EP3.

29 TON

Switching Frequency-Setting Input. An external resistor between the input power source and TONsets the switching period (tSW = 1/fSW) according to the following equation:

where CTON = 16.26pF and VFB = VREFIN under normal operating conditions. If the TON currentdrops below 10μA, the MAX15035 shuts down and enters a high-impedance state. TON is highimpedance in shutdown.

30 BSTBoost Flying Capacitor Connection. Connect to an external 0.1μF capacitor as shown in Figure 1. TheMAX15035 contains an internal boost switch/diode (Figure 2).

t C R kV

VSW TON TONFB

OUT= +( ) ⎛

⎝⎜⎞⎠⎟

6 5. Ω

DYNAMIC OUTPUT-VOLTAGE TRANSITION(PWM MODE)

MAX15035 toc28

40µs/divC. INDUCTOR CURRENT, 10A/divD. LX, 10V/div

-6A0

1.5V

1.5V1.05V

1.05V

12V

0

A

B

C

D

A. REFIN, 500mV/divB. VOUT, 200mV/div

IOUT = 2A

DYNAMIC OUTPUT-VOLTAGE TRANSITION(SKIP MODE)

MAX15035 toc29

40µs/divC. INDUCTOR CURRENT, 10A/divD. LX, 10V/div

10A

0

1.5V

1.5V

1.05V

1.05V

12V

0

A

B

C

D

A. REFIN, 500mV/divB. VOUT, 200mV/div

IOUT = 2A


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______________________________________________________________________________________ 11

Pin Description (continued)

PIN NAME FUNCTION

32 FB

Feed b ack V ol tag e S ense C onnecti on. C onnect d i r ectl y to the p osi ti ve ter m i nal of the outp ut cap aci tor sfor outp ut vol tag es l ess than 2V as show n i n Fi g ur e 1. For fixed-output voltages greater than 2V,connect REFIN to REF and use a resistive divider to set the output voltage (Figure 6). FB senses theoutput voltage to determine the on-time for the high-side switching MOSFET.

33 ILIMCurrent-Limit Threshold Adjustment. The current-limit threshold is 0.05 times (1/20) the voltage atILIM. Connect ILIM to a resistive divider (from REF) to set the current-limit threshold between 20mVand 100mV (with 0.4V to 2V at ILIM).

34 REFIN

External Reference Input. REFIN sets the feedback regulation voltage (VFB = VREFIN) of theMAX15035 using a resistor-divider connected between REF and AGND. The MAX15035 includes aninternal window comparator to detect REFIN voltage transitions, allowing the controller to blankPGOOD and the fault protection.

35 REF2V Reference Voltage. Bypass to analog ground using a 1nF ceramic capacitor. The reference cansource up to 50μA for external loads.

36 SKIP

Pulse-Skipping Control Input. This four-level input determines the mode of operation under normalsteady-state conditions and dynamic output-voltage transitions:VDD (5V) = Forced-PWM operationREF (2V) = Pulse-skipping mode (with forced-PWM during transitions)Open (3.3V) = Ultrasonic mode (without forced-PWM during transitions)GND = Pulse-skipping mode (without forced-PWM during transitions)

37 VCC5V Analog Supply Voltage. Internally connected to VDD through an internal 20Ω resistor. Bypass VCCto analog ground using a 1μF ceramic capacitor.

38 PGOOD

Open-Drain Power-Good Output. PGOOD is low when the output voltage is more than 200mV (typ)below or 300mV (typ) above the target voltage (VREFIN), during soft-start, and soft-shutdown. Afterthe soft-start circuit has terminated, PGOOD becomes high impedance if the output is in regulation.PGOOD is blanked—forced high-impedance state—when a dynamic REFIN transition is detected.

EP1(41)

AGNDExposed Pad 1/Analog Ground. Internally connected to the controller’s ground plane and substrate.Connect directly to ground.

EP2(42)

LXExposed Pad 2/Inductor Connection. Internally connected to drain of the low-side MOSFET andsource of the high-side MOSFET (Figure 2). Connect LX to the switched side of the inductor asshown in Figure 1.

EP3(43)

INExposed Pad 3/Power MOSFET Input Power Source. Internally connected to drain of the high-sideMOSFET (Figure 2).

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12 ______________________________________________________________________________________

C11μF

ENOFFON

VDD

VCC

C21μF

PGOOD

R10100kΩ

REFINR2

54.9kΩ

R440.2kΩ

RT60.4kΩ

NTC10kΩ

B = 3435

R549.4kΩ

BST

LX

CBST0.1μF L1

COUT

TON

R397.6kΩ

GND/OPEN/REF/VDD

PGND

FB

ILIM

HILO

REF

RTON200kΩ

AGND

AGND

PWR

PWRPWR

AGND

AGND

AGND

2

34

36

4

37

18–26, EP3

30

6–15

AGND

38

SKIP

35

32

33

OUTPUT1.05V/1.50V15A (MAX)

INPUT7V TO 24V

5V BIASSUPPLY

MAX15035

CIN

PWR

C31000pF

R149.9kΩ

5, 16, EP2

29

IN

3, 28, EP1

PWR

Figure 1. MAX15035 Standard Application Circuit

SEE TABLE 1 FOR COMPONENT SELECTION.

Table 1. Component Selection for Standard ApplicationsVOUT = 1.5V/1.05V AT 15A

(FIGURE 1)VOUT = 3.3V AT 6A

(FIGURE 6)VOUT = 1.5V/1.05V AT 10A

(FIGURE 1)COMPONENT

VIN = 7V to 20VTON = 200kΩ (300kHz)



Input Capacitor(3x) 10μF, 25V

Taiyo Yuden TMK432BJ106KM(2x) 10μF, 25V

Taiyo Yuden TMK432BJ106KM(3x) 10μF, 25V

Taiyo Yuden TMK432BJ106KM

Output Capacitor(2x) 330μF, 6mΩ, 2V

Panasonic EEFSX0D331XR(1x) 330μF, 18mΩ, 4VSANYO 4TPE330MI

(1x) 470μF, 7mΩ, 2.5VSANYO 2R5TPLF470M7

Inductor1.0μH, 5.3mΩ, 27.5A

Vishay IHLP4040DZER1R01.5μH, 14mΩ, 9A

NEC TOKIN MPLC1040L3R30.47μH, 3.7mΩ, 15ACooper FP3-R47-R

Table 2. Component SuppliersSUPPLIER WEBSITE

AVX Corp. www.avxcorp.com

BI Technologies www.bitechnologies.com

Cooper Bussmann www.cooperet.com

KEMET Corp. www.kemet.com

Murata Mfg. Co., Ltd. www.murata.com

NEC TOKIN Corp. www.nec-tokin.com

Panasonic Corp. www.panasonic.com

SUPPLIER WEBSITE

Pulse Engineering www.pulseeng.com

SANYO NA Corp. www.sanyo.com

Sumida Corp. www.sumida.com

Taiyo Yuden www.t-yuden.com

TDK Corp. www.component.tdk.com

TOKO America, Inc. www.tokoam.com

V i shay www.vishay.com

Wür th E l ectr oni k Gm b H & C o. KG www.we-online.com

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______________________________________________________________________________________ 13

Standard Application CircuitThe MAX15035 standard application circuit (Figure 1)generates a 1.5V or 1.05V output rail for general-purposeuse. See Table 1 for component selections. Table 2 liststhe component suppliers.

Detailed DescriptionThe MAX15035 step-down controller is ideal for low-duty-cycle (high-input voltage to low-output voltage)applications. Maxim’s proprietary Quick-PWM pulse-width modulator in the MAX15035 is specificallydesigned for handling fast-load steps while maintaininga relatively constant operating frequency and inductoroperating point over a wide range of input voltages.The Quick-PWM architecture circumvents the poorload-transient timing problems of fixed-frequency, cur-rent-mode PWMs while also avoiding the problemscaused by widely varying switching frequencies in con-ventional constant-on-time (regardless of input voltage)pulse-frequency modulation (PFM) control schemes.

+5V Bias Supply (VCC/VDD)The MAX15035 requires an external 5V bias supply inaddition to the input. See Figure 6 for an optional 5Vbias generation circuit.

The 5V bias supply powers both the PWM controllerand internal gate-drive power, so the maximum currentdrawn is determined by:

IBIAS = IQ + fSWQG = 2mA to 20mA (typ)

The MAX15035 includes a 20Ω resistor between VDDand VCC, simplifying the PCB layout requirement.

Free-Running Constant-On-Time PWMController with Input Feed-Forward

The Quick-PWM control architecture is a pseudo-fixed-frequency, constant on-time, current-mode regulatorwith voltage feed-forward (Figure 2). This architecturerelies on the output filter capacitor’s ESR to act as a current-sense resistor, so the output ripple voltage pro-vides the PWM ramp signal. The control algorithm issimple: the high-side switch on-time is determined sole-ly by a one-shot whose pulse width is inversely propor-tional to input voltage and directly proportional tooutput voltage. Another one-shot sets a minimum off-time (200ns typ). The on-time one-shot is triggered ifthe error comparator is low, the low-side switch currentis below the valley current-limit threshold, and the mini-mum off-time one-shot has timed out.

On-Time One-ShotThe heart of the PWM core is the one-shot that sets thehigh-side switch on-time. This fast, low-jitter, adjustableone-shot includes circuitry that varies the on-time inresponse to input and output voltage. The high-sideswitch on-time is inversely proportional to the input volt-age as sensed by the TON input, and proportional tothe feedback voltage as sensed by the FB input:

On-Time (tON) = tSW (VFB/VIN)

where tSW (switching period) is set by the resistance(RTON) between TON and IN. This algorithm results in anearly constant switching frequency despite the lack ofa fixed-frequency clock generator. Connect a resistor(RTON) between TON and IN to set the switching periodtSW = 1/fSW:

where CTON = 16.26pF. When used with unity-gain feed-back (VOUT = VFB), a 96.75kΩ to 303.25kΩ correspondsto switching periods of 167ns (600kHz) to 500ns(200kHz), respectively. High-frequency (600kHz) opera-tion optimizes the application for the smallest compo-nent size, trading off efficiency due to higher switchinglosses. This may be acceptable in ultra-portable deviceswhere the load currents are lower and the controller ispowered from a lower voltage supply. Low-frequency(200kHz) operation offers the best overall efficiency atthe expense of component size and board space.

For continuous conduction operation, the actual switchingfrequency can be estimated by:

where VDIS is the sum of the parasitic voltage drops inthe inductor discharge path, including synchronous recti-fier, inductor, and PCB resistances; VCHG is the sum ofthe resistances in the charging path, including the high-side switch, inductor, and PCB resistances; and tON isthe on-time calculated by the MAX15035.

Power-Up Sequence (POR, UVLO)The MAX15035 is enabled when EN is driven high andthe 5V bias supply (VDD) is present. The referencepowers up first. Once the reference exceeds its UVLOthreshold, the internal analog blocks are turned on andmasked by a 50μs one-shot delay in order to allow thebias circuitry and analog blocks enough time to settleto their proper states. With the control circuitry reliablypowered up, the PWM controller may begin switching.

fV V

t V VSWFB DIS

ON IN CHG=

+−( )

t C R kV

VSW TON TONFB

OUT= +( ) ⎛

⎝⎜⎞⎠⎟

6 5. Ω

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14 ______________________________________________________________________________________

DYNAMIC OUTPUTTRANSITION DETECTION

BST

LX

ONE-SHOTTRIGQ

tOFF(MIN)ON-TIME

COMPUTEFB

TON

ERRORAMPLIFIER

EA + 0.3V

ONE-SHOT

TRIG QtON

ZERO CROSSING

FB

EAPGOOD

EN

PGOODAND FAULT

PROTECTION

SKIP

EA - 0.2V

BLANK

REFIN

VALLEY CURRENT LIMIT

PGND

S

RQ

S

R

Q

IN

SOFT-START/SOFT-STOP

QUAD-LEVEL

DECODEBLANK

VCC

VDD

2VREF

FAUL

T

INTEGRATOR(CCV)

REF

ILIM

MAX15035

IN

Figure 2. MAX15035 Block Diagram

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______________________________________________________________________________________ 15

Power-on reset (POR) occurs when VCC rises aboveapproximately 3V, resetting the fault latch and prepar-ing the controller for operation. The VCC UVLO circuitryinhibits switching until VCC rises above 4.25V. The con-troller powers up the reference once the systemenables the controller, VCC exceeds 4.25V, and EN isdriven high. With the reference in regulation, the con-troller ramps the output voltage to the target REFIN volt-age with a 1.2mV/μs slew rate:

The soft-start circuitry does not use a variable currentlimit, so full output current is available immediately.PGOOD becomes high impedance approximately200μs after the target REFIN voltage has been reached.The MAX15035 automatically uses pulse-skipping modeduring soft-start and uses forced-PWM mode duringsoft-shutdown, regardless of the SKIP configuration.

For automatic startup, the input voltage should be pre-sent before VCC. If the controller attempts to bring theoutput into regulation without the input voltage present,the fault latch trips. The controller remains shut downuntil the fault latch is cleared by toggling EN or cyclingthe VCC power supply below 0.5V.

If the VCC voltage drops below 4.25V, the controllerassumes that there is not enough supply voltage tomake valid decisions. To protect the output from over-voltage faults, the controller shuts down immediatelyand forces a high impedance on LX.

ShutdownWhen the system pulls EN low, the MAX15035 enterslow-power shutdown mode. PGOOD is pulled lowimmediately, and the output voltage ramps down with a1.2mV/μs slew rate:

Slowly discharging the output capacitors by slewingthe output over a long period of time (typically 0.5ms to2ms) keeps the average negative inductor current low(damped response), thereby preventing the negativeoutput-voltage excursion that occurs when the con-troller discharges the output quickly by permanentlyturning on the low-side MOSFET (underdampedresponse). This eliminates the need for the Schottkydiode normally connected between the output andground to clamp the negative output-voltage excursion.After the controller reaches the zero target, theMAX15035 shuts down completely—the drivers are dis-abled (high impedance on LX)—the reference turns off,and the supply currents drop to about 0.1μA (typ).

When a fault condition—output UVP or thermal shut-down—activates the shutdown sequence, the protectioncircuitry sets the fault latch to prevent the controller fromrestarting. To clear the fault latch and reactivate thecontroller, toggle EN or cycle VCC power below 0.5V.

The MAX15035 automatically uses pulse-skipping modeduring soft-start and uses forced-PWM mode duringsoft-shutdown, regardless of the SKIP configuration.

Modes of OperationUltrasonic Mode (SSKKIIPP = Open = 3.3V)

Leaving SKIP unconnected activates a unique pulse-skipping mode with a minimum switching frequency of18kHz. This ultrasonic pulse-skipping mode eliminatesaudio-frequency modulation that would otherwise bepresent when a lightly loaded controller automaticallyskips pulses. In ultrasonic mode, the controller automati-cally transitions to fixed-frequency PWM operation whenthe load reaches the same critical conduction point(ILOAD(SKIP)) that occurs when normally pulse skipping.

An ultrasonic pulse occurs when the controller detectsthat no switching has occurred within the last 33μs.Once triggered, the ultrasonic controller turns on thelow-side MOSFET to induce a negative inductor current(Figure 3). After the inductor current reaches the nega-tive ultrasonic current threshold, the controller turns offthe low-side MOSFET and triggers a constant on-time.

tVmV s

VV msSHDN

FB FB= =1 2 1 2. .μ

tVmV s

VV msSTART

FB FB= =1 2 1 2. .μ

ON-TIME (tON)ISONIC

ZERO-CROSSINGDETECTION

0

33μs (typ)

INDUCTORCURRENT

Figure 3. Ultrasonic Waveform

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16 ______________________________________________________________________________________

When the on-time expires, the controller re-enables thelow-side MOSFET until the controller detects that theinductor current drops below the zero-crossing thresh-old. Starting with a negative inductor current pulsegreatly reduces the peak output voltage when com-pared to starting with a positive inductor current pulse.

The output voltage at the beginning of the ultrasonicpulse determines the negative ultrasonic currentthreshold, resulting in the following equation:

where VFB > VREFIN.

Forced-PWM Mode (SSKKIIPP = VDD)The low-noise, forced-PWM mode (SKIP = VDD) dis-ables the zero-crossing comparator, which controls thelow-side switch on-time. This forces the low-side gate-drive waveform to constantly be the complement of thehigh-side gate-drive waveform, so the inductor currentreverses at light loads while LX maintains a duty factorof VOUT/VIN. The benefit of forced-PWM mode is tokeep the switching frequency fairly constant. However,forced-PWM operation comes at a cost: the no-load 5Vbias current remains between 10mA to 50mA, depend-ing on the switching frequency.

The MAX15035 automatically always uses forced-PWMoperation during shutdown, regardless of the SKIPconfiguration.

Automatic Pulse-Skipping Mode (SSKKIIPP = GND or REF)In skip mode (SKIP = GND or 3.3V), an inherent auto-matic switchover to PFM takes place at light loads. Thisswitchover is affected by a comparator that truncatesthe low-side switch on-time at the inductor current’szero crossing. The zero-crossing comparator thresholdis set by the differential across LX to PGND.

DC output-accuracy specifications refer to the thresholdof the error comparator. When the inductor is in continu-ous conduction, the MAX15035 regulates the valley ofthe output ripple, so the actual DC output voltage ishigher than the trip level by 50% of the output ripplevoltage. In discontinuous conduction (SKIP = GND andIOUT < ILOAD(SKIP)), the output voltage has a DC regu-lation level higher than the error-comparator thresholdby approximately 1.5% due to slope compensation.

When SKIP is pulled to GND, the MAX15035 remains inpulse-skipping mode. Since the output is not able tosink current, the timing for negative dynamic output-volt-age transitions depends on the load current and outputcapacitance. Letting the output voltage drift down is typically recommended to reduce the potential for audi-ble noise since this eliminates the input current surgeduring negative output-voltage transitions. See Figures4 and 5.

V I V VISONIC L REFIN FB= × = −( ) ×0 006 0 7. .

REFIN

PGOOD

INTERNALPWM CONTROL

BLANK HIGH-Z

SKIP

LX

SET TO REF + 300mVOVP

OUTPUTVOLTAGE INTERNAL TARGET

ACTUAL VOUT

BLANK HIGH-Z

NO PULSES: VOUT > VTARGET

DYNAMIC REFIN WINDOW

DYNAMIC TRANSITION WHEN SKIP# = GND

EA TARGET + 300mV

Figure 4. Dynamic Transition when SKIP = GND

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______________________________________________________________________________________ 17

Valley Current-Limit ProtectionThe current-limit circuit employs a unique “valley” current-sensing algorithm that senses the inductor current through the low-side MOSFET. If the currentthrough the low-side MOSFET exceeds the valley current-limit threshold, the PWM controller is notallowed to initiate a new cycle. The actual peak currentis greater than the valley current-limit threshold by anamount equal to the inductor ripple current. Therefore,the exact current-limit characteristic and maximum loadcapability are a function of the inductor value and inputvoltage. When combined with the undervoltage protec-tion circuit, this current-limit method is effective inalmost every circumstance.

In forced-PWM mode, the MAX15035 also implementsa negative current limit to prevent excessive reverseinductor currents when VOUT is sinking current. Thenegative current-limit threshold is set to approximately120% of the positive current limit.

Integrated Output VoltageThe MAX15035 regulates the valley of the output ripple,so the actual DC output voltage is higher than the slope-compensated target by 50% of the output ripple voltage.

Under steady-state conditions, the MAX15035’s internalintegrator corrects for this 50% output ripple-voltageerror, resulting in an output voltage that is dependentonly on the offset voltage of the integrator amplifier pro-vided in the Electrical Characteristics table.

Dynamic Output VoltagesThe MAX15035 regulates FB to the voltage set at REFIN.By changing the voltage at REFIN (Figure 1), theMAX15035 can be used in applications that requiredynamic output-voltage changes between two setpoints. For a step-voltage change at REFIN, the rate ofchange of the output voltage is limited either by theinternal 9.45mV/μs slew-rate circuit or by the componentselection—inductor current ramp, the total outputcapacitance, the current limit, and the load during thetransition—whichever is slower. The total output capaci-tance determines how much current is needed tochange the output voltage, while the inductor limits thecurrent ramp rate. Additional load current may slowdown the output voltage change during a positive REFINvoltage change, and may speed up the output voltagechange during a negative REFIN voltage change.

REFIN

PGOOD

INTERNALPWM CONTROL

BLANK HIGH-Z

SKIP

LX

SET TO REF +300mVOVP

OUTPUTVOLTAGE INTERNAL EA TARGET = ACTUAL VOUT

BLANK HIGH-Z

DYNAMIC REFIN WINDOW

DYNAMIC TRANSITION WHEN SKIP = REF

SKIPPWMPWM

EA TARGET + 300mV EA TARGET + 300mV

Figure 5. Dynamic Transition when SKIP = REF

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18 ______________________________________________________________________________________

Output Voltages Greater than 2VAlthough REFIN is limited to a 0 to 2V range, the out-put-voltage range is unlimited since the MAX15035 uti-lizes a high-impedance feedback input (FB). By addinga resistive voltage-divider from the output to FB to ana-log ground (Figure 6), the MAX15035 supports outputvoltages above 2V. However, the controller also usesFB to determine the on-time, so the voltage-dividerinfluences the actual switching frequency, as detailedin the On-Time One-Shot section.

Internal IntegrationAn integrator amplifier forces the DC average of the FBvoltage to equal the target voltage. This internal amplifi-er integrates the feedback voltage and provides a fineadjustment to the regulation voltage (Figure 2), allowingaccurate DC output-voltage regulation regardless of thecompensated feedback ripple voltage and internalslope-compensation variation. The integrator amplifierhas the ability to shift the output voltage by ±55mV (typ).

The MAX15035 disables the integrator by connecting theamplifier inputs together at the beginning of all downwardREFIN transitions done in pulse-skipping mode. The inte-grator remains disabled until 20μs after the transition iscompleted (the internal target settles) and the output is inregulation (edge detected on the error comparator).

Power-Good Outputs (PGOOD)and Fault Protection

PGOOD is the open-drain output that continuouslymonitors the output voltage for undervoltage and over-voltage conditions. PGOOD is actively held low in shut-down (EN = GND), and during soft-start andsoft-shutdown. Approximately 200μs (typ) after the soft-start terminates, PGOOD becomes high impedance aslong as the feedback voltage is above the UVP thresh-old (REFIN - 200mV) and below the OVP threshold(REFIN + 300mV). PGOOD goes low if the feedbackvoltage drops 200mV below the target voltage (REFIN)or rises 300mV above the target voltage (REFIN), or theSMPS controller is shut down. For a logic-level PGOOD

C11μF

ENOFFON

VDD

VCC

C21μF

PGOOD

R10100kΩ

REFIN

IN

LX

CBST0.1μF L1

COUT

TON

GND/OPEN/REF/VDD

BST

PGND

FB

REF

ILIM REF

RTON332kΩ

R549.4kΩ

R449.9kΩ

AGND

AGND

PWR

PWR PWR

AGND

AGNDPWR

2

34

36

4

37

29

18–26, EP3

30

5, 16, EP2

6–15

AGND

3, 28, EP1

38

SKIP

35

32

33

OUTPUT3.3V

INPUT7V TO 24V

5V BIASSUPPLY

MAX15035

CIN

PWR

R720kΩ

R613.0kΩ

AGND

C31000pF

5VBIAS7V TO 15V

INPUT

OPTIONAL5.6V

1kΩ

Figure 6. High Output-Voltage Application Using a Feedback Divider


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______________________________________________________________________________________ 19

output voltage, connect an external pullup resistorbetween PGOOD and VDD. A 100kΩ pullup resistorworks well in most applications. Figure 7 shows thepower-good and fault-protection circuitry.

Overvoltage Protection (OVP)When the internal feedback voltage rises 300mV abovethe target voltage and OVP is enabled, the OVP compara-tor immediately forces LX low, pulls PGOOD low, sets thefault latch, and disables the SMPS controller. Toggle ENor cycle VCC power below the VCC POR to clear the faultlatch and restart the controller.

Undervoltage Protection (UVP)When the feedback voltage drops 200mV below thetarget voltage (REFIN), the controller immediately pullsPGOOD low and triggers a 200μs one-shot timer. If thefeedback voltage remains below the undervoltage faultthreshold for the entire 200μs, the undervoltage faultlatch is set and the SMPS begins the shutdownsequence. When the internal target voltage dropsbelow 0.1V, the MAX15035 forces a high impedance onLX. Toggle EN or cycle VCC power below VCC POR toclear the fault latch and restart the controller.

Thermal-Fault Protection (TSHDN)The MAX15035 features a thermal fault-protection cir-cuit. When the junction temperature rises above+160°C, a thermal sensor activates the fault latch, pullsPGOOD low, shuts down the controller, and forces a

high impedance on LX. Toggle EN or cycle VCC powerbelow VCC POR to reactivate the controller after thejunction temperature cools by 15°C.

Quick-PWM Design ProcedureFirmly establish the input voltage range and maximumload current before choosing a switching frequency andinductor operating point (ripple-current ratio). The prima-ry design trade-off lies in choosing a good switching fre-quency and inductor operating point, and the followingfour factors dictate the rest of the design:

• Input Voltage Range: The maximum value(VIN(MAX)) must accommodate the worst-case inputsupply voltage. The minimum value (VIN(MIN)) mustaccount for the lowest input voltage after drops dueto connectors, fuses, and battery selector switches. Ifthere is a choice at all, lower input voltages result inbetter efficiency.

• Maximum load current: There are two values toconsider. The peak load current (ILOAD(MAX))determines the instantaneous component stressesand filtering requirements, and thus drives outputcapacitor selection, inductor saturation rating, andthe design of the current-limit circuit. The continu-ous load current (ILOAD) determines the thermalstresses and thus drives the selection of inputcapacitors, MOSFETs, and other critical heat-con-tributing components.

OVP ENABLED

POWER-GOOD

FAULT

TARGET- 200mV

TARGET+ 300mV

FAULTLATCH

FB

OVPSOFT-STARTCOMPLETE

EN

UVP

ONE-SHOT

200μs

IN

CLK

OUT

POWER-GOOD AND FAULT PROTECTION

Figure 7. Power-Good and Fault Protection

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20 ______________________________________________________________________________________

• Switching frequency: This choice determines thebasic trade-off between size and efficiency. Theoptimal frequency is largely a function of maximuminput voltage due to MOSFET switching losses thatare proportional to frequency and VIN2. The opti-mum frequency is also a moving target, due torapid improvements in MOSFET technology that aremaking higher frequencies more practical.

• Inductor operating point: This choice providestrade-offs between size vs. efficiency and transientresponse vs. output noise. Low inductor values pro-vide better transient response and smaller physicalsize, but also result in lower efficiency and higheroutput noise due to increased ripple current. Theminimum practical inductor value is one that causesthe circuit to operate at the edge of critical conduc-tion (where the inductor current just touches zerowith every cycle at maximum load). Inductor valueslower than this grant no further size-reduction bene-fit. The optimum operating point is usually foundbetween 20% and 50% ripple current.

Inductor SelectionThe switching frequency and operating point (% ripplecurrent or LIR) determine the inductor value as follows:

Find a low-loss inductor having the lowest possible DCresistance that fits in the allotted dimensions. Ferritecores are often the best choice, although powderediron is inexpensive and can work well at 200kHz. Thecore must be large enough not to saturate at the peakinductor current (IPEAK):

Transient ResponseThe inductor ripple current impacts transient-responseperformance, especially at low VIN - VOUT differentials.Low inductor values allow the inductor current to slewfaster, replenishing charge removed from the output fil-ter capacitors by a sudden load step. The amount ofoutput sag is also a function of the maximum duty factor,

which can be calculated from the on-time and minimumoff-time. The worst-case output sag voltage can bedetermined by:

where tOFF(MIN) is the minimum off-time (see the ElectricalCharacteristics table).

The amount of overshoot due to stored inductor energywhen the load is removed can be calculated as:

Setting the Valley Current LimitThe minimum current-limit threshold must be highenough to support the maximum load current when thecurrent limit is at the minimum tolerance value. The val-ley of the inductor current occurs at ILOAD(MAX) minushalf the inductor ripple current (ΔIL); therefore:

where ILIMIT(LOW) equals the minimum current-limitthreshold voltage divided by 0.006.

The valley current-limit threshold is precisely 1/20 thevoltage seen at ILIM. Connect a resistive divider fromREF to ILIM to analog ground (AGND) to set a fixed val-ley current-limit threshold. The external 400mV to 2Vadjustment range corresponds to a 20mV to 100mV val-ley current-limit threshold. When adjusting the current-limit threshold, use 1% tolerance resistors and a dividercurrent of approximately 5μA to 10μA to prevent signifi-cant inaccuracy in the valley current-limit tolerance.

The MAX15035 uses the low-side MOSFET’s on-resis-tance as the current-sense element (RSENSE =RDS(ON)). A good general rule is to allow 0.5% addi-tional resistance for each degree celsius of tempera-ture rise, which must be included in the design marginunless the design includes an NTC thermistor in theILIM resistive voltage-divider to thermally compensatethe current-limit threshold.

I II

LIMIT LOW LOAD MAXL

( ) ( )> −Δ2

VI L

C VSOARLOAD MAX

OUT OUT≈

( )Δ ( )2

2

V

V tV

t

SAG

OUT SW

INOFF

=( ) ⎛

⎝⎜⎞⎠⎟

+L ILOAD(MAX)2

Δ (MMIN

OUT OUTIN OUT SW

INC V

V V tV

)⎡

⎣⎢

⎤

⎦⎥

−( )⎛

⎝⎜⎞

⎠⎟−2 ttOFF MIN( )

⎡

⎣⎢⎢

⎤

⎦⎥⎥

I II

PEAK LOAD MAXL= +( )

Δ2

LV V

f I LIRVV

IN OUT

SW LOAD MAX

OUT

IN=

−⎛

⎝⎜

⎞

⎠⎟

⎛

⎝⎜⎞

( ) ⎠⎠⎟

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______________________________________________________________________________________ 21

Foldback Current LimitIncluding an additional resistor between ILIM and theoutput automatically creates a current-limit threshold thatfolds back as the output voltage drops (see Figure 8).The foldback current limit helps limit the inductor currentunder fault conditions, but must be carefully designed toprovide reliable performance under normal conditions.The current-limit threshold must not be set too low, or thecontroller will not reliably power up. To ensure the con-troller powers up properly, the minimum current-limitthreshold (when VOUT = 0V) must always be greater thanthe maximum load during startup (which at least consistsof leakage currents), plus the maximum current requiredto charge the output capacitors:

ISTART = COUT x 1mV/μs + ILOAD(START)

Output Capacitor SelectionThe output filter capacitor must have low enough equiv-alent series resistance (ESR) to meet output ripple andload-transient requirements. Additionally, the ESRimpacts stability requirements. Capacitors with a highESR value (polymers/tantalums) do not need additionalexternal compensation components.

In core and chipset converters and other applicationswhere the output is subject to large-load transients, theoutput capacitor’s size typically depends on how muchESR is needed to prevent the output from dipping toolow under a load transient. Ignoring the sag due tofinite capacitance:

In low-power applications, the output capacitor’s sizeoften depends on how much ESR is needed to maintainan acceptable level of output ripple voltage. The outputripple voltage of a step-down controller equals the totalinductor ripple current multiplied by the output capaci-tor’s ESR. The maximum ESR to meet ripple require-ments is:

where fSW is the switching frequency.

RV f L

V V VVESR

IN SW

IN OUT OUTRIPPLE≤

× ×−( )

⎡

⎣⎢⎢

⎤

⎦⎥⎥

R RV

IESR PCBSTEP

LOAD MAX+( ) ≤

Δ ( )

C11μF

ENOFFON

VDD

VCC

C21μF

PGOOD

R10100kΩ

REFINR2

54.9kΩ

BST

LX

CBST0.1μF L1

COUT

TON

R397.6kΩ

GND/OPEN/REF/VDD

PGND

FB

HILO

REF

ILIM REF

RTON200kΩ

R549.4kΩ

R8100kΩ

R449.9kΩ

AGND

AGND

PWR

PWRPWR

AGND

AGNDPWR

AGND

AGND

2

34

36

4

37

29

18–26, EP3

30

5, 16, EP2

6–15

AGND

38

SKIP

35

32

33

OUTPUT1.50V 10A1.05V 7A

INPUT7V TO 24V

5V BIASSUPPLY

MAX15035

CIN

PWR

C31000pF

R149.9kΩ

3, 28, EP1

IN

Figure 8. Standard Application with Foldback Current-Limit Protection


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22 ______________________________________________________________________________________

With most chemistries (polymer, tantalum, aluminumelectrolytic), the actual capacitance value requiredrelates to the physical size needed to achieve low ESRand the chemistry limits of the selected capacitor tech-nology. Ceramic capacitors provide low ESR, but thecapacitance and voltage rating (after derating) aredetermined by the capacity needed to prevent VSAGand VSOAR from causing problems during load tran-sients. Generally, once enough capacitance is added tomeet the overshoot requirement, undershoot at the ris-ing load edge is no longer a problem (see the VSAG andVSOAR equations in the Transient Response section).Thus, the output capacitor selection requires carefullybalancing capacitor chemistry limitations (capacitancevs. ESR vs. voltage rating) and cost. See Figure 9.

Output Capacitor Stability ConsiderationsFor Quick-PWM controllers, stability is determined by thein-phase feedback ripple relative to the switching frequen-cy, which is typically dominated by the output ESR. Theboundary of instability is given by the following equation:

where COUT is the total output capacitance, RESR is thetotal ESR of the output capacitors, RPCB is the parasiticboard resistance between the output capacitors andfeedback sense point, and RCOMP is the effective resis-tance of the DC- or AC-coupled current-sense compen-sation (see Figure 11).

For a standard 300kHz application, the effective zerofrequency must be well below 95kHz, preferably below50kHz. With these frequency requirements, standardtantalum and polymer capacitors already commonlyused have typical ESR zero frequencies below 50kHz,allowing the stability requirements to be achieved with-out any additional current-sense compensation. In thestandard application circuit (Figure 1), the ESR neededto support a 15mVP-P ripple is 15mV/(10A x 0.3) =5mΩ. Two 330μF, 9mΩ polymer capacitors in parallelprovide 4.5mΩ (max) ESR and 1/(2π x 330μF x 9mΩ) =53kHz ESR zero frequency. See Figure 10.

fR C

R R R R

SW

EFF OUT

EFF ESR PCB COMP

π π≥

= + +

12

CIN

L1

COUT

PWR

PWR

PWR

BST

LX

IN

PGND

FB

AGND

AGND2fSW

RESRCOUT1

≥

STABILITY REQUIREMENT

MAX15035

OUTPUT

INPUT

Figure 9. Standard Application with Output Polymer or Tantalum

PCB PARASITIC RESISTANCE-SENSERESISTANCE FOR EVALUATION

OUTPUT VOLTAGE REMOTELYSENSED NEAR POINT OF LOAD

FEEDBACK RIPPLE IN PHASE WITH INDUCTOR CURRENT

CIN

L1

COUT

PWR

PWR

PWRRCOMP100Ω

PWR

BST

LX

IN

DH

PGND

FB

GND

AGND

2fSWRESRCOUT

1≥

fSWAND RCOMPCCOMP

1≥

STABILITY REQUIREMENT

MAX15035

CLOAD

PWR

OUTPUT

INPUT

CCOMP0.1μF

Figure 10. Remote-Sense Compensation for Stability and Noise Immunity

OPTION B: AC-COUPLED CURRENT-SENSE COMPENSATION

OPTION A: DC-COUPLED CURRENT-SENSE COMPENSATIONDC COMPENSATION<> FEWER COMPENSATION COMPONENTS<> CREATES OUTPUT LOAD LINE<> LESS OUTPUT CAPACITANCE REQUIRED FOR TRANSIENT RESPONSE

AC COMPENSATION<> NOT DEPENDENT ON ACTUAL DCR VALUE<> NO OUTPUT LOAD LINE

CIN

L

COUTRSENA

RSENB

CSENPWR

PWR

PWR

PWR

BST

LX

IN

PGND

FB

GND

AGND STABILITY REQUIREMENT

MAX15035

OUTPUT

INPUT


CIN

L

COUTRSEN

RCOMP

CCOMP

CSEN

PWR

PWR

PWR

PWR

BST

LX

IN

PGND

FB

GND

AGNDSTABILITY REQUIREMENT

MAX15035

OUTPUT

INPUT


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______________________________________________________________________________________ 23

Ceramic capacitors have a high-ESR zero frequency,but applications with sufficient current-sense compen-sation may still take advantage of the small size, lowESR, and high reliability of the ceramic chemistry. Usingthe inductor DCR, applications using ceramic outputcapacitors may be compensated using either a DCcompensation or AC compensation method (Figure 11).

The DC-coupling requires fewer external compensationcapacitors, but this also creates an output load line thatdepends on the inductor’s DCR (parasitic resistance).Alternatively, the current-sense information may be AC-coupled, allowing stability to be dependent only on theinductance value and compensation components andeliminating the DC load line.

Figure 11. Feedback Compensation for Ceramic Output Capacitors

LR R C

Cf

ANDLOASENA SENB SEN

OUTSW||( )

⎛

⎝⎜

⎞

⎠⎟ ≥ 1

2DDLINE

R RR R

SENB DCR

SENA SENB=

+

LR C

Cf

ANDR CfSEN SEN

OUTSW

COMP COMP⎛

⎝⎜⎞

⎠⎟≥ ≥1

21

SSW

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24 ______________________________________________________________________________________

When only using ceramic output capacitors, outputovershoot (VSOAR) typically determines the minimumoutput capacitance requirement. Their relatively lowcapacitance value may allow significant output over-shoot when stepping from full-load to no-load condi-tions, unless designed with a small inductance valueand high switching frequency to minimize the energytransferred from the inductor to the capacitor duringload-step recovery.

Unstable operation manifests itself in two related butdistinctly different ways: double pulsing and feedback-loop instability. Double pulsing occurs due to noise onthe output or because the ESR is so low that there isnot enough voltage ramp in the output voltage signal.This “fools” the error comparator into triggering a newcycle immediately after the minimum off-time periodhas expired. Double pulsing is more annoying thanharmful, resulting in nothing worse than increased out-put ripple. However, it can indicate the possible pres-ence of loop instability due to insufficient ESR. Loopinstability can result in oscillations at the output afterline or load steps. Such perturbations are usuallydamped, but can cause the output voltage to riseabove or fall below the tolerance limits.

The easiest method for checking stability is to apply avery fast zero-to-max load transient and carefullyobserve the output voltage-ripple envelope for over-shoot and ringing. It can help to simultaneously monitorthe inductor current with an AC current probe. Do notallow more than one cycle of ringing after the initialstep-response under/overshoot.

Input Capacitor SelectionThe input capacitor must meet the ripple currentrequirement (IRMS) imposed by the switching currents.The IRMS requirements may be determined by the fol-lowing equation:

The worst-case RMS current requirement occurs whenoperating with VIN = 2VOUT. At this point, the aboveequation simplifies to IRMS = 0.5 x ILOAD.

For most applications, nontantalum chemistries (ceramic,aluminum, or OS-CON) are preferred due to their resis-tance to inrush surge currents typical of systems with amechanical switch or connector in series with the input.If the Quick-PWM controller is operated as the secondstage of a two-stage power-conversion system, tanta-lum input capacitors are acceptable. In either configu-ration, choose an input capacitor that exhibits less than+10°C temperature rise at the RMS input current foroptimal circuit longevity.

Minimum Input-Voltage Requirementsand Dropout Performance

The output voltage-adjustable range for continuous-conduction operation is restricted by the nonadjustableminimum off-time one-shot. For best dropout perfor-mance, use the slower (200kHz) on-time settings. Whenworking with low-input voltages, the duty-factor limitmust be calculated using worst-case values for on- andoff-times. Manufacturing tolerances and internal propa-gation delays introduce an error to the on-times. Thiserror is greater at higher frequencies. Also, keep inmind that transient response performance of buck reg-ulators operated too close to dropout is poor, and bulkoutput capacitance must often be added (see the VSAGequation in the Quick-PWM Design Procedure section).

The absolute point of dropout is when the inductor cur-rent ramps down during the minimum off-time (ΔIDOWN)as much as it ramps up during the on-time (ΔIUP). Theratio h = ΔIUP/ΔIDOWN is an indicator of the ability toslew the inductor current higher in response toincreased load, and must always be greater than 1. Ash approaches 1, the absolute minimum dropout point,the inductor current cannot increase as much duringeach switching cycle and VSAG greatly increasesunless additional output capacitance is used.

A reasonable minimum value for h is 1.5, but adjustingthis up or down allows trade-offs between VSAG, outputcapacitance, and minimum operating voltage. For agiven value of h, the minimum operating voltage can becalculated as:

where VDROOP is the voltage-positioning droop, VCHG isthe parasitic voltage drop in the charge path, andtOFF(MIN) is from the Electrical Characteristics table. Theabsolute minimum input voltage is calculated with h = 1.

If the calculated VIN(MIN) is greater than the required min-imum input voltage, reduce the operating frequency oradd output capacitance to obtain an acceptable VSAG. Ifoperation near dropout is anticipated, calculate VSAG tobe sure of adequate transient response.

Dropout design example:

VOUT = 3.3V

fSW = 300kHz

tOFF(MIN) = 350ns

VDROOP = 0V

VCHG = 150mV (10A load)

h = 1.5

VV V V

h t fIN MIN

OUT DROOP CHG

OFF MIN SW( )

( )=

− +− ×( )1

⎛⎛

⎝⎜⎜

⎞

⎠⎟⎟

II

VV V VRMS

LOAD

INOUT IN OUT=

⎛

⎝⎜⎞

⎠⎟−( )

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______________________________________________________________________________________ 25

Calculating again with h = 1 gives the absolute limit ofdropout:

Therefore, VIN must be greater than 3.52V, even withvery large output capacitance, and a practical input volt-age with reasonable output capacitance would be 3.74V.

Applications InformationPCB Layout Guidelines

Careful PCB layout is critical to achieve low switchinglosses and clean, stable operation. The switchingpower stage requires particular attention. If possible,mount all the power components on the top side of theboard with their ground terminals flush against oneanother. Follow these guidelines for good PCB layout:

1) Keep the high-current paths short, especially at theground terminals. This is essential for stable, jitter-free operation.

2) Connect all analog grounds to a separate solidcopper plane, which connects to the AGND pin ofthe Quick-PWM controller. This includes the VCCbypass capacitor, REF bypass capacitors, REFINcomponents, and feedback compensation/dividers.

3) Keep the power traces and load connections short.This is essential for high efficiency. The use of thick

copper PCB (2oz vs. 1oz) can enhance full-loadefficiency by 1% or more. Correctly routing PCBtraces is a difficult task that must be approached interms of fractions of centimeters, where a singlemilliohms of excess trace resistance causes a mea-surable efficiency penalty.

4) Keep the power plane—especially LX—away fromsensitive analog areas (REF, REFIN, FB, ILIM).

Layout Procedure1) Place the power components first, with ground ter-

minals adjacent (CIN and COUT). If possible, makeall these connections on the top layer with wide,copper-filled areas.

2) Make the DC-DC controller ground connections asshown in Figure 1. This diagram can be viewed ashaving four separate ground planes: input/outputground, where all the high-power components go;the power ground plane, where the PGND pin andVDD bypass capacitor go; the controller’s analogground plane where sensitive analog components,the controller’s AGND pin, and VCC bypass capaci-tor go. The controller’s AGND plane must meet thePGND plane only at a single point directly beneaththe IC. This point must also be very close to the out-put capacitor ground terminal.

3) Connect the output power planes (VCORE and sys-tem ground planes) directly to the output filtercapacitor positive and negative terminals with multi-ple vias. Place the entire DC-DC converter circuitas close to the load as is practical.

VV V mV

ns kHzIN MIN( ).

( . )= − +

− × ×⎡ 3 3 0 1501 1 0 350 300⎣⎣⎢

⎤⎦⎥

= 3 52. V

VV V mV

ns kHzIN MIN( ).

( . )= − +

− × ×⎡ 3 3 0 1501 1 5 350 300⎣⎣⎢

⎤⎦⎥

= 3 74. V

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26 ______________________________________________________________________________________

Chip InformationTRANSISTOR COUNT: 7169

PROCESS: BiCMOS

Package InformationFor the latest package outline information and land patterns, goto www.maxim-ic.com/packages.

PACKAGE TYPE PACKAGE CODE DOCUMENT NO.

40 TQFN T4066-MCM 21-0177

http://pdfserv.maxim-ic.com/package_dwgs/21-0177.PDF

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Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses areimplied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 27

© 2008 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.

SPRINGER

Revision History

REVISION NUMBER

REVISION DATE

DESCRIPTIONPAGES

CHANGED

0 5/08 Initial release —

1 7/08 Modified Figure 1, Tables 1 and 2. 12

2 10/08 Updated Pin Description, Figure 1, and Detailed Description.11, 12, 13, 16,

18–21, 24

15a step-down regulator with internal switches

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